JP2000152620A - Switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To miniaturize a switching power supply, reduce its weight, and improve power conversion efficiency by varying a switching frequency according to the level of a secondary side DC output voltage and hence compositively controlling the conduction angle between the resonance impedance and the switching element of a primary side parallel resonance circuit, and performing the constant-voltage control of the secondary side output voltage. SOLUTION: When a switching frequency is to be varied, a period when a switching element Q1 is turned off is set to constant and a period when it is turned on is controlled variably in a circuit. More specifically, by performing operation so that the switching frequency is controlled variably as a constant- voltage control operation, it is regarded that resonance impedance control for switching output is made and at the same time the conduction angle control of the switching element at the switching period is made. Then, this complex control operation is achieved by a set of control circuit systems. By forming a switching converter in a voltage resonance system, a power conversion efficiency can be improved and miniaturized and its weight can be reduced.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. These switching converters are also called hard switching power supplies because the switching operation waveform is rectangular. On the other hand, switching power supply circuits using various resonant converters, which are called soft switching power supplies, are also known. The resonance type converter can easily obtain high power conversion efficiency, and can realize lower noise than the hard switching power supply because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 7 is a circuit diagram showing a configuration example of a power supply circuit as a hard switching power supply employing a so-called RCC (ringing choke converter) system. The power supply circuit shown in FIG.
Load power of 0.5 W or less, or as a so-called standby power supply provided separately from the main power supply.
The configuration corresponding to the condition of W or less is adopted.

In the power supply circuit shown in FIG. 1, a full-wave circuit comprising a bridge rectifier circuit Di and a smoothing capacitor Ci serves as a rectifying and smoothing circuit for receiving a commercial AC power supply AC (AC input voltage VAC) to obtain a DC input voltage. A rectifier circuit is provided, and generates a rectified smoothed voltage Ei corresponding to a level approximately one time of the AC input voltage VAC, and supplies the same as a DC input voltage.

[0005] The switching element Q1 obtains a switching output by intermittently receiving the DC input voltage.
In this case, for example, a bipolar transistor is used as the switching element Q1. The collector of switching element Q1 is connected to the positive terminal (rectified smoothing voltage Ei line) of smoothing capacitor Ci via the serial connection of primary winding N1 of converter transformer CVT. The base of the switching element Q1 is connected to a rectified smoothed voltage Ei via a starting resistor RS.
By being connected to the line, the base current at the time of startup is supplied from the rectified smoothed voltage Ei. Further, a series connection circuit of a base current limiting resistor RB, a diode D4, and a driving winding NB is connected to a base of the switching element Q1. The end of the drive winding NB is connected to the primary side ground. A capacitor CB is connected in parallel with the diode D4. These base current limiting resistor RB, diode D4, drive winding NB, and capacitor CB form a self-excited oscillation drive circuit that oscillates and drives switching element Q1 in a self-excited manner. The emitter of the switching element Q1 is connected to the primary side ground via the resistor R7.

[0006] The converter transformer CVT is provided for transmitting the switching output obtained on the primary side to the secondary side, and is wound with a primary winding N1 and a secondary winding N2 as shown in the figure. . In this case, the drive winding NB for self-excited oscillation is also wound around the primary side.

The secondary winding N2 of the converter transformer CVT
Is connected to a half-wave rectifier circuit composed of a rectifier diode DO1 and a smoothing capacitor CO1, as shown in the figure, and generates and outputs a secondary-side DC output voltage EO1. The secondary side DC output voltage EO1 is supplied to a load (not shown) and is input as a detection voltage to a control circuit 1 for constant voltage control.

The control circuit 1 has a photocoupler PC to insulate the secondary side and the primary side in a DC manner. In the control circuit 1 on the secondary side, a voltage value obtained by dividing the secondary side DC output voltage EO1 by the resistors R3 and R4 is input to the detection input of the detection element Q3. One end of the detection element Q3 is connected to the secondary DC output voltage EO1 via a series connection of the resistor R1 and the photodiode PD of the photocoupler PC, and the other end is grounded to the secondary ground. A series connection circuit of a capacitor C11 and a resistor R2 is connected in parallel to the resistor R4. A series connection circuit of a capacitor C12 and a resistor R5 is also connected in parallel between a connection point between the resistors R4 and R3 and a connection point between the detection element Q3 and the photodiode PD. In the primary side control circuit 1, a photocoupler PC
Are provided. The collector of the phototransistor PD is connected to a half-wave rectifier circuit (D3, C3) for rectifying and smoothing the alternating voltage excited by the drive winding NB, so that the low-voltage DC obtained by the half-wave rectifier circuit is obtained. Voltage is supplied as operating power. The emitter of the phototransistor PD is a transistor Q as an amplifier.
Connected to the base of 4. Also, the phototransistor P
A series connection circuit of a resistor R8 and a Zener diode ZD is inserted between the emitter of D and the connection point between the drive winding NB and the diode D4, as shown in the figure. The collector of transistor Q4 is connected to the base of switching element Q1, and the emitter is grounded to the primary side ground. The base of the transistor Q4 is connected to a resistor R6 ///
It is grounded to the primary side ground through the resistor R7 from the parallel connection circuit of the capacitor C13.

The reset circuit 10 includes a resistor R
RS // CRS parallel connection circuit and diode DRS
Are connected in series and provided in parallel with the primary winding N1. The snubber circuit 11 is formed by connecting a series connection circuit of a capacitor Csn and a resistor Rsn in parallel between the collector of the switching element Q1 and the primary side ground. The reset circuit 10 and the snubber circuit 11 are required for suppressing a spike voltage generated when the switching element Q1 is turned off.

The switching operation of the RCC according to the above configuration is well known and will not be described in detail here. However, after the switching element Q1 is turned on by the current applied through the starting resistor RS, the switching operation is performed. Is started, magnetic energy is stored in the primary winding N1 of the converter transformer when the switching element Q1 is on, and the magnetic energy stored in the primary winding N1 is discharged to the secondary side when the switching element Q1 is off. By repeating this operation, an output voltage is obtained on the secondary side.

In the control circuit 1, the secondary side DC output voltage EO1
The amount of current that conducts the detection element Q3 is changed in accordance with the current. The photocoupler PC variably controls the base current that flows through the transistor Q4 on the primary side in accordance with the amount of current that conducts the detection element Q3. This changes the collector current of the transistor Q4. Since the collector of the transistor Q4 is connected to the base of the switching element Q1, a change in the collector current of the transistor Q4 causes the self-excited oscillation drive circuit to switch the switching element Q1.
The base current (the amount of drive current) flowing to one base changes. As a result, the ON time of the switching element Q1 is varied, and as a result, the switching frequency is variably controlled. The constant voltage control is performed by such an operation. In the configuration shown in FIG. 7, a constant voltage effect is obtained by controlling the switching frequency to increase with an increase in the AC input voltage or a decrease in the load power. The control range is set to a wide range of fs = 25 KHz to 250 KHz due to the low control sensitivity.

FIG. 8 is a waveform chart showing the operation of the main part of the power supply circuit shown in FIG. The voltage Vcp (between the collector and the primary side ground) of the switching element Q1 is shown in FIG.
As shown in (a), during the period TON when the switching element Q1 is on, it is at the 0 level, and during the period TOFF when it is off.
Has a waveform from which a rectangular pulse can be obtained. This FIG.
As can be seen from the pulse waveform shape of the voltage Vcp shown in (a), when the switching element Q1 is turned off, the leakage inductance component of the converter transformer CVT and
A spike voltage is generated by the distributed capacitance (capacitance) between the windings wound around the converter transformer CVT. The reset circuit 10 and the snubber circuit 11 described above suppress the waveform portion as the spike voltage. It is provided in. The collector current Icp flowing into the collector of the switching element Q1 in response to the switching operation of the switching element Q1 has a waveform shown in FIG.
The operation flows during the period TON. The rectified current flowing from the secondary winding N2 into the rectifier diode DO1 flows during the period TOFF when the switching element Q1 is turned off,
This corresponds to the switching operation of C.

As in the case of the power supply circuit shown in FIG. 7, a light load condition in which the load power is 50 W or less, or a load power of 0.
As another example of the power supply circuit corresponding to the condition of 5 W or less, FIG. 9 shows a switching power supply circuit that can be configured based on the invention proposed by the present applicant. In FIG. 9, the same components as those in FIG. 7 are denoted by the same reference numerals, and the description of the same components will be omitted.

The power supply circuit shown in FIG. 9 includes a self-excited current resonance type converter using the rectified smoothed voltage Ei as an operation power supply. The switching converter of this power supply circuit is connected in such a manner that two switching elements Q1 and Q2 are half-bridge-coupled as shown in the figure, and then inserted between the connection point on the positive side of the smoothing capacitor Ci and the ground. I have. Starting resistors RS1, RS2 are provided between the collectors and bases of the switching elements Q1, Q2, respectively.
Is inserted. A clamp diode D is provided between each base and emitter of the switching elements Q1 and Q2.
D1 and DD2 are inserted. A resonance capacitor CB1 and a base current limiting resistor RB are provided between the base of the switching element Q1 and the collector of the switching element Q2.
1. A series connection circuit composed of the drive winding NB1 is inserted.
The resonance capacitor CB1 forms a series resonance circuit for self-excited oscillation together with its own capacitance and an inductance LB1 of the drive winding NB1 described below, thereby determining the switching frequency of the switching element Q1. Similarly,
A resonance capacitor CB2 and a base current limiting resistor R are provided between the base of the switching element Q2 and the primary side ground.
A series connection circuit composed of B2 and a drive winding NB2 is inserted, and forms a series resonance circuit for self-excited oscillation together with a resonance capacitor CB2 and an inductance LB2 of the drive winding NB2.
The switching frequency of the switching element Q2 is determined.

Further, capacitors Cc1 and Cc2 for partial resonance are connected in parallel between the respective collectors and emitters of the switching elements Q1 and Q2. The partial resonance capacitors Cc1, Cc2 are connected to the switching elements Q1,
Although provided to absorb the switching noise of Q2, here, when the switching elements Q1 and Q2 are turned off, the switching frequency is variably controlled by a constant voltage control operation performed as described later. It also has an action for obtaining a zero voltage switching operation. Thereby, the switching loss is reduced.

Drive transformer PRT (Power Regulati
ng Transformer) is provided to drive the switching elements Q1 and Q2 and to perform constant voltage control by variably controlling the switching frequency. In this case, the drive windings NB1 and NB2 and the resonance current detection are performed. Winding N
D is wound, and a control winding N is provided for each of these windings.
C is an orthogonal saturable reactor wound in the orthogonal direction. One end of the drive winding NB1 of the drive transformer PRT is connected to the base of the switching element Q1 via a series connection of a resonance capacitor CB1 and a resistor RB1, and the other end is connected to the emitter of the switching element Q1. One end of the drive winding NB2 is grounded, and the other end is connected to the base of the switching element Q2 through a series connection of a resonance capacitor CB2 and a resistor RB2. The drive winding NB1 and the drive winding NB2 are wound so that voltages of opposite polarities are generated.

Insulated converter transformer PIT (Power Is
olation Transformer) is the switching element Q1, Q2
Is transmitted to the secondary side. In this case, one end of the primary winding N1 of the insulating converter transformer PIT is
It is connected to the contact point (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND, and the other end is grounded to the primary side earth via the series resonance capacitor C1. , A switching output is obtained.

In the above connection form, the series resonance capacitor C1 and the primary winding N1 are connected in series. However, the insulation converter transformer including the capacitance of the series resonance capacitor C1 and the primary winding N1 (series resonance winding). A leakage resonance (leakage inductance) component of the PIT forms a series resonance circuit for making the operation of the switching converter a current resonance type.

On the secondary side of the insulating converter transformer PIT in this figure, an alternating voltage of a switching cycle is excited in the secondary winding N2 by the switching output supplied to the primary winding N1. The secondary winding N2 is provided with taps, and rectifier diodes DO1, DO2,
By connecting DO3 and the smoothing capacitors CO1 and CO2 as shown in the figure, a full-wave rectifier circuit consisting of [rectifier diodes DO1, DO2 and smoothing capacitor CO1] and [rectifier diode D
O3, a smoothing capacitor CO2].
A set of rectifier circuits is provided. [Rectifier diode DO
1, DO2, smoothing capacitor CO1], generates a DC output voltage EO1 by full-wave rectification operation, and supplies power to a subsequent stage load (not shown). Further, the DC output voltage EO1 is also branched and input to the control circuit 1 and used as a detection voltage of the control circuit 1. In this case, a half-wave rectifier circuit composed of [rectifier diode DO3, smoothing capacitor CO2] generates a DC output voltage by a half-wave rectification operation, and is used as an operation power source of control circuit 1 described later. In this case, for protection in the case of a load short circuit, for example, I.sub.0 is applied between the DC output voltage EO1 and the load.
A fuse IL such as a C link is inserted.

The control circuit 1 has, for example, a secondary side output voltage EO1
DC current whose level varies according to the level of
Control winding NC of drive transformer PRT as control current
To perform constant voltage control as described later. The control circuit 1 shown in this figure inputs a voltage obtained by dividing the secondary output voltage EO1 by the resistors R3 and R4 to the detection element Q3. The cathode of the detection element Q3 is connected to the positive electrode of the smoothing capacitor CO2 through the serial connection of the control winding NC, and the anode is connected to the secondary side ground.
A series connection circuit of a capacitor C12 and a resistor R5 is inserted between the positive electrode of the smoothing capacitor CO1 and the connection point of the resistors R3 and R4, and the cathode of the detection element Q3 and the resistors R3 and R4 are connected.
A series connection circuit of a capacitor C11 and a resistor R2 is inserted between the four connection points.

The switching operation of the power supply circuit according to the above configuration is as follows. First, when a commercial AC power supply is turned on, the switching elements Q1 and Q2 are activated, for example, via starting resistors RS1 and RS2.
2 is supplied with a starting current. For example, if the switching element Q1 is turned on first,
The switching element Q2 is controlled to be turned off.
As an output of the switching element Q1, a resonance current flows through the resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1, and when the resonance current becomes zero, the switching element Q2 is turned on and the switching element Q1 is turned on. It is controlled to be off. Then, a resonance current flows in the opposite direction through the switching element Q2. Or later,
A self-excited switching operation in which the switching elements Q1 and Q2 are turned on alternately is started. As described above, by using the terminal voltage of the smoothing capacitor Ci as the operating power source, the switching elements Q1 and Q2 alternately open and close,
A drive current close to the resonance current waveform is supplied to the primary winding N1 of the insulating converter transformer PIT, and an alternating output is obtained at the secondary winding N2.

The constant voltage control by the drive transformer PRT is performed as follows. For example, in the configuration of the control circuit 1 described above, assuming that the secondary output voltage EO1 fluctuates so as to increase, the level of the control current flowing through the control winding NC also depends on the increase in the secondary output voltage EO. Is controlled to be higher. The control current tends to approach a saturation state in the drive transformer PRT due to a change in magnetic flux generated in the drive transformer PRT, and acts to reduce the inductance of the drive windings NB1 and NB2. The switching frequency is controlled so as to increase by changing the condition of the self-excited oscillation circuit. In this power supply circuit, the switching frequency is set in a frequency region higher than the resonance frequency of the series resonance circuit of the series resonance capacitor C1 and the primary winding N1 (upper side control). Then, the switching frequency is separated from the resonance frequency of the series resonance circuit. This allows
The resonance impedance of the series resonance circuit with respect to the switching output increases. By increasing the resonance impedance in this way, the primary winding N1 of the primary side series resonance circuit
As a result, the secondary side output voltage is suppressed, and constant voltage control is achieved (switching frequency control method).

FIG. 10 is a waveform chart showing the operation of the main part of the power supply circuit having the configuration shown in FIG. 10A to 10D show the maximum load power (Poma).
x), operating waveforms of the respective units at the time of the minimum guaranteed AC input voltage (VAC min), and FIGS. 10 (e) to 10 (h) show the minimum load power (Pomin) and the maximum guaranteed AC input voltage (VAC max), respectively. FIG. 10 (a) to time
The operation waveform of the same part as (d) is shown.

When the switching element Q2 performs a switching operation, the collector-emitter voltage Vcp obtained between the collector and the emitter of the switching element Q2 becomes
As shown in FIGS. 10A and 10E, the switching element Q
In the period TON in which 2 is on, a level of 0 is obtained, and in the period TOFF in which it is off, a rectangular waveform is obtained. As can be seen by comparing the collector-emitter voltage Vcp shown in FIGS. 10A and 10E, the maximum load power (Pomax) and the minimum guaranteed AC input voltage (VAC m
In), when the minimum load power (Pomin) and the maximum guaranteed AC input voltage (VAC max), the switching frequency is controlled to be higher by the above-described constant voltage control operation. At this time, the switching element Q
The collector current Icp flowing through the collector of FIG.
(B) As shown in FIG.
When turned OFF, a waveform having a 0 level is obtained. The switching output current (primary-side series resonance current) I1 flowing through the primary winding N1 and the series resonance capacitor C1 is shown in FIG.
(C) As shown in (g), the current waveform substantially corresponds to the switching frequency. Here, when the switching frequency is low, the collector current Icp and the primary side series resonance current I1 have a sinusoidal waveform corresponding to the current resonance type as shown in FIGS. 10B and 10C. However, as the switching frequency increases, the waveform becomes closer to a sawtooth wave as shown in FIGS. The switching element Q2 has a waveform that is 180 ° shifted from the waveforms shown in FIGS. 10 (a) to 10 (c) and FIGS. 10 (e) to 10 (g). On the secondary side, the rectifier diode DO2 is turned on substantially at the timing of the period TON during which the switching element Q2 is turned on, so that the rectified current I2 flowing from the secondary winding N2 to the rectifier diode DO2 is as shown in FIG. d) The waveform shown in (h) is obtained. The operation of the rectifier diode DO1 is also shown in FIG.
The waveform shown in (h) is shifted by 180 °.

[0025]

The RC shown in FIG.
In the power supply circuit of the C system, the switching frequency is variably controlled as the constant voltage control operation as described above. However, since the control sensitivity for the constant voltage control is low, as described above, the variable range of the switching frequency fs is changed. Is fs = 25
A relatively wide range of KHz to 250 KHz. For this reason, when the switching frequency is lowered at the time of the minimum load power, the switching loss is increased and the power conversion efficiency is significantly reduced. Further, in order to suppress the spike voltage when the switching element is turned off, the power loss is increased by connecting the reset circuit 10 and the snubber circuit 11. Further, as shown in FIG. 8A, the alternating voltage generated by the switching operation is a rectangular pulse, and switching noise occurs at the time of turn-on and at the time of turn-off. For this reason, for example, the condition of the load power that can be put to practical use as a power supply for video equipment is about 1 W or less, and is actually limited to the use as a standby power supply with a load power of about 0.5 W or less. .

On the other hand, in the switching power supply circuit shown in FIG. 9, a current resonance type converter having a half-bridge coupling and having a partial resonance capacitor connected between the collector and the emitter of the two switching elements is provided. This realizes a so-called zero volt (zero voltage) switching operation when the switching element is turned off. Therefore, compared to the power supply circuit shown in FIG. 7, the noise is lower and the power conversion efficiency is improved.

However, also in the switching power supply circuit shown in FIG. 9, the reactive power becomes large when the switching frequency becomes low at the time of the minimum load power, and the power conversion efficiency becomes large, for example, about 60%. descend. Since the power supply circuit shown in FIG. 9 employs a self-excited current resonance type converter in which two switching elements are half-bridge-coupled, a switching circuit system having two self-oscillation driving circuits is provided. Need to be formed, the number of components increases accordingly, and there is a limit in reducing the size and weight of the power supply board. Further, the circuit configuration shown in FIG. 9 does not have a load short-circuit protection function. That is,
When the load is short-circuited, the control current (the amount of direct current) flowing through the control winding NC of the orthogonal control transformer PRT becomes almost zero, but this causes the switching frequency to drop to almost the lower limit in the control range, and the primary The primary side series resonance current I1 flowing in the side series resonance circuit also tends to increase. In this state, the heat generated by the switching loss in the switching elements Q1 and Q2 increases to a non-negligible level, and in some cases, may be destroyed due to thermal runaway. Therefore, for example, as shown in FIG. 9, it is necessary to insert a fuse IL into the secondary-side DC output voltage line so that the secondary-side DC output voltage and the load are disconnected when the load is short-circuited. This also causes an increase in circuit board size and a reduction in power conversion efficiency.

[0028]

SUMMARY OF THE INVENTION In view of the above-mentioned problems, the present invention has been made to reduce the size and weight of a switching power supply circuit corresponding to relatively light load conditions such as a load power of 50 W or less, and to reduce the power consumption. The purpose is to improve the conversion efficiency.

Therefore, a rectifying / smoothing means for receiving a commercial AC power supply, generating a rectified smoothed voltage having a level corresponding to the same level as that of the commercial AC power supply, and outputting the rectified smoothed voltage as a DC input voltage; The required coupling coefficient can be obtained by the absence of the converter converter, and the primary winding and the secondary winding wound around the core are in the polarity operation mode. A switching means for intermittently inputting the input voltage by a switching element and outputting the voltage to the primary winding of the insulated converter transformer, and at least a leakage inductance component including the primary winding of the insulated converter transformer and a capacitance of a parallel resonant capacitor. A primary-side parallel resonance circuit formed to make the operation of the switching means a voltage resonance type; DC output voltage generation configured to be able to generate a secondary DC output voltage corresponding to almost equal to the input voltage level by inputting the alternating voltage obtained to the secondary winding of the Means for controlling the resonance impedance of the primary side parallel resonance circuit and the conduction angle of the switching element in a complex manner by varying the switching frequency of the switching element according to the level of the secondary side DC output voltage. The switching power supply circuit is configured to include a constant voltage control unit configured to perform constant voltage control on the side output voltage.

According to the above configuration, the primary side is provided with a voltage resonance type converter, the isolation converter transformer is loosely coupled, and the secondary side generates a secondary side DC output voltage by a half-wave rectifier circuit. To supply power to the load. Further, in the configuration of the constant voltage control according to the above configuration, by changing the switching frequency according to the secondary side output voltage level, the resonance impedance of the primary side parallel resonance circuit and the conduction angle of the switching element are simultaneously controlled. As a result, the control sensitivity is improved by the composite control operation.

[0031]

FIG. 1 is a circuit diagram showing the configuration of a switching power supply circuit according to an embodiment of the present invention. In this figure, the same parts as those of the power supply circuit shown in FIGS. 7 and 9 are denoted by the same reference numerals, and the description of the same components will be omitted here.

For the primary side of the power supply circuit shown in FIG.
A self-excited voltage resonance type switching converter having one switching element Q1 is provided. In this case, a high withstand voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1. In this case, the base of the switching element Q1 is connected to the positive electrode side of the smoothing capacitor Ci1 (rectified smoothing voltage 2Ei) through a series connection of the base current limiting resistor RB and the starting resistor RS, so that the base current at the time of starting is increased. Is obtained from the rectifying and smoothing line. A base current limiting resistor RB, a resonance capacitor CB, and a detection drive winding NB are provided between the base of the switching element Q1 and the temporary ground.
The self-excited oscillation driving resonance circuit (self-excited oscillation driving circuit) composed of the series connection circuit of FIG. Switching element Q1
After being driven by the starting current, the switching is driven by the driving current applied to the base from the self-excited oscillation driving circuit. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a damper current flowing when the switching element Q1 is off. The collector of the switching element Q1 is connected to one end of the primary winding N1 of the insulation converter transformer PIT, and the emitter is grounded, so that the switching output of the switching element Q1 is transmitted to the primary winding N1.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Is the capacitance of itself and the leakage inductance L1 on the primary winding N1 side of the isolated converter transformer PIT.
Thus, a primary side parallel resonance circuit for forming the switching operation of the switching element Q1 into a voltage resonance type is formed. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vcr across the resonance capacitor Cr actually becomes a sinusoidal pulse waveform due to the action of the parallel resonance circuit, and the voltage resonance type operation is performed. Is obtained.

The orthogonal control transformer PRT shown in FIG.
Is a saturable reactor on which a detection winding ND, a drive winding NB, and a control winding NC are wound. The structure of the orthogonal control transformer PRT is, as shown in FIG.
Two U-shaped cores 201, 2 having two magnetic legs
The two-dimensional core 200 is formed so that the ends of the magnetic legs 02 are joined to each other. Then, the detection winding N is applied to the predetermined two magnetic legs of the three-dimensional core 200 in the same winding direction.
D, the drive winding NB is wound, and the control winding NC is wound in a direction orthogonal to the detection winding ND and the drive winding NB.

In this case, the detection winding ND of the orthogonal control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching of the switching element Q1 is performed. The output is transmitted to the detection winding ND via the primary winding N1.
In the orthogonal control transformer PRT, the drive winding NB is excited by the switching output obtained from the detection winding ND, and an alternating voltage is generated in the drive winding NB. This alternating voltage becomes a source of a driving voltage in the self-excited oscillation driving circuit.

The control circuit 1 shown in FIG. 3 operates to change the level of the control current (DC current) flowing through the control winding NC in accordance with the level of the input secondary-side DC output voltage EO1. Note that the internal configuration thereof may be substantially the same as the configuration of the control circuit 1 shown in FIG. 9, for example.

By the operation of the control circuit 1, the level of the control current (DC current) flowing through the control winding NC is changed in accordance with the change in the secondary-side DC output voltage level, so that it is wound around the orthogonal control transformer PRT. The inductance LB of the driven winding NB is variably controlled. Thereby, the resonance condition of the series resonance circuit in the self-excited oscillation drive circuit for the switching element Q1 formed including the inductance LB of the drive winding NB changes. This is an operation of varying the switching frequency of the switching element Q1, as will be described later with reference to FIG. 5, and this operation has the effect of stabilizing the secondary-side DC output voltage.

As shown in FIG. 3, the insulating converter transformer PIT according to the present embodiment includes an EE-type core 100 in which E-type cores 101 and 102 made of, for example, a ferrite material are combined so that their magnetic legs face each other. With respect to the center magnetic leg of the EE type core 100, actually, a primary winding N1 and a secondary winding N2 (using a divided bobbin in which a winding portion is divided into a primary side and a secondary side). And N2A) are wound separately. In this case, no gap is formed with respect to the center magnetic leg, so that a loose coupling state is obtained to a certain required saturation state. The coupling coefficient k in this case is, for example, k ≒ 0.90.

In this case, the insulation converter transformer PIT
In, an alternating voltage is excited in the secondary winding N2 by the switching output transmitted to the primary winding N1. In the case of the present embodiment, a tap is provided for the secondary winding N2 as shown in the figure, and an anode of a rectifier diode DO1 connected in series to the tap output is connected. The cathode of the rectifier diode DO1 is a smoothing capacitor C
The positive terminal of O1 is connected to the positive terminal, and the negative terminal of the smoothing capacitor CO1 is connected to the secondary side ground. In other words, a half-wave rectification is performed by inputting the alternating voltage obtained in the tap winding of the secondary winding N2 and performing a half-wave rectification to obtain the secondary-side DC output voltage EO1 by a set of [rectifying diode DO1, smoothing capacitor CO1]. A rectifier circuit is formed. The secondary-side DC output voltage EO1 is supplied to a load (not shown) and is branched and input as a detection voltage of the control circuit 1 described above. Also, by connecting the anode of the rectifier diode DO2 to the end of the secondary winding N2 at the beginning of the winding and connecting the cathode to the positive electrode of the smoothing capacitor CO2, the set of [rectifying diode DO1, smoothing capacitor CO1] can be obtained. Is formed. This [rectifier diode DO1, smoothing capacitor CO1]
Is a secondary-side DC output voltage E
O2 is generated, and in this case, is supplied as the operating power of the control circuit 1.

Here, in the isolated converter transformer PIT having the structure shown in FIG. 3, the polarities (winding directions) of the primary winding N1 and the secondary winding N2 and the rectifier diodes (DO1, D0).
O2), the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 is + M (additive mode; forward method) and -M (depolarization). Mode; flyback method). For example, the mutual inductance is + M in the operation when the connection form shown in FIG. 4A is adopted, and the mutual inductance is -M in the operation when the connection form shown in FIG. 4B is adopted.
In the circuit shown in this figure, the polarities of the primary winding N1 and the secondary winding N2 are in an additive polarity mode.

In this embodiment, a secondary side parallel resonance capacitor C2 is provided for the secondary winding N2. Thereby, a parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary side parallel resonance capacitor C2. The alternating voltage excited by the secondary winding N2 by this parallel resonance circuit becomes a resonance voltage. That is, a voltage resonance operation is obtained on the secondary side. That is, in the power supply circuit of the present embodiment, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is also formed by the secondary winding N2 and the parallel resonance capacitor C2. Parallel resonance circuit is provided. In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

As described above, when the secondary side parallel resonance capacitor C2 is provided for the secondary winding N2, the electric power during the rectification operation on the secondary side is generated by the resonance operation of the secondary side parallel resonance circuit. Because of the increase, the load power that can be handled can be increased. For example, in the case of the power supply circuit according to the present embodiment, it becomes possible to cope with a load power of 1 W to 50 W by inserting the secondary side parallel resonance capacitor C2. However, in the case of a load power of 1 W or less, such as when using a standby power supply, the load power can be adjusted by not inserting the secondary-side parallel resonance capacitor C2.

FIG. 5 is a waveform chart showing the operation of the main part of the power supply circuit having the structure described with reference to FIGS. 5A to 5E show the maximum load power (Po).
max = 50W), the minimum guaranteed AC input voltage (VAC mi)
5 shows the operation waveforms of the respective parts when n = 80 V).
(F) to (j) respectively show the minimum load power (Pomi)
n (Po = 0)), the maximum guaranteed AC input voltage (VAC ma)
5 shows operation waveforms of the same part as in FIGS. 5A to 5E at time x). 5 (b), (c), (d), and (e), the waveforms indicated by broken lines are the secondary-side parallel resonance capacitors C.
The operation obtained in the case of the configuration in which 2 is omitted is shown.
5 (b), (c), (d) and (e), the waveforms shown by solid lines are the operations when the secondary parallel resonance capacitor C2 is connected.

When the switching element Q1 performs a switching operation, the resonance voltage Vcr obtained between the collector and the emitter of the switching element Q1 is as shown in FIGS.
As shown in the figure, a waveform having a rectangular pulse shape is obtained during the period TON when the switching element Q1 is on and during the period TOFF when the switching element Q1 is off. FIG.
(A) As can be seen by comparing the resonance voltage Vcr shown in (f), the minimum load power (Pomin = 0) with respect to the maximum load power (Pomax = 50 W) and the minimum guaranteed AC input voltage (VAC min = 80 V). ), When the maximum guaranteed AC input voltage (VAC max = 288 V), the switching frequency is controlled to be higher by the above-described constant voltage control operation. At this time, the collector current Icp flowing through the collector of the switching element Q1 is as shown in FIG.
As shown in (g), a waveform which flows into the period TON and becomes 0 level during the period TOFF is obtained. The primary winding N1
The switching output current I1 flowing through
(C) As shown in (h), the alternating current substantially corresponds to the switching frequency, and a waveform close to a sine wave is obtained by the action of the primary side parallel resonance circuit.
In addition, the rectification operation on the secondary side in the present embodiment is described in FIG.
As described with reference to the above, the additional polarity mode (forward method) is set. This is shown as an operation in which the rectifier diodes DO1 and DO2 are turned on and the rectified current I2 flows substantially corresponding to the period TON during which the switching element Q1 is turned on, as shown in FIGS. ing. Also, the secondary-side parallel resonance voltage V generated in the secondary-side parallel resonance circuit
2 is when the rectifier diodes DO1, DO2 are not conducting (off)
Has a sine wave in the direction of negative polarity and a rectifier diode D
When O1 and DO2 are conducting (on), the waveform has a positive polarity and is clamped at the level of the DC output voltage EO (EO1, EO2).

In the case where the parallel resonance capacitor C2 is not inserted, the maximum load power Pomax = 50
W, when the minimum guaranteed AC input voltage VAC = 80 V,
The switching operation waveforms (Icp, I1, V2, I2) change as shown by the broken lines in (b) to (e).

As can be seen from the above description, the power supply circuit of the present embodiment includes a voltage resonance type converter (parallel resonance circuit) on the primary side, and a parallel resonance circuit and a half-wave rectification circuit on the secondary side. And a composite resonance type switching converter provided with: FIG. 6 shows the switching frequency fs and the secondary side DC output voltage EO in the present embodiment.
(E01). In this case, the horizontal axis indicates the switching frequency, and the vertical axis indicates the level of the secondary side DC output voltage EO. As can be seen from the resonance curve shown by the solid line, in the power supply circuit of the present embodiment shown in FIG. 1, the secondary side DC output voltage EO1 is changed to a predetermined level (for example, in response to a load change or a change in DC input voltage). For example, to stabilize the switching frequency fs at about 100 V
The control is performed in a range of Δ100 KHz from Hz to 200 KHz. On the other hand, for example, in the case of the power supply circuit shown in FIG. 7, in order to make the secondary DC output voltage constant, the switching frequency fs is set to, for example, 2 as described above.
It is necessary to control within a range of Δ225 KHz from 5 KHz to 250 KHz.

The reason why the control range of the switching frequency is reduced in the present embodiment as described above is as follows. In the present embodiment, the control circuit 1
As described above, the switching voltage of the switching element Q1 is variably controlled by the operation of the constant voltage control circuit system including the quadrature control transformer. This operation is also shown in FIG. 5. For example, as can be seen by comparing the waveforms (Vcr, Icp) of FIGS. 5 (a) and 5 (b) with FIGS. In making this change, this circuit is configured such that the period TOFF during which the switching element Q1 is off is fixed, and the period TON during which the switching element Q1 is on is variably controlled. That is, in the present embodiment, as the constant voltage control operation, the resonance frequency control for the switching output is performed by operating to variably control the switching frequency, and at the same time, the conduction angle control of the switching element in the switching cycle ( PWM control)
Can also be seen as doing. This complex control operation is realized by a set of control circuit systems. Actually, when the operation waveforms shown in FIGS. 5A and 5B correspond to Pomax and VAC = 80 V, FIG.
Po = 0 and VAC = 28 corresponding to the operation waveform shown in (g).
At 8 V, the period TON is shortened according to the switching frequency. Accordingly, the amount of current I1 flowing from the smoothing capacitor Ci to the voltage resonance type converter also increases as shown in FIG.
(C) → the transition is limited as shown in FIG. 4 (h), thereby improving the control sensitivity.

In FIG. 6, the switching frequency fs and the parallel resonance frequency fo1 of the primary side parallel resonance circuit and the parallel resonance frequency fo2 of the secondary side parallel resonance circuit with respect to the switching frequency fs are shown.
Here, for example, as shown in the figure, the parallel resonance frequency fo1 and the series resonance frequency fo2 are set to 80 KHz.
If the inductance and the capacitance are selected so as to be equal in the vicinity, the switching frequency control operation (constant voltage control operation) operates so as to simultaneously control the resonance impedance of the two sets of parallel resonance circuits, and the secondary side An operation of variably controlling the output voltage is obtained.
Such an operation also greatly improves the control sensitivity. By improving the control sensitivity as described above, the substantial control range can be expanded in the power supply circuit according to the present embodiment. This, as mentioned earlier,
The variable width of the switching frequency can be made smaller than before.

The pulses obtained during the period TOFF as the resonance voltage Vcr shown in FIGS. 5A and 5F are parallel to the DC input voltage (rectified smoothed voltage Ei) on the primary side of the voltage resonance type converter. This is caused by the action of the impedance of the resonance circuit. Assuming that the level Lvcr of the pulse of the resonance voltage Vcr is represented by Ei as the rectified smoothed voltage level and TOFF and TON as the lengths of the OFF period and the ON period within one switching cycle of the switching element Q1, respectively, Lvcr = Ei ｛ 1+ (π / 2) · (TON / TOFF)｝ (Expression 1) Here, it is assumed that the power supply circuit of the present embodiment is shared by the AC 100 V system and the AC 200 V system as a commercial AC power supply. In this case, AC100V system (VAC = 80
V) and the AC 200 V system (VAC = 288 V), the DC input voltage (rectified smoothed voltage Ei) is 110 V to 400 V
About 3.6 times. In the constant voltage control of the present embodiment, as described above, the period TOFF during which the switching element Q1 is off is constant, and the switching frequency is varied by variably controlling the period TON during which the switching element Q1 is on. That is, the higher the DC input voltage (rectified smoothed voltage Ei), the shorter the period TON. If such an operation is made to correspond to the above (Equation 1), the pulse level Lvcr of the resonance voltage Vcr is obtained even if the rectified smoothed voltage Ei has a 3.6-fold change width between the AC 100 V system and the AC 200 V system. Is not proportional to the rise of the rectified smoothed voltage Ei, and the rate of increase is suppressed. Actually, the pulse level Lvcr of the resonance voltage Vcr with respect to a change of VAC = 80 V to 288 V (that is, a change of the rectified smoothing voltage Ei) is Lvcr = 550 Vp as shown in FIGS. ~ 715
Vp, which is suppressed to an increase rate of about 1.3 times. Therefore, for the switching element Q1 and the parallel resonance capacitor Cr to which the pulse of the resonance voltage Vcr is applied, for example, a 900V withstand voltage product may be selected. As a result, inexpensive switching element Q1 and parallel resonance capacitor Cr can be selected. In particular, for switching element Q1 which is a bipolar transistor, saturation voltage VCE (SAT) and storage time t
Good characteristics such as STG, falling time tf, and current amplification factor hFE can be selected.

Also, the orthogonal control transformer PRT provided in the power supply circuit of the present embodiment is provided with a control winding NC, a detection winding ND, and a drive winding NB as shown in FIG. However, for example, if the same wire as that used as the control winding NC is used for the detection winding ND and the drive winding NB, the parts management and the manufacturing process are simplified, and the manufacturing efficiency is improved.

Further, in the power supply circuit of the present embodiment, since the secondary side parallel resonance circuit is provided, even when the load is short-circuited, the secondary side parallel resonance voltage of the secondary side parallel resonance circuit is maintained by the parallel resonance operation of the secondary side parallel resonance circuit. The state where V2 occurs is obtained. For this reason, the secondary DC voltage EO1 is, for example, 15 V
To 10V, but for the control circuit 1,
The supply of the secondary side output voltage can be maintained. Therefore, when the load is short-circuited, the DC current flowing through the control winding NC of the orthogonal control transformer PRT is maintained by short-circuiting the error amplifying IC in the control circuit 1 to reduce the switching frequency. Suppress. As a result,
The increase in the primary side series resonance current I1 and the collector current Icp flowing through the collector of the switching element Q1 are also suppressed, and thermal runaway of the switching element Q1 is also avoided. In other words, the power supply circuit according to the present embodiment has a load short-circuit protection function in the circuit, so that a stable switching operation can be continued even when the load is short-circuited. For this reason, in the power supply circuit according to the present embodiment, it is also possible to eliminate a protection component such as an IC link fuse.

As an experimental result of the actual power supply circuit shown in FIG. 1, when the maximum load power Pomax = 50 W and the AC input voltage VAC = 100 V, a power conversion efficiency of about 90% is obtained, and the minimum load power Pomin = Under the conditions of 10 W and AC input voltage VAC = 240 V, a power conversion efficiency of about 80% is obtained. In particular, a result is obtained that the power conversion efficiency at the minimum load power is improved by about 20% as compared with the conventional case. ing.

In the above embodiment, a self-excited type voltage resonance type converter is provided on the primary side. However, the present invention uses, for example, an IC (integrated circuit) instead of the self-excited oscillation drive circuit. Oscillation and drive circuits are provided.
It is also possible to adopt a configuration in which the switching element of the voltage resonance type converter is driven by the drive circuit.
In this case, as the constant voltage control, the drive signal waveform generated by the oscillation / drive circuit is variably controlled according to the secondary output voltage level. As the control, FIG.
In response to the operation shown in (f), the drive signal is set such that the period TOFF during which the switching element is turned off is constant and the period TON during which the switching element is turned on is shortened in accordance with an increase in the secondary-side output voltage level. What is necessary is just to generate a waveform.
By such control, the same operation as that described with reference to FIG. 5 is obtained as the power supply circuit. When such a separately excited configuration is employed, the orthogonal control transformer PRT is omitted.

In the case of adopting the separately excited configuration as described above, a single bipolar transistor (BJT)
Instead of the switching element Q1
It is possible to employ a Darlington circuit in which a stone bipolar transistor (BJT) is connected in Darlington. Furthermore, one bipolar transistor (BJT)
MOS-FE instead of the switching element Q1
It is possible to use T (MOS field effect transistor; metal oxide semiconductor), IGBT (insulated gate bipolar transistor), or SIT (static induction thyristor), and these Darlington circuits or any of the above elements When is used as a switching element, it is possible to further increase the efficiency. When these elements are used as switching elements, although not shown here, the configuration of the drive circuit is changed so as to conform to the characteristics of the elements to be actually used instead of the switching element Q1. If a MOS-FET is used as a switching element, a configuration in which voltage driving is performed by a separately-excited system may be employed.

[0055]

As described above, according to the present invention, as a switching power supply circuit corresponding to a relatively light load having a load power of about 50 W or less, a voltage resonance type switching converter is provided on the primary side and an insulation converter is provided. The mutual inductance of the primary winding and the secondary winding of the transformer is set to the polarity operation mode (+ M; forward system). And, by providing a half-wave rectifier circuit on the secondary side, by the half-wave rectification operation in the polarity operation mode,
An alternating voltage (excitation voltage) secondary-side DC output voltage obtained in the secondary winding is obtained.

Also, as a configuration for constant voltage control for stabilizing the secondary output voltage, the primary side switching frequency is changed according to the secondary output voltage level, so that the resonance impedance in the power supply circuit is reduced. The conduction angle of the switching element is controlled in a complex manner.

The following can be said from the above configuration.
Since the switching converter of the present invention is of the voltage resonance type, a switching operation with lower noise than that of the RCC type switching power supply is realized. This allows RCC
There is no need to provide a reset circuit or a snubber circuit to suppress the spike voltage as in the switching power supply of the system. Therefore, when the present invention is compared with the RCC switching power supply, the power conversion efficiency is greatly improved. Further, the power conversion efficiency at the time of maximum load power is significantly improved as compared with a current resonance type converter whose power conversion efficiency is relatively high in terms of characteristics.

Also in terms of circuit scale, the power supply circuit of the present invention is a voltage resonance type converter when compared with a current resonance type converter formed by half-bridge coupling of two switching elements, for example. Thus, since it is possible to obtain a configuration in which almost the same load power is obtained by one switching element, the number of components can be reduced accordingly, and it is possible to promote reduction in size and weight and cost of the circuit.

Also, by changing the switching frequency, the resonance impedance with respect to the switching output and the conduction angle of the switching element are controlled in a complex manner, and the constant voltage control is performed by this operation.
As a result, the control sensitivity is improved and the controllable range is expanded, so that it is possible to stabilize the secondary-side output voltage in a control range with a narrower switching frequency than in the related art. Such a reduction in the control range of the switching frequency can be achieved by reducing the number of windings wound around a transformer forming a power supply circuit,
It also contributes to the miniaturization of various component elements. As a configuration of the constant voltage control circuit of the present invention, when the circuit system for driving the switching element is a self-excited type, a quadrature control transformer having a control winding, a drive winding, and a detection winding wound thereon. However, if the drive winding and the detection winding are wound by the same wire as the control winding, the manufacturing becomes easier and the manufacturing efficiency is improved.

Further, by connecting a secondary side parallel resonance capacitor to the secondary winding in parallel to form a parallel resonance circuit, the secondary side half-wave rectifier circuit becomes an alternating output which becomes the resonance output of this parallel resonance circuit. Since the secondary DC output voltage is obtained by inputting the voltage, the load power increases. That is, it is possible to cope with a load power more than required only by inserting the secondary-side parallel resonance capacitor. Conversely, the secondary side parallel resonance capacitor may be omitted under the condition that it is sufficient to correspond to a required load power or less, for example, corresponding to an application such as a standby power supply. In the present invention, the secondary side parallel resonance capacitor is used. Can be adjusted according to the load power condition depending on the presence or absence of the insertion.

Further, a parallel resonance circuit is provided on the secondary side so that a parallel resonance voltage can be obtained on the secondary side even when the load is short-circuited. A stable switching operation in which the switching frequency does not decrease is obtained, and an increase in power loss in the switching element is suppressed. That is, a protection function at the time of load short circuit is provided. Therefore, in the power supply circuit of the present invention, it is not necessary to insert an IC link fuse or the like into the secondary side output. This also promotes improvement in power conversion efficiency and reduction in size and weight of the circuit.

The switching element can be constituted by a Darlington circuit formed with a bipolar transistor, or a MOS field effect transistor, an insulated gate bipolar transistor, or an electrostatic induction thyristor. For example, the power conversion efficiency can be further improved as compared with the case where the switching means is formed by a single bipolar transistor.

As described above, according to the present invention, the power supply circuit having the voltage resonance type converter on the primary side and corresponding to a relatively light load can promote the improvement of characteristics such as low cost, small size and light weight, and power conversion efficiency. Is what is done.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a perspective view showing a structure of an orthogonal control transformer provided in the power supply circuit shown in FIG.

3 is a perspective view showing a structure of an insulating converter transformer provided in the power supply circuit shown in FIG.

FIG. 4 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 5 is a waveform chart showing an operation of a main part in the power supply circuit shown in FIG.

FIG. 6 is an explanatory diagram illustrating a relationship between a switching frequency and a secondary-side DC output voltage in the power supply circuit illustrated in FIG. 1;

FIG. 7 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

8 is a waveform chart showing an operation of a main part in the power supply circuit shown in FIG.

FIG. 9 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 10 is a waveform chart showing an operation of a main part in the power supply circuit shown in FIG. 9;

[Explanation of Signs] 1 Control circuit, Ci, CO1, CO2 smoothing capacitor, C
r Parallel resonant capacitor, C2 secondary parallel resonant capacitor, Di bridge rectifier circuit, DO1, DO2 rectifier diode, PIT isolation converter transformer, PRT orthogonal control transformer, NC control winding, ND detection winding, NB
Drive winding, Q1 switching element

 ──────────────────────────────────────────────────続 き Continued on the front page F-term (reference) EE72 FD01 FD24 FG03 FG07 VV03

Claims (9)

[Claims] 1. A rectifying / smoothing means for receiving a commercial AC power supply, generating a rectified smoothed voltage having a level corresponding to the same level as the commercial AC power supply level, and outputting the rectified smoothed voltage as a DC input voltage; Insulating converter transformer in which the required coupling coefficient is obtained so that the polarity of the primary winding and the secondary winding wound on the core is in the polarity operation mode, Switching means for intermittently outputting a DC input voltage by a switching element to output to a primary winding of the insulating converter transformer; at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a parallel resonant capacitor A primary-side parallel resonance circuit that makes the operation of the switching means a voltage resonance type; DC output voltage configured to be able to generate a secondary side DC output voltage corresponding to almost equal to the input voltage level by performing the half-wave rectification operation by inputting the alternating voltage obtained to the secondary winding of the transformer. Generating means for controlling the resonance impedance of the primary-side parallel resonance circuit and the conduction angle of the switching element in a complex manner by varying the switching frequency of the switching element according to the level of the secondary DC output voltage So that
A constant-voltage control unit configured to perform constant-voltage control on the secondary-side output voltage. 2. A secondary-side parallel resonance capacitor is connected in parallel to a secondary winding of the insulated converter transformer when the load power condition exceeds a required level. A secondary-side parallel resonance circuit is formed by the leakage inductance component of the winding and the capacitance of the secondary-side parallel resonance capacitor. The switching power supply circuit according to claim 1, wherein the switching power supply circuit is omitted. 3. The switching means is formed to include at least a series resonance circuit formed by connecting a drive winding and a resonance capacitor in series, and the switching element is self-excited based on a resonance output of the series resonance circuit. The constant voltage control means includes: the detection winding and the drive winding connected in series to a primary winding of the insulating converter transformer; and the detection winding. And a quadrature control transformer as a saturable reactor around which a drive winding and a control winding whose winding direction is orthogonal are provided, and a variable control current is provided in accordance with the level of the secondary DC output voltage. 2. The switching frequency can be variably controlled by flowing through the control winding to change the inductance of the drive winding. Switching power supply circuit according. 4. The switching power supply circuit according to claim 3, wherein the detection winding and the drive winding use the same wire as that used for the control winding. 5. The switching means includes a separately-excited drive circuit for driving the switching element in a separately-excited manner, and the constant-voltage control means includes a switch for controlling the switching element in accordance with a level of the secondary-side DC output voltage. The switching power supply circuit according to claim 1, wherein the switching frequency is variably controllable by variably controlling the ON period while keeping the OFF period constant. 6. The switching power supply circuit according to claim 1, wherein said switching means is configured to use a Darlington circuit formed with a bipolar transistor as one switching element. 7. The switching power supply circuit according to claim 1, wherein said switching means includes a MOS field effect transistor as a switching element. 8. The switching power supply circuit according to claim 1, wherein said switching means includes an insulated gate bipolar transistor as a switching element. 9. The switching power supply circuit according to claim 1, wherein said switching means is formed with an electrostatic induction thyristor as a switching element.