JP2000125555A - Voltage-resonance type switching power supply circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To improve power conversion efficiency and to miniaturize and a circuit high in weight. SOLUTION: A boost circuit (coil winding N3 for boosting that is formed by winding up primary coil winding N1 of an insulation converter transformer PIT, a diode DB for boosting, a smoothing capacitor CiB for generating a boost voltage, and coil winding Nr to be controlled of a cross-type control transformer PRT) is provided as a voltage-resonance type switching power supply circuit which corresponds to relatively high-load conditions, exceeding 150 W in an AC input voltage AC 100 V system to obtain a boost smoothing voltage, so that the maximum load power can be increased. Furthermore, at a secondary side, a secondary side DC output voltage is obtained by a full-wave rectifying circuit (a set of [rectifying diodes D01 and D02 and a smoothing capacitor C01] and a set of [rectifying diodes D03 and D04 and a smoothing capacitor C02]), where a parallel resonance capacitor C2 is connected to increase the maximum load power. A primary side is formed to be an unmagnified voltage rectifying circuit according to the full-wave rectifying circuit instead of the double-voltage rectification circuit.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a voltage resonance type switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a so-called soft switching power supply circuit, a switching power supply circuit having a voltage resonance type switching converter is known. The voltage resonance type switching converter has a low noise because a smooth waveform is obtained for the switching output voltage pulse and the switching output current flowing into the insulating converter transformer, and can be configured with a relatively small number of parts.

FIG. 10 is a circuit diagram showing an example of a voltage resonance type switching power supply circuit which can be constructed based on the invention proposed by the present applicant. In the switching power supply circuit shown in this figure, for example, a commercial AC power supply in Japan or the United States is a so-called AC 100 V system, and corresponds to a condition where the maximum load power is 150 W or more.

The switching power supply circuit shown in FIG. 1 is provided with a so-called voltage doubler rectification circuit comprising rectification diodes Di1, Di2 and smoothing capacitors Ci1, Ci2 as a rectification and smoothing circuit for rectifying and smoothing the AC power supply AC. Can be In this voltage doubler rectifier circuit, assuming that a DC input voltage corresponding to, for example, one time the peak value of the AC input voltage VAC is Ei, a DC input voltage 2Ei that is approximately twice that of the DC input voltage Ei.
Generate For example, if the AC input voltage VAC is 144V, the DC input voltage 2Ei is about 400V. As described above, the reason why the voltage doubler rectifier circuit is employed as the rectifier / smoothing circuit is that the AC input voltage is 100 V AC as described above.
The system is designed to cope with a relatively heavy load condition in which the maximum load power is 150 W or more. That is, by increasing the DC input voltage to twice the normal value, the amount of current flowing into the subsequent switching converter is suppressed, and the reliability of the components forming the switching power supply circuit is ensured. . In the voltage doubler rectifier circuit shown in this figure, an inrush current limiting resistor Ri is inserted in the rectified current path, for example, so as to suppress the inrush current flowing into the smoothing capacitor when the power is turned on. I have.

The voltage-resonant type switching converter shown in FIG. 1 has a self-excited type provided with a single switching element Q1. In this case, a high withstand voltage bipolar transistor (BJT;
Junction type transistor). The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci1 (rectified smoothed voltage 2Ei) via a starting resistor RS so that a base current at the time of starting can be obtained from the rectified smoothing line. A self-excited oscillation resonance circuit including an inductor LB, a drive winding NB, a resonance capacitor CB, and a damping resistor RB is connected in series between the base of the switching element Q1 and the temporary ground. In this case, the drive winding NB is connected to an insulation converter transformer PIT (Power Isol
ation Transformer) and the inductor LB
At the same time, a required inductance for setting the switching frequency is obtained. A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary ground) of the smoothing capacitor Ci forms a path for a damper current flowing when the switching element Q1 is off. The collector of the switching element Q1 is an isolated converter transformer PI
T is connected to one end of the primary winding N1 and the emitter is grounded.

A parallel resonance capacitor Cr is connected in parallel between the collector and the emitter of the switching element Q1. This parallel resonance capacitor Cr
Is the combined capacitance (L1, LR) obtained by connecting the primary winding N1 of the isolated converter transformer PIT described later and the controlled winding NR of the orthogonal control transformer PRT (Power Regulating Transformer) in series. Thereby, a parallel resonance circuit of the voltage resonance type converter is formed. Although the detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vd across the resonance capacitor Cr is obtained by the action of the parallel resonance circuit.
cr is actually a sinusoidal pulse waveform so that a voltage resonance type operation can be obtained.

The insulated converter transformer PIT is for transmitting the switching output of the switching element Q1 to the secondary side. In this case, the insulated converter transformer PIT is used.
One end of the primary winding N1 of T is connected to the collector of the switching element Q1, and the other end is connected in series with the controlled winding NR of the orthogonal control transformer PRT as shown in the figure.

On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a parallel resonance circuit is formed by connecting the secondary side parallel resonance capacitor C2 to the secondary winding N2 in parallel. With this parallel resonance circuit,
The alternating voltage excited in the secondary winding N2 is a resonance voltage,
This resonance voltage is supplied to two sets of half-wave rectifier circuits, a half-wave rectifier circuit including a rectifier diode DO1 and a smoothing capacitor CO2, and a half-wave rectifier circuit including a rectifier diode DO2 and a smoothing capacitor CO2. Then, the DC output voltages EO1 and EO2 are obtained by these two sets of half-wave rectifier circuits. The rectifier diodes DO1 and DO2 forming the half-wave rectifier circuit use a high-speed rectifier to rectify the alternating voltage in the switching cycle.

The control circuit 1 compares, for example, a DC voltage output on the secondary side with a reference voltage and outputs a DC current corresponding to the error.
The control winding N of the orthogonal control transformer PRT is used as the control current.
This is the error amplifier that supplies C. In this case, the DC voltage output EO1 is input to the control circuit 1 as the detection voltage, and the DC output voltage EO2 is input as the operation power supply.

For example, when the DC output voltage EO2 on the secondary side fluctuates due to the fluctuation of the AC input voltage VAC or the minimum load power, the control circuit 1 controls the control current flowing through the control winding NC to 10 mA to 40 mA, for example. Vary in range.
Thus, the inductance LR of the controlled winding NR is changed in a range of, for example, 0.1 mH to 0.6 mH. Here, FIG. 16 shows an inductance DC superposition characteristic of the orthogonal control transformer PRT. In this figure,
The horizontal axis represents the current IR flowing through the controlled winding NR, and the vertical axis represents the inductance LR of the controlled winding NR. The parameter is a control current IC flowing through the control winding NC. In this case, the inductance LR tends to increase as the level of the control current IC decreases.

Since the controlled winding NR forms a parallel resonance circuit for obtaining a voltage-resonant switching operation as described above, this parallel resonance circuit is not controlled for a fixed switching frequency. The resonance condition of the circuit is changed. At both ends of the parallel connection circuit of the switching element Q1 and the parallel resonance capacitor Cr, a sinusoidal resonance pulse is generated by the action of the parallel resonance circuit corresponding to the OFF period of the switching element Q1, but the resonance condition of the parallel resonance circuit , The width of the resonance pulse is variably controlled. That is, PWM (P
(ulse Width Moduration) control operation is obtained. The PWM control of the width of the resonance pulse means the control of the off period of the switching element Q1, which in other words means variably controlling the on period of the switching element Q1 under the condition of a fixed switching frequency. . By variably controlling the ON period of the switching element Q1 in this manner, the switching output transmitted from the primary winding N1 forming the parallel resonance circuit to the secondary side changes, and the DC output voltage ( The output levels of EO1, EO2) are also changed. As a result, the secondary DC voltage (EO1, EO2) can be made constant. Note that such a constant voltage control method is hereinafter referred to as an inductance control method.

FIG. 11 is a circuit diagram showing another example of a voltage resonance type switching power supply circuit which can be configured based on the invention proposed by the present applicant. Note that FIG.
In FIG. 10, the same parts as those in FIG. In the power supply circuit shown in this figure, an example is shown in which the controlled winding NR of the orthogonal control transformer PRT is provided on the secondary side. In the case shown in this figure, the controlled winding NR
Are connected in series between the end of the secondary winding N2 and the anode of the rectifier diode DO2.

In the circuit shown in FIG. 11, a winding N3 is inserted in series between the tap output of the secondary winding N2 and the anode of the rectifier diode DO1. In this case, the winding N3 is connected to the orthogonal control transformer PRT.
It is wound in a winding direction orthogonal to the control winding NC.

Thus, even if the controlled winding NR of the orthogonal control transformer PRT is provided on the secondary side,
Constant voltage control is achieved by the same operation as described above.

Here, the structure of the insulated converter transformer PIT provided in the power supply circuit shown in FIGS. 10 and 11 is shown in cross section by FIG. Insulation converter transformer P
IT is, for example, E type cores CR1 and C made of ferrite material.
An EE-type core is formed by combining R2 with their magnetic legs facing each other. At this time, no gap is formed in the center magnetic leg as shown in the figure. Then, the primary winding N1 (and the driving winding N
B), the secondary winding N2 is wound in a divided state. Thereby, the primary winding N1 and the secondary winding N2
With, a state of loose coupling (for example, coupling coefficient k ≒ 0.9) is obtained.

In the isolated converter transformer PIT, the polarities (winding direction) of the primary winding N1 and the secondary winding N2 are set.
And the connection of the rectifier diodes DO (DO1, DO2), the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes + M and -M. There is. For example, when the connection configuration shown in FIG. 13A is employed, the mutual inductance is + M, and when the connection configuration shown in FIG. 13B is employed, the mutual inductance is -M.

Next, the structure of the orthogonal control transformer PRT provided in the power supply circuit shown in FIG. 10 or 11 is shown in FIG.
Is shown in a perspective view. The orthogonal control transformer PR shown in FIG.
T has cores CR11 and CR12 made of a ferrite material having four magnetic legs.
Are assembled such that their magnetic legs face each other to form a core body. In this case, the core body is 23 ×
23 × 23 (mm) size, each magnetic leg is 7 × 7m
m. The control winding NC is wound around the two magnetic legs of the four magnetic legs, and the controlled winding is performed using the two magnetic legs so that the winding direction is orthogonal to the control winding NC. The line NR is wound. Further, among the four magnetic legs, for example, the gaps G are formed on the opposing surfaces of the two pairs of magnetic legs shown in the figure, so that the saturable reactor is formed.

FIG. 15 shows operation waveforms of the power supply circuit shown in FIGS. 10 and 11 according to the switching cycle.
FIGS. 15A and 15B respectively show the switching element Q1 of the power supply circuit shown in FIG. 10, the voltage Vcr across the parallel resonance capacitor Cr, and the collector of the switching element Q1 via the controlled winding NR and the primary winding N1. 2 shows the switching current I1 flowing through the switch. FIGS. 15C and 15D respectively show the switching element Q1 of the power supply circuit shown in FIG. 11, the voltage Vcr across the parallel resonance capacitor Cr, and the primary winding N1.
Shows the switching current I2 flowing through the collector of the switching element Q1 through the switching element I1. The periods TON and TOFF indicate periods when the switching element Q1 is turned on and off, respectively.

As can be seen from this figure, FIG. 10 and FIG.
In any of the power supply circuits shown in FIG. 1, during a period TOFF when the switching element Q1 is turned off, a resonance voltage Vcr of about 1800 V is generated at a peak. This is because the impedance of the parallel resonance circuit on the primary side of the voltage resonance converter acts on the DC input voltage of 2Ei obtained by the voltage doubler rectification.

[0020]

The above-mentioned FIG.
In the voltage resonance type converter having the configuration described with reference to FIG. 16, since the AC input voltage VAC corresponds to the AC 100 V system and the maximum load power is 150 W or more, the DC input voltage of the level of 2Ei is obtained by the voltage doubler rectification method. I have to. Therefore, as shown in FIG. 15, a resonance voltage Vcr of 1800 V is generated across the switching element Q1 and the parallel resonance capacitor Cr when the switching Q1 is off. Therefore, for the switching element Q1 and the parallel resonance capacitor Cr, it is required to select a high withstand voltage product of 1800V. In this case, especially for the switching element Q1, the saturation voltage VCE (SAT), the accumulation time tSTG, the fall time tf are large, and the current amplification factor hFE is small, so that the switching loss and the drive power increase, and accordingly the power supply circuit Power loss increases.

In the configuration shown in FIGS. 10 and 11,
Although the DC output voltages (EO1, EO2) are obtained by the half-wave rectifier circuit, the rectifier diodes (DO1, DO) of the half-wave rectifier circuit are obtained.
In the non-conducting state of 2), as shown in FIGS. 16D and 16I, a secondary side voltage V2 of about 2 to 3.5 times the DC output voltage EO (EO1, EO2) is applied. You. Therefore, the rectifier diodes (DO1, DO) used in the half-wave rectifier circuit
As for 2), for example, assuming that the DC output voltage is about 135 V, a high withstand voltage product of about 500 V is required, which increases the forward voltage drop VF and the reverse recovery time trr, which causes an increase in power loss. For example, if the switching frequency is increased, it is possible to reduce the size of various component elements. However, it is difficult to set the switching frequency fs to be high due to the above-described circumstances.
It has been found that when s is equal to or higher than 50 KHz, the power conversion efficiency is rapidly reduced. Further, as described above, a voltage doubler rectifier circuit is required to obtain a highly reliable DC input voltage, so that two relatively large smoothing capacitors are required and the substrate area is increased.

[0022]

SUMMARY OF THE INVENTION In view of the above-mentioned problems, the present invention provides a voltage-resonant switching power supply circuit, for example, capable of accommodating a maximum load power of 150 W in an AC 100 V system, thereby improving power conversion efficiency, and An object is to reduce the size and weight of the substrate area.

Therefore, as the voltage resonance type switching power supply circuit, rectifying and smoothing means for generating a rectified and smoothed voltage of a level corresponding to the input commercial AC power supply level by full-wave rectification, and a required coupling which is loosely coupled. An insulating converter transformer having a gap formed so as to obtain a coefficient and provided for transmitting the primary output to the secondary side, and a primary winding of the insulating converter transformer intermittently intermitting a rectified and smoothed voltage output from the rectifying and smoothing means. A switching means adapted to output to a line, a primary resonance circuit formed by at least an inductance component including a primary winding of an insulating converter transformer and a capacitance of a resonance capacitor to make the operation of the switching means a voltage resonance type; Inductor component of secondary winding of converter transformer and secondary parallel resonance Enter a secondary-side resonant circuit by the capacitance of the capacitors formed at the secondary side, an alternating voltage obtained at the secondary winding of the insulating converter transformer,
DC output voltage generating means for obtaining a secondary-side DC output voltage by full-wave rectification; a controlled winding provided to function as an inductance component of a primary-side resonance circuit; a controlled winding and its winding direction Is equipped with an orthogonal control transformer wound with a control winding that is orthogonal to the control winding, and a variable control current flows through the control winding according to the level of the DC output voltage to change the inductance of the controlled winding. By letting
A constant voltage control unit configured to perform constant voltage control on the secondary DC output voltage; and a boost voltage generated by using a switching output of the switching unit on the rectified smoothed voltage, and a boost rectified smoothed voltage. And boost means having a configuration capable of controlling the boost rectification smoothed voltage to be constant by changing the inductance of the controlled winding by including the controlled winding. is there. The boost means is formed by winding up a primary winding of an insulating converter transformer to obtain a switching output, and a boost rectifying diode for rectifying an alternating voltage obtained in the boost winding. And a controlled winding connected in series or parallel to the boost winding, and a rectified output obtained by the series connection circuit being charged to charge both ends thereof. A boosting smoothing capacitor provided to generate a boost voltage, wherein the boosting smoothing capacitor is formed by connecting a negative electrode side to a positive electrode side of a smoothing capacitor forming a rectifying smoothing means. It was decided to.

According to the above configuration, the insulation converter transformer is loosely coupled, and on the secondary side, a secondary-side DC output voltage is generated by the full-wave rectifier circuit to supply power to the load. In other words, a required load condition is handled by providing a full-wave rectifier circuit on the secondary side, and accordingly, the primary side is not a voltage doubler rectifier circuit, but one time of the AC input voltage level. Is provided with a full-wave rectifier circuit that generates a rectified smoothed voltage corresponding to. In addition, a boost circuit for obtaining a boosted smoothed voltage by superimposing a boosted voltage on the rectified smoothed voltage on the primary side is provided. This will increase the apparent DC input voltage level to the switching converter. Further, the boost circuit is obtained by rectifying and smoothing the alternating voltage obtained in the boost winding obtained by winding up the primary winding. In other words, there is a relation of being connected to the primary side winding forming the primary side series resonance circuit. The stabilization control of the secondary DC output voltage is performed by varying the inductance of the controlled winding of the orthogonal control transformer connected to the primary winding. The width (ie, the on / off period of the switching means) is also controlled according to the variable inductance of the controlled winding.

[0025]

FIG. 1 shows an example of the configuration of a voltage resonance type switching converter according to a first embodiment of the present invention. In the switching power supply circuit shown in this figure, similar to the case of FIG. 10 or FIG.
A self-excited voltage resonance type switching converter using one switching element (bipolar transistor) is employed. In this figure, FIG. 10 or FIG.
The same parts as those described above are denoted by the same reference numerals and description thereof will be omitted.

In the power supply circuit according to the embodiment shown in FIG. 1, a full-wave rectifying / smoothing circuit comprising a bridge rectifier circuit Di and a smoothing capacitor Ci serves as a rectifying / smoothing circuit for receiving an AC input voltage VAC to obtain an AC input voltage. A rectifier circuit is provided to generate a rectified smoothed voltage Ei corresponding to a level that is one-time the AC input voltage VAC. That is, in the present embodiment, a voltage doubler rectifier circuit is not provided as in the related art. In this specification, the rectified and smoothed voltage E corresponding to one time the level of the AC input voltage VAC is used.
The full-wave rectifier circuit that generates i is also referred to as an “equal voltage rectifier circuit”.

Further, in the insulation converter transformer PIT of the present embodiment, a winding N3 is provided by winding up the primary winding N1 in addition to the primary winding N1, the secondary winding N2 and the driving winding NB. I have. The winding N3 forms a boost circuit described later. This insulation converter transformer P
As shown in FIG. 2, the IT is provided with an EE-type core in which E-type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other. , The primary winding N1 using the bobbin B
(And N3, NB) and the secondary winding N2 are wound separately. In the present embodiment, a gap G is formed for the center magnetic leg as shown in the figure. As a result, for example, a loose coupling is achieved by a coupling coefficient smaller than that of the insulating converter transformer PIT shown in FIG. 12 as a conventional example, so that a saturated state is hardly obtained. Coupling coefficient k in this case
Is set to, for example, k５0.85.

In the present embodiment, the boost voltage V
A boost circuit is provided for generating B and superimposing it on the rectified smoothed voltage Ei to obtain a boost smoothed voltage EB. This boost circuit is an isolated converter transformer PI
It comprises a boost winding N3 formed by winding up a primary winding N1 of T, a boost diode DB, a smoothing capacitor CiB for generating a boost voltage, and a controlled winding NR of a quadrature control transformer PRT. . The end of the winding N3 is connected to the positive electrode of the smoothing capacitor CiB via the series connection of the controlled winding NR. The negative electrode of the smoothing capacitor CiB is connected to the positive electrode (Ei line) of the smoothing capacitor Ci. The boost diode DB has an anode connected to a connection point (Ei line) between the negative electrode of the smoothing capacitor CiB and the positive electrode of the smoothing capacitor Ci, and a cathode connected to the connection point between the primary winding N1 and the winding N3. .

According to such a connection form, the switching output voltage obtained in the winding N3 is rectified by the boost diode DB and smoothed by the smoothing capacitor CiB, so that the boost voltage VB is applied across the smoothing capacitor CiB. The boost circuit to generate is formed.
In the present embodiment, the voltage resonance type switching converter including the switching element Q1 performs switching using the boost smoothed voltage EB obtained by superimposing the boost voltage VB on the rectified smoothed voltage Ei as an operation power supply. To be. That is, in the present embodiment,
The boost circuit increases the apparent DC input voltage level to be supplied to the voltage resonance type switching converter.

As the primary side parallel resonance circuit in this configuration, a combined inductance (L1 + L3 + L) obtained by connecting the primary winding N1, the winding N3 and the controlled winding NR in series.
R) and the capacitance of the parallel resonance capacitor Cr.

On the secondary side of the power supply circuit of the present embodiment, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, DO3, DO4 and the smoothing capacitors CO1, CO2 are provided. By connecting as shown in the figure,
A set of [rectifier diodes DO1, DO2, smoothing capacitor C01] and [rectifier diodes DO3, DO4, smoothing capacitor C0]
O2], two sets of full-wave rectifier circuits are provided.
A full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] generates a DC output voltage EO1, and a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. I do. That is, in the present embodiment, a full-wave rectifier circuit is provided to obtain a DC output voltage on the secondary side. In this case, the DC output voltage EO1 and the DC output voltage EO2 are branched and input to the control circuit 1. The DC output voltage EO1 is used as a detection voltage in the control circuit 1, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

As described above with reference to FIG. 13, in the isolated converter transformer PIT, the mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 becomes + M and -M. It has been described that there are cases where the secondary winding N2
Rectifier diode D when the alternating voltage obtained at
The operation in which the rectified current flows through O1 and DO3 can be regarded as a + M operation mode (forward mode). Conversely, when the alternating voltage obtained in the secondary winding N2 has a negative polarity, it flows through the rectified diodes DO2 and DO4. The rectified current can be considered to be in the -M operation mode (flyback method). That is, in the present embodiment, each time the alternating voltage obtained in the secondary winding becomes positive / negative, the mutual inductance operates in the + M / -M mode.

Further, in the power supply circuit having the above-described configuration, as described above, the boost circuit provided on the primary side generates the boost voltage VB across the smoothing capacitor CiB. A boosted smoothing voltage EB obtained by superimposing the boosted voltage VB on the rectified smoothed voltage Ei is obtained at both ends of the series-connected smoothing capacitor CiB-smoothing capacitor Ci. ,

(Equation 1) Can be represented by Here, the relationship of L3 = 0.1 × L1 is obtained as the inductance of the winding N3 and the primary winding N1, and LR = about the inductance LR of the controlled winding NR and the inductance L1 of the primary winding N1. Assuming that the relationship of 0.9 × L1 to 0.1 × L1 is obtained, the boost smoothing voltage EB changes in the range of approximately 2Ei to 1.2Ei by the above (Equation 1).

FIG. 3 shows the relationship between the rectified smoothed voltage Ei and the boost smoothed voltage EB with respect to the AC input voltage VAC. In this figure, the horizontal axis is the AC input voltage VAC
The vertical axis indicates the level of the rectified smoothed voltage Ei or the boosted smoothed voltage EB. Here, as the AC input voltage AC 100 V system, the commercial AC power supply in the area A is in the range of AC80 to 120, and the AC1
The characteristics under both conditions of the range of 00 to 140 are shown.

As shown in the figure, the rectified and smoothed voltage Ei obtained at both ends of the smoothing capacitor Ci rises as the AC input voltage VAC rises. On the other hand, the boosted smoothed voltage EB obtained by superimposing the boosted voltage VB is variable in the range of 1.9 Ei to 1.5 Ei as described above. At the time of the maximum load power Pomax and the time of the minimum load power Pomin,
It is controlled to be substantially constant. This is due to the fact that the controlled winding NR of the orthogonal control transformer PRT for constant voltage control is inserted in the boost circuit.

FIG. 4 is a waveform diagram showing the operation of the main part of the power supply circuit according to the present embodiment during the switching cycle.
FIGS. 4A, 4B, and 4C show resonance voltages obtained at both ends of the parallel connection of the switching element Q1 and the parallel resonance capacitor Cr when the AC input voltage VAC is the minimum (80V AC) and the maximum load power Pomax = 160W. Vcr, primary winding N
1 shows a current I1 flowing through 1 and a rectified current IDB flowing through a boost diode DB. In addition, FIGS.
(F) shows that the AC input voltage VAC is maximum (140V AC),
The resonance voltage Vcr at the time of the minimum load power Pomin = 0W,
The current I1 and the rectified current IDB are shown. Also, the period TO
N and TOFF indicate periods when the switching element Q1 is turned on and off, respectively. As you can see from this figure,
The pulse waveform appearing as the resonance voltage Vcr shown in FIG. 4A is a voltage resonance type operation, and has a smooth waveform shape close to a sine wave instead of a rectangular waveform. In the case of the present embodiment, the resonance voltage Vcr is the combined inductance of the primary winding N1, the winding N3, and the controlled winding NR (L1 + L3 +
Due to the resonance action of the primary side parallel resonance circuit composed of LR) and the capacitance of the parallel resonance capacitor Cr, the switching element Q1 appears during the OFF period TOFF.

As the control circuit 1 of the present embodiment, a control current of a level corresponding to the fluctuation of the DC output voltage EO1 is supplied to the control winding NC of the orthogonal transformer PRT, so that the control winding NR It operates to change the inductance LR. In the connection mode of the present embodiment, the inductance LR
Forms a primary-side parallel resonance circuit, and operates so as to make the secondary-side DC output voltage constant by the above-described inductance control method using the variable inductance LR. That is, as shown in FIGS. 4A and 4D, as a result of variably controlling the pulse width of the resonance voltage Vcr (ie, TOFF period), the TON period of the switching element Q1 is variably controlled. In addition, in the present embodiment, by providing the inductance LR in the boost circuit, the boost smoothing voltage EB is controlled so as to be constant by the change in the inductance LR as shown in FIG. This also operates to keep the secondary DC output voltage constant.

In this way, the stabilization of the boosted smoothing voltage EB in the boost circuit and the stabilization control of the secondary side DC voltage are achieved by a combination of the inductance control method.

Therefore, in the case of the present embodiment, based on the above (Equation 1), the inductance L3 of the winding N3 is
This is only required to be 1/10 of the inductance L1 of the primary winding N1. As a result, the number of turns of the winding N3 can be reduced accordingly, and in an actual operation, the current flowing from the winding N3 to the boost circuit can be reduced to about half of the current I1 flowing to the primary winding N1. . This also reduces the amount of current flowing through the controlled winding NR in the boost circuit. As a result, in the present embodiment, the size of the insulating converter transformer PIT and the orthogonal control transformer PRT can be reduced as compared with the conventional configuration. Also, since the control sensitivity is improved by using the boost control circuit in combination with the stabilization of the boost smoothing voltage EB and the inductance control method, the constant voltage control range can be expanded and the ripple of the secondary output voltage can be suppressed. is there. Further, the amount of current to be passed through the controlled winding NR can be reduced with the improvement of the control sensitivity. Therefore, in the present embodiment, the size of the orthogonal control transformer PRT can be made smaller than before.

In the present embodiment, the inductance L3 of the winding N3 and the inductance LR of the controlled winding NR are determined.
Experiments have shown that if a large value is selected for, it is possible to easily generate a boost smoothed voltage VB that is twice or more the rectified smoothed voltage Ei. For this reason, FIG. 1 employs a form of a power supply circuit for inputting a commercial AC power supply. However, the present embodiment is also applicable to a configuration in which a low-voltage DC voltage from a battery is input instead of the rectified smoothed voltage Ei. This configuration is effective.

Further, in the present embodiment, by increasing the apparent AC input voltage by the boost circuit as described above, almost the same as the conventional case where a rectified smoothed voltage is obtained by voltage doubler rectification. It is possible to cope with the maximum load power. In addition to this, when the secondary side parallel resonance circuit is formed by providing the secondary side parallel resonance capacitor C2 as in the power supply circuit of the present embodiment, for example, the operation of the secondary side parallel resonance circuit Since power is supplied to the load side, the maximum load power is further increased as compared with the case where the secondary side parallel resonance capacitor C2 is not provided. At this time, if the rectifying / smoothing circuit connected to the secondary parallel resonance circuit is a half-wave rectifying circuit, a rectified current cannot be obtained when the alternating voltage generated in the secondary winding N2A has a negative polarity. The load power increase rate is about 50%. On the other hand, when a full-wave rectifier circuit is connected to the secondary-side parallel resonance circuit as in the present embodiment, as described above, the mutual inductance is in both operation modes of + M / -M. A rectified current flows alternately. In other words, since the rectified output is obtained during the period in which the alternating voltage is both the positive electrode and the negative electrode, the power supplied to the load increases accordingly, and the rate of increase of the maximum load power also improves. For example, compared to a configuration in which the secondary parallel resonance capacitor C2 is not provided and a half-wave rectifier circuit is connected, in the present embodiment, the maximum load power is increased by about twice.

By increasing the maximum load power as described above, in the present embodiment, the rectifying and smoothing circuit that generates the DC input voltage (rectified and smoothed voltage) adopts the double voltage rectification method to reduce the load power. No need to cover. As a result, in the present embodiment, as described with reference to FIG. 1, for example, a configuration of a normal equal-voltage rectifier circuit using a bridge rectifier circuit can be adopted. Thus, for example, in the present embodiment, the rectified smoothed voltage Ei at the time of the AC input voltage VAC = 144 V becomes about 200 V. Resonant voltage V
The cr is generated when the switching element Q1 is turned off by the action of the primary parallel resonance circuit on the rectified smoothed voltage Ei. In the present embodiment, the rectified smoothed voltage Ei is double-voltage rectified as described above. It will be about 1/2 of the time. However, in the present embodiment, the boost smoothed voltage EB is superimposed on the rectified smoothed voltage Ei by superimposing the boosted voltage VB.
Is generated, the resonance voltage Vcr becomes equal to the boost smoothed voltage E.
Although it depends on the level of B, the resonance voltage V
As shown in FIG. 4A, cr can be suppressed from the conventional 1800 V to about 1500 V at the maximum load power as shown in FIG. Therefore, in the present embodiment, it is sufficient to select a withstand voltage product of 1500 V for the switching element Q1 and the parallel resonance capacitor Cr.

Further, as described above, the full-wave rectifier circuit is provided on the secondary side so that the rectified current flows during both the positive and negative periods of the alternating voltage of the secondary winding N2A. , The secondary side resonance voltage V2 is suppressed to the same level as the rectified smoothed voltage Ei in both the positive and negative periods. Thereby, the rectifier diodes (DO1 to D04) forming the secondary-side full-wave rectifier circuit include:
It suffices to select a withstand voltage product substantially corresponding to the level of the rectified smoothing voltage Ei.

As described above, in the present embodiment, the switching element Q1, the parallel resonance capacitor Cr, and the rectifier diode forming the full-wave rectifier circuit on the secondary side have a lower breakdown voltage than those to be provided in the conventional example. Can be used, so that the element becomes less expensive. Therefore, for example, the switching element Q1 and a rectifying diode forming a full-wave rectifier circuit on the secondary side have improved characteristics (for the switching element Q1, the saturation voltage VCE (SAT ), Good characteristics such as accumulation time tSTG, fall time tf, and current amplification factor hFE, and rectifier diodes having good characteristics such as forward voltage drop VF and reverse recovery time trr). Therefore, reduction of power loss is promoted accordingly. That is, it is possible to improve the power conversion efficiency at a lower cost or almost the same cost as the conventional one. Further, by improving the power conversion efficiency, for example, a radiator plate or the like which has been required conventionally becomes unnecessary.

If the switching frequency is set higher than that of the prior art as a configuration of the voltage resonance type converter, the above-mentioned various components can be reduced in size and weight. Here, if the winding N3 is selected such that the boost voltage VB is optimized according to the maximum load power to be actually handled, further reduction in size and weight of various components can be realized.
Furthermore, from the viewpoint of reducing the size and weight of the power supply circuit, the conventional configuration including the voltage doubler rectifier circuit for generating the DC input voltage requires two sets of rectifier diodes and a smoothing capacitor. However, in the present embodiment, for example, since a full-wave rectifier circuit using a normal bridge rectifier circuit is used, a set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. The cost and the size of parts can be reduced.

FIG. 5 is a circuit diagram showing a configuration example of a voltage resonance type switching power supply circuit according to a second embodiment of the present invention. In this figure, the same parts as those in FIG.

In the power supply circuit shown in FIG.
As switching element Q1, bipolar transistors (BJT) Q11 and Q12, clamp diodes DD1 and DD
2. A Darlington circuit formed by connecting the resistors R11 and R12 as shown in the figure is provided. In this case, as the connection mode of the Darlington circuit, the collector of the transistor Q11 is connected to the collector of the transistor Q12, the emitter of the transistor Q11 is connected to the emitter of the transistor Q12, and the emitter of the transistor Q12 is grounded. . Further, the anode of the damper diode DD1 is connected to the emitter of the transistor Q11, and the cathode of the damper diode DD1 is connected to the transistor Q11 via the resistor R11.
Connected to the base. The anode of the damper diode DD2 is connected to the emitter of the transistor Q12, and the cathode is connected to the collector of the transistor Q12. The resistor R12 is connected in parallel between the base and the emitter of the transistor Q12. In the Darlington circuit thus formed, the base of the transistor Q11 is connected to the switching element Q shown in each of the above embodiments.
1 and the collector contacts of the transistors Q11 and Q12 become equivalent to the collector of the switching element Q1. Further, the emitter of the transistor Q12 is equivalent to the emitter of the switching element Q1.

In this case, the self-excited oscillation circuit for driving the switching element in a self-excited manner is omitted.
Instead, a configuration in which the oscillation / drive circuit 2 is provided and switching driving by separately excited is performed is employed. For this reason, in the present embodiment, the isolated converter transformer PIT
Are provided with windings N4A and N4B. And winding N
A DC voltage of +12 V is generated by a half-wave rectifier circuit including 4A, rectifier diode D1, and capacitor CA.
A DC voltage of -12 V is generated by a half-wave rectifier circuit composed of a winding N4B, a rectifier diode D2, and a capacitor CB. The DC voltage of +12 V and −12 V is supplied to the oscillation / drive circuit 22 as an operation power supply.

The oscillation / drive circuit 2 is activated by an activation resistor RS, and generates an oscillation signal having a cycle having a required switching frequency fs (for example, fs = 100 KHz). Then, by using an operating power supply of +12 V / −12 V, the oscillation signal is converted into a positive (on) / negative (off) switching drive current for each switching cycle and output to the base terminal of the switching element Q 1. . As a result, the switching element Q1 is driven to perform a switching operation at a required switching frequency. When a Darlington circuit is employed for the switching element Q1 as in the present embodiment, higher power conversion efficiency can be obtained than when the switching element Q1 is a single bipolar transistor, for example.

The circuit shown in FIG. 5 employs a configuration in which the controlled winding NR of the orthogonal control transformer PRT is provided in the boost circuit by being connected in parallel with the winding N3. In this connection configuration, the current I2 to flow in the boost circuit is shunted according to the ratio between the inductances LR and L3 of the controlled winding NR and the winding N3.
The boost smoothing voltage EB obtained by the boost circuit having the circuit configuration shown in FIG. 7 is also stabilized in accordance with the controlled winding NR inductance LR, as in the first embodiment. The pulse width control (switching element on / off timing control) operation can also be obtained.

FIG. 6 shows a configuration as a modification of the present embodiment. In this figure, only the isolated converter transformer PRT and the circuit on the secondary side are extracted and shown, and the configuration of the other parts is the same as that of FIG. 1 or FIG. In FIG. 6, a full-wave voltage rectification system is employed as a rectifier circuit for obtaining a DC output voltage EO on the secondary side. That is, a voltage doubler rectifier circuit formed by two smoothing capacitors CO11 and CO12 and rectifier diodes DO1 and DO2 is provided for the secondary winding N2A.
In this circuit, during the period in which the alternating voltage obtained in the secondary winding N2A is positive, a rectified current flows through the route of the secondary winding N2A → the rectifier diode DO1 → the smoothing capacitor CO11 → the secondary winding N2A, and the smoothing capacitor CO11. Charge the battery. In addition, during the period when the alternating voltage obtained in the secondary winding N2A is negative, the secondary winding N
A rectified current flows through the path of 2A → smoothing capacitor CO12 → rectifier diode DO2 → secondary winding N2A to charge the smoothing capacitor CO12. This operation is repeated every cycle. By this operation, the smoothing capacitor CO1
1, a DC voltage level corresponding to the voltage level obtained at the secondary winding N2A is obtained at CO12. Therefore, both ends of the series-connected smoothing capacitors CO11-CO12 are
A DC voltage corresponding to twice the voltage level obtained at the secondary winding N2A is obtained, and this DC voltage becomes the DC output voltage EO. In such a configuration, for example, in order to obtain a DC output voltage EO of the same level as the DC output voltage EO1 in the power supply circuits of the first and second embodiments described above, the number of turns of the secondary winding N2A Can be reduced, for example, to about 1/2 of the number of turns of the secondary winding N2 in the first and second embodiments (shown as N2A = N2 / 2 in the figure). Although not shown, a full-wave rectifier circuit using a bridge rectifier circuit may be provided to obtain a secondary-side DC output voltage.

In the description of the embodiments above,
The case where a single bipolar transistor (BJT) or a Darlington circuit having two bipolar transistors is employed as the switching element Q1 has been described as an example, but the embodiment of the present invention will be described below. The switching element can be employed instead of the switching element Q1.

FIG. 7 shows a MOS-FET (MOS type field effect transistor) instead of the switching element Q1.
An example using a metal oxide semiconductor is shown. MO
When an S-FET is used, a Zener diode ZD for forming a current path during the switching-off period is connected in parallel in the direction shown in the drawing between the drain and the source. That is, the anode is connected to the source of the MOS-FET, and the cathode is connected to the drain of the Zener diode ZD. In this case, the base, the collector, and the emitter of the switching element Q1 shown in the above embodiment are respectively the gate, the drain, and the MOS-FET.
Will be replaced by the source.

FIG. 8 shows that the switching element Q1 is replaced with
An example using an IGBT (insulated gate bipolar transistor) is shown. A diode D for forming a current path during switching off is connected in parallel between the collector and the emitter of the IGBT. Here, the anode and cathode of the diode D are I
It is connected to the collector and emitter of the GBT.
In this circuit, the base, collector and emitter of the switching element Q1 shown in each of the above embodiments are respectively
Replaced by the gate, collector and emitter of the IGBT.

FIG. 9 shows an example in which the switching element Q1 is replaced by
An example using an SIT (static induction thyristor) is shown. Even between the collector and emitter of this SIT,
A diode D for forming a current path during switching off is connected in parallel, and an anode and a cathode of the diode D are connected to a cathode and an anode of the SIT, respectively. In this circuit, the base, collector and emitter of the switching element Q1 shown in each of the above embodiments are replaced with the gate, anode and cathode of the SIT, respectively.

In this embodiment, even if any of the configurations shown in FIGS. 7 to 9 is employed, it is possible to further increase the efficiency. In the case where the configuration shown in FIGS. 7 to 9 is employed, although not shown here, the switching element Q
The configuration of the drive circuit is changed to conform to the configuration to be adopted instead of 1. Further, the details of the configuration shown in each of the above-described embodiments may be changed in accordance with actual use conditions and the like.

[0057]

As described above, the present invention provides, as a voltage resonance type switching power supply circuit corresponding to a relatively high load condition of, for example, 150 W or more in an AC input voltage AC 100 V system, a boost voltage for a rectified smoothed voltage. A boost circuit (boost means) for superimposing and obtaining a boosted smoothed voltage is provided, so that the maximum load power that can be handled is, for example, a rectified smoothed voltage corresponding to twice the AC input voltage level by a voltage doubler rectifier circuit. Is substantially equal to the case of Furthermore, by making the insulating converter transformer loosely coupled, an operation mode (+ M / -M) in which the mutual inductances of the primary winding and the secondary winding have opposite polarities is obtained. In addition, by providing the full-wave rectifier circuit on the secondary side, the secondary-side DC output voltage is obtained from the alternating voltage obtained in the secondary winding. That is, as a result of supplying power to the load by the full-wave rectifier circuit on the secondary side, in the present invention, the maximum load power that can be handled is smaller than in the conventional case where the secondary-side DC output voltage is obtained by the half-wave rectifier circuit. Can be improved. Along with this, the primary side is not a voltage doubler rectifier circuit, but a full-wave rectifier circuit is used to input a rectified smoothed voltage at a level corresponding to the AC input voltage level. It will be able to respond.

For example, conventionally, when the above conditions are satisfied, it is necessary to obtain a rectified smoothed voltage corresponding to twice the AC input voltage level by a voltage doubler rectifier circuit. It is necessary to select a withstand voltage product corresponding to the switching voltage generated according to the rectified smoothing voltage level for the parallel resonance capacitor. In the related art, a DC output voltage is generated by a half-wave rectifier circuit on the secondary side.
Since a voltage about twice to 3.5 times is applied, a withstand voltage product corresponding to this voltage level has been selected.

On the other hand, in the present invention, since the switching voltage depending on the rectified smoothing voltage level is １ ／ of the conventional one, the switching element and the resonance capacitor on the primary side have much lower withstand voltage than the conventional one. Component elements can be used. On the secondary side, an alternating voltage (resonance voltage) obtained in a secondary winding forming a parallel resonance circuit
Is rectified by a full-wave rectifier circuit to obtain a DC output voltage. As a result, after increasing the maximum load power that can be handled, the voltage applied to the rectifier diode is suppressed to approximately the same level as the rectification smoothed voltage level. Things can be selected. As a result, it is possible to reduce costs for the switching element, the primary-side parallel resonance capacitor, the secondary-side rectifier diode, and the like. In addition, if the switching element and the secondary rectifier diode having improved characteristics are selected, the power conversion efficiency can be improved. Further, if the characteristics of the switching element are improved, it becomes easy to increase the switching frequency. Therefore, if the switching frequency is increased, it is possible to reduce the size and weight of circuit components around the switching element. . Further, since a circuit for obtaining a rectified smoothed voltage from a commercial AC power supply is a normal full-wave rectifier circuit, for example, a normal set of a block-type smoothing capacitor and a bridge rectifier diode can be employed. However, cost and circuit scale can be reduced.

According to the present invention, the controlled winding of the orthogonal control transformer is inserted into the boost circuit, and the on-period of the switching element (ie, the resonance voltage) is changed due to the change in the inductance of the controlled winding. Pulse width)
Is variably controlled to perform constant voltage control of the secondary side output voltage, while the boost smoothing voltage is also controlled to be constant. Thus, the control sensitivity is improved, and the effect of expanding the constant voltage control range and reducing the ripple component of the DC output voltage is obtained. Also, for example, it is possible to increase the minimum load power. In addition, since the on / off period of the switching operation, which is an operation as an inductance control method, is controlled after the boost circuit is provided, it is possible to increase the minimum load power. For this reason, in the present invention, it is possible to reduce the current to be passed through the controlled winding, for example, as compared with the conventional case where the controlled winding is inserted into the primary side parallel resonance circuit. This makes it possible to reduce the size of the orthogonal control transformer.
The boost circuit includes a boost winding, a controlled winding, and a series connection circuit in which a boost rectifier diode is connected in series, and a boost that obtains a boost voltage by charging a rectified output of the series connection circuit. And a smoothing capacitor. Further, as for the boosting smoothing capacitor, a rectified smoothed voltage (DC input voltage) is obtained by a normal full-wave rectification method instead of a voltage doubler rectification method.
An inexpensive and small-sized product can be adopted as long as a pressure-resistant product of a sufficient level can be selected.

Further, as the present invention, as the rectifier circuit provided on the secondary side, a full-wave rectifier circuit using a bridge rectifier circuit, a full-wave rectifier circuit of a center tap system, and a double voltage full-wave rectifier circuit are used. Although it is possible to have a certain degree of freedom as a circuit form, the number of turns of the secondary winding should be reduced to about 1/2 of the conventional one, especially when a voltage doubler full-wave rectifier circuit is adopted. Also becomes possible.

The switching means can be constituted by a Darlington circuit formed with a bipolar transistor, or a MOS type field effect transistor, an insulated gate bipolar transistor, or an electrostatic induction thyristor. For example, the power conversion efficiency can be further improved as compared with the case where the switching means is formed by a single bipolar transistor.

As described above, according to the present invention, the cost, size, and weight of the power supply circuit having the voltage resonance type converter and the improvement of the power conversion efficiency are promoted.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to a first embodiment of the present invention.

FIG. 2 is a cross-sectional view illustrating a configuration of an insulating converter transformer of the power supply circuit according to the present embodiment.

FIG. 3 shows an AC input voltage and a DC input voltage (rectified smoothed voltage,
FIG. 5 is an explanatory diagram showing a relationship with a boost smoothed voltage.

FIG. 4 is a waveform diagram illustrating an operation of a main part of the power supply circuit according to the first embodiment.

FIG. 5 is a circuit diagram showing a configuration example of a power supply circuit according to a second embodiment of the present invention.

FIG. 6 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 7 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 8 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 9 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.

FIG. 10 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 11 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.

FIG. 12 is a circuit diagram showing a configuration of an insulating converter transformer of a power supply circuit as a conventional example.

FIG. 13 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 14 is a perspective view showing a configuration of an orthogonal control transformer provided in a power supply circuit as a conventional example.

FIG. 15 is a waveform chart showing an operation of a main part of the power supply circuit shown in FIGS. 10 and 11;

FIG. 16 is an explanatory diagram showing inductance superposition characteristics of a constant voltage control system in the power supply circuits shown in FIGS. 10 and 11;

[Explanation of symbols]

1 control circuit, 2 oscillation circuit, 3 drive circuit, C
i, CiB smoothing capacitor, Cr parallel resonance capacitor, C2 (secondary side) parallel resonance capacitor, Di, DO
Bridge rectifier circuit, DO1, DO2, DO3, DO4 rectifier diode, PIT isolated converter transformer, N1 primary winding, N2, N2A secondary winding, PRT orthogonal control transformer, NC control winding, NR controlled winding, Q1 Switching element, diode for DB boost, N3 (for boost) winding

Claims (8)

[Claims] 1. A rectifying / smoothing means for generating a rectified / smoothed voltage having a level corresponding to an input commercial AC power supply level by full-wave rectification, and a gap formed so as to obtain a required coupling coefficient which is loosely coupled. An insulating converter transformer provided for transmitting a primary output to a secondary side; and a rectifying and smoothing voltage output from the rectifying and smoothing means being intermittently output to a primary winding of the insulating converter transformer. Switching means, at least a primary-side resonance circuit formed by an inductance component including a primary winding of the insulated converter transformer and a capacitance of a resonance capacitor to make the operation of the switching means a voltage resonance type; and Of the secondary winding and the capacitance of the secondary side parallel resonant capacitor And a secondary-side resonance circuit formed on the secondary side, and an alternating voltage obtained in a secondary winding of the insulating converter transformer, and a DC output voltage generation for obtaining a secondary-side DC output voltage by full-wave rectification. Means, a controlled winding provided so as to function as an inductance component of the primary side resonance circuit, and a control winding whose winding direction is orthogonal to the controlled winding. An orthogonal control transformer having a variable control current flowing through the control winding according to the level of the DC output voltage to change the inductance of the controlled winding.
Constant voltage control means configured to perform constant voltage control on the secondary side DC output voltage; and superimposing a boost voltage generated using the switching output of the switching means on the rectified smoothed voltage, thereby boosting A boost configured to obtain a rectified smoothed voltage and including the above-described controlled winding, so as to be configured to be able to control the boost rectified smoothed voltage to be constant by a change in inductance of the controlled winding. Means, wherein the boost means is formed by winding up a primary winding of the insulated converter transformer to obtain a switching output, and a boost for rectifying an alternating voltage obtained in the boost winding. And a series connection circuit formed by connecting a rectifier diode in series with the boost winding. The controlled winding connected to a column, and a boosting smoothing capacitor provided so as to generate a boost voltage at both ends thereof by charging a rectified output obtained by the series connection circuit, A voltage resonance type switching power supply circuit formed by connecting a negative electrode side of a smoothing capacitor for use to a positive electrode side of a smoothing capacitor forming the rectifying / smoothing means. 2. The voltage resonance type switching power supply circuit according to claim 1, wherein said DC output voltage generation means is configured as a full-wave rectification circuit having a bridge rectification circuit. 3. The voltage-resonant switching power supply circuit according to claim 1, wherein said DC output voltage generating means is provided with a rectifier circuit based on a center tap full-wave rectification method. 4. The voltage resonance type switching power supply circuit according to claim 1, wherein said DC output voltage generation means includes a voltage doubler full-wave rectifier circuit. 5. The voltage resonance type switching power supply circuit according to claim 1, wherein said switching means is a Darlington circuit formed with a bipolar transistor. 6. The voltage resonance type switching power supply circuit according to claim 1, wherein said switching means is a MOS field effect transistor. 7. The voltage resonance type switching power supply circuit according to claim 1, wherein said switching means is an insulated gate bipolar transistor. 8. The voltage resonance type switching power supply circuit according to claim 1, wherein said switching means is an electrostatic induction thyristor.