JP4218089B2 - Switching power supply circuit - Google Patents

Description

により表すことができる。
そして、巻線Ｎ3及び一次巻線Ｎ1のインダクタンスとしてＬ3＝Ｌ1の関係が得られるようにし、整流平滑電圧Ｅｉ、ブースト用ダイオードＤBの降下電圧ＶF、及びスイッチング素子Ｑ1の飽和電圧Ｖ(SAT)についてＥｉ≫ＶF，Ｖ(SAT)の関係が成立しているとすると、ブースト平滑電圧ＥBは上記（数１）に基づいて、
【数２】

により示されることになる。この場合、図６に示す電源回路では、例えば、直交型トランスＰＲＴの被制御巻線ＮRのインダクタンスＬRを、０．１×Ｌ1〜１．２×Ｌ1の範囲で変化させることで、ブースト平滑電圧ＥBについて、ほぼＥｉ〜２Ｅｉ（Ｅｉは、平滑コンデンサＣｉの両端に得られる整流平滑電圧レベルに相当する）の範囲で可変することが可能とされる。
【００４６】
また、本実施の形態の電源回路の接続形態では、ブースト用の整流ダイオードＤBを介して、一次巻線Ｎ1と平滑コンデンサＣｉの正極間に対して被制御巻線ＮRが直列に挿入されているものと見ることが出来るので、一次側の並列共振回路は、被制御巻線ＮRのインダクタンスＬRを含んで形成されることになる。従って、この場合には、図１に示した電源回路と同様に、共振電圧Ｖｃｒのパルス幅を制御することによって二次側出力の安定化が行われる。
【００４７】
本実施の形態では、スイッチング素子Ｑ1を備えて成る電圧共振形スイッチングコンバータは、上述のブースト平滑電圧ＥBを動作電源としてスイッチング動作を行うようにされる。つまり、本実施の形態では、ブースト回路によって、電圧共振形スイッチングコンバータに供給すべき見かけ上の直流入力電圧レベルを上昇させているものである。このブースト回路の動作によって直流入力電圧レベルを引き上げていることで、本実施の形態では、倍電圧整流回路によって直流入力電圧を得る構成（即ち従来例）の場合とほぼ同等の最大負荷電力に対応することが可能となる。そして、先の各実施の形態と同様にして、二次側直列共振動作を利用した倍電圧整流回路により直流出力電圧を得るようにされていることで、結果的には２００Ｗ程度の最大負荷電力に対応することが可能となる。
なお、本実施の形態においても、先の第１の実施の形態にて説明した効果は同様に得られるものである。
【００４８】
図７は、本発明の第４の実施の形態としての、複合共振形スイッチングコンバータを備えた電源回路の構成を示す回路図であり、図１、図４、図６及び図１２、図１３と同一部分には同一符号を付して説明を省略する。
【００４９】
この図に示す電源回路においては、一次巻線としてはセンタータップが設けられて、一次巻線Ｎ1A（インダクタンスＬ1A），Ｎ1B（インダクタンスＬ1B）に分割される。このセンタータップ端子は、直交型制御トランスＰＲＴの被制御巻線ＮR（インダクタンスＬR）の直列接続を介して整流平滑電圧Ｅｉのラインと接続される。
【００５０】
この図に示す電圧共振形コンバータは、２石のスイッチング素子Ｑ1，Ｑ2を備えた他励式の構成を採っている。この場合、スイッチング素子Ｑ1,Ｑ2には、ＭＯＳ−ＦＥＴが採用されている。
この場合、スイッチング素子Ｑ1 のドレインは一次巻線Ｎ1Aの端部と接続され、ソースは一次側アースに接地される。また、スイッチング素子Ｑのドレイン−ソース間には、スイッチングオフ時の帰還電流の経路を形成するダンパーダイオードＤD1、及び並列共振コンデンサＣｒ1が並列に接続される。
スイッチング素子Ｑ2 のドレインは一次巻線Ｎ1Bの端部と接続され、ソースは一次側アースに接地される。また、スイッチング素子Ｑ2のドレイン−ソース間に対しても、ダンパーダイオードＤD2及び並列共振コンデンサＣｒ2が並列に接続される。
【００５１】
並列共振コンデンサＣｒ1は、自身のキャパシタンスと、インダクタンスＬ1A，インダクタンスＬRの合成インダクタンスとによって、スイッチング素子Ｑ1のスイッチング動作を電圧共振形とする並列共振回路を形成される。
同様にして、並列共振コンデンサＣｒ2は、自身のキャパシタンスと、インダクタンスＬ1B，インダクタンスＬRの合成インダクタンスとによって、スイッチング素子Ｑ2のスイッチング動作を電圧共振形とする並列共振回路を形成する。
【００５２】
この接続形態では、スイッチング素子Ｑ1，Ｑ2がそれぞれ電圧共振形のスイッチング動作を行うようにされた、二系統のスイッチング回路系が備えられることになる。なお、上記スイッチング素子Ｑ1，Ｑ2のスイッチング出力点は、一次巻線Ｎ1のセンタータップとなる。
【００５３】
発振ドライブ回路２は、起動抵抗ＲSにより起動されるようになっており、例えばスイッチング周波数ｆｓ＝１００ＫＨｚに対応する周波数信号を発振出力すると共に、この周波数信号に基づいてスイッチング駆動信号（駆動電圧）を生成して、スイッチング素子Ｑ1，Ｑ2のゲートに印加する。この際、スイッチング素子Ｑ1，Ｑ2のゲートに印加するスイッチング駆動信号としては、互いに反転した逆極性の信号である。
【００５４】
上記発振ドライブ回路２によってスイッチング素子Ｑ1，Ｑ2が駆動されると、スイッチング素子Ｑ1，Ｑ2は、例えばスイッチング周波数ｆｓ＝１００ＫＨｚで、交互にオン／オフとなるタイミングでスイッチング動作を行うことになる。
つまり、スイッチング素子Ｑ1，Ｑ2の両者によるスイッチング動作は、いわゆるプッシュプル動作となり、例えば１石のスイッチング素子による、いわゆるシングルエンド動作の場合よりも大きな電力を負荷側に対して供給することができる。これにより、本実施の形態では最大負荷電力２００Ｗ以上の重負荷の条件に対応する。
【００５５】
また、この場合には直交型制御トランスＰＲＴの被制御巻線ＮRは一次巻線に対して接続されることから、例えば図１に示した電源回路と同様に、インダクタンス制御方式として、スイッチング素子と並列共振コンデンサの並列接続回路の両端に得られる共振電圧をＰＷＭ制御することで定電圧制御を行うようにされる。
【００５６】
図７に示す構成によっても、複合形共振コンバータとされたことによる効果、及び等倍電圧整流回路によって直流入力電圧を得たことによるよる効果が、先の各実施の形態と同様に得られるものである。
【００５７】
これまでの実施の形態の説明においては、スイッチング素子Ｑ1（及びＱ2）として、１石のバイポーラトランジスタ（ＢＪＴ）、或いはＭＯＳ−ＦＥＴを採用した場合を例に挙げていたが、本発明の実施の形態としては、以降示すようなスイッチング回路或いはスイッチング素子をスイッチング素子Ｑ1（Ｑ2）に代えて採用することも可能である。
図８は、スイッチング素子Ｑ1（Ｑ2）に代えて、２本のバイポーラトランジスタＱ11，Ｑ12をダーリントン接続した回路を使用する例を示している。
この場合の接続形態としては、トランジスタＱ11のコレクタとトランジスタＱ12のコレクタを接続し、トランジスタＱ11のエミッタをトランジスタＱ12のエミッタと接続し、トランジスタＱ12のエミッタをアースに接地している。また、ダンパーダイオードＤD1のアノードをトランジスタＱ11のエミッタと接続し、ダンパーダイオードＤD1のカソードを抵抗Ｒ11を介してトランジスタＱ11のベースに接続している。ダンパーダイオードＤD2のアノードは、トランジスタＱ12のエミッタに接続され、カソードはトランジスタＱ12のコレクタに接続されている。抵抗Ｒ12は、トランジスタＱ12のベース−エミッタ間に対して並列に接続されている。このようにして形成したダーリントン接続回路においては、トランジスタＱ11のベースが先の各実施の形態に示したスイッチング素子Ｑ1（Ｑ2）のベースと等価となり、トランジスタＱ11，Ｑ12のコレクタ接点がスイッチング素子Ｑ1（Ｑ2）のコレクタと等価となる。また、トランジスタＱ12のエミッタがスイッチング素子Ｑ1（Ｑ2）のエミッタと等価となる。
【００５８】
また、第１〜第３の実施の形態に対応する図１、図４、図６の電源回路のようにして、１つのスイッチング素子Ｑ1によりシングルエンド動作を行う構成を採る場合にも、図９に示すようにして、バイポーラトランジスタのスイッチング素子Ｑ1に代えて、ＭＯＳ−ＦＥＴ（ＭＯＳ型電界効果トランジスタ；金属酸化膜半導体）を使用することができる。ＭＯＳ−ＦＥＴを用いる場合、ドレイン−ソース間に対して、スイッチングオフ時の帰還電流の経路を形成するためのツェナーダイオードＺＤが図に示す方向により並列に接続される。つまり、アノードがＭＯＳ−ＦＥＴのソースと接続され、カソードがツェナーダイオードＺＤのドレインと接続される。この場合、先の各実施の形態に示したスイッチング素子Ｑ1のベース、コレクタ、エミッタは、それぞれ、ＭＯＳ−ＦＥＴのゲート、ドレイン、ソースに置き換わることになる。また、ＭＯＳ−ＦＥＴを用いる場合には、図７に示したような他励式の構成を採るなどして、電流駆動ではなく、電圧駆動する必要がある。
【００５９】
図１０は、スイッチング素子Ｑ1（Ｑ2）に代えて、ＩＧＢＴ（絶縁ゲートバイポーラトランジスタ）を使用した例が示されている。ＩＧＢＴのコレクタ−エミッタ間に対しては、スイッチングオフ時の帰還電流の経路を形成するためのダイオードＤが並列に接続される。ここでは、ダイオードＤのアノード、カソードはそれぞれＩＧＢＴのコレクタ，エミッタに対して接続されている。
この回路では、先の各実施の形態に示したスイッチング素子Ｑ1（Ｑ2）のベース、コレクタ、エミッタは、それぞれ、ＩＧＢＴのゲート、コレクタ、エミッタに置き換わる。
【００６０】
図１２は、スイッチング素子Ｑ1（Ｑ2）に代えて、ＳＩＴ（静電誘導サイリスタ）を使用した例が示されている。このＳＩＴのコレクタ−エミッタ間に対しても、スイッチングオフ時の帰還電流の経路を形成するためのダイオードＤが並列に接続され、ダイオードＤのアノード、カソードがそれぞれＳＩＴのカソード，アノードに対して接続される。
上記図８〜図１２に示す何れの構成を採った場合にも、本実施の形態では更なる高効率化を図ることが可能になる。なお、図８〜図１２に示す構成を採る場合、ここでは図示しないが、実際にスイッチング素子Ｑ1（Ｑ2）に代えて採られるべき構成に適合するようにして、その駆動回路の構成を変更して構わないものである。
また、実施の形態として上記各図に示した構成の細部は、実際の使用条件等に応じて変更されて構わない。
【００６１】
【発明の効果】
以上説明したように本発明は、スイッチング電源回路として、一次側に電圧共振形スイッチングコンバータを備えたうえで、絶縁コンバータトランスを疎結合とすることで、一次巻線と二次巻線の相互インダクタンスが互いに逆極性となる動作モード（＋Ｍ／−Ｍ）が得られるようにしている。更に、二次側においては、二次巻線に二次側直列共振回路を直列に接続して直列共振回路を形成して、この直列共振回路を利用した倍電圧全波整流回路を備えることで、二次巻線に得られる交番電圧（励起電圧）の二倍に対応する二次側直流出力電圧を得るようにされる。
【００６２】
上記のようにして倍電圧全波整流回路によって負荷に電力供給をする結果、本発明では、例えば従来のように半波整流回路により等倍の二次側直流出力電圧を得る場合よりも、対応可能な最大負荷電力を向上させることが可能になる。
そしてこれに伴い、一次側は倍電圧整流回路ではなく、通常の全波整流回路により交流入力電圧レベルに対応するレベルの整流平滑電圧を入力するように構成しても、充分に上記した条件に対応することができることになる。
【００６３】
以上の構成から次のようなことが言える。
例えば従来においては、上記の条件に対応する場合には、倍電圧整流回路により交流入力電圧レベルの２倍に対応する整流平滑電圧を得る必要があり、このため、スイッチング素子や一次側の並列共振コンデンサには、整流平滑電圧レベルに応じて発生するスイッチング電圧に応じた耐圧品を選定する必要があった。
また、従来においては、二次側において半波整流回路により直流出力電圧を生成するようにしていたことで、整流ダイオードの非導通期間において整流平滑電圧の２．５倍〜３．５倍程度の電圧が印加されるため、この電圧レベルに応じた耐圧品を選定していた。
【００６４】
これに対して本発明では、整流平滑電圧レベルに依存するスイッチング電圧が従来の１／２となることから、スイッチング素子や一次側の共振コンデンサについて、従来の１／２程度の耐圧品を用いることができる。
また、二次側においては、前述したように、倍電圧全波整流回路が設けられるが、ここで倍電圧全波整流回路は、交番電圧が正／負の両期間で整流動作を行う全波整流動作である結果、整流ダイオードに印加される電圧は整流平滑電圧レベルとほぼ同等に抑制されるため、二次側の整流ダイオードについても従来より耐圧の低いものを選定することができる。
これによって、先ずスイッチング素子、一次側の並列共振コンデンサ、及び二次側整流ダイオード等にかかるコストを削減することができる。また、スイッチング素子及び二次側整流ダイオードの特性の向上したものを選定して、スイッチング周波数を高く設定することも容易に可能となり、これによって、電力変換効率の向上が図られることになる。また、スイッチング素子周辺の回路部品の小型・軽量化を図ることも可能になるものである。
また、前述のように、商用交流電源から整流平滑電圧を得る回路が通常の等倍電圧整流回路とされたことで、例えば通常の１組のブロック型の平滑コンデンサとブリッジ整流ダイオードを採用することができるので、この点でも、コストの削減及び回路規模の縮小が図られる。更には、被制御巻線の巻数が削減されて、定電圧制御に用いる直交型制御トランスの小型軽量化及び低コスト化も図られるものである。
【００６５】
更に本発明として、二次側に設けられる整流回路については倍電圧全波整流回路が採用されることで、例えば等倍電圧整流回路が備えられる場合と同等レベルの直流出力電圧を得ようとすれば、二次巻線の巻数を従来の１／２程度にまで少なくすることが可能になる。
また、本発明の構成では、従来よりも狭い被制御巻線のインダクタンス可変範囲によって二次側出力の安定化動作を確保できるので、その分被制御巻線の巻数を削減して直交型制御トランスの小型軽量化及び低コスト化を図ることが可能である。
【００６６】
更には、スイッチング手段としてプッシュプル動作を行う構成を採れば、比較的簡略な回路構成によっても、対応可能な負荷電力を更に増加させることが可能になる。
【００６７】
また、スイッチング素子としては、バイポーラトランジスタを備えて形成されるダーリントン回路、又はＭＯＳ型電界効果トランジスタ、絶縁ゲートバイポーラトランジスタ、又は静電誘導サイリスタにより構成することが可能であり、この場合には、例えば１石のバイポーラトランジスタにより上記スイッチング手段を形成する場合よりも、更に電力変換効率を向上させることが可能となる。
【００６８】
このように本発明では、電圧共振形コンバータを備えた電源回路の低コスト化、小型軽量化、及び電力変換効率等の諸特性の向上が促進されるものである。
【図面の簡単な説明】
【図１】本発明の第１の実施の形態の電源回路の構成例を示す回路図である。
【図２】本実施の形態の電源回路の絶縁コンバータトランスの構成を示す断面図である。
【図３】第１の実施の形態の電源回路の要部の動作を示す波形図である。
【図４】本発明の第２の実施の形態の電源回路の構成例を示す回路図である。
【図５】図４に示す回路の二次側直列共振周波数に対するスイッチング周波数、及び二次側直流出力電圧との関係を示す説明図である。
【図６】本発明の第３の実施の形態の電源回路の構成例を示す回路図である。
【図７】本発明の第４の実施の形態の電源回路の構成例を示す回路図である。
【図８】本発明の実施の形態の変形例としての構成を示す回路図である。
【図９】本発明の実施の形態の変形例としての構成を示す回路図である。
【図１０】本発明の実施の形態の変形例としての構成を示す回路図である。
【図１１】本発明の実施の形態の変形例としての構成を示す回路図である。
【図１２】従来例としての電源回路の構成を示す回路図である。
【図１３】従来例としての電源回路の構成を示す回路図である。
【図１４】従来例としての絶縁コンバータトランスの構成を示す断面図である。
【図１５】相互インダクタンスが＋Ｍ／−Ｍの場合の各動作を示す説明図である。
【図１６】図１２及び図１３に示す電源回路における要部の動作を示す波形図である。
【符号の説明】
１ 制御回路、２ 発振・ドライブ回路、Ｃｉ，ＣｉB 平滑コンデンサ、Ｃｒ 並列共振コンデンサ、Ｃｓ1，Ｃｓ2 二次側直列共振コンデンサ、Ｄｉ ブリッジ整流回路、ＤO1，ＤO2，ＤO3，ＤO4 整流ダイオード、ＰＩＴ 絶縁コンバータトランス、ＰＲＴ 直交型制御トランス、ＮC 制御巻線、ＮR 被制御巻線、Ｑ1，Ｑ2 スイッチング素子、ＤB ブースト用ダイオード、Ｎ3 （ブースト用）巻線[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a switching power supply circuit provided as a power source for various electronic devices.
[0002]
[Prior art]
As a so-called soft switching power supply circuit, a switching power supply circuit including a voltage resonance type switching converter is known. The voltage resonance type switching converter has low noise because a smooth waveform can be obtained with respect to the switching output pulse voltage and the switching output current flowing into the insulating converter transformer, and can be configured with a relatively small number of parts. Has a merit.
[0003]
The circuit diagram of FIG. 12 shows an example of a voltage resonance type switching power supply circuit that can be configured based on the invention previously proposed by the present applicant.
In the switching power supply circuit shown in this figure, for example, a commercial AC power supply in Japan or the United States is a so-called AC100V system, and the maximum load power corresponds to a condition of 150 W or more.
[0004]
In the switching power supply circuit shown in this figure, a so-called voltage doubler rectifier circuit comprising rectifier diodes Di1, Di2 and smoothing capacitors Ci1, Ci2 is provided as a rectifying / smoothing circuit for rectifying and smoothing commercial AC power supply AC. In this voltage doubler rectifier circuit, for example, when a DC input voltage corresponding to one time the peak value of the AC input voltage VAC is Ei, a DC input voltage 2Ei approximately twice that is generated. For example, if the AC input voltage VAC = 144V, the DC input voltage 2Ei is about 400V.
As described above, the voltage doubler rectifier circuit is adopted as the rectifying / smoothing circuit, as described above, in response to a relatively heavy load condition in which the AC input voltage is AC100V and the maximum load power is 150 W or more. Because. That is, by making the DC input voltage twice that of the normal one, the amount of current flowing into the subsequent switching converter is suppressed, and the reliability of the components forming the switching power supply circuit is ensured. .
For the voltage doubler rectifier circuit shown in this figure, an inrush current limiting resistor Ri is inserted in the rectified current path so as to suppress an inrush current flowing into the smoothing capacitor when the power is turned on, for example. Yes.
[0005]
The voltage resonance type switching converter in this figure has a self-excited configuration including one switching element Q1. In this case, a high breakdown voltage bipolar transistor (BJT; junction transistor) is employed as the switching element Q1.
The base of the switching element Q1 is connected to the positive side of the smoothing capacitor Ci1 (rectified smoothing voltage 2Ei) via the starting resistor RS so that the base current at the time of starting can be obtained from the rectifying smoothing line. A resonant circuit for self-excited oscillation comprising an inductor LB, a detection drive winding NB, a resonant capacitor CB, and a damping resistor RB is connected in series between the base of the switching element Q1 and the temporary ground. In this case, the detection drive winding NB is wound around an insulating converter transformer PIT (Power Isolation Transformer), and a required inductance for setting the switching frequency is obtained together with the inductor LB.
A clamp diode DD inserted between the base of the switching element Q1 and the negative electrode (primary side ground) of the smoothing capacitor Ci forms a path for a damper current that flows when the switching element Q1 is turned off. The collector of the switching element Q1 is connected to one end of the primary winding N1 of the insulating converter transformer PIT, and the emitter is grounded.
[0006]
A parallel resonant capacitor Cr is connected in parallel between the collector and emitter of the switching element Q1. This parallel resonant capacitor Cr has a combined inductance (capacitance obtained by series connection of its own capacitance, a primary winding N1 of an insulating converter transformer PIT described later, and a controlled winding N R of an orthogonal control transformer PRT (Power Regulating Transformer)). L1 and LR) form a parallel resonance circuit of a voltage resonance type converter. Although detailed description is omitted here, when the switching element Q1 is turned off, the voltage Vcr across the resonance capacitor Cr is actually a sinusoidal pulse waveform due to the action of the parallel resonance circuit. Can be obtained.
[0007]
The insulating converter transformer PIT is for transmitting the switching output of the switching element Q1 to the secondary side. In this case, one end of the primary winding N1 of the insulating converter transformer PIT is connected to the collector of the switching element Q1. The other end is connected in series with the controlled winding NR of the orthogonal control transformer PRT as shown in the figure.
[0008]
On the secondary side of the insulating converter transformer PIT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, a secondary resonant circuit is formed by connecting a secondary side parallel resonant capacitor C2 in parallel to the secondary winding N2. By this parallel resonance circuit, the alternating voltage excited in the secondary winding N2 is a resonance voltage, and this resonance voltage is generated from a half-wave rectifier circuit including a rectifier diode DO1 and a smoothing capacitor CO2, and from the rectifier diode DO2 and the smoothing capacitor CO2. Are supplied to two sets of half-wave rectifier circuits. The two sets of half-wave rectifier circuits provide DC output voltages EO1 and EO2, respectively.
The rectifier diodes DO1 and DO2 forming the half-wave rectifier circuit use a high-speed type for rectifying the alternating voltage of the switching period.
[0009]
The control circuit 1 is, for example, an error amplifier that compares a DC voltage output on the secondary side with a reference voltage and supplies a DC current corresponding to the error to the control winding NC of the orthogonal control transformer PRT as a control current. In this case, a DC voltage output E01 is input to the control circuit 1 as a detection voltage, and a DC output voltage E02 is input as an operating power supply.
[0010]
For example, when the secondary side DC output voltage EO2 fluctuates with fluctuations in the AC input voltage VAC or load power, the control current flowing through the control winding NC is changed by the control circuit 1 in the range of, for example, 10 mA to 40 mA. . As a result, the inductance LR of the controlled winding NR is changed in the range of 0.1 mH to 0.6 mH, for example.
[0011]
Since the controlled winding NR forms a parallel resonance circuit for obtaining a voltage resonance type switching operation as described above, the resonance of the parallel resonance circuit is fixed with respect to a fixed switching frequency. Conditions are made to change. At both ends of the parallel connection circuit of the switching element Q1 and the parallel resonance capacitor Cr, a sinusoidal resonance pulse is generated by the action of the parallel resonance circuit corresponding to the OFF period of the switching element Q1, but the resonance condition of the parallel resonance circuit The width of the resonance pulse is variably controlled by changing. That is, a PWM (Pulse Width Moduration) control operation for the resonance pulse is obtained. The PWM control of the resonance pulse width is control of the off period of the switching element Q1. In other words, this means that the on period of the switching element Q1 is variably controlled under the condition of a fixed switching frequency. . In this way, the ON period of the switching element Q1 is variably controlled, so that the switching output transmitted from the primary winding N1 forming the parallel resonant circuit to the secondary side changes, and the secondary side DC output voltage ( The output level of EO1, EO2) is also changed. As a result, the secondary side DC voltage (EO1, EO2) is made constant. Such a constant voltage control method is hereinafter referred to as an inductance control method.
[0012]
The circuit diagram of FIG. 13 shows another example of a voltage resonance type switching power supply circuit that can be configured based on the invention previously proposed by the present applicant. In FIG. 13, the same parts as those in FIG. 12 are denoted by the same reference numerals, and description of parts having the same configuration is omitted.
In the power supply circuit shown in this figure, an example is shown in which the controlled winding of the orthogonal control transformer PRT is provided on the secondary side.
In this case, as the controlled winding of the orthogonal control transformer PRT, two controlled windings NR and NR1 are wound and provided, and the controlled winding NR is connected to the end of the secondary winding N2. The rectifier diodes DO1 are connected so as to be inserted in series between the anodes. The controlled winding NR1 is inserted in series between the tap output of the secondary winding N2 and the anode of the rectifier diode D02. In such a connection form, the secondary side parallel resonant circuit is formed including the inductance components of the controlled windings NR and NR1.
[0013]
As described above, when the controlled winding (NR, NR1) of the orthogonal control transformer PRT is provided on the secondary side, the inductance of the controlled winding NR can be varied as an inductance control method. Thus, the pulse width of the resonance voltage V2 of the secondary side parallel resonant capacitor C2, that is, the conduction angle of the secondary side rectifier diode, is variably controlled. Thus, constant voltage control is achieved by controlling the output level obtained on the secondary side.
[0014]
Here, the structure of the insulating converter transformer PIT provided in the power supply circuit shown in FIGS. 12 and 13 is shown in cross section in FIG.
The insulating converter transformer PIT is formed with an EE type core in which E type cores CR1 and CR2 made of, for example, a ferrite material are combined so that their magnetic legs face each other. At this time, no gap is formed in the central magnetic leg as shown in the figure. The central magnetic leg is wound by using the bobbin B in a state where the primary winding N1 and the secondary winding N2 are divided. Thereby, a state of loose coupling (for example, coupling coefficient k≈0.9) is obtained between the primary winding N1 and the secondary winding N2.
[0015]
Further, in the insulating converter transformer PIT, the inductance L1 of the primary winding N1 and the second winding N1 depend on the relationship between the polarity (winding direction) of the primary winding N1 and secondary winding N2 and the connection of the rectifier diodes DO (DO1, DO2). The mutual inductance M with the inductance L2 of the next winding N2 may be + M (additional polarity mode) or -M (depolarization mode).
For example, the mutual inductance is + M in the operation when the connection configuration shown in FIG. 15A is adopted, and the mutual inductance is −M in the operation when the connection configuration shown in FIG. 15B is adopted.
[0016]
FIG. 16 shows an operation waveform according to the switching period of the power supply circuit shown in FIG. In this figure, periods TON and TOFF indicate periods during which the switching element Q1 is turned on and off, respectively, and periods DON and DOFF indicate periods during which the secondary side rectifier diode DO1 is turned on and off, respectively.
As the resonance voltage Vcr generated at both ends of the switching element Q1 / parallel resonance capacitor Cr, as shown in FIG. 16 (a), a waveform that becomes a sinusoidal pulse in the period TOFF in which the switching element Q1 is off is obtained. The operation of the switching converter is a voltage resonance type. The level of the pulse of the resonance voltage Vcr is about 1800 V at the peak as shown in the figure. This is due to the fact that the impedance of the parallel resonance circuit on the primary side of the voltage resonance type converter acts on the 2Ei DC input voltage obtained by voltage doubler rectification.
Further, as a secondary side operation, the secondary side rectifier diode D01 can obtain an operation in which a rectified current flows in a period DON substantially corresponding to the period TON of the switching element, by the waveform shown in FIG. That is, the operation is based on + M (additive polarity mode) described with reference to FIG. In this regard, substantially the same operation timing can be obtained on the rectifier diode D02 side.
As a result of this rectification operation, the resonant voltage V2 obtained across the secondary parallel resonant capacitor C2 is a DC output during the period DOFF when the rectifier diode D01 is off, as shown in FIG. Equivalent to the DC output voltage EO (EO1, EO2) during the period DON when the rectifier diode DO1 is turned on when a sine wave voltage with a peak level of about 2 to 3.5 times the voltage EO (EO1, EO2) is applied. The voltage level is applied.
[0017]
[Problems to be solved by the invention]
By the way, in the voltage resonance type converter having the configuration described with reference to FIGS. 12 to 16, since the AC input voltage VAC corresponds to the AC 100V system and the maximum load power is 150 W or more, 2Ei is obtained by the voltage doubler rectification method. A DC input voltage of a level is obtained. For this reason, as shown in FIG. 16A, a resonance voltage Vcr of 1800 V is generated at both ends of the switching element Q1 and the parallel resonance capacitor Cr when the switching Q1 is turned off.
For this reason, it is required to select a high withstand voltage product of 1800 V for the switching element Q1 and the parallel resonant capacitor Cr. Therefore, the switching element Q1 and the parallel resonant capacitor Cr are correspondingly enlarged. In particular, when a high breakdown voltage product is selected for the switching element Q1, the saturation voltage VCE (SAT), the accumulation time tSTG, the fall time tf are large, and the current amplification factor hFE is small. It becomes difficult. If the switching frequency is low, the switching loss and the drive power increase, so that the power loss as a power supply circuit increases accordingly.
[0018]
In addition, since the DC input voltage of 2Ei level is obtained by the voltage doubler rectification method, the number of turns of the primary winding N1 correspondingly increases corresponding to this DC input voltage level. In the circuit configurations shown in FIG. 12 and FIG. 13, the constant voltage control is performed by the inductance control method by providing the orthogonal control transformer PRT as described above, but the number of turns of the primary winding N1 is large. In order to satisfy the control range as the constant voltage control, it is necessary to increase the number of turns of the controlled winding NR to expand the inductance variable range of the controlled winding NR. For this reason, the orthogonal control transformer PRT is also enlarged and expensive.
[0019]
12 and 13, the DC output voltages (EO1, EO2) are obtained by the half-wave rectifier circuit. When the rectifier diodes (DO1, DO2) of the half-wave rectifier circuit are not conductive, As shown in FIG. 16B, a resonance voltage V2 that is about 2 to 3.5 times the DC output voltage EO (EO1, EO2) is applied. For this reason, as the rectifier diodes (DO1, DO2) used in the half-wave rectifier circuit, for example, a high withstand voltage product of about 500V is required assuming that the DC output voltage is about 135V, and the forward voltage drop VF and the reverse recovery time trr accordingly. Becomes a factor that increases power loss. For example, if the switching frequency is increased, it is possible to reduce the size of various component elements. However, due to the circumstances described above, it is difficult to set the switching frequency fs high. For example, when fs = 50 KHz or more, power is rapidly increased. It has been found that conversion efficiency is reduced.
Furthermore, as described above, a voltage doubler rectifier circuit is required to obtain a high-level DC input voltage and ensure reliability, so that two relatively large smoothing capacitors are required, and the board area is also increased. growing.
[0020]
[Means for Solving the Problems]
Therefore, in consideration of the above-described problems, the present invention provides, for example, a voltage resonance type switching power supply circuit corresponding to a condition where the maximum load power is 150 W in an AC100V system, and improves the power conversion efficiency, reduces the board area in size and weight, and reduces the power consumption. The purpose is to reduce costs.
[0021]
Therefore, a rectifying / smoothing means that inputs a commercial AC power supply, generates a rectified and smoothed voltage having a level corresponding to the commercial AC power supply level, and outputs it as a DC input voltage, and a required coupling that is loosely coupled. A gap is formed so that a coefficient is obtained, and an insulation converter transformer provided for transmitting the primary side output to the secondary side and a switching element are provided, and the primary winding of the insulation converter transformer is provided by intermittently connecting the DC input voltage. A primary side resonance circuit formed by switching means configured to output to a line, at least a leakage inductance component including a primary winding of an insulating converter transformer, and a capacitance of a resonance capacitor, and operating the switching means as a voltage resonance type And a secondary series resonant capacitor connected in series to the secondary winding of the isolated converter transformer A secondary side series resonant circuit that forms a series resonant circuit by the leakage inductance component of the secondary winding of the insulating converter transformer and the capacitance of the secondary side series resonant capacitor, and the secondary side with respect to the rectified current path A secondary side that is formed by inserting a series resonant capacitor and that receives the alternating voltage obtained in the secondary winding of the isolated converter transformer and performs a double voltage full-wave rectification operation, corresponding to almost twice the input voltage level DC output voltage generating means configured to obtain a DC output voltage, a controlled winding connected to the primary winding or secondary winding of the insulating converter transformer, the controlled winding and its winding An orthogonal type control transformer is wound with a control winding whose directions are orthogonal to each other, and a variable control current is passed through the control winding in accordance with the level of the secondary side DC output voltage to control the controlled winding. line It was possible to configure the switching power supply circuit and a constant voltage control means arranged to perform constant voltage control by changing the inductance.
[0022]
According to the above configuration, the isolation converter transformer is loosely coupled, and on the secondary side, the secondary side DC output voltage is generated by the secondary side series resonance circuit and the voltage doubler full wave rectifier circuit to supply power to the load. To be. In other words, the required load conditions are basically handled by providing a voltage doubler full-wave rectifier circuit on the secondary side. Along with this, the primary side is not a voltage doubler rectifier circuit but an AC input. A full-wave rectifier circuit that generates a rectified and smoothed voltage corresponding to one time of the voltage level is provided.
[0023]
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 shows an example of the configuration of a switching power supply circuit as a first embodiment of the present invention. In the power supply circuit shown in this figure, a self-excited voltage resonance switching converter with a single switching element (bipolar transistor) is provided on the primary side as in the case of FIGS. 12 and 13 described above. It is done. In this figure, the same parts as those in FIG. 12 and FIG.
[0024]
In the power supply circuit as the present embodiment shown in this figure, a full-wave rectifier circuit including a bridge rectifier circuit Di and a smoothing capacitor Ci is used as a rectifier / smoothing circuit for obtaining an AC input voltage by inputting the AC input voltage VAC. And a rectified and smoothed voltage Ei corresponding to a level that is one time the AC input voltage VAC. That is, in the present embodiment, a voltage doubler rectifier circuit is not provided as in the prior art.
In this specification, a full-wave rectifier circuit that generates a rectified and smoothed voltage Ei corresponding to one level of the AC input voltage VAC is also referred to as an “equal voltage rectifier circuit”.
[0025]
Further, in the self-excited oscillation drive circuit of the voltage resonance converter shown in this figure, the insertion point of the base current limiting resistor RB is inserted between the base of the switching element Q1 and the inductor LB as shown in FIG. Although different from the circuit shown in FIG. 13, a self-oscillation driving circuit for the switching element Q1 is formed in the same manner as described above with reference to FIGS.
[0026]
As shown in FIG. 2, the insulating converter transformer PIT provided in the power supply circuit according to the present embodiment includes an EE type core in which, for example, ferrite type E cores CR1 and CR2 are combined so that their magnetic legs face each other. The primary winding N1 and the secondary winding N2d are respectively divided using the divided bobbin B in which the winding portion is divided on the primary side and the secondary side with respect to the central magnetic leg of the EE type core. Wrapped in the state. In this embodiment, a gap G is formed with respect to the central magnetic leg as shown in the figure. The gap G can be formed by forming the center magnetic legs of the E-type cores CR1 and CR2 shorter than the two outer magnetic legs.
As a result, for example, as a conventional example, a loose coupling with a smaller coupling coefficient than that of the insulating converter transformer PIT shown in FIG. 13 is achieved, so that a saturation state is hardly obtained. In this case, the coupling coefficient k is, for example, k≈0.85.
[0027]
On the secondary side of the power supply circuit according to the present embodiment, a secondary winding N2d having a different number of turns from the conventional one is provided as will be described later. One end of the secondary winding N2d is connected to the secondary side ground, and the other end is connected to a connection point between the anode of the rectifier diode D01 and the cathode of the rectifier diode D02 through a series connection of the series resonant capacitor Cs1. . The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. The negative side of the smoothing capacitor CO1 is connected to the secondary side ground.
The secondary winding N2d is provided with a tap, and this tap terminal is connected to the connection point between the anode of the rectifier diode D03 and the cathode of the rectifier diode D04 through a series connection of the series resonant capacitor Cs2. The cathode of the rectifier diode D03 is connected to the positive electrode of the smoothing capacitor C02, and the anode of the rectifier diode D04 is connected to the secondary side ground. The negative side of the smoothing capacitor CO2 is connected to the secondary side ground.
As a result, in such a connection form, a combination of [series resonant capacitor Cs1, rectifier diodes DO1, DO2, smoothing capacitor CO1] and [series resonant capacitor Cs2, rectifier diodes DO3, DO4, smoothing capacitor CO2] are combined. Thus, two sets of voltage doubler full wave rectifier circuits are provided. Here, the series resonance capacitor Cs1 forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1 and DO2 by its own capacitance and the leakage inductance component of the secondary winding N2d. The series resonance capacitor Cs2 The series resonance circuit corresponding to the on / off operation of the rectifier diodes D03 and D04 is formed by its own capacitance and the leakage inductance component of the secondary winding N2d.
That is, the power supply circuit of the present embodiment is provided with a parallel resonance circuit for making the switching operation a voltage resonance type on the primary side, and a series resonance for obtaining a voltage doubler full-wave rectification operation on the secondary side. A circuit is provided. In the present specification, the switching converter configured to operate with the resonance circuits provided on the primary side and the secondary side as described above is also referred to as a “composite resonance switching converter”.
[0028]
Here, the double voltage full wave rectification operation by the set of [the series resonance capacitor Cs1, the rectifier diodes DO1, DO2, and the smoothing capacitor CO1] is as follows.
FIG. 3 is a waveform diagram showing the operation of the main part of the power supply circuit having the configuration shown in FIG. When a switching output is obtained in the primary winding N1 by the switching operation on the primary side, this switching output is excited in the secondary winding N2d. This operation is performed as a switching output current I1 obtained in the primary winding N1 in FIG. 3G and a rectification current IC2 flowing in the rectification current path of the rectification diodes DO1 and DO2 from the secondary winding N2d in FIG. Indicated.
During the period T1 when the rectifier diode D01 is turned off and the rectifier diode D02 is turned on, the secondary winding operates in the depolarization mode in which the polarity of the primary winding N1 and the secondary winding N2d is -M. By the series resonance action of the leakage inductance of the line N2d and the series resonant capacitor Cs1, an operation of charging the series resonant capacitor Cs1 with the rectified current IC2 rectified by the rectifier diode DO2 is obtained. FIGS. 3E and 3F show the rectified current ID3 of the rectifier diode D02 and the voltage VD3 across the rectifier diode D02, respectively.
In the period T2 in which the rectifier diode D02 is turned off and the rectifier diode D01 is turned on to perform the rectification operation, the primary winding N1 and the secondary winding N2d are in the additive mode in which the polarity is + M. The smoothing capacitor CO1 is charged while the series resonance occurs in which the potential of the series resonance capacitor Cs1 is added to the voltage induced in the winding N2d. FIGS. 3C and 3D respectively show the rectified current ID2 of the rectifier diode D01 and the voltage VD2 across the rectifier diode D01.
As described above, the rectifying operation is performed using both the additional mode (+ M; forward operation) and the depolarization mode (−M; flyback operation). A DC output voltage EO1 corresponding to approximately twice the induced voltage of the secondary winding N2d is obtained. The double voltage full-wave rectifier circuit consisting of a set of [series resonant capacitor Cs2, rectifier diodes D03, D04, smoothing capacitor C02] can cope with almost twice the induced voltage of the secondary winding N2d by the same operation. DC output voltage EO2 is obtained.
[0029]
As described above with reference to FIG. 2, the structure for obtaining the above-described voltage doubler full-wave rectification operation is formed by forming a gap G with respect to the insulating converter transformer PIT to achieve loose coupling with a required coupling coefficient. Further, this is realized by obtaining a state in which it is difficult to become saturated. For example, when the gap G is not provided for the insulating converter transformer PIT as in the prior art, it is highly possible that the insulating converter transformer PIT becomes saturated during the flyback operation and the operation becomes abnormal. It is difficult to hope that the voltage doubler rectification operation as shown in FIG.
[0030]
Further, the constant voltage control according to the present embodiment has the same configuration as that of FIG. 12, that is, the controlled winding NR is connected in series with the primary winding N1, so that the primary voltage control is performed. This is performed by controlling the resonance pulse of the parallel resonance voltage Vcr on the side (the OFF period of the switching element Q1).
[0031]
According to the above configuration, the secondary side DC output voltage is obtained by performing voltage doubler full-wave rectification using the state in which the mutual inductance is in the + M and -M operation modes. The supplied power is also increased, and the maximum load power is increased.
[0032]
Further, by increasing the maximum load power as described above, in the present embodiment, the rectifying / smoothing circuit that generates the DC input voltage does not need to cover the load power by adopting the double voltage rectification method. Therefore, as described with reference to FIG. 1, it is possible to adopt a normal equal voltage rectifier circuit configuration using, for example, a bridge rectifier circuit.
As a result, for example, the rectified and smoothed voltage Ei when the AC input voltage VAC = 144 V is about 200 V. The resonant voltage Vcr obtained at both ends of the parallel connection circuit of the switching element Q1 / secondary parallel resonant capacitor Cr shown in FIG. 3A is obtained by the primary parallel resonant circuit acting on the rectified smoothing voltage Ei. The switching element Q1 is generated when the switching element Q1 is off. In the present embodiment, the rectified smoothing voltage Ei is approximately ½ that of the double voltage rectification as described above. The resonance voltage Vcr (1800 V) generated in the power supply circuit as the conventional example shown in FIG. That is, the resonance voltage Vcr is suppressed to about 900 V at the peak.
Therefore, in the present embodiment, a 900V withstand voltage product may be selected for the switching element Q1 and the parallel resonant capacitor Cr.
[0033]
Further, on the secondary side, a double voltage full-wave rectifier circuit that performs rectification operation in both periods in which the excitation voltage of the secondary winding N2d is positive and negative is provided. Like the waveform of the both-end voltages VD2 and VD3 of the rectifier diodes D01 and D02 shown in f), the both-end voltages of the secondary-side rectifier diode in the off period are suppressed to a level equivalent to the secondary-side DC output voltage EO. It will be. As a result, as the rectifier diode forming the secondary side voltage doubler rectifier circuit, a withstand voltage product corresponding to the level of the secondary side DC output voltage EO may be selected.
[0034]
Further, by obtaining the secondary side DC output voltage by the voltage doubler rectifier circuit, for example, obtain a level equivalent to the secondary side DC output voltage obtained by the equal voltage rectifier circuit (half-wave rectifier circuit), for example. If so, the number of turns of the conventional winding N2d in this embodiment is ½. This reduction in the number of turns leads to a reduction in size and weight and cost of the insulating converter transformer PIT.
[0035]
Furthermore, in the present embodiment, the switching element Q1, the parallel resonant capacitor Cr, and the secondary side rectifier diode can be used with a lower withstand voltage than what should be conventionally provided. For this reason, the characteristics of the switching element Q1 and the bridge rectifier circuit D0 are improved without considering the cost increase (for the switching element Q1, the saturation voltage VCE (SAT), the accumulation time tSTG, the fall time tf In the case of a rectifier diode, one having good characteristics such as forward voltage drop VF and reverse recovery time trr can be selected.
By improving the characteristics as described above, in this embodiment, the switching frequency can be set higher than the conventional one, and accordingly, the reduction of power loss and the reduction in size and weight of various components are promoted. In other words, it is possible to improve various characteristics such as power conversion efficiency as compared with the prior art, and to promote reduction in size and weight and cost.
[0036]
Furthermore, from the viewpoint of reducing the size and weight of the power supply circuit, the conventional configuration including the voltage doubler rectifier circuit for generating the DC input voltage requires two sets of rectifier diodes and smoothing capacitors, respectively. However, in the present embodiment, for example, a full-wave rectifier circuit using a normal bridge rectifier circuit is used, and therefore, a set of block type smoothing capacitors and bridge rectifier diodes can be employed. Reduction and miniaturization of parts are achieved.
[0037]
Furthermore, since the number of turns of the primary winding N1 is reduced as compared with the prior art because of the equal voltage rectifier circuit, as a result, the inductance variable range of the controlled winding of the orthogonal control transformer PRT can be reduced as compared with the conventional case. Even if it is narrow, constant voltage control is performed stably. That is, the number of turns of the controlled winding can be reduced as compared with the conventional one, and the orthogonal control transformer PRT can be reduced in size and weight and cost.
[0038]
As an actual experimental result, for example, in the configuration of the conventional power supply circuit shown in FIGS. 12 and 13, when the AC input voltage VAC = 100 V ± 20%, the switching frequency fs = 50 KHz and the maximum load power that can be handled is 180 W. On the other hand, in the power supply circuit of the present embodiment shown in FIG. 1, when the AC input voltage VAC = 100 V ± 20%, the switching frequency fs = 100 KHz and the maximum load power that can be handled is 160 W. While the frequency is doubled, it corresponds to a substantially equal maximum load power.
[0039]
FIG. 4 shows a configuration example of a switching power supply circuit according to the second embodiment of the present invention. In this figure, the same parts as those in FIGS. 1, 12 and 13 are denoted by the same reference numerals, and the description thereof is omitted.
[0040]
The power supply circuit as the composite resonance type switching converter shown in this figure has a configuration in which a controlled winding of an orthogonal control transformer PRT is provided on the secondary side.
In the case of the present embodiment, two controlled windings NR and NRA are wound in the orthogonal control transformer PRT. The controlled winding NR is inserted in series between one end of the secondary winding N2d and the secondary side series resonant capacitor Cs1, so that the capacitance of the secondary side series resonant capacitor Cs1 and the secondary winding N2d A series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1 and DO2 is formed by the combined inductance of the leakage inductance component and the inductance of the controlled winding NR.
The controlled winding NRA is inserted in series between the tap output of the secondary winding N2d and the secondary side series resonance capacitor Cs2, so that the capacitance of the secondary side series resonance capacitor Cs2 and the secondary winding N2d are increased. A series resonance circuit corresponding to the on / off operation of the rectifier diodes D03 and D04 is formed by the combined inductance of the leakage inductance component of the current and the inductance of the controlled winding NRA.
Also in this embodiment, the insulating converter transformer PRT adopts the configuration described with reference to FIG.
[0041]
Even in such a configuration, the secondary side DC output voltages EO1 and EO2 can be obtained as the secondary side DC output voltages EO1 and EO2 by obtaining the voltage doubler rectification operation including the secondary side series resonance operation as in the first embodiment. A level corresponding to twice the excitation voltage obtained at winding N2d is obtained.
[0042]
In this case, the controlled winding of the orthogonal control transformer PRT is provided on the secondary side, so that the constant voltage control is performed by the inductance control method described with reference to the power supply circuit shown in FIG. . That is, constant voltage control is performed by variably controlling the conduction angle of the secondary side rectifier diode in accordance with the variable inductance of the controlled winding.
FIG. 5 shows the relationship between the secondary side series resonance frequency and the secondary side DC output voltage EO (E01, E02) in the second embodiment. In this case, the horizontal axis represents frequency and the vertical axis represents the level of the secondary side DC output voltage EO.
In this figure, the secondary side series resonance frequencies fo1 and fo2 are shown together with the switching frequency fs. The secondary side series resonance frequency fo1 is a frequency obtained when the maximum AC input voltage (Vacmax) and the minimum load power (Pomin) are obtained. The secondary side series resonance frequency fo2 is the minimum AC input voltage (Vacmin) and the maximum load power. This is the frequency obtained at (Pomax). In this case, the secondary side series resonance frequency is set so as to be in a frequency region (lower side) lower than the switching frequency fs. In this embodiment, based on such characteristics, the inductance of the controlled winding is varied (that is, the secondary side series resonance frequency is varied) so that the DC output voltage EO is constant. The control circuit 1 operates.
Even with the configuration as the second embodiment, the same effects as those of the first embodiment described above can be obtained.
[0043]
FIG. 6 is a circuit diagram showing a configuration of a switching power supply circuit having a composite switching converter as a third embodiment of the present invention, and is the same as FIG. 1, FIG. 4, FIG. 12, and FIG. Are denoted by the same reference numerals and description thereof is omitted.
In the insulating converter transformer PIT according to the present embodiment, in addition to the primary winding N1, secondary winding N2d, and drive winding NB, a winding N3 is provided so as to wind up the primary winding N1. This insulating converter transformer PIT is also formed with a gap with respect to the central magnetic leg as shown in FIG. 2 so that loose coupling with a required coupling coefficient is obtained on the primary side and the secondary side. ing.
[0044]
In the present embodiment, the end of the winding N3 formed so as to wind up the primary winding N1 of the insulating converter transformer PIT is connected to the positive electrode of a smoothing capacitor CiB for boost voltage generation described later. The negative electrode of the smoothing capacitor CiB is connected to the positive electrode (Ei line) of the smoothing capacitor Ci.
In the present embodiment, a boost diode DB is provided. The boost diode DB has an anode connected to a connection point (Ei line) between the negative electrode of the smoothing capacitor CiB and the positive electrode of the smoothing capacitor Ci, and a cathode connected via a series connection of the controlled winding N R of the orthogonal control transformer PRT. Thus, the connection is made to the connection point between the primary winding N1 and the winding N3.
According to such a connection form, the switching output voltage obtained in the winding N3 is rectified by the boost diode DB and smoothed by the smoothing capacitor CiB, thereby generating the boost voltage VB at both ends of the smoothing capacitor CiB. A circuit will be formed. However, as described above, the controlled winding NR is inserted in series in this boost circuit.
[0045]
In the configuration including the boost circuit as described above, the boost smoothing voltage EB in which the boost voltage VB is superimposed on the rectified smoothing voltage Ei is obtained at both ends of the smoothing capacitor CiB and the smoothing capacitor Ci connected in series. However, this boost smoothing voltage EB is
[Expression 1]

Can be represented by
Then, the relationship of L3 = L1 is obtained as the inductance of the winding N3 and the primary winding N1, and the rectified and smoothed voltage Ei, the drop voltage VF of the boost diode DB, and the saturation voltage V (SAT) of the switching element Q1. If the relationship of Ei >> VF, V (SAT) is established, the boost smooth voltage EB is based on the above (Equation 1),
[Expression 2]

Will be shown. In this case, in the power supply circuit shown in FIG. 6, for example, the boost smoothing voltage is changed by changing the inductance LR of the controlled winding NR of the orthogonal transformer PRT in a range of 0.1 × L1 to 1.2 × L1. EB can be varied in a range of approximately Ei to 2Ei (Ei corresponds to the rectified and smoothed voltage level obtained at both ends of the smoothing capacitor Ci).
[0046]
In the connection form of the power supply circuit of the present embodiment, the controlled winding NR is inserted in series between the primary winding N1 and the positive electrode of the smoothing capacitor Ci via the boost rectifier diode DB. Therefore, the primary side parallel resonant circuit is formed including the inductance LR of the controlled winding NR. Therefore, in this case, the secondary side output is stabilized by controlling the pulse width of the resonance voltage Vcr as in the power supply circuit shown in FIG.
[0047]
In the present embodiment, the voltage resonance type switching converter including the switching element Q1 performs a switching operation using the boost smooth voltage EB as an operation power supply. That is, in the present embodiment, the apparent DC input voltage level to be supplied to the voltage resonance type switching converter is raised by the boost circuit. Since the DC input voltage level is raised by the operation of this boost circuit, the present embodiment supports the maximum load power almost equivalent to the configuration in which the DC input voltage is obtained by the voltage doubler rectifier circuit (that is, the conventional example). It becomes possible to do. As in the previous embodiments, the DC output voltage is obtained by the voltage doubler rectifier circuit using the secondary side series resonance operation. As a result, the maximum load power of about 200 W is obtained. It becomes possible to cope with.
In the present embodiment, the effects described in the first embodiment can be obtained similarly.
[0048]
FIG. 7 is a circuit diagram showing a configuration of a power supply circuit including a composite resonance type switching converter according to a fourth embodiment of the present invention, and FIG. 1, FIG. 4, FIG. 6, FIG. 12, FIG. The same parts are denoted by the same reference numerals and description thereof is omitted.
[0049]
In the power supply circuit shown in this figure, a center tap is provided as a primary winding and is divided into primary windings N1A (inductance L1A) and N1B (inductance L1B). This center tap terminal is connected to the line of the rectified and smoothed voltage Ei through a serial connection of the controlled winding NR (inductance LR) of the orthogonal control transformer PRT.
[0050]
The voltage resonance type converter shown in this figure has a separately-excited configuration having two switching elements Q1 and Q2. In this case, MOS-FETs are employed for the switching elements Q1, Q2.
In this case, the drain of the switching element Q1 is connected to the end of the primary winding N1A, and the source is grounded to the primary side ground. Further, between the drain and source of the switching element Q, a damper diode DD1 and a parallel resonance capacitor Cr1 that form a feedback current path at the time of switching off and a parallel resonance capacitor Cr1 are connected in parallel.
The drain of the switching element Q2 is connected to the end of the primary winding N1B, and the source is grounded to the primary side ground. Further, a damper diode DD2 and a parallel resonant capacitor Cr2 are connected in parallel between the drain and source of the switching element Q2.
[0051]
The parallel resonance capacitor Cr1 forms a parallel resonance circuit in which the switching operation of the switching element Q1 is a voltage resonance type by its own capacitance and the combined inductance of the inductance L1A and the inductance LR.
Similarly, the parallel resonance capacitor Cr2 forms a parallel resonance circuit in which the switching operation of the switching element Q2 is a voltage resonance type by its own capacitance and the combined inductance of the inductance L1B and the inductance LR.
[0052]
In this connection form, two switching circuit systems are provided in which the switching elements Q1 and Q2 each perform a voltage resonance type switching operation. The switching output point of the switching elements Q1, Q2 is the center tap of the primary winding N1.
[0053]
The oscillation drive circuit 2 is activated by the activation resistor RS, and for example, oscillates and outputs a frequency signal corresponding to the switching frequency fs = 100 KHz, and generates a switching drive signal (drive voltage) based on this frequency signal. It is generated and applied to the gates of the switching elements Q1, Q2. At this time, the switching drive signals applied to the gates of the switching elements Q1 and Q2 are signals having opposite polarities that are inverted from each other.
[0054]
When the switching elements Q 1 and Q 2 are driven by the oscillation drive circuit 2, the switching elements Q 1 and Q 2 perform a switching operation at an on / off timing alternately, for example, at a switching frequency fs = 100 KHz.
That is, the switching operation by both of the switching elements Q1 and Q2 is a so-called push-pull operation, and for example, larger power can be supplied to the load side than in the so-called single-end operation by a single switching element. Thereby, in this Embodiment, it corresponds to the conditions of the heavy load more than the maximum load electric power 200W.
[0055]
In this case, since the controlled winding NR of the orthogonal control transformer PRT is connected to the primary winding, as in the power supply circuit shown in FIG. Constant voltage control is performed by PWM control of the resonance voltage obtained at both ends of the parallel connection circuit of the parallel resonance capacitor.
[0056]
Even with the configuration shown in FIG. 7, the effect obtained by the combined resonance converter and the effect obtained by obtaining the DC input voltage using the equal voltage rectifier circuit can be obtained in the same manner as in the previous embodiments. It is.
[0057]
In the description of the embodiments so far, the case where a single bipolar transistor (BJT) or a MOS-FET is adopted as the switching element Q1 (and Q2) is taken as an example. As a form, it is also possible to employ a switching circuit or a switching element as described below instead of the switching element Q1 (Q2).
FIG. 8 shows an example in which a circuit in which two bipolar transistors Q11 and Q12 are connected in a Darlington connection is used in place of the switching element Q1 (Q2).
As a connection form in this case, the collector of the transistor Q11 and the collector of the transistor Q12 are connected, the emitter of the transistor Q11 is connected to the emitter of the transistor Q12, and the emitter of the transistor Q12 is grounded. The anode of the damper diode DD1 is connected to the emitter of the transistor Q11, and the cathode of the damper diode DD1 is connected to the base of the transistor Q11 via the resistor R11. The damper diode DD2 has an anode connected to the emitter of the transistor Q12 and a cathode connected to the collector of the transistor Q12. The resistor R12 is connected in parallel to the base and emitter of the transistor Q12. In the Darlington connection circuit thus formed, the base of the transistor Q11 is equivalent to the base of the switching element Q1 (Q2) shown in each of the previous embodiments, and the collector contacts of the transistors Q11 and Q12 are connected to the switching element Q1 ( Equivalent to the collector of Q2). The emitter of the transistor Q12 is equivalent to the emitter of the switching element Q1 (Q2).
[0058]
Further, when the single-end operation is performed by one switching element Q1 as in the power supply circuits of FIGS. 1, 4, and 6 corresponding to the first to third embodiments, FIG. As shown in FIG. 5, a MOS-FET (MOS type field effect transistor; metal oxide semiconductor) can be used in place of the switching element Q1 of the bipolar transistor. When the MOS-FET is used, a Zener diode ZD for forming a feedback current path at the time of switching off is connected in parallel in the direction shown in the drawing between the drain and the source. That is, the anode is connected to the source of the MOS-FET, and the cathode is connected to the drain of the Zener diode ZD. In this case, the base, collector, and emitter of the switching element Q1 described in each of the previous embodiments are replaced with the gate, drain, and source of the MOS-FET, respectively. Further, when the MOS-FET is used, it is necessary to perform voltage driving instead of current driving by adopting a separately excited configuration as shown in FIG.
[0059]
FIG. 10 shows an example in which an IGBT (insulated gate bipolar transistor) is used instead of the switching element Q1 (Q2). A diode D for forming a feedback current path when switching off is connected in parallel between the collector and the emitter of the IGBT. Here, the anode and cathode of the diode D are connected to the collector and emitter of the IGBT, respectively.
In this circuit, the base, collector, and emitter of the switching element Q1 (Q2) shown in each of the previous embodiments are replaced with the gate, collector, and emitter of the IGBT, respectively.
[0060]
FIG. 12 shows an example in which an SIT (electrostatic induction thyristor) is used in place of the switching element Q1 (Q2). Also between the collector and emitter of this SIT, a diode D for forming a feedback current path when switching off is connected in parallel, and the anode and cathode of the diode D are connected to the cathode and anode of the SIT, respectively. Is done.
Even when any of the configurations shown in FIGS. 8 to 12 is adopted, the present embodiment can further increase the efficiency. When the configuration shown in FIGS. 8 to 12 is adopted, although not shown here, the configuration of the drive circuit is changed so as to be adapted to the configuration to be actually used instead of the switching element Q1 (Q2). It does not matter.
Further, the details of the configuration shown in each of the above drawings as an embodiment may be changed according to actual use conditions and the like.
[0061]
【The invention's effect】
As described above, according to the present invention, as a switching power supply circuit, a voltage resonance type switching converter is provided on the primary side, and the insulating converter transformer is loosely coupled, so that the mutual inductance of the primary winding and the secondary winding is reduced. The operation modes (+ M / −M) in which the polarities are opposite to each other are obtained. Further, on the secondary side, a secondary side series resonance circuit is connected in series to the secondary winding to form a series resonance circuit, and a voltage doubler full-wave rectifier circuit using this series resonance circuit is provided. The secondary side DC output voltage corresponding to twice the alternating voltage (excitation voltage) obtained in the secondary winding is obtained.
[0062]
As a result of supplying power to the load by the voltage doubler full-wave rectifier circuit as described above, the present invention can cope with, for example, the case of obtaining a secondary side DC output voltage of the same magnification by a half wave rectifier circuit as in the prior art. It is possible to improve the maximum possible load power.
As a result, the primary side is not a voltage doubler rectifier circuit, but a normal full-wave rectifier circuit is configured to input a rectified and smoothed voltage having a level corresponding to the AC input voltage level. It will be possible to respond.
[0063]
The following can be said from the above configuration.
For example, conventionally, when the above conditions are met, it is necessary to obtain a rectified and smoothed voltage corresponding to twice the AC input voltage level by the voltage doubler rectifier circuit. For this reason, the switching element and the primary side parallel resonance are required. As the capacitor, it was necessary to select a withstand voltage product corresponding to the switching voltage generated according to the rectified smoothing voltage level.
Further, in the prior art, the DC output voltage is generated by the half-wave rectifier circuit on the secondary side, so that the rectified smoothing voltage is about 2.5 to 3.5 times the rectified smoothing voltage in the non-conduction period of the rectifier diode. Since a voltage is applied, a pressure-resistant product corresponding to this voltage level has been selected.
[0064]
On the other hand, in the present invention, the switching voltage depending on the rectified and smoothed voltage level is ½ that of the conventional one. Therefore, the conventional withstand voltage product of about ½ is used for the switching element and the primary side resonance capacitor. Can do.
On the secondary side, as described above, a voltage doubler full wave rectifier circuit is provided. Here, the voltage doubler full wave rectifier circuit is a full wave that performs rectification operation in both periods where the alternating voltage is positive / negative. As a result of the rectifying operation, the voltage applied to the rectifying diode is suppressed to be approximately equal to the rectified smoothing voltage level, so that a secondary rectifying diode having a lower withstand voltage can be selected.
As a result, it is possible to reduce costs for the switching element, the primary side parallel resonant capacitor, the secondary side rectifier diode, and the like. In addition, it is possible to easily select a switching element and a secondary side rectifier diode with improved characteristics and set a high switching frequency, thereby improving power conversion efficiency. It is also possible to reduce the size and weight of the circuit components around the switching element.
In addition, as described above, the circuit for obtaining the rectified and smoothed voltage from the commercial AC power supply is a normal equal voltage rectifier circuit, so that, for example, a normal set of block-type smoothing capacitors and bridge rectifier diodes are employed. Therefore, also in this respect, cost reduction and circuit scale reduction can be achieved. Furthermore, the number of turns of the controlled winding is reduced, and the orthogonal control transformer used for constant voltage control can be reduced in size and weight and cost can be reduced.
[0065]
Further, according to the present invention, a double voltage full-wave rectifier circuit is adopted for the rectifier circuit provided on the secondary side, so that, for example, it is possible to obtain a DC output voltage of the same level as when an equal voltage rectifier circuit is provided. For example, the number of turns of the secondary winding can be reduced to about ½ of the conventional number.
Further, in the configuration of the present invention, the stabilization operation of the secondary output can be ensured by the inductance variable range of the controlled winding which is narrower than before, so that the number of turns of the controlled winding can be reduced and the orthogonal control transformer It is possible to achieve a reduction in size and weight and cost.
[0066]
Furthermore, if a configuration that performs a push-pull operation as the switching means is adopted, it is possible to further increase the load power that can be handled with a relatively simple circuit configuration.
[0067]
Further, the switching element can be configured by a Darlington circuit formed with a bipolar transistor, or a MOS field effect transistor, an insulated gate bipolar transistor, or an electrostatic induction thyristor. The power conversion efficiency can be further improved as compared with the case where the switching means is formed by a single bipolar transistor.
[0068]
As described above, according to the present invention, the power circuit including the voltage resonance converter can be reduced in cost, reduced in size and weight, and improved in various characteristics such as power conversion efficiency.
[Brief description of the drawings]
FIG. 1 is a circuit diagram showing a configuration example of a power supply circuit according to a first embodiment of the present invention.
FIG. 2 is a cross-sectional view showing a configuration of an insulating converter transformer of the power supply circuit according to the present embodiment.
FIG. 3 is a waveform diagram showing an operation of a main part of the power supply circuit according to the first embodiment.
FIG. 4 is a circuit diagram showing a configuration example of a power supply circuit according to a second embodiment of the present invention.
5 is an explanatory diagram showing a relationship between a switching frequency with respect to a secondary side series resonance frequency and a secondary side DC output voltage of the circuit shown in FIG. 4; FIG.
FIG. 6 is a circuit diagram showing a configuration example of a power supply circuit according to a third embodiment of the present invention.
FIG. 7 is a circuit diagram showing a configuration example of a power supply circuit according to a fourth embodiment of the present invention.
FIG. 8 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.
FIG. 9 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.
FIG. 10 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.
FIG. 11 is a circuit diagram showing a configuration as a modification of the embodiment of the present invention.
FIG. 12 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.
FIG. 13 is a circuit diagram showing a configuration of a power supply circuit as a conventional example.
FIG. 14 is a cross-sectional view showing a configuration of an insulating converter transformer as a conventional example.
FIG. 15 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.
16 is a waveform diagram showing an operation of a main part in the power supply circuit shown in FIGS. 12 and 13. FIG.
[Explanation of symbols]
1. Control circuit, 2. Oscillation / drive circuit, Ci, CiB smoothing capacitor, Cr parallel resonance capacitor, Cs1, Cs2 secondary side series resonance capacitor, Di bridge rectification circuit, D01, D02, D03, D04 Rectifier diode, PIT Insulation converter transformer , PRT Orthogonal control transformer, NC control winding, NR controlled winding, Q1, Q2 switching element, DB boost diode, N3 (boost) winding

Claims (9)

Rectifying and smoothing means for inputting a commercial AC power supply, generating a rectified and smoothed voltage having a level corresponding to the same magnification as the level of the commercial AC power supply, and outputting it as a DC input voltage;
An insulating converter transformer provided to form a gap so as to obtain a required coupling coefficient to be loosely coupled, and to transmit the primary side output to the secondary side;
Switching means comprising a switching element, wherein the DC input voltage is intermittently output to the primary winding of the insulating converter transformer;
A primary side resonance circuit which is formed by at least a leakage inductance component including a primary winding of the insulating converter transformer and a capacitance of a resonance capacitor, and the operation of the switching means is a voltage resonance type;
By connecting a secondary side series resonant capacitor in series with the secondary winding of the insulation converter transformer, the leakage inductance component of the secondary winding of the insulation converter transformer and the capacitance of the secondary series resonance capacitor A secondary side series resonant circuit that forms a series resonant circuit with
The secondary side series resonant capacitor is inserted into the rectified current path, and the alternating voltage obtained at the secondary winding of the isolation converter transformer is input to perform a double voltage full wave rectification operation. DC output voltage generating means having a secondary side rectifier diode configured to obtain a secondary side DC output voltage corresponding to approximately twice the level;
A controlled winding connected between the secondary winding of the insulating converter transformer and the secondary side series resonant capacitor, and a controlled winding in which the controlled winding and the winding direction thereof are orthogonal to each other. An orthogonal control transformer around which a wire is wound;
The conduction angle of the secondary rectifier diode is variably controlled by flowing a control current having a magnitude corresponding to the level of the secondary DC output voltage to the control winding to change the inductance of the controlled winding. A constant voltage control circuit ,
Switching power supply circuit comprising a. It said switching means, the switching power supply circuit according to claim 1, further comprising a self-oscillation driving circuit for performing a switching operation by self-excited. Said switching means, the switching power supply circuit according to claim 1, further comprising a driving circuit for performing a switching operation by a separately excited. The switching power supply circuit according to claim 1, wherein the switching unit includes a bipolar transistor as a switching element. 2. The switching power supply circuit according to claim 1, wherein the switching means is configured to use a Darlington circuit formed with a bipolar transistor as one switching element. It said switching means, the switching power supply circuit according to claim 1, further comprising a MOS field effect transistor as a switching element. The switching power supply circuit according to claim 1, wherein the switching means includes an insulated gate bipolar transistor as a switching element. The switching power supply circuit according to claim 1, wherein the switching unit includes an electrostatic induction thyristor as a switching element. It said switching means being two sets, switching power supply circuit according to claim 1 so that the switching operation by the push-pull is achieved.

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