JP2001119943A - Switching power circuit - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reduce power loss in a switching element, without increasing the scale of the circuit. SOLUTION: Clamp circuits 2A to 2D are composed of Schottky barrier diodes DSBD1 to DSBD4 and resistors RD1 to RD4. As a result, the waveform of drive current IB supplied to switching elements Q1 to Q4 becomes ideal, for examples, a current waveform with increased reverse current at turn-off of the switching elements Q1 to Q4. Therefore, the accumulation time t stg and a fall time tf of the switching elements Q1 to Q4 are shortened, thereby reducing power loss by the switching elements Q1 to Q4.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a switching power supply circuit provided as a power supply for various electronic devices.

[0002]

2. Description of the Related Art As a switching power supply circuit, a circuit employing a switching converter of a type such as a flyback converter or a forward converter is widely known. Since these switching converters have a rectangular switching operation waveform, there is a limit in suppressing switching noise. Also, due to its operating characteristics,
It has been found that there is a limit in improving the power conversion efficiency. Therefore, the present applicant has previously proposed various switching power supply circuits using various resonance type converters.
The resonance type converter can easily obtain high power conversion efficiency and realize low noise because the switching operation waveform is sinusoidal. It also has the advantage that it can be configured with a relatively small number of parts.

FIG. 7 is a circuit diagram showing one configuration example of a switching power supply circuit that can be configured based on the invention proposed by the present applicant. This power supply circuit employs a self-excited current resonance type converter.

In the switching power supply circuit shown in FIG. 1, a commercial AC power supply AC (AC input voltage VAC) is rectified and smoothed by a rectifying and smoothing circuit including a bridge rectifier circuit Di and a smoothing capacitor Ci.
A DC input voltage corresponding to one time the peak value of AC is generated. In the circuit shown in this figure, a rush current limiting resistor Ri is inserted in the line of the commercial AC power supply AC, for example, to suppress a rush current flowing into the smoothing capacitor Ci when the power is turned on. .

The switching converter of this power supply circuit has four switching elements Q1, Q2, Q
A self-excited current resonance type converter in which 3 and Q4 are full-bridge-coupled. The switching elements Q1 and Q2 are connected in series via the respective collectors and emitters between the positive electrode of the smoothing capacitor Ci and the ground. Switching elements Q3 and Q4 are also connected in the same manner as described above. In this case, a bipolar transistor (BJT: junction type transistor) is employed as the switching elements Q1 to Q4.

As a drive circuit system for the switching element Q1, a starting resistor RS1 is inserted between the collector and the base, and a clamp diode DD1 having a reverse recovery time trr lengthened to a predetermined length between the base and the emitter as described later. Has been inserted. In this case, the cathode of the clamp diode DD1 is connected to the base of the switching element Q1, and the anode is connected to the emitter of the switching element Q1. Also, with respect to the base of the switching element Q1,
A drive circuit for self-excited oscillation (resonant capacitor CB1-drive winding NB1) is connected via a base current limiting resistor RB1.
In this case, the resonance capacitor CB1 forms a series resonance circuit with its own capacitance and the inductance LB1 of the drive winding NB1. Similarly, switching element Q2
, Q3, and Q4, respectively, by the same connection configuration as [starting resistors (RS2, RS3, RS4), clamp diodes (DD2, DD3, DD4), and base current limiting resistors (R
B2, RB3, RB4), resonance capacitors (CB2, CB3, CB)
4), drive windings (NB2, NB3, NB4)] to form a drive circuit system.

[0007] Drive transformer PRT (Power Regulati
ng Transformer) is provided to drive the switching elements Q1 to Q4 and to perform constant voltage control as described later. Drive transformer PRT shown in this figure
Correspond to the drive windings NB1 to N
B4 and the resonance current detection winding ND formed by winding up the drive winding NB1 are wound, and the control winding NC is wound so that these windings are orthogonal to the winding direction. It is configured as a saturated reactor.

One end of the drive winding NB1 is connected to the base of the switching element Q1 via a series connection of the resonance capacitor CB1 and the base current limiting resistor RB1, and the other end is connected to the collector of the switching element Q2. One end of the drive winding NB2 is connected to the base of the switching element Q2 via a series connection of a resonance capacitor CB2 and a base current limiting resistor RB2, and the other end is grounded. In this case, the drive winding NB1 and the drive winding NB2 are wound so that voltages of opposite polarities are generated. Similarly,
One end of the drive winding NB3 is connected to a switching element Q via a series connection of a resonance capacitor CB3 and a base current limiting resistor RB3.
3 and the other end is connected to the collector of the switching element Q4. One end of the drive winding NB4 is connected to the base of the switching element Q4 via a series connection of a resonance capacitor CB4 and a base current limiting resistor RB4, and the other end is grounded. Also in this case, the driving winding NB3 and the driving winding NB4 are wound so that voltages of opposite polarities are generated.

One end of the resonance current detecting winding ND is connected to a connection point (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2, and the other end is an insulation converter transformer PIT described later.
Is connected to one end of the primary winding N1. The number of turns (number of turns) of the resonance current detection winding ND is, for example, about 1T (turn).

[0010] Insulated converter transformer PIT (Power Is
olation Transformer) is composed of switching elements Q1 to Q4
Is transmitted to the secondary side. In this case, one end of the primary winding N1 of the insulating converter transformer PIT is
It is connected to the contact point (switching output point) between the emitter of the switching element Q1 and the collector of the switching element Q2 via the resonance current detection winding ND. The primary winding N1
Is connected to the contact point (switching output point) between the emitter of the switching element Q3 and the collector of the switching element Q4 via a series resonance capacitor C1 composed of, for example, a film capacitor. In this case, the series resonance capacitor C1 and the primary winding N1 are connected in series, but the capacitance of the series resonance capacitor C1 and the leakage inductance of the insulating converter transformer PIT including the primary winding N1 (series resonance winding) ( The leakage inductance L1) component forms a series resonance circuit for making the operation of the switching converter a current resonance type. As a result, the switching outputs of the switching elements Q1 to Q4 are supplied to a series resonance circuit including the primary winding N1 of the isolation converter transformer PIT and the series resonance capacitor C1.

On the secondary side of the insulating converter transformer PIT in FIG. 1, a center tap is provided for the secondary winding N2, and the rectifier diodes DO1, DO2, D
By connecting O3, DO4 and the smoothing capacitors CO1, CO2 as shown in the figure, a set of [rectifier diodes DO1, DO2, smoothing capacitor CO1] and a set of [rectifier diodes DO3, DO4, smoothing capacitor CO2] are obtained. Two sets of full wave rectifier circuits are provided. The full-wave rectifier circuit composed of [rectifier diodes DO1, DO2, smoothing capacitor CO1] has a DC output voltage EO1.
And a full-wave rectifier circuit composed of [rectifier diodes DO3, DO4, smoothing capacitor CO2] generates a DC output voltage EO2. That is, in the circuit shown in this figure, a full-wave rectifier circuit is provided to obtain a DC output voltage on the secondary side. In this case, the DC output voltage EO1 and the DC output voltage EO2 are also branched and input to the control circuit 1. In the control circuit 1, the DC output voltage EO1 is used as a detection voltage, and the DC output voltage EO2 is used as an operation power supply of the control circuit 1.

The control circuit 1 supplies a DC current whose level varies in accordance with, for example, the level of the DC voltage output E01 on the secondary side to the control winding NC of the drive transformer PRT as a control current, as will be described later. The constant voltage control is performed as described above.

As a switching operation of the power supply circuit having the above-described configuration, for example, a set of switching elements [Q1, Q4] and a set of switching elements [Q2, Q3] alternately perform on / off operations. For example, when a commercial AC power supply is turned on, a starting current (base current) is supplied to the bases of the switching elements Q1 to Q4 via the starting resistors RS1 to RS4, but temporarily the switching elements [Q1, Q4]. Is turned on first, the set of switching elements [Q2, Q3] is controlled to be turned off. The output of the switching element [Q1, Q4] is the collector-emitter of the switching element Q1 → the resonance current resonance current detection winding ND → the primary winding N1 → the series resonance capacitor C1 → the collector-emitter of the switching element Q4 → the primary side ground. At this time, the set of switching elements [Q2, Q3] turns on and the set of switching elements [Q1, Q4] turns off near the point where the resonance current flowing through the primary side series resonance circuit becomes zero. Is controlled so that And the switching elements [Q2, Q3]
, A resonance current flows in the opposite direction to the previous one. Thereafter, a self-excited switching operation in which a set of switching elements [Q1, Q4] and [Q2, Q3] is turned on alternately is started. As described above, the switching elements [Q1, Q4] and [Q2] are used with the terminal voltage of the smoothing capacitor Ci as the operating power supply.
, Q3] alternately open and close,
A drive current close to the resonance current waveform is supplied to the primary winding N1 of the insulation converter transformer PIT, and the secondary winding N1
2 to get the alternating output.

The constant voltage control by the drive transformer PRT is performed as follows. For example, if the secondary output voltage EO1 fluctuates due to fluctuations in the AC input voltage and load power, the control circuit 1 varies the level of the control current flowing through the control winding NC in accordance with the fluctuation in the secondary output voltage EO1. Control. The drive current PR
Under the influence of the magnetic flux generated in T, the state of the saturation tendency in the drive transformer PRT changes, and the drive windings NB1 to NB4
Of the self-excited oscillation circuit, thereby controlling the switching frequency to change. In the power supply circuit shown in this figure, the switching frequency is set in a frequency range higher than the resonance frequency of the series resonance circuit of the series resonance capacitor C1 and the primary winding N1, but when the switching frequency increases, for example, The switching frequency is set apart from the resonance frequency. Thereby, the resonance impedance of the primary side series resonance circuit with respect to the switching output increases. By increasing the resonance impedance in this way, the drive current supplied to the primary winding N1 of the primary side series resonance circuit is suppressed, and as a result, the secondary side output voltage is suppressed, and the constant voltage Control will be achieved. Hereinafter, the constant voltage control method using such a method is referred to as a “switching frequency control method”.

FIG. 8 shows an example of a switching operation waveform of the primary-side current resonance type converter in the switching power supply circuit having the configuration shown in FIG. A resonance voltage VB shown in FIG. 8E is generated in the drive winding NB2. And
Due to the resonance voltage VB, a resonance current flows to the base of the switching element Q2 via a series connection of the resonance capacitor CB2 and the base current limiting resistor RB2. This resonance current is combined with a clamp current ID2 as shown in FIG. 8 (f) flowing from the clamp diode DD2, for example, so that the base of the switching element Q2 is
A drive current IB2 flows as shown in FIG. With such a drive current IB2, the switching element Q2 is turned on during the period T.
When it is ON, the collector of the switching element Q2 has a collector current IC1 according to the waveform shown in FIG.
Flows. When the period TOFF is reached, the driving current IB2
(FIG. 8 (c)) becomes the 0 level, and the switching element Q2 is also turned off (non-conductive). As a result, the period T
Collector of switching element Q2 at ON, TOFF
A waveform shown in FIG. 8A is obtained as the emitter-to-emitter voltage VC1.

A waveform shown in FIG. 8D is obtained as the base-emitter voltage VBE of the switching element Q2 during the above-mentioned periods TON and TOFF. In response to this, a clamp current ID2 flowing through the clamp diode DD2 is obtained. Has a waveform as shown in FIG.

The period T during which the switching element Q2 is turned on
In the ON state, the region of the positive current shown in FIG.
) Corresponds to the region of the forward bias current of the drive current IB2. In the same period TON, a period (T22) in which the drive current IB2 becomes the reverse bias current is the accumulation time tstg of the switching element Q2. In this period T22, the transistor Q2 operates even if the drive current IB2 becomes the reverse bias current. The on state will be maintained.

In the power supply circuit shown in FIG. 7, since the clamp diode DD2 is connected between the base and the emitter of the switching element Q2, the switching element Q2
Is turned off, the clamp diode DD2 becomes conductive, and a negative series resonance current flows through the clamp diode DD2 → the base current limiting resistor RB2 → the resonance capacitor CB2 → the drive winding NB2. And then the period TON
Starts immediately, the damper period starts, and the charging / discharging energy of the series resonance capacitor C1 is changed to the damper current (I
D2) flows through the clamp diode DD2 → the base of the switching element Q2 → the collector. When the damper period ends, the clamp diode DD2 enters the region of the reverse recovery time trr. In a period (T32) corresponding to the reverse recovery time trr, the clamp diode DD2
Are conducting. When the drive current IB2 increases during the reverse recovery time trr and a current amount sufficient to turn on the switching element Q2 is obtained, the switching element Q2 is turned on. However, during the period (T32) corresponding to the reverse recovery time trr, even if the drive current (base current) IB2 in FIG. 8C is inverted to the positive polarity, the base of the switching element Q2 is actually applied. Since no current is supplied, the switching element Q2 does not turn on immediately, and thereafter, at a certain timing, when the base current level of the switching element Q2 reaches a level sufficient to make the switching element Q2 conductive. Switching element Q
2 transitions to the ON state. That is, the period TON
In the period (T32) corresponding to the reverse recovery time trr of the transistor Q2, the switching element Q2 is turned off, and at the end of the period T32, the forward drive current IB2 starts flowing to the base of the switching element Q2. The switching element Q2 is turned on. Although not shown, the operation of the switching element Q3 is almost the same as the operation of the switching element Q2 described above.

On the other hand, the operation waveform of the switching element Q1 is shown as a waveform whose phase is shifted by 180 degrees from the operation waveform of the switching element Q2 described above, and the driving current IB1 of the switching element Q1 is shown in FIG. ). In this case, for example, during a period in which the switching element Q1 is turned on (a period corresponding to the period TOFF of the switching element Q2), a period (T21) in which the drive current IB1 becomes a reverse bias current is the accumulation time tstg of the switching element Q1, In this period T21, the transistor Q1
Means that the ON state is maintained even if the drive current IB1 becomes the reverse bias current. Also, the switching element Q1
Since the clamp diode DD1 is also connected between the base and the emitter, when the period in which the switching element Q1 is turned on starts, the clamp diode DD1 immediately becomes the damper period like the clamp diode DD2, and the damper period. After that, it becomes a region of the reverse recovery time trr. Then, a period (T) corresponding to the reverse recovery time trr
In 31), the clamp diode DD1 is conductive. Although not shown, the operation of the switching element Q4 is almost the same as the operation of the switching element Q1 described above.

In such a power supply circuit of the push-pull operation, if the set of the switching elements [Q1, Q4] and the set of the switching elements [Q2, Q3] are simultaneously turned on, an abnormal operation will occur. In the power supply circuit,
For example, after the set of switching elements [Q2, Q3] (or the set of switching elements [Q1, Q4]) is turned off, the set of switching elements [Q1, Q4], (or the set of switching elements [Q2, Q3] Q2) Is turned on, the switching elements [Q1, Q4]
A period is provided in which both the set of switching elements [Q2, Q3] are turned off. For example, at the timing when the set of switching elements [Q2, Q3] turns on → off and the set of switching elements [Q1, Q4] turns off → on, the clamp diode DD1 (DD4) connected to the switching element Q1 (Q4). Reverse recovery time trr
T31 corresponding to the switching element [Q1, Q4]
And the set of switching elements [Q2, Q3] are in the OFF state, and the switching elements [Q1, Q3]
4] turns on → off, switching elements [Q2, Q3
Is turned on from off to on, a period T corresponding to the reverse recovery time trr of the clamp diode DD2 (DD3) connected to the switching element Q2 (Q3).
The period 32 is a period during which both the set of switching elements [Q1, Q4] and the set of switching elements [Q2, Q3] are in the off state.

For this reason, in the power supply circuit shown in FIG. 7, the clamp diodes DD1 to DD4 are set so that the set of switching elements [Q1, Q4] and the set of switching elements [Q2, Q3] are not simultaneously turned on. Recovery time trr and the storage time tst of the switching elements Q1 to Q4.
g was sorted out. The selection range in this case corresponds to, for example, the accumulation time tstg of the switching element Q2 shown in FIG. 8C due to, for example, variations in the clamp diodes DD1 to DD4 and the switching elements Q1 to Q4 and fluctuations in the switching frequency. Period T22
Even if the period T31 corresponding to the reverse recovery time trr of the clamp diode DD1 shown in FIG. 8G overlaps, the accumulation time tstg of the overlapping switching element Q2
And the reverse recovery time trr of the clamp diode DD1 is set within a range in which the relationship of trr> tstg can be maintained, thereby preventing the switching elements Q1 and Q2 from being simultaneously turned on. .

However, even if the selection of the switching elements Q1 to Q4 and the clamp diodes DD1 to DD4 as described above is performed, the reverse recovery time trr of the clamp diodes DD1 to DD4 and the accumulation time tstg of the switching elements Q1 to Q4 do not increase. Of the switching elements Q1 to Q4, the reliability of the normal on / off operation of the switching elements Q1 to Q4 depends on the clamp diodes DD1 to DD4 within the selected range.
The reverse recovery time trr of DD4 and the switching elements Q1 to
Since the variation of the accumulation time tstg of Q4 is balanced, it is difficult to select the switching elements Q1 to Q4 under optimum driving conditions.

Therefore, a switching power supply circuit in which such a point is improved has been previously proposed by the present applicant. FIG. 9 is a circuit diagram showing one configuration example of a switching power supply circuit that can be configured based on the invention previously proposed by the present applicant. The same parts as those in FIG. 7 are denoted by the same reference numerals, and description thereof will be omitted.

The power supply circuit shown in this figure is different from the power supply circuit shown in FIG. 7 in that each drive winding NB1, NB2, NB3, NB4
With respect to the resonance capacitors CD1, CD2, CD3,
CD4 is connected in parallel, and is different in that a parallel resonance circuit composed of each of the drive windings NB1 to NB4 and the resonance capacitors CD1 to CD4 is formed. In such a circuit configuration, the switching element Q1 is [drive winding NB1 // resonant capacitor CD1]
Is driven by a self-excited oscillation by a parallel resonance circuit formed by the parallel resonance circuit, and its switching frequency is also set by these parallel resonance circuits. Similarly, the switching elements Q2, Q3, and Q4 are respectively connected to the [drive winding NB2 //
Resonant capacitor CD2], [Drive winding NB3 // Resonant capacitor CD3], [Drive winding NB4 // Resonant capacitor CD4]
Is driven by self-excited oscillation by a parallel resonance circuit formed by the parallel resonance circuit, and its switching frequency is also set by each of these parallel resonance circuits. In this case, each of the resonance capacitors CB1 to CB4 also functions as a DC blocking capacitor.

The low-speed clamp diode D inserted between the base and the emitter of the switching element Q1
Instead of D1, a series connection circuit of a clamp diode DD11, which is a small-capacity high-speed diode, and a zener diode DZ1 is inserted. Similarly, clamp diodes DD22, DD33 and DD44, which are small-capacity high-speed diodes, and a Zener diode DZ are provided between the bases and emitters of the switching elements Q2, Q3 and Q4.
2. Each series connection circuit with DZ3 and DZ4 is inserted.
In this case, the anode of the Zener diode DZ1 is connected to the base of the switching element Q1, and the cathode of the Zener diode DZ1 is connected to the cathode of the clamp diode DD11. Then, the anode of the clamp diode DD11 is connected to the emitter of the switching element Q1. Similarly, each series connection circuit connected between the base and the emitter of the switching elements Q2, Q3, Q4 is a series connection circuit comprising a Zener diode DZ1 and a clamp diode DD11 provided for the switching element Q1. The connection form is similar. The clamp diodes DD11 to DD44 are respectively Zener diodes D11 to D44.
It has a function as a reverse current blocking diode for preventing a current flowing in the forward direction (anode → cathode) of Z1 to DZ4.

With such a configuration, for example, the clamp diodes DD11 to DD44 are originally provided for supplying a clamp current when the switching elements Q1 to Q4 are turned off. By inserting the zener diodes DZ1 to DZ4 in the directions shown, the clamp current is prevented from flowing through the clamp diodes DD11 to DD44. In this case, the level (amplitude) of the drive voltage of the self-excited oscillation drive circuit that drives the switching elements Q1 to Q4 can be determined by the Zener voltage of the Zener diodes DZ1 to DZ4, but the drive voltage level is increased. With this setting, the negative-level base current (the current for extracting the accumulated charge of the base) when the switching elements Q1 to Q4 are turned off is also increased. As a result, the accumulation time of the switching elements Q1 to Q4 during the switching operation can be reduced.

For example, in a configuration in which a switching operation is performed by a switching element of a bipolar transistor as a current resonance type switching converter, if the switching frequency fs is set to be increased to about 100 kHz, the switching element caused by the variation in the storage time of the switching element is set. It has been found that the conduction time difference between them can be so large that it cannot be ignored. Therefore, if the zener diodes DZ1 to DZ4 are inserted as in the configuration shown in FIG. 9, the accumulation time tstg of the switching elements Q1 to Q4 becomes shorter,
As shown in the power supply circuit shown in FIG.
There is no need to manage the reverse recovery time trr of D4. That is, as a result, it is not necessary to select the storage time tstg of the switching elements Q1 to Q4 and the reverse recovery time trr of the clamp diodes DD11 to DD44.

FIG. 10 shows an example of a switching operation waveform of the primary-side current resonance type converter in the switching power supply circuit having the configuration shown in FIG. Note that the operation waveforms shown in this figure are those of the main part of the switching element Q2. In this case, the parallel resonance circuit including the drive winding NB2 and the parallel resonance capacitor CD2 performs self-excited oscillation, thereby generating the parallel resonance voltage VA shown in FIG. The parallel resonance voltage VA is, for example, as shown in FIG.
The waveform amplified from the resonance voltage VB when the parallel resonance capacitor CD2 is not connected in parallel to the drive winding NB2 shown in (e) is obtained. Then, by the parallel resonance voltage VA, a self-excited oscillation circuit is connected to the base of the switching element Q2 through a series connection of the base current limiting resistor RB2 and the resonance capacitor CB2 as shown in FIG. The resulting drive current IB flows. This FIG.
The peak level IBH of the forward base current and the peak level IBL of the reverse base current of the drive current IB shown in FIG.
Is larger than the peak level IBH of the forward base current and the peak level IBL of the reverse base current of the drive current IB shown in FIG. 8C. In this case, in the period TON, the switching element Q2 is turned on, and a collector current IC1 having a waveform as shown in FIG. In addition, period TOFF
Then, the drive current IB becomes the 0 level, and the switching element Q2 is also turned off (non-conductive). In addition, the base of the switching element Q2 in the above-mentioned periods TON and TOFF.
The waveform shown in FIG. 10C is obtained as the emitter-to-emitter voltage VBE. In response to this, the current I flowing through the series connection circuit of the Zener diode DZ2 and the clamp diode DD22 is obtained.
As shown in FIG. 10 (e), Z actually has a waveform in which a clamp current slightly flows in the period TOFF.

[0029]

However, the power supply circuit shown in FIG. 9 is different from the power supply circuit shown in FIG. 7 in that resonance capacitors CD1 to CD4 are provided for the respective drive windings NB1 to NB4.
Are connected in parallel, and clamp diodes DD11 to DD44 and a series connection circuit of zener diodes DZ1 to DZ4 need to be inserted between the base and emitter of each switching element Q1 to Q4. There is a disadvantage that the score increases.

Also, in each of the power supply circuits shown in FIGS. 7 and 9, as shown in FIGS. 8 (c) and 10 (b), the forward direction of the switching elements Q1 to Q4 flows in the forward direction. The absolute value of the peak level IBH of the base current is equal to the absolute value of the peak level IBL of the reverse base current flowing in the negative direction. in this case,
The reverse base current for turning off each of the switching elements Q1 to Q4 is small, and each of the switching elements Q1 to Q4 is turned off.
It is assumed that the power loss at the falling time tf of Q4 is large. Therefore, when the switching frequency of each of the switching elements Q1 to Q4 is increased, the power loss in the falling time tf of each of the switching elements Q1 to Q4 increases accordingly, and each of the switching elements Q1 to Q4 increases.
It has been considered difficult to increase the switching frequency of Q4 to Q4. If the switching frequency is increased, each of the switching elements Q1 to Q1
4, the power conversion efficiency (ηAC → DC) was reduced due to the power loss due to the falling time tf.

[0031]

In view of the above-mentioned problems, the present invention is configured as a switching power supply circuit as follows. In other words, a gap is formed between the rectifying / smoothing means that receives a commercial AC power supply, generates a rectified smoothed voltage, and outputs the rectified smoothed voltage as a DC input voltage, so that a required coupling coefficient that is loosely coupled is obtained. Converter that is provided to transmit power to the secondary side, and is formed by full-bridge coupling of four switching elements, so that the DC input voltage is intermittently output to the primary winding of the isolated converter transformer Switching means, a switching means, forming a switching means, a self-excited oscillation drive circuit for switchingly driving the switching element in a self-excited manner, at least a leakage inductance component including a primary winding of an insulating converter transformer, and a primary winding. Formed by the capacitance of the series resonant capacitor connected in series, And a primary side series resonance circuit for the quenching operation and resonant. Also, the leakage inductance component of the secondary winding of the insulating converter transformer,
A secondary-side resonance circuit formed by a capacitance inserted on the secondary side and a secondary-side resonance circuit. DC output voltage generating means configured to generate the secondary DC output voltage of the above, and performing constant voltage control by variably controlling the switching frequency of the switching means according to the level of the secondary DC output voltage. Constant voltage control means. And
At least, a current in a negative direction is supplied to the self-excited oscillation drive circuit during a period in which the four switching elements are turned off, each being formed by a series connection of a diode element and a resistance element having no reverse recovery time. It is configured to include a clamp circuit provided in each of the four switching elements so as to flow.

According to the above configuration, the clamp circuit is formed by a diode element having a characteristic that has no reverse recovery time and a resistor connected in series. By configuring the clamp circuit in this way, the driving current in the reverse direction flowing when each switching element is turned off can be increased without increasing the circuit configuration, so that the accumulation time of the switching element can be reduced. At the same time, it is possible to eliminate the reverse recovery time as a diode element for the clamp function.

[0033]

FIG. 1 is a circuit diagram showing a configuration of a power supply circuit according to an embodiment of the present invention. Note that, in this figure, the same parts as those in FIGS. In the power supply circuit shown in this figure, while each of the switching elements Q1, Q2, Q3, Q4 is off, each self-excited oscillation drive circuit [(CB1-NB1), (CB2) of each of the switching elements Q1, Q2, Q3, Q4 is provided. −NB2),
(CB3-NB3), (CB4-NB4)], a clamp circuit 2 for supplying a clamp current which is a negative level current.
A, 2B, 2C, and 2D are provided. The clamp circuit 2A includes a Schottky barrier diode DSBD1 and a resistor RD1.
Are connected in series. Similarly, the clamp circuits 2B, 2C, and 2D form a series connection circuit including Schottky barrier diodes DSBD2, DSBD3, and DSBD4 and resistors RD2, RD3, and RD4 in the same connection form.

Here, the Schottky barrier diode D
SBD1 to DSBD4 have a characteristic that the reverse recovery time trr = 0. The anode side of the Schottky barrier diode DSBD1 is connected to the emitter of the switching element Q1 through a series connection with the resistor RD1, and the cathode is connected to the base of the switching element Q1 so as to be inserted into the circuit. Similarly, the anode side of each Schottky barrier diode DSBD2, DSBD3, DSBD4 is connected to the emitter of each switching element Q2, Q3, Q4 via a series connection with each of the resistors RD2, RD3, RD4, respectively. Barrier diodes DSBD2, DSBD
3, the cathode side of DSBD4 is connected to each switching element Q2, Q3
, Q4 to be connected to the circuit.

That is, in the present embodiment, instead of the low-speed clamp diodes DD1 to DD4 used in the power supply circuit shown in FIG. 7, a series connection of Schottky barrier diodes DSBD1 to DSBD4 and resistors RD1 to RD4 is performed. A configuration in which clamp circuits 2A to 2D composed of circuits are provided is employed.

On the secondary side of the power supply circuit according to the present embodiment, a full-wave rectifier circuit is provided for obtaining the DC output voltage EO1 and the output output voltage EO2. As shown in FIG. 5, the insulated converter transformer PIT of the present embodiment includes, for example, E-type cores CR1 and CR made of a ferrite material.
An EE-type core in which the magnetic legs 2 are opposed to each other is provided, and a primary winding N1 and a secondary winding N2 are provided to the center magnetic leg of the EE-type core by using a divided bobbin B. Are wound separately. A gap G is formed for the center magnetic leg as shown in the figure. As a result, loose coupling with a required coupling coefficient can be obtained.

The gap G can be formed by making the central magnetic legs of the E-shaped cores CR1 and CR2 shorter than the two outer magnetic legs. As the coupling coefficient k, for example, k ≒
A loosely coupled state of 0.85 is obtained, so that a saturated state is hardly obtained.

On the secondary side of the insulating converter transformer PRT, an alternating voltage induced by the primary winding N1 is generated in the secondary winding N2. In this case, for the secondary winding N2,
By connecting the secondary parallel resonance capacitor C2 in parallel, a parallel resonance circuit is formed by the leakage inductance L2 of the secondary winding N2 and the capacitance of the secondary parallel resonance capacitor C2. The alternating voltage excited by the secondary winding N2 by this parallel resonance circuit becomes a resonance voltage. That is, a voltage resonance operation is obtained on the secondary side.

That is, in this power supply circuit, the primary side is provided with a parallel resonance circuit for making the switching operation a voltage resonance type, and the secondary side is provided with a full-wave rectification operation (voltage resonance operation).
And a parallel resonance circuit for obtaining In the present specification, such a switching converter configured to operate with a resonance circuit provided on the primary side and the secondary side is also referred to as a “composite resonance type switching converter”.

Incidentally, the insulation converter transformer PIT
, The polarity (winding direction) of the primary winding N1 and the secondary winding N2 and the rectifier diode DO (DO1, DO2 / DO3, DO4)
, The mutual inductance M between the inductance L1 of the primary winding N1 and the inductance L2 of the secondary winding N2 may be + M or -M.

For example, when the connection form shown in FIG. 6A is employed, the mutual inductance is + M, and when the connection form shown in FIG. 6B is employed, the mutual inductance is -M. If this is made to correspond to the operation of the secondary side shown in FIG. 1, [rectifier diodes DO1, DO2, smoothing capacitor C
O1], for example, when the alternating voltage obtained in the secondary winding N2 has a positive polarity, a rectification current flows through the rectification diode DO1 to perform + M operation mode (forward mode). When the alternating voltage obtained in the secondary winding N2 has a negative polarity, the rectifier diode DO1 is turned off and no rectified current flows. That is, in this power supply circuit, the rectifying operation is performed in a mode where the mutual inductance is + M.

In such a configuration, the power is supplied to the load side increased by the action of the primary side parallel resonance circuit and the secondary side parallel resonance circuit, and the power supplied to the load side is also increased. The rate of increase in load power is also improved. This is because, as described above with reference to FIG. 5, the gap G is formed in the insulating converter transformer PIT and loose coupling is performed with a required coupling coefficient, thereby obtaining a state that is more difficult to be saturated. It is realized.

FIG. 2 shows an example of a switching operation waveform of the power supply circuit of FIG. 1 having such a configuration. Note that the operation waveforms shown in this figure are those of the main part of the switching element Q2. Here, the switching element Q
The waveforms of the collector-emitter voltage VC1 of the switching element Q2 and the collector current IC1 of the switching element Q2 obtained by the switching operation of FIG.
The waveforms shown in FIG. 8A and FIG.
The waveforms are the same as those shown in FIG. 8A and FIG.

In the periods TON and TOFF, the base-emitter voltage VBE of the switching element Q2 is shown in FIG.
The waveform shown in (d) is obtained, and in response to this, the current ID flowing through the Schottky barrier diode DSBD2 has a waveform as shown in FIG. 2 (e). In this case, the Schottky barrier diode DSBD2 is connected to the switching element Q2
During the period TOFF during which the switch is off, the self-excited oscillation drive circuit (CB2-
NB2) is made to conduct a clamp current flowing in the negative direction, and when a period TON during which the switching element Q2 is turned on is started, the damper current flows through the base-collector of the switching element Q2 in a short damper period. However, since the Schottky barrier diode DSBD2 has the reverse recovery time trr = 0, after the end of the damper period, the phenomenon that the reverse recovery current flows due to the reverse recovery time trr disappears, and the shot is immediately started. The key barrier diode DSBD2 is turned off.

By eliminating the period as the reverse recovery time trr, the operation of drawing the resonance current of the self-excited oscillation drive circuit to the primary side ground via the clamp diode DD2 is eliminated. Therefore, the drive current IB shown in FIG. 2C is supplied to the base of the switching element Q2 as a forward current from the time when the damper period ends and the damper current becomes 0 level, and The element Q2 is turned on. That is, the switching element Q2 is turned on from the start of the period TON shown in FIG.

The operation waveforms of the drive current IB shown in FIG. 2C include the absolute value of the peak level IBL of the reverse base current of the drive current IB during the period corresponding to the accumulation time tstg and the absolute value of the forward base current. When comparing the absolute value of the peak level IBH, the absolute value of the peak level IBL of the reverse base current of the drive current IB and the peak level IBH of the forward base current shown in FIG. 8C and FIG. In this embodiment, the peak level IBL of the reverse base current is about twice as large as the peak level IBH of the forward base current. The waveform of the forward drive current IB shown in FIG.
The waveform is flatter than the waveform of the drive current IB shown in FIGS. 8C and 10B.

This is because the Schottky barrier diode D
This is due to the fact that the forward voltage drop of SBD2 is smaller than that of the slow clamp diode DD2, which allows the stored carriers when the switching element Q2 is turned off to disappear more quickly. It also acts to shorten the accumulation time tstg of the switching element Q2 and the fall time tf.

Although not shown, the operation waveform of the main part of the switching element Q3 is shown by the same operation waveform as the operation waveform of the switching element Q2 described above, and the main part of the switching elements Q1 and Q4. Has a phase of 1 with the operation waveform of the switching element Q2.
The waveforms are different from each other by 80 degrees but are shown by almost the same waveform.

The power supply circuit according to this embodiment has a reverse recovery time tr instead of the clamp diodes DD1 to DD4.
Schottky barrier diodes DSBD1 to r = 0
The use of DSBD4 eliminates the cause of variation in the reverse recovery time trr. Then, as described above, the accumulation time t
The reverse base current (drive current IB) during the period corresponding to stg
) Increases about twice, it is possible to reduce the power loss due to the falling time tf. This also makes it possible to realize a higher switching frequency. Furthermore, the storage time tstg and the current amplification factor hFE for the switching elements Q1 to Q4
The range of ranking is expanded. Thereby, for example, improvement in manufacturing efficiency such as parts management is achieved.

Further, the power supply circuit according to the present embodiment
It is not necessary to connect the resonance capacitors CD1 to CD4 in parallel to the drive windings NB1 to NB4 as in the power supply circuit shown in FIG. 1 and instead of the series connection circuit of the clamp diodes DD11 to DD44 and the Zener diodes DZ1 to DZ4. A clamp circuit 2A comprising a series connection circuit of Schottky barrier diodes DSBD1 to DSBD4 and resistors RD1 to RD4 between the base and emitter of each of the switching elements Q1 to Q4.
.About.2D, the number of components can be reduced accordingly, and the circuit size can be reduced.

In the power supply circuit of the present embodiment, the switching elements Q1 to Q4 are provided with the base current limiting resistors RB1 to RB4, which are current limiting resistors, in order to obtain optimum driving conditions. , Clamp circuit 2
If the resistance values of the resistors RD1 to RD4 constituting the A to 2D are increased to obtain the optimum driving conditions, the base current limiting resistors RB1 to RB4 can be eliminated. Also in this case, the circuit size can be reduced by reducing the number of components.

However, for example, the base current limiting resistors RB1 to RB
B4 is deleted (for example, RB1 to RB4 = 0), and the clamp circuit 2
If the resistance values of the resistors RD1 to RD4 constituting A to 2D are increased to obtain the optimum driving conditions, the power loss in the resistors RD1 to RD4 increases as the load power increases. For example, as the resistors RD1 to RD4 become larger, it becomes difficult to arrange other components around the resistors RD1 to RD4 as the resistors RD1 to RD4 generate heat. For this reason, the circuit scale increases.

To avoid this, for example, in the case of the power supply circuit according to the present embodiment, the base current limiting resistors RB1 to RB4 should not be deleted when the maximum load power PoMAX is 150 W or more. In this case, heat generated due to power loss is generated by the base current limiting resistors RB1 to RB4 and the resistors RD1 to RD.
Since the heat is distributed to D4, the heat generated by each resistor is also reduced. Therefore, it is possible to prevent the resistors RD1 to RD4 from increasing in size, thereby preventing the circuit size from increasing.

Here, as a comparison between the power supply circuit shown in FIG. 1 and the power supply circuit shown in FIG. 7, under the circuit configuration shown in FIG. 7, a maximum load power of 300 W and an AC input voltage VAC = 100
When set to V, the base current limiting resistor RB (RB1 to RB
4) Assuming that 1 Ω / 2 W, the input power was 337 W, and the result was that the power conversion efficiency η (AC → DC) = 89%. On the other hand, under the configuration shown in FIG. 1, each base current limiting resistor RB (RB1 to RB4) = 0.68Ω / 1
W, each resistance RD (RD1 to RD4) = 4.7Ω / 2W, the input power = 331.5W, and the power conversion efficiency η (AC →
DC) = 90.5%. That is, FIG.
In this embodiment, the power loss is reduced by about 5.5 W in comparison with the circuit shown in FIG.

Next, FIG. 3 shows a configuration of a power supply circuit according to another embodiment of the present invention. In this figure, the same parts as those in FIG. The power supply circuit shown in this figure has a base current limiting resistor RB (R
B1 to RB4) and delete each of the clamp circuits 3A, 3B, 3
C, 3D, ferrite bead inductor LD
The difference from the clamp circuits 2A to 2D shown in FIG. 1 lies in that 1, LD2, LD3, and LD4 are added. In this case, the ferrite bead inductors LD1 to LD4 are configured to be inserted in series with respective series-connected circuits of Schottky barrier diodes DSBD1 to DSBD4 and resistors RD1 to RD4. Thus, even when the power supply circuit of the present embodiment is configured, the same operation waveform as that shown in FIG. 2 is obtained as the operation waveform. As such clamp circuits 3A to 3D, the resistance of each of the resistors RD1 to RD4 is 4.7Ω, and each of the ferrite bead inductors LD1 to
0.5 μH can be selected as the inductance of LD4.

Further, as shown in FIG. 4 as a clamp circuit 4, a connection form in which a ferrite bead inductor LD is inserted in parallel with the resistor RD, and such a clamp circuit 4 is connected to each of the switching elements Q1, Q2, Q3, The same effect can be obtained by a configuration in which Q4 is connected as clamp circuits 4A, 4B, 4C and 4D. Note that the clamp circuits 3A to 3D shown as provided in the power supply circuit shown in FIG. 3 and the clamp circuit 4 shown in FIG.
Can be naturally applied to the power supply circuit shown in FIG. 1 as the clamp circuits 4A to 4D. Conversely, the clamp circuits 2A to 2D shown as provided in the power supply circuit shown in FIG. 1 may be applied to the power supply circuit shown in FIG.

As a configuration of the secondary side of the power supply circuit shown in this figure, one end of the secondary winding N2 is connected to the secondary side ground, and the other end is connected to the rectifier diode via the series connection of the series resonance capacitor CS. It is connected to the connection point between the anode of DO1 and the cathode of rectifier diode DO2. The cathode of the rectifier diode DO1 is connected to the positive electrode of the smoothing capacitor CO1, and the anode of the rectifier diode DO2 is connected to the secondary side ground. The negative side of the smoothing capacitor CO1 is connected to the secondary side ground.

As a result, in such a connection form, a voltage doubler full-wave rectifier circuit comprising a set of [series resonance capacitor CS, rectifier diodes DO1, DO2, smoothing capacitor CO1] is provided. Here, the series resonance capacitor CS
Forms a series resonance circuit corresponding to the on / off operation of the rectifier diodes DO1 and DO2 by its own capacitance and the leakage inductance component of the secondary winding N2. That is, the power supply circuit of the present embodiment is provided with a series resonance circuit on the primary side for making the switching operation a current resonance type, and a series resonance circuit for obtaining the voltage doubler full-wave rectification operation on the secondary side. A complex resonance type switching converter provided with a circuit is employed.

Here, the above-mentioned [series resonance capacitor CS,
The voltage doubler full-wave rectification operation by the combination of the rectifier diodes DO1, DO2 and the smoothing capacitor CO1] is as follows. When a switching output is obtained on the primary winding N1 by the switching operation on the primary side, the switching output is excited by the secondary winding N2. During the period in which the rectifier diode DO1 is turned off and the rectifier diode DO2 is turned on, the rectifier diode DO1 operates in the depolarization mode in which the polarity (mutual inductance M) of the primary winding N1 and the secondary winding N2 is -M. ,
The rectifier diode D is formed by the leakage inductance of the secondary winding N2 and the series resonance action of the series resonance capacitor CS.
The operation of charging the series resonant capacitor CS with the rectified current rectified by O2 is obtained.

During the period in which the rectifier diode DO2 is turned off and the rectifier diode DO1 is turned on to perform the rectification operation, the polarity (mutual inductance M) of the primary winding N1 and the secondary winding N2 is + M. The operation becomes an additive polarity mode, in which the smoothing capacitor CO1 is charged in a state where series resonance occurs in which the potential of the series resonance capacitor CS is added to the voltage induced in the secondary winding N2. As described above, the rectifying operation is performed by using both the adding polarity mode (+ M; forward operation) and the decreasing polarity mode (-M; flyback operation).
In the smoothing capacitor CO1, a DC output voltage EO1 corresponding to almost twice the induced voltage of the secondary winding N2 is obtained.

According to the above configuration, in the present embodiment, the secondary side DC output voltage is obtained by performing the double voltage full-wave rectification utilizing the state in which the mutual inductance is in the operation mode of + M and -M. I have to. That is, since the electromagnetic energy due to the primary side current resonance action and the secondary side current resonance action is simultaneously supplied to the load side, the power supplied to the load side further increases accordingly, and the maximum load It is also possible to significantly increase the power.

Further, since the secondary DC output voltage is obtained by the voltage doubler full-wave rectifier circuit, for example, a level equivalent to the secondary DC output voltage obtained by the equal voltage rectifier circuit is obtained. Then, as the secondary winding N2,
Only one half of the conventional number of turns is required. This reduction in the number of turns leads to a reduction in the size and weight of the insulating converter transformer PIT and a reduction in cost. In this case, a secondary winding N2A is wound independently of the secondary winding N2, a center tap is grounded to the secondary winding N2A, and a rectifier diode DO3 is connected to the secondary winding N2A. , DO4 and smoothing capacitor CO2
Is connected to generate a DC output voltage EO2.

The configuration of the secondary-side resonance circuit of the power supply circuit according to the present embodiment is merely an example, and is not limited to the configuration described above.

[0064]

As described above, according to the present invention, there is no reverse recovery time trr of a Schottky diode or the like for a switching power supply circuit formed by connecting four switching elements in full bridge. The diode element and the resistance element form a clamp circuit to be provided in the self-excited resonance converter.
When the clamp circuit is configured in this manner, the reverse current of the drive current flowing when the switching element is turned off can be increased without increasing the circuit scale. It can be a current waveform. This can reduce the power loss of the switching element and improve the power conversion efficiency.

Further, according to the present invention, since the accumulation time and the fall time of each switching element can be shortened, the margin for the accumulation time of each switching element and the variation of the current amplification factor is expanded, and the optimum driving condition is accordingly increased. Margins are also expanded. In addition, the rank range when selecting each switching element is expanded, so that the working efficiency can be improved. Further, it is not necessary to select the reverse recovery time of the clamp diode.

Furthermore, since the fall time of each switching element is shortened, the switching frequency of the switching element can be further increased, so that the size of the series resonance capacitor and the insulating converter transformer can be reduced. It is also possible to reduce the weight.

Further, in the clamp circuit according to the present invention, even when a ferrite bead inductor is inserted in series or in parallel with a resistance element, for example, the drive current supplied to the switching element has an almost ideal current waveform. Therefore, power loss in the switching element can be reduced and power conversion efficiency can be improved.

Further, according to the present invention, if a current limiting resistor element for limiting current is not inserted between the self-excited oscillation driving circuit for driving the switching element and the control input of the switching element, the circuit scale becomes larger. Can be reduced in size. In the switching power supply circuit of the present invention, a current limiting resistor element is inserted between the self-excited oscillation drive circuit and the control input of the switching element under the condition that the load power to be handled is equal to or more than a predetermined value. ing. In this case, the heat generated due to the increase in the load power is dispersed by the resistance element of the clamp circuit and the current limiting resistance element without increasing the size of the resistance element of the clamp circuit and in the vicinity of the resistance element. Since it is possible to arrange other components in the circuit, it is possible to avoid an increase in circuit scale.

[Brief description of the drawings]

FIG. 1 is a circuit diagram illustrating a configuration example of a power supply circuit according to an embodiment of the present invention.

FIG. 2 is a waveform chart showing an operation of a main part of the power supply circuit of the present embodiment.

FIG. 3 is a circuit diagram illustrating a configuration example of a power supply circuit according to another embodiment of the present invention.

FIG. 4 is a diagram illustrating another configuration example of the clamp circuit according to the present embodiment;

FIG. 5 is a cross-sectional view showing a structure of the insulating converter transformer according to the present embodiment.

FIG. 6 is an explanatory diagram showing each operation when the mutual inductance is + M / −M.

FIG. 7 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 8 is a waveform chart showing an operation of a main part of a power supply circuit according to the prior art.

FIG. 9 is a circuit diagram showing a configuration of a power supply circuit as a prior art.

FIG. 10 is a waveform diagram showing an operation of a main part of a power supply circuit according to the prior art.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 Control circuit, 2A-2D 3A-3D4 Clamp circuit, Ci smoothing capacitor, Q1-Q4 switching element, PIT insulation converter transformer, PRT orthogonal type control (drive) transformer, C1 primary side series resonance capacitor, C2 secondary side parallel Resonant capacitor, CS secondary side series resonant capacitor, NC control winding, NB1 to NB4 drive winding, DSBD1 to DSBD4 Schottky barrier diode, RD1 to RD4 resistor, LD1 to LD4 ferrite bead inductor

Claims (6)

[Claims] A gap is formed between a rectifying / smoothing means that receives a commercial AC power supply to generate a rectified smoothed voltage and outputs the rectified smoothed voltage as a DC input voltage, and a required coupling coefficient that is loosely coupled. An insulated converter transformer provided to transmit the primary side output to the secondary side is formed by full-bridge coupling of four switching elements, and the DC input voltage is intermittently connected to the primary winding of the insulated converter transformer. A switching unit configured to output, a self-excited oscillation drive circuit that forms the switching unit and performs switching driving of the switching element in a self-excited manner, and at least a leakage inductance component including a primary winding of the insulating converter transformer. And the capacitance of a series resonant capacitor connected in series with the primary winding. A secondary series resonance circuit that makes the switching operation of the switching element a resonance type, a leakage inductance component of a secondary winding of the insulating converter transformer, and a capacitance inserted into the secondary side. A side resonance circuit is formed including the secondary side resonance circuit, and is configured to generate a predetermined secondary side DC output voltage by inputting an alternating voltage obtained in a secondary winding of the insulating converter transformer. DC output voltage generating means, and constant voltage control means for performing constant voltage control by variably controlling the switching frequency of the switching means according to the level of the secondary DC output voltage. Formed by a series connection of a diode element and a resistance element having no reverse recovery time. Re in the period in which the OFF, to flow the negative direction current to said self-excited oscillation driving circuit,
And a clamping circuit provided for each of the four switching elements. 2. The switching power supply circuit according to claim 1, wherein a ferrite bead inductor is inserted into the clamp circuit. 3. The switching power supply circuit according to claim 2, wherein said ferrite bead inductor is connected in series with said resistance element. 4. The switching power supply circuit according to claim 2, wherein said ferrite bead inductor is connected in parallel to said resistance element. 5. The switching power supply circuit according to claim 1, wherein a current limiting resistance element for limiting a current is not inserted between the self-excited oscillation driving circuit and a control input of the switching element. 6. A current for limiting current between the self-excited oscillation drive circuit and a control input of the switching element under a condition that a load power to be handled by the switching power supply circuit is equal to or more than a predetermined value. 2. The switching power supply circuit according to claim 1, wherein a limiting resistance element is inserted.