WO2024066480A1 - 双模式电荷控制方法 - Google Patents

双模式电荷控制方法 Download PDF

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Publication number
WO2024066480A1
WO2024066480A1 PCT/CN2023/099861 CN2023099861W WO2024066480A1 WO 2024066480 A1 WO2024066480 A1 WO 2024066480A1 CN 2023099861 W CN2023099861 W CN 2023099861W WO 2024066480 A1 WO2024066480 A1 WO 2024066480A1
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WIPO (PCT)
Prior art keywords
resonant
voltage
load
charge control
dual
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PCT/CN2023/099861
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English (en)
French (fr)
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WO2024066480A9 (zh
Inventor
彭柏瑞
林建璋
林祐任
黄嘉熊
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台达电子工业股份有限公司
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Publication of WO2024066480A1 publication Critical patent/WO2024066480A1/zh
Publication of WO2024066480A9 publication Critical patent/WO2024066480A9/zh

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0032—Control circuits allowing low power mode operation, e.g. in standby mode
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33538—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only of the forward type
    • H02M3/33546—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only of the forward type with automatic control of the output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33571—Half-bridge at primary side of an isolation transformer

Definitions

  • the present invention relates to a charge control method, and more particularly to a dual-mode charge control method.
  • the main power structure of the traditional series resonant circuit converter includes a switch network, an LC series resonant tank, and a rectifier, and the control structure includes a feedback voltage and current detection circuit (sensor), a compensator, a pulse generator (PWM generator), and a switch gate drive circuit (gate driver). It has the advantages of zero voltage switching, low electromagnetic interference noise, and a wide operating frequency.
  • the main disadvantages of this circuit converter are that the compensation is difficult to control, resulting in poor dynamic response, and the low-frequency DC voltage gain is low and the low-frequency ripple cannot be suppressed.
  • FIG1 is a waveform diagram of the charge control method of the prior art.
  • the control method shown in FIG1 is a bang-bang charge control (BBCC) control method.
  • the BBCC control method can also be called a hysteretic charge control method, which aims to improve the dynamic response speed and low-frequency DC voltage gain of the traditional LLC.
  • the technical means used are: two limit lines are generated by feedback of the relationship between the output voltage compensation control and the input voltage, and the two limit lines generated are compared with the sensed resonant voltage to control the power switch.
  • the upper limit line (upper limit line) is generated by the output voltage compensation calculation result
  • the lower limit line (lower limit line) is designed to be generated by subtracting the upper limit line value from the input voltage sensing result.
  • the positions of the two limit lines are calculated, and the current output power can be known by the distance between the two limit lines. For example, when the two limit lines are close to each other, the output power is low (light load), and conversely, when the two limit lines are far away from each other, the output power is high (heavy load).
  • the problem with the BBCC control method is that when the load is light or no-load, the two control lines will be very close to each other or even exchanged, and coupled with feedback signal noise, interference or voltage divider resistor errors, the circuit will be unstable or the control signal will oscillate, which can also be called subharmonic oscillation.
  • this method adds a monostable trigger to avoid entering the SR control error zone when the two limit lines are exchanged (that is, the SR trigger cannot accept the simultaneous input of high level at the S and R pins), if there is a rapid load change, it may cause the monostable trigger to fail to be correctly triggered, resulting in the above-mentioned control instability and oscillation.
  • the hybrid hysteresis control method shown in Figure 1 aims to improve the efficiency and stability of the BBCC control method under light load or no load.
  • the technical means used are: generating two limit lines by feeding back the output voltage compensation calculation result and its complementary value, and using a capacitor voltage divider circuit, two precision current sources and a differential comparator to compare the two limit lines generated with the resonant voltage generated by the capacitor voltage divider, thereby controlling the power switch.
  • the current load status can also be obtained.
  • the hybrid hysteresis control method limits the compensator output result to be always greater than the center value of the resonant voltage, so the two limit lines will not produce an exchange result, and the monostable trigger can be discarded to increase the stability of the control.
  • the two added precision current sources can provide slope compensation to improve the stability of the circuit, but the addition of the slope compensation circuit will reduce the circuit control frequency, so the advantages of the charge control method cannot be fully utilized.
  • the hybrid hysteresis control method will greatly reduce the control resolution, so it is not suitable for large wattage control.
  • both the BBCC control method and the hybrid hysteresis control method use a resonant circuit sensing method for voltage sensing, and both light load and no load control use two (limit) line control, and the sensing circuit does not need to be isolated.
  • the BBCC control method requires the use of a monostable trigger, while the hybrid hysteresis control method does not require the use of a monostable trigger.
  • the BBCC control method is lower, while the hybrid hysteresis control method is higher than the BBCC control method.
  • the purpose of the present invention is to provide a dual-mode charge control method to solve the problems of the prior art.
  • the dual-mode charge control method proposed in the present invention is used to control the output voltage of a load coupled to a circuit having a resonant tank.
  • the dual-mode charge control method includes the steps of: detecting the input voltage of the resonant tank, the resonant current of the resonant tank, the output current of the load, and the output voltage of the load; judging the current state of the load according to the output current, and performing single-line charge control when the load is judged to be in a light load or no-load state; in the single-line charge control, compensating the output voltage to generate an upper critical voltage, and the resonant current is calculated by a resettable integrator to obtain the resonant voltage; comparing the resonant voltage with the upper critical voltage to generate a first control signal, wherein when the resonant voltage is greater than or equal to the upper critical voltage, the resettable integrator is reset; generating a second control signal complementary to the first control signal through a pulse on-
  • the current status of the load is determined according to the output current, and when the load is in a heavy load or normal state, the single-line charge control is switched to the dual-line charge control.
  • the resettable integrator when the resonant voltage increases and reaches an upper threshold voltage, the resettable integrator is reset and the on-time length of the current first control signal is captured.
  • resetting the resettable integrator is performed by generating a control signal through a pulse width modulation generator to control the resettable integrator to return the integrated value of the sensed resonant current to zero.
  • the output voltage is compensated by voltage and slope.
  • the resonant current is sensed by an isolated current transformer connected in series with a resettable integrator.
  • the on-time length of the first power switch is the same as the on-time length of the second power switch; the off-time length of the first power switch is also the same as the off-time length of the second power switch.
  • the steps are further included: subtracting the upper threshold voltage from the result of sensing the input voltage to generate a lower threshold voltage; comparing the isolated sensed resonant voltage with the upper threshold voltage and the lower threshold voltage to generate a first control signal and a second control signal; providing the first control signal and the second control signal to control the resonant voltage respectively.
  • the first power switch and the second power switch of the circuit are further included: subtracting the upper threshold voltage from the result of sensing the input voltage to generate a lower threshold voltage; comparing the isolated sensed resonant voltage with the upper threshold voltage and the lower threshold voltage to generate a first control signal and a second control signal; providing the first control signal and the second control signal to control the resonant voltage respectively.
  • the resonant bridge arm switch control signal is combined with the signal of the resonant frequency of the resonant tank to control the synchronous rectification bridge arm switch.
  • the resonant current is operated through a current transformer connected in series with a resettable integrator to obtain an isolated resonant voltage.
  • the resettable integrator when the resonant voltage is greater than or equal to the upper critical voltage, the resettable integrator is reset and the on-time length of the first control signal is recorded.
  • a single-line charge control method i.e., a dual-mode charge control method
  • a dual-mode charge control method is added to the dual-line charge control method to increase the light-load and no-load stability of the resonant circuit, and the single limit line and dual-line charge control can be freely switched according to the load conditions.
  • FIG1 is a waveform diagram of a charge control method in the prior art.
  • FIG. 2 is a waveform diagram of the dual-mode charge control method of the present invention.
  • FIG3 is a waveform diagram of the dual-line charge control under normal load or heavy load of the present invention.
  • FIG4 is a waveform diagram of single-line charge control under light load or no load according to the present invention.
  • FIG. 5 is a circuit diagram of a circuit structure of a converter with a resonant circuit according to the present invention.
  • FIG. 6 is a block diagram of a control architecture of a converter with a resonant circuit according to the present invention.
  • FIG. 7 is a block diagram showing the operation of the resonant circuit and the control unit of the present invention.
  • FIG8 is a flow chart of single-line charge control of the dual-mode charge control method of the present invention.
  • FIG. 9 is a flow chart of dual-line charge control of the dual-mode charge control method of the present invention.
  • the main purpose of the present invention is that the known two-line charge control has the advantages of high frequency response and high and low frequency voltage gain, but this control method will have oscillation problems when the resonant circuit operates under light load or no load conditions. Therefore, the present invention is based on the two-line charge control method, and adds a single-line charge control method (that is, a dual-mode charge control method) to increase the stability of the resonant circuit under light load and no load. Furthermore, the control method of the present invention can freely switch between single limit line and two-line charge control according to the load condition.
  • the light load (or no load) or heavy load (or general state) recorded in the present invention can be easily determined by those with ordinary knowledge in the technical field, so it will not cause misunderstanding or ambiguity to this definition.
  • the technical content and detailed description of the present invention are described as follows with the accompanying drawings.
  • FIG8 is a flow chart of the single-line charge control of the dual-mode charge control method of the present invention.
  • FIG5 and FIG6 are respectively a circuit diagram of the circuit architecture of the converter with a resonant circuit of the present invention and a block diagram of the control architecture, and refer to FIG7, which is a block diagram of the operation of the resonant circuit and the control unit of the present invention.
  • the dual-mode charge control method of the present invention is used to control the output voltage of a load coupled to a resonant circuit with a resonant tank.
  • the converter with a resonant circuit of the present invention is used to receive a DC input, and through a resonant circuit, a rectifier, and a charge control method, controls a stable output DC voltage to supply the load.
  • the resonant circuit includes an isolation transformer, and the resonant tank is composed of a resonant inductor Lr, a resonant capacitor Cr, and an excitation inductor Lm of the isolation transformer, and is formed on the primary side of the isolation transformer.
  • the front stage of the resonant circuit is a conversion stage, and the rear stage of the resonant circuit is a rectification stage.
  • the conversion stage is a conversion circuit, having a first switch Q1 and a second switch Q2 coupled in series, and the resonant tank is coupled between the common point of the first switch Q1 and the second switch Q2 and the ground point.
  • the rectification stage is a synchronous rectification circuit, having a first synchronous rectification switch SR1 and a second synchronous rectification switch SR2 coupled to the secondary side of the isolation transformer.
  • the output power control method of the converter with a resonant circuit includes the following steps: first, detecting the input voltage Vin of the resonant tank, the resonant current Ir of the resonant tank, the output current Io of the load, and the output voltage Vo of the load (S10).
  • ⁇ A shown in FIG5 is the detection of the input voltage Vin of the resonant tank
  • ⁇ B shown in FIG5 is the detection of the resonant current Ir of the resonant tank
  • ⁇ C shown in FIG5 is the detection of the output current Io of the load
  • ⁇ D shown in FIG5 is the detection of the output voltage Vo of the load.
  • the input voltage Vin, the resonant current Ir, the output current Io, and the output voltage Vo are detected respectively by the sensing circuit. It is worth mentioning that the sensing of the resonant current needs to be realized by using an isolated current transformer in series with a resettable integrator. In this way, the isolation transformer is used as a current transformer to isolate the controller electrical circuit from the primary side of the resonant converter circuit, which is conducive to the integration of the controller electrical circuit and the subsequent circuit.
  • the load state is determined according to the output current Io, that is, when the load is determined to be light load or no load, the load state is determined to be
  • the single-line charge control is switched to a dual-line charge control (as shown in the left half of FIG. 2 ) (S20). That is, the output current sensing result is used to determine whether the load is in a normal state (not a light load or no load state) or a light load or no load state.
  • the resonant circuit is currently required to operate in a dual-limit charge control or a single-line charge control through mode switching to control the resonant circuit.
  • step (S20) In the single-line charge control when the load is judged to be light-loaded or no-loaded in step (S20), refer to FIG. 4, which is a waveform diagram of the single-line charge control under light-loaded or no-loaded conditions of the present invention, the output voltage Vo is compensated to generate the upper threshold voltage VTH, and the resonant current Ir is calculated by connecting the current transformer in series with the resettable integrator to obtain the isolated resonant voltage Vr (S30). Then, the resonant voltage Vr is compared with the upper threshold voltage VTH to generate the first control signal SQ1.
  • the resettable integrator When the resonant voltage Vr is greater than or equal to the upper threshold voltage VTH, the length of time the first control signal SQ1 is turned on is captured and the resettable integrator is reset (S40). That is, when the resonant circuit operates under the single-line charge control, a signal is generated through a pulse width modulation (PWM) module to control the resettable integrator, thereby returning the sensed resonant current (signal) to zero, which means that the integral value of the resettable integrator can be cleared through the PWM module, thereby resetting the resettable integrator. Therefore, under light-loaded or no-loaded conditions, when the resonant voltage Vr reaches the upper threshold voltage VTH, the resettable integrator is reset through the pulse width modulation generator.
  • PWM pulse width modulation
  • the dual-mode charge control method of the present invention does not need to use a monostable trigger, and can avoid the upper threshold voltage VTH flipping to a low-level threshold voltage under light load or no-load conditions to cause output signal errors. Furthermore, it can avoid the divergence of the resonant voltage Vr signal during single-line charge control, causing abnormal control signals of the first power switch Q1 and the second power switch Q2.
  • a second control signal SQ2 complementary to the first control signal SQ1 is generated by a pulse on-time duplicator according to the recorded on-time length of the first control signal SQ1 (S50).
  • the first control signal SQ1 and the second control signal SQ2 are provided to control the first power switch Q1 and the second power switch Q2 of the resonant circuit respectively. Therefore, in the same cycle, the on-time length of the first power switch Q1 is the same as the on-time length of the second power switch Q2; the off-time length of the first power switch Q1 is the same as the off-time length of the second power switch Q2.
  • the resonant circuit operates in single-line charge control, and the sensed output voltage Vo is subjected to voltage and slope compensation to generate an upper threshold voltage VTH, which is compared with the result of the sensed resonant current circuit, that is, the resonant current Ir is operated through a resettable integrator to obtain a resonant voltage Vr, so that the resonant voltage Vr is compared with the upper threshold voltage VTH.
  • a first control signal SQ1 is generated, which is a pulse width modulation (PWM) control command for controlling the first power switch Q1 (i.e., the upper arm switch), and then, the PWM control signal (first control signal SQ1) is generated through a pulse on-time replicator to generate a complementary PWM control signal (second control signal SQ2) to control the second power switch Q2 (i.e., the lower arm switch) to achieve high frequency response and increase low frequency voltage gain.
  • PWM pulse width modulation
  • the charge control means is to perform single-line charge control under light load.
  • the upper threshold voltage VTH is obtained by the output voltage Vo through the compensation circuit;
  • the resonant voltage Vr is obtained by the resonant current Ir through Ksen and the resettable integrator;
  • the control signal SQ1 of the first power switch Q1 is to convert the resonant voltage Vr
  • the control signal SQ2 of the second power switch Q2 is generated by passing the resonant voltage Vr and the upper threshold voltage VTH through the first comparator, the pulse width modulation limiter, and then through the pulse conduction time replicator to generate a complementary signal, and then through the pulse width modulation generator.
  • the rectifier bridge arm switch is controlled by a signal combination of the resonant bridge arm switch (i.e., the first power switch Q1 and the second power switch Q2) control signal and the resonant frequency of the resonant tank, that is, the resonant bridge arm switch (i.e., the first power switch Q1 and the second power switch Q2) control signal is compensated by the resonant frequency of the resonant tank, so that the converter efficiency can be improved by using synchronous rectification.
  • the resonant bridge arm switch i.e., the first power switch Q1 and the second power switch Q2
  • FIG3 is a waveform diagram of the dual-line charge control under normal load or heavy load of the present invention.
  • FIG9 is a flow chart of the dual-line charge control of the dual-mode charge control method of the present invention.
  • the steps of the dual-line charge control include, first, the sensed output voltage Vo is subjected to the voltage and slope compensation result to generate the upper threshold voltage VTH, and the sensed input voltage Vin is subtracted from the compensation result to generate the lower threshold voltage VTL (S21).
  • the above two limit lines are compared with the isolated sensing resonant current circuit result (i.e., the resonant voltage Vr), and the comparison result will generate the first control signal SQ1 and the second control signal SQ2 (S22). And the first control signal SQ1 and the second control signal SQ2 respectively control the first power switch Q1 and the second power switch Q2 of the resonant circuit (S23).
  • the final PWM control command must be transmitted to the PWM module to generate a pulse control signal for controlling the first power switch Q1 and the second power switch Q2 and the first synchronous rectifier switch SR1 and the second synchronous rectifier switch SR2.
  • a single-line charge control method i.e., a dual-mode control method is added to increase the light-load and no-load stability of the resonant circuit.
  • the resonant circuit includes an isolation transformer, and the resonant tank is composed of a resonant inductor, a resonant capacitor, and the magnetizing inductor of the isolation transformer, and is formed on the primary side of the isolation transformer.
  • the isolation transformer is used as a current transformer to isolate the controller electrical circuit from the primary side of the resonant converter circuit, which is conducive to the integration of the controller electrical circuit and the subsequent circuit.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

一种双模式电荷控制方法，用以对耦接于具有谐振槽的电路的负载进行输出电压控制，该方法包括：检测谐振槽的输入电压、谐振槽的谐振电流、负载的输出电流以及负载的输出电压；根据输出电流判断目前负载状况，当负载在轻载或无载状态时，进行单线电荷控制；在单线电荷控制中，补偿输出电压以产生上临界电压，且谐振电流通过可复位积分器运算以获得谐振电压；比较谐振电压与上临界电压以产生第一控制信号，其中当谐振电压大于或等于上临界电压时，重置可复位积分器；通过脉冲导通时间复制器产生与第一控制信号互补的第二控制信号；提供第一控制信号与第二控制信号分别控制谐振电路的第一功率开关与第二功率开关。

Description

双模式电荷控制方法 技术领域
本发明是有关一种电荷控制方法，特别涉及一种双模式电荷控制方法。
背景技术
传统串联式谐振电路转换器(LLC converter)的主电源结构包括开关网络(switch network)、LC串联谐振槽、整流器(rectifier)，而控制结构包括反馈电压及电流检测线路(sensor)、补偿器(compensator)、脉冲产生器(PWM generator)以及开关栅极驱动线路(gate driver)。其具备有零电压切换、低电磁干扰噪声与操作频率宽广等优点。然而，补偿不易控制而造成动态响应差、低频直流电压增益低而无法抑制低频涟波，则为该电路转换器的主要缺点。
请参见图1所示，其为现有技术的电荷控制方法的波形示意图。图1所示的控制方法为双线电荷控制(bang-bang charge control,BBCC)控制方法。BBCC控制方法亦可称为迟滞电荷控制(hysteretic charge control)方法，其目的在于改善传统LLC动态响应速度与低频直流电压增益，所使用的技术手段为：通过反馈输出电压补偿控制与输入电压之间的关系来产生两条限制线，并将产生出的两条限制线与感测的谐振电压做比较，进而控制功率开关。如图1所示的形示意图，其中上面的一条限制线(上限制线)是由输出电压补偿计算结果所产生，而下面的一条限制线(下限制线)是设计为与输入电压感测结果减去上限制线数值所产生。计算出两条限制线的位置，并且通过两限制线之间距离可以得知目前输出功率的大小，例如：两限制线互相靠近时输出功率较低(轻载)，反之两限制线互相远离时输出功率较高(重载)。
然而，BBCC控制方法存在的问题为：当负载较轻或是无载状态时，两条线制线会非常靠近彼此甚至交换，再加上反馈信号噪声、干扰或分压电阻的误差等情况，使得电路产生不稳定现象或是控制信号振荡的情况，此现象亦可称之为次谐波振荡。该方法虽有加入单稳态触发器以避免在两限制线交换情况下进入SR控制误区(是指SR触发器不能接受S端及R端引脚同时输入高电平的情况)，但是，如遇快速的负载变化则可能导致无法正确触发单稳态触发器，因而产生上述控制不稳定与振荡的现象。
图1所示的混合迟滞控制方法目的在于改善BBCC控制法在轻载或无载下的效率及稳定性，所使用的技术手段为：通过反馈输出电压补偿计算结果与其互补值来产生两条限制线，并且使用电容分压式的分压线路、两个精密电流源以及差分比较器，将产生出的两条限制线与电容分压出的谐振电压做比较，进而控制功率开关。通过两条限制线之间距离 亦可得到目前负载状况。混合迟滞控制方法限制补偿器输出结果恒大于谐振电压的中心值，因此两限制线不会产生交换的结果，进而可将单稳态触发器舍弃以增加控制的稳定性。另外所增加的两个精密电流源，可提供斜率补偿以提高电路的稳定性，但加入斜率补偿电路会降低电路控制频，因而无法完全发挥电荷控制方式的优势。混合迟滞控制方法会大幅减少控制分辨率，因此不适合较大瓦数的控制。
综上，BBCC控制法与混合迟滞控制法皆采用电压感测的谐振电路感测方式，并且在轻载与无载下控制皆采用两(限制)线控制，并且感测电路均无须隔离。此外，BBCC控制法需要使用单稳态触发器，而混合迟滞控制法无需使用单稳态触发器。再者，以控制复杂度而言，BBCC控制法较低，而混合迟滞控制法较BBCC控制法为高。
为此，如何设计出一种双模式电荷控制方法，解决现有技术所存在的问题与技术瓶颈，乃为本公开发明人所研究的重要课题。
发明内容
本发明的目的在于提供一种双模式电荷控制方法，解决现有技术的问题。
为实现前揭目的，本发明所提出的双模式电荷控制方法用以对耦接于具有谐振槽的电路的负载进行输出电压控制。该双模式电荷控制方法包括步骤：检测谐振槽的输入电压、谐振槽的谐振电流、负载的输出电流以及负载的输出电压；根据输出电流判断负载目前状态，当判断负载为轻载或无载状态时，进行单线电荷控制；在单线电荷控制中，补偿输出电压以产生上临界电压，且谐振电流通过可复位积分器运算以获得谐振电压；比较谐振电压与上临界电压以产生第一控制信号，其中当谐振电压大于或等于上临界电压时，重置可复位积分器；通过脉冲导通时间复制器产生与第一控制信号互补的第二控制信号；以及提供第一控制信号与第二控制信号分别控制谐振电路的第一功率开关与第二功率开关。
在一实施例中，根据输出电流判断负载目前状况，当负载为重载或一般状态时，将单线电荷控制切换为双线电荷控制。
在一实施例中，当谐振电压增加以至于触及到上临界电压时，重置可复位积分器，并且抓取目前第一控制信号的导通时间长度。
在一实施例中，重置可复位积分器是通过脉冲宽度调制产生器产生控制信号以控制可复位积分器，将感测的谐振电流的积分值归零。
在一实施例中，输出电压是以电压与斜率进行补偿。
在一实施例中，谐振电流的感测是通过隔离式比流器串接可复位积分器所实现。
在一实施例中，在同一周期中，第一功率开关的导通时间长度与第二功率开关的导通时间长度相同；第一功率开关的关断时间长度与第二功率开关的断关时间长度亦相同。
在一实施例中，其中在双线电荷控制中，还包括步骤：将感测输入电压的结果减去上述的上临界电压以产生下临界电压；比较隔离感测的谐振电压与上临界电压、下临界电压以产生第一控制信号与第二控制信号；提供第一控制信号与第二控制信号分别控制谐振电 路的第一功率开关与第二功率开关。
在一实施例中，谐振桥臂开关控制信号与谐振槽的谐振频率的信号联集进行同步整流桥臂开关的控制。
在一实施例中，谐振电流通过比流器串接可复位积分器运算以获得隔离后的谐振电压。
在一实施例中，当谐振电压大于或等于上临界电压时，重置可复位积分器，并且记录第一控制信号的导通时间长度。
通过所提出的双模式电荷控制方法，基于双线电荷控制方式下，加入单线电荷控制方法(即为双模式电荷控制方法)以增加谐振电路于轻载与无载稳定度，并且依据负载状况可自由切换单限制线与双线电荷控制。
为了能更进一步了解本发明为实现预定目的所采取的技术、手段及技术效果，请参阅以下有关本发明的详细说明与附图，相信本发明的目的、特征与特点，当可由此得一深入且具体的了解，然而附图仅提供参考与说明用，并非用来对本发明加以限制者。
附图说明
图1：为现有技术的电荷控制方法的波形示意图。
图2：为本发明双模式电荷控制方法的波形示意图。
图3：为本发明一般负载或重载下的双线电荷控制的波形示意图。
图4：为本发明轻载或无载下的单线电荷控制的波形示意图。
图5：为本发明具谐振电路的转换器的电路架构的电路图。
图6：为本发明具谐振电路的转换器的控制架构的方框图。
图7：为本发明谐振电路与控制单元操作的方框示意图。
图8：为本发明双模式电荷控制方法的单线电荷控制的流程图。
图9：为本发明双模式电荷控制方法的双线电荷控制的流程图。
附图标记说明：
Q1：第一功率开关
Q2：第二功率开关
SR1：第一同步整流开关
SR2：第二同步整流开关
Vin：输入电压
Vo：输出电压
Ir：谐振电流
Io：输出电流
Vr：谐振电压
SQ1：第一控制信号
SQ2：第二控制信号
VTH：上临界电压
VTL：下临界电压
S10～S60：步骤
S21～S23：步骤
具体实施方式
兹有关本发明的技术内容及详细说明，配合附图说明如下。
本发明的主要目的为，已知双线电荷控制具有高频率响应与高低频电压增益的优点，但此控制法在谐振电路操作于轻载或无载状况下会有振荡的问题，因此本发明基于双线电荷控制方式下，加入单线电荷控制方法(即为双模式电荷控制方法)以增加谐振电路于轻载与无载稳定度，再者，本发明控制方法依据负载状况可自由切换单限制线与双线电荷控制。附带一提，本发明所记载的轻载(或无载)或重载(或一般状态)为所属技术领域中具通常知识者可轻易判知，因此并不会对此定义造成误解或模糊不明。兹有关本发明的技术内容及详细说明，配合附图说明如下。
请参见图8所示，其为本发明双模式电荷控制方法的单线电荷控制的流程图。配合参见图5与图6，其是分别为本发明具谐振电路的转换器的电路架构的电路图与控制架构的方框图，以及配合参见图7，其为本发明谐振电路与控制单元操作的方框示意图。本发明的双模式电荷控制方法，用以对耦接于具有谐振槽的谐振电路的负载进行输出电压控制。本发明的具谐振电路的转换器是用以接收直流输入，经谐振电路、整流器，并通过电荷控制手段控制稳定的输出直流电压，以供给负载所需。如图5所示，该谐振电路包括隔离变压器，且谐振槽由谐振电感Lr、谐振电容Cr以及隔离变压器的激磁电感Lm所组成，且形成在隔离变压器的一次侧上。
谐振电路的前级为转换级，谐振电路的后级为整流级。转换级为转换电路，具有串联耦接的第一开关Q1与第二开关Q2，且谐振槽耦接于第一开关Q1与第二开关Q2的共接点与接地点之间。整流级为同步整流电路，具有耦接于隔离变压器的二次侧上的第一同步整流开关SR1与第二同步整流开关SR2。
该具谐振电路的转换器的输出功率控制方法包括步骤：首先，检测谐振槽的输入电压Vin、谐振槽的谐振电流Ir、负载的输出电流Io以及负载的输出电压Vo(S10)。图5所示的○A即为检测谐振槽的输入电压Vin；图5所示的○B即为检测谐振槽的谐振电流Ir；图5所示的○C即为检测负载的输出电流Io；图5所示的○D即为检测负载的输出电压Vo。通过感测电路各别检测输入电压Vin、谐振电流Ir、输出电流Io与输出电压Vo，值得一提，谐振电流的感测需使用隔离式比流器串接可复位积分器(resettable integrator)所实现。借此，利用隔离变压器作为比流器，使控制器电气回路与谐振转换器电路一次侧隔离，如此有利于控制器电气回路与后级电路整合。
然后，根据输出电流Io判断负载的状态，意即当判断负载为轻载或无载状态时，进行 单线电荷控制(如图2所示的右半边)，或判断负载为重载或一般状态(即非前述轻载或无载状态)时，将单线电荷控制切换为双线电荷控制(如图2所示的左半边)(S20)。意即，利用输出电流感测结果判断负载为一般状态(非轻载与非无载状态)或者是轻载与无载状态。换言之，在判断负载的状态后，经由模式切换来控制谐振电路目前需操作于双限制电荷控制或者是单线电荷控制。
在步骤(S20)若判断负载为轻载或无载状态时的单线电荷控制中，配合参见图4所示，其为本发明轻载或无载状态下的单线电荷控制的波形示意图，补偿输出电压Vo以产生上临界电压VTH，且谐振电流Ir通过比流器串接可复位积分器运算以获得隔离后的谐振电压Vr(S30)。然后，比较谐振电压Vr与上临界电压VTH以产生第一控制信号SQ1。其中当谐振电压Vr大于等于上临界电压VTH时，抓取第一控制信号SQ1导通的时间长度并且重置可复位积分器(S40)。意即，当谐振电路操作在单线电荷控制下，需经由脉冲宽度调制(PWM)模块产生信号控制可复位积分器，进而将感测的谐振电流(信号)归零，意即可通过PWM模块清除可复位积分器的积分值，进而重置可复位积分器。因此，在轻载或无载状态下，当谐振电压Vr顶到上临界电压VTH时，会通过脉冲宽度调制产生器重置可复位积分器。
借此，本发明双模式电荷控制方法不需要使用单稳态触发器，并且可避免在轻载或无载状态下，上临界电压VTH翻转为低电平的临界电压而造成输出信号错误，再者可避免单线电荷控制时，谐振电压Vr信号发散，造成第一功率开关Q1与第二功率开关Q2的控制信号异常。
然后，通过所记录第一控制信号SQ1导通时间长度经由脉冲导通时间复制器(duplicator)产生与第一控制信号SQ1互补的第二控制信号SQ2(S50)。最后，提供第一控制信号SQ1与第二控制信号SQ2分别控制谐振电路的第一功率开关Q1与第二功率开关Q2。因此，在同一周期中，第一功率开关Q1的导通时间长度与第二功率开关Q2的导通时间长度相同；第一功率开关Q1的关断时间长度与第二功率开关Q2的断关时间长度相同。
具体地，在轻载或无载状态下，谐振电路操作于单线电荷控制，将感测输出电压Vo经过电压与斜率补偿结果产生上临界电压VTH，经过与感测谐振电流电路结果做比较，即谐振电流Ir通过可复位积分器运算以获谐振电压Vr，使得谐振电压Vr与上临界电压VTH进行比较。根据比较结果将产生第一控制信号SQ1，其为脉冲宽度调制(PWM)控制命令，用以控制第一功率开关Q1(即上臂开关)，然后，再将此PWM控制信号(第一控制信号SQ1)通过脉冲导通时间复制器来产生出互补的PWM控制信号(第二控制信号SQ2)控制第二功率开关Q2(即下臂开关)，以达到高频率响应与增加低频电压增益。
如图6与图7所示，电荷控制手段是在轻载下进行单线电荷控制。其中，上临界电压VTH为将输出电压Vo经补偿电路获得；谐振电压Vr为将谐振电流Ir经Ksen与可复位积分器(resettable integrator)获得；第一功率开关Q1的控制信号SQ1是将谐振电压Vr 与上临界电压VTH经第一比较器、脉冲宽度调制限制器后，再经脉冲宽度调制产生器产生；第二功率开关Q2的控制信号SQ2是将谐振电压Vr与上临界电压VTH经第一比较器、脉冲宽度调制限制器后，再经脉冲导通时间复制器产生互补信号，再经脉冲宽度调制产生器产生。
对于同步整流控制，是以以谐振桥臂开关(意即第一功率开关Q1与第二功率开关Q2)控制信号与谐振槽谐振频率的信号联集进行整流器桥臂开关的控制，意即谐振桥臂开关(意即第一功率开关Q1与第二功率开关Q2)控制信号通过谐振槽谐振频率进行补偿，如此通过同步整流的使用，可提升转换器效率。
相较于轻载或无载状态的重载或一般负载状态下，谐振电路操作于双线电荷控制，请参见图3所示，其为本发明一般负载或重载下的双线电荷控制的波形示意图。请配合参见图9所示，其为本发明双模式电荷控制方法的双线电荷控制的流程图。双线电荷控制的步骤包括，首先，将感测输出电压Vo经过电压与斜率补偿结果产生上临界电压VTH，且将感测输入电压Vin减去补偿结果来产生下临界电压VTL(S21)。然后，将上述两限制线与隔离感测谐振电流电路结果(即谐振电压Vr)做比较，比较结果将产生出第一控制信号SQ1与第二控制信号SQ2(S22)。并且第一控制信号SQ1与第二控制信号SQ2分别控制谐振电路的第一功率开关Q1与第二功率开关Q2(S23)。
无论在一般负载或轻载与无载情况下，都须将最终PWM控制命令传送给PWM模块来产生控制第一功率开关Q1与第二功率开关Q2以及第一同步整流开关SR1与第二同步整流开关SR2的脉冲控制信号。
综上所述，本发明是具有以下的特征与优点：
1、基于双线电荷控制方式下，加入单线电荷控制方法(即为双模式控制方法)以增加谐振电路于轻载与无载稳定度。
2、依据负载状况可自由切换单限制线与双线电荷控制。
3、维持双线电荷控制频率响应快的特性，并增加该控制方法在轻载下的稳定性及控制分辨率。
4、谐振电路包括隔离变压器，且谐振槽由谐振电感、谐振电容以及隔离变压器的激磁电感所组成，且形成在隔离变压器的一次侧上。利用隔离变压器作为比流器，使控制器电气回路与谐振转换器电路一次侧隔离，如此有利于控制器电气回路与后级电路整合。
5、舍弃BBCC控制法中单稳态触发器(mono-stable)以提升控制稳定性。
以上所述，仅为本发明优选具体实施例的详细说明与附图，而本发明的特征并不局限于此，并非用以限制本发明，本发明的所有范围应以下述的权利要求为准，凡合于本发明权利要求的构思与其类似变化的实施例，皆应包含于本发明的范围中，任何本领域技术人员在本发明的领域内，可轻易思及的变化或修饰皆可涵盖在以下本公开的权利要求。

Claims (11)

  1. 一种双模式电荷控制方法，用以对耦接于具有一谐振槽的一电路的一负载进行输出电压控制，该控制方法包括步骤：
    检测该谐振槽的一输入电压、该谐振槽的一谐振电流、该负载的一输出电流以及该负载的一输出电压；
    根据该输出电流判断该负载目前状况，当该负载为轻载或无载状态时，进行单线电荷控制；
    在单线电荷控制中，补偿该输出电压以产生一上临界电压，且该谐振电流通过一可复位积分器运算以获得一谐振电压；
    比较该谐振电压与该上临界电压以产生一第一控制信号，其中当该谐振电压大于或等于该上临界电压时，重置该可复位积分器；
    通过一脉冲导通时间复制器产生与该第一控制信号互补的一第二控制信号；以及
    提供该第一控制信号与该第二控制信号分别控制该谐振电路的一第一功率开关与一第二功率开关。
  2. 如权利要求1所述的双模式电荷控制方法，其中根据该输出电流判断该负载目前状况，当该负载为重载或一般状态时，将单线电荷控制切换为双线电荷控制。
  3. 如权利要求1所述的双模式电荷控制方法，其中当该谐振电压增加以至于触及到该上临界电压时，重置该可复位积分器，并且抓取目前该第一控制信号的导通时间长度。
  4. 如权利要求3所述的双模式电荷控制方法，其中重置该可复位积分器是通过一脉冲宽度调制产生器产生一控制信号以控制该可复位积分器，将隔离感测的该谐振电流的积分值归零。
  5. 如权利要求1所述的双模式电荷控制方法，其中该输出电压是以电压与斜率进行补偿。
  6. 如权利要求1所述的双模式电荷控制方法，其中该谐振电流的感测是通过一隔离式比流器串接该可复位积分器所实现。
  7. 如权利要求1所述的双模式电荷控制方法，其中在同一周期中，该第一功率开关的导通时间长度与该第二功率开关的导通时间长度相同；该第一功率开关的关断时间长度与该第二功率开关的断关时间长度相同。
  8. 如权利要求2所述的双模式电荷控制方法，其中在双线电荷控制中，还包括步骤：
    将感测输入电压的结果减去该上临界电压补偿该输入电压以产生一下临界电压；
    比较隔离感测的该谐振电压与该上临界电压、该下临界电压以产生该第一控制信号与该第二控制信号；以及
    提供该第一控制信号与该第二控制信号分别控制该谐振电路的该第一功率开关与该 第二功率开关。
  9. 如权利要求1所述的双模式电荷控制方法，其中该第一功率开关与该第二功率开关的控制信号与该谐振槽的谐振频率的信号联集进行同步整流桥臂开关的控制。
  10. 如权利要求1所述的双模式电荷控制方法，其中该谐振电流通过一比流器串接该可复位积分器运算以获得隔离后的该谐振电压。
  11. 如权利要求1所述的双模式电荷控制方法，其中当该谐振电压大于或等于该上临界电压时，重置该可复位积分器，并且记录该第一控制信号的导通时间长度。
PCT/CN2023/099861 2022-09-29 2023-06-13 双模式电荷控制方法 WO2024066480A1 (zh)

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