WO2023238376A1 - Impedance converter - Google Patents

Impedance converter Download PDF

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Publication number
WO2023238376A1
WO2023238376A1 PCT/JP2022/023436 JP2022023436W WO2023238376A1 WO 2023238376 A1 WO2023238376 A1 WO 2023238376A1 JP 2022023436 W JP2022023436 W JP 2022023436W WO 2023238376 A1 WO2023238376 A1 WO 2023238376A1
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Prior art keywords
line
signal line
ground layer
signal
distance
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PCT/JP2022/023436
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French (fr)
Japanese (ja)
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美和 武藤
史人 中島
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日本電信電話株式会社
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Priority to PCT/JP2022/023436 priority Critical patent/WO2023238376A1/en
Publication of WO2023238376A1 publication Critical patent/WO2023238376A1/en

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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
    • H01P5/00—Coupling devices of the waveguide type
    • H01P5/02—Coupling devices of the waveguide type with invariable factor of coupling

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  • Microstrip lines, coplanar lines, grounded coplanar lines, etc. are used as typical structures for propagating high-speed signals on high-frequency transmission line substrates.
  • a transmission line is constructed by forming a ground plane of a planar conductive layer on one side of a dielectric substrate, and forming a strip-shaped line on the other side of the dielectric substrate.
  • the characteristic impedance of these transmission lines is determined by the width and thickness of the signal line, the dielectric constant and thickness of the dielectric substrate, and the distance between the signal line and the ground pattern.
  • the characteristic impedance of the high-frequency circuit and the load circuit or signal source must be matched in order to efficiently transmit power and signals at the connection part. It is necessary to do so.
  • an impedance converter is used in which the characteristic impedance is different at both ends of the line (see Non-Patent Document 1).
  • the line width is gradually changed in a tapered shape.
  • the spacing d1 between the substrate connection pads 103 also increases, resulting in a problem that the size of the impedance converter increases.
  • FIG. 14B when the width of the signal lines 102 increases and the interval d2 between the signal lines 102 decreases, there is a problem in that crosstalk noise between the signal lines 102 increases.
  • Crosstalk noise between the signal lines 102 is caused by displacing electrons on the other signal line 102 when a signal pulse is transmitted by one signal line 102. Therefore, as the interval between the signal lines 102 becomes smaller, the amount of displacement of electrons on the other signal line 102 becomes larger, and the crosstalk noise also becomes larger. As described above, with conventional impedance converters, it is difficult to simultaneously improve line density and reduce crosstalk noise between lines, and it is difficult to apply them to high-density packaging.
  • the present invention was made to solve the above problems, and an object of the present invention is to provide an impedance converter that can both improve line density and reduce crosstalk noise between lines.
  • the second microstrip line connects the dielectric substrate, the first ground layer, and a signal line in a direction crossing the signal propagation direction.
  • second signal lines formed in the dielectric substrate so as to be arranged alternately with the first signal lines along the dielectric substrate, and the first grounded coplanar line is arranged along the dielectric substrate. and a second signal line formed on both sides of the first signal line, on the same layer as the first signal line, and above the second signal line.
  • the second grounded coplanar line includes the dielectric substrate, the second signal line, and the third ground layer formed on both sides of the second signal line; and the second ground layer formed above the second signal line.
  • the first microstrip line and the second microstrip line are arranged alternately along the direction intersecting the signal propagation direction on one side of the impedance converter, and the first microstrip line and the second microstrip line are arranged alternately along the direction crossing the signal propagation direction, and
  • the width of the first and second signal lines can be adjusted.
  • 1A-1C are a plan view and a cross-sectional view of an impedance converter of the present invention.
  • 2A-2C are cross-sectional views of impedance converters of the present invention.
  • FIG. 3 is a cross-sectional view of a granded coplanar line.
  • 4A and 4B are diagrams showing a model of a conventional impedance converter.
  • 5A and 5B are diagrams showing a model of an impedance converter according to a first embodiment of the present invention.
  • FIG. 6 is a diagram showing simulation results of backward crosstalk of the impedance converter according to the first embodiment of the present invention and the conventional impedance converter.
  • FIG. 1A-1C are a plan view and a cross-sectional view of an impedance converter of the present invention.
  • 2A-2C are cross-sectional views of impedance converters of the present invention.
  • FIG. 3 is a cross-sectional view of a granded coplanar line.
  • 4A and 4B are
  • FIG. 7 is a diagram showing simulation results of forward crosstalk of the impedance converter according to the first embodiment of the present invention and the conventional impedance converter.
  • 8A to 8C are a plan view and a cross-sectional view of an impedance converter according to a second embodiment of the present invention.
  • 9A to 9C are cross-sectional views of an impedance converter according to a second embodiment of the present invention.
  • 10A and 10B are diagrams showing a model of an impedance converter according to a second embodiment of the present invention.
  • FIG. 11 is a diagram showing simulation results of backward crosstalk of the impedance converter according to the second embodiment of the present invention and the conventional impedance converter.
  • FIG. 12 is a diagram showing simulation results of forward crosstalk of the impedance converter according to the second embodiment of the present invention and the conventional impedance converter.
  • 13A to 13C are a plan view and a cross-sectional view showing the structure of a conventional impedance converter.
  • 14A and 14B are plan views illustrating problems with conventional impedance converters.
  • FIG. 1A is a plan view of the impedance converter of the present invention
  • FIG. 1B is a sectional view taken along line AA' of the impedance converter of FIG. 1A
  • FIG. 1C is a sectional view taken along line BB' of the impedance converter of FIG. 1A
  • . 2A is a sectional view taken along line CC' of the impedance converter in FIG. 1A
  • FIG. 2B is a sectional view taken along line DD' of the impedance converter in FIG. 1A
  • FIG. 2C is a sectional view taken along line EE' of the impedance converter in FIG. 1A.
  • the first grounded coplanar line 3 includes a dielectric substrate 10, a signal line 12, and a signal line 13 on both sides of the signal line 12 on the same layer as the signal line 12 (on the surface of the dielectric substrate 10).
  • the second grounded coplanar line 4 includes a dielectric substrate 10, a signal line 13, a ground layer 15 formed on both sides of the signal line 13, and a ground layer 15 formed above the signal line 13. It is composed of a layer 14.
  • ground layers 14 formed on both sides of the signal line 12 are connected to the ground layer 11 via vias 22.
  • Ground layers 15 formed on both sides of the signal line 13 are connected to the ground layer 11 via vias 23. Further, the ground layer 14 and the ground layer 15 are connected to each other via a via 24.
  • the dielectric constant of the dielectric substrate 10 is ⁇ 1.
  • h1 be the distance between the signal line 12 and the ground layer 15 and the distance between the signal line 13 and the ground layer 14 in the thickness direction of the dielectric substrate 10.
  • the distance between the signal line 13 and the ground layer 11 in the thickness direction of the dielectric substrate 10 and the distance between the ground layer 14 and the ground layer 11 are assumed to be h2.
  • the distance from the longitudinal center line of the lower surface of the signal line 12 to the longitudinal center line of the upper surface of the signal line 13 is h3.
  • the distance between the signal line 12 and the ground layer 14 in the direction perpendicular to the signal propagation direction is g1
  • the distance between the signal line 13 and the ground layer 15 in the direction perpendicular to the signal propagation direction is g2.
  • the present invention is applicable to high-density packaging, and the widths w1 and w2 of the signal lines 12 and 13 and the distance between the center lines of the signal line 12 and the ground layer 14 (the distance between the signal line 13 and the ground layer 15) This makes it possible to continuously change the characteristic impedance while maintaining the centerline distance (distance between center lines) g3.
  • one end (input side) of the impedance converter of this embodiment has an input impedance Z i and the other end (output side) has an output impedance Z o (Z i >Z o ).
  • the left end is the input side and the right end is the output side.
  • the distance between the signal line 12 and the ground layers 14 on both sides, and the distance between the signal line 13 and the ground layers 15 on both sides gradually become smaller from the input side to the output side.
  • the characteristic impedance of the transmission line also gradually decreases from Z i to Z o .
  • the width of the signal line 202 is a
  • the distance between the ground layers 203 on the surface is b
  • the distance between the signal line 202 and the ground layer 201 (thickness of the dielectric substrate 200) is h.
  • ⁇ 0 is the spatial impedance
  • ⁇ r is the dielectric constant of the dielectric substrate 200. If the distance (ba)/2 between the signal line 202 and the ground layer 203 becomes smaller, the characteristic impedance Z 0 becomes smaller according to equation (1). Therefore, the microstrip line according to this embodiment forms an impedance converter having a large characteristic impedance on the input side and a small characteristic impedance on the output side.
  • FIG. 4A, FIG. 4B, FIG. 5A, and FIG. 5B are diagrams showing models of impedance converters by the electromagnetic field simulator Sonnet (registered trademark)-EM.
  • 4A and 4B are perspective views of a model of a conventional impedance converter
  • FIGS. 5A and 5B are perspective views of a model of an impedance converter of this embodiment.
  • 100 is a dielectric substrate
  • 101 is a ground layer
  • 102 is a signal line.
  • the thickness a1 of the signal line 12 and the ground layer 14, the thickness a2 of the signal line 13 and the ground layer 15, and the thickness a of the signal line 102 are all 2 ⁇ m, and the characteristic impedance of the conventional and this embodiment is equal to 50 ⁇ . Ta.
  • the width W of the conventional signal line 102 was 6 ⁇ m, the interval G between the signal lines 102 was 6 ⁇ m, and the distance h between the signal line 102 and the ground layer 101 was 4 ⁇ m.
  • the widths w1 and w2 of the signal lines 12 and 13 are 6 ⁇ m, the distance g1 between the signal line 12 and the ground layer 14 is 3 ⁇ m, the distance g2 between the signal line 13 and the ground layer 15 is 3 ⁇ m, and the signal line 12 and the ground layer 15 are
  • the interval h1 (the interval between the signal line 13 and the ground layer 14) was set to 8 ⁇ m.
  • the number of conventional signal lines 102 and the total number of signal lines 12 and 13 in this embodiment are both five.
  • p1 is the input port of one of the two adjacent signal lines 102 of a conventional impedance converter
  • p2 is the output port of one of the signal lines 102.
  • Port p3 is an input port of the other signal line 102
  • port p4 is an output port of the other signal line 102.
  • p1 is the input port of the signal line 13
  • p2 is the output port of the signal line 13
  • p3 is an input port of the signal line 12
  • p4 is an output port of the signal line 12.
  • S31 is the voltage ratio between port p1 and port p3 when a signal is applied to port p1, and represents backward (near end) crosstalk.
  • S41 is the voltage ratio between port p1 and port p4, and represents forward (far end) crosstalk.
  • 6 and 7 are diagrams showing the simulation results of S31 and S41, respectively, and are expressed in decibels to make the differences easier to understand. 600 in FIG. 6 shows the backward crosstalk of the conventional impedance converter, and 601 shows the backward crosstalk of the impedance converter of this embodiment. Further, 700 in FIG. 7 shows the forward crosstalk of the conventional impedance converter, and 701 shows the forward crosstalk of the impedance converter of this embodiment.
  • the backward crosstalk of the impedance converter of this embodiment is smaller than the backward crosstalk of the conventional impedance converter, and is particularly smaller by about 20 dB over a wide range from 0 GHz to 100 GHz.
  • the forward crosstalk of the impedance converter of this embodiment is smaller than that of the conventional impedance converter, and is particularly smaller by about 25 to 60 dB over a wide range of 0 GHz to 100 GHz. I understand.
  • the characteristic impedance can be continuously changed while maintaining g3, and it is applicable to high-density packaging.
  • the distance h3 between the signal lines 12 and 13 by making the distance h3 between the signal lines 12 and 13 larger than the distance g3 between the center lines of the signal line 12 and the ground layer 14 (distance between the center lines of the signal line 13 and the ground layer 15), Crosstalk noise between the signal lines 12 and 13 can be reduced. Further, the distance g1 between the signal line 12 and the ground layer 14 and the distance h1 between the signal line 12 and the ground layer 15 (the distance between the signal line 13 and the ground layer 14) should be made smaller than the distance h3 between the signal lines 12 and 13. Accordingly, crosstalk noise can be further reduced.
  • the characteristic impedance on the output side of the impedance converter is made small, but it is also possible to form an impedance converter with a small characteristic impedance on the input side.
  • the right end may be set as the input side and the left end may be set as the output side in FIGS. 1A to 1C.
  • the tapered portions 140, 150 become gradually narrower from the input side to the output side.
  • FIGS. 1A to 1C and 2A to 2C the case where the total number of signal lines 12 and 13 provided in parallel is five is described. Since the ground layers 14 are provided on both sides of the signal line 12, the number of ground layers 14 is one more than the number of signal lines 12. Since the ground layers 15 are provided on both sides of the signal line 13, the number of ground layers 15 is one more than the number of signal lines 13. It goes without saying that the total number of signal lines 12 and 13 is not limited to five.
  • FIG. 8A is a plan view of an impedance converter according to a second embodiment of the present invention
  • FIG. 8B is a sectional view taken along line AA' of the impedance converter of FIG. 8A
  • FIG. 8C is a B-- It is a sectional view taken along the line B'
  • 9A is a sectional view taken along line CC' of the impedance converter in FIG. 8A
  • FIG. 9B is a sectional view taken along line DD' of the impedance converter in FIG. 8A
  • FIG. 9C is a sectional view taken along line EE' of the impedance converter in FIG. 8A.
  • FIG. 8A to 8C and 9A to 9C the same components as those in FIGS. 1A to 1C and FIGS. 2A to 2C are denoted by the same reference numerals.
  • the impedance converter of this embodiment has a microstrip line structure on the input side, and a structure in which grounded coplanar lines and microstrip lines are alternately arranged on the output side.
  • the first microstrip line 1 includes a dielectric substrate 10, a ground layer 11, and a signal line 12.
  • the second microstrip line 2 includes a dielectric substrate 10, a ground layer 11, and a signal line 13.
  • the first microstrip line 1 and the second microstrip line 2 are arranged alternately along the direction perpendicular to the signal propagation direction (the left-right direction in FIGS. 8A to 8C). It is located in
  • the grounded coplanar line 3a is connected to the dielectric substrate 10, the signal line 12, on the same layer as the signal line 12 (on the surface of the dielectric substrate 10), on both sides of the signal line 12, and above the signal line 13. It is composed of a ground layer 14 made of a conductor and a ground layer 11 formed on the back surface of the dielectric substrate 10.
  • the third microstrip line 5 includes a dielectric substrate 10, a signal line 13, and a ground layer 14.
  • the signal line 12 is formed on the surface of the dielectric substrate 10, which is the grounded coplanar line 3a
  • the signal line 13 is formed on the surface of the dielectric substrate 10, which is the third grounded coplanar line 3a.
  • Microstrip lines 5 are alternately arranged along a direction perpendicular to the signal propagation direction.
  • the ground layers 14 on both sides of the signal line 12 are formed to face the signal line 13 with a dielectric material in between.
  • the position of the center line of the ground layer 14 coincides with the position of the center line of the signal line 13 in the direction perpendicular to the signal propagation direction (horizontal direction in FIGS. 9A to 9C).
  • the ground layer 14 is connected to the ground layer 11 via a via 22. Similar to the first embodiment, the ground layer 14 has a tapered portion 140 whose width gradually increases along the signal propagation direction, and a constant width formed so as to be continuous with the widest portion of the tapered portion 140. It has a rectangular portion 141. By providing the ground layers 14 on both sides of the signal line 12, it is possible to realize a structure in which the distance g1 between the signal line 12 and the ground layer 14 gradually changes.
  • a second microstrip line 2 is configured by the dielectric substrate 10, the signal line 13, and the ground layer 11, and on the output side, the dielectric substrate 10, the signal line 13, and the ground layer 11 are configured.
  • 14 constitutes the third microstrip line 5.
  • the microstrip lines 2 and 5 form an impedance converter having a large characteristic impedance on the input side and a small characteristic impedance on the output side.
  • FIGS. 10A and 10B are diagrams showing a model of the impedance converter of this example by the electromagnetic field simulator Sonnet-EM. Models of conventional impedance converters are shown in FIGS. 4A and 4B.
  • the width W of the conventional signal line 102 was 6 ⁇ m
  • the interval G between the signal lines 102 was 6 ⁇ m
  • the distance h between the signal line 102 and the ground layer 101 was 4 ⁇ m.
  • the widths w1 and w2 of the signal lines 12 and 13 were 6 ⁇ m
  • the distance g1 between the signal line 12 and the ground layer 14 was 2.5 ⁇ m
  • the distance h1 between the signal line 13 and the ground layer 14 was 3 ⁇ m.
  • the number of conventional signal lines 102 and the total number of signal lines 12 and 13 in this embodiment are both five.
  • FIGS. 11 and 12 are diagrams showing the simulation results of S31 and S41, respectively, and are expressed in decibels to make the differences easier to understand.
  • 800 in FIG. 11 indicates backward crosstalk of the conventional impedance converter
  • 801 indicates backward crosstalk of the impedance converter of this embodiment.
  • 900 in FIG. 12 indicates forward crosstalk of the conventional impedance converter
  • 901 indicates forward crosstalk of the impedance converter of this embodiment.
  • the backward crosstalk of the impedance converter of this embodiment is smaller than the backward crosstalk of the conventional impedance converter, and is particularly smaller by about 0 to 8 dB over a wide range of 0 GHz to 100 GHz.
  • the forward crosstalk of the impedance converter of this embodiment is smaller than that of the conventional impedance converter, and is particularly smaller by about 0 to 15 dB over a wide range of 0 GHz to 100 GHz. I understand.
  • crosstalk noise between the signal lines 12 and 13 is reduced by making the distance h3 between the signal lines 12 and 13 larger than the distance g3 between the center lines of the signal line 12 and the ground layer 14. I can do it. Further, crosstalk noise can be further reduced by making the distance g1 between the signal line 12 and the ground layer 14 and the distance h1 between the signal line 13 and the ground layer 14 smaller than the distance h3 between the signal lines 12 and 13. can.
  • FIGS. 8A to 8C and FIGS. 9A to 9C the case where the total number of signal lines 12 and 13 provided in parallel is five has been described. Since the ground layers 14 are provided on both sides of the signal line 12, the number of ground layers 14 is one more than the number of signal lines 12. It goes without saying that the total number of signal lines 12 and 13 is not limited to five. Even in a configuration in which the first embodiment and this embodiment are arbitrarily combined, the essential effects of the present invention are not impaired.
  • the impedance converter of the present invention includes a first microstrip line and a second microstrip line alternately arranged with the first microstrip line along a direction intersecting the signal propagation direction. , a first grounded coplanar line connected to the first microstrip line and a grounded coplanar line alternately arranged with the first grounded coplanar line along a direction intersecting the signal propagation direction, and the first grounded coplanar line connected to the first microstrip line; a second grounded coplanar line connected to a second microstrip line, and the first microstrip line includes a dielectric substrate and a first ground layer formed on the back surface of the dielectric substrate.
  • the second microstrip line includes the dielectric substrate, the first ground layer, and the signal propagation direction. and second signal lines formed in the dielectric substrate so as to be arranged alternately with the first signal line along the intersecting direction, and the first grounded coplanar line includes: the dielectric substrate, the first signal line, the same layer as the first signal line, on both sides of the first signal line, and above the second signal line.
  • a second ground layer is formed on the same layer as the second signal line, and a second ground layer is formed on both sides of the second signal line so as to be located below the first signal line.
  • the second grounded coplanar line includes the dielectric substrate, the second signal line, and the third ground layer formed on both sides of the second signal line. It is composed of a ground layer and the second ground layer formed above the second signal line.
  • the second ground layer includes a first tapered portion whose width gradually changes along the signal propagation direction, and a thickest portion of the first tapered portion. a first rectangular part with a constant width formed so as to be continuous with the first rectangular part, and the third ground layer has a second tapered part with a width that gradually changes along the signal propagation direction; and a second rectangular portion having a constant width formed so as to be continuous with the thickest portion of the second tapered portion.
  • the impedance converter of the present invention includes a first microstrip line and a second microstrip line alternately arranged with the first microstrip line along a direction intersecting the signal propagation direction. , a grounded coplanar line connected to the first microstrip line, and a grounded coplanar line alternately arranged with the grounded coplanar line along a direction intersecting the signal propagation direction, and connected to the second microstrip line.
  • a third microstrip line connected thereto, the first microstrip line includes a dielectric substrate, a first ground layer formed on the back surface of the dielectric substrate, and a surface of the dielectric substrate.
  • the second microstrip line includes the dielectric substrate, the first ground layer, and the first signal line formed along the direction intersecting the signal propagation direction. and second signal lines formed in the dielectric substrate so as to be arranged alternately with the first signal line, and the grounded coplanar line is configured to connect the dielectric substrate and the first signal line.

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Abstract

A microstrip line (1) is composed of a dielectric board (10), a ground layer that is formed on the back surface of the board (10), and a signal line (12). A microstrip line (2) is composed of the board (10), the ground layer that is formed on the back surface of the board (10), and a signal line (13). A grounded coplanar line (3) is composed of the board (10), a line (12), ground layers (14) formed on both sides of the line (12), and ground layers (15) formed on both sides of the line (13). A grounded coplanar line (4) is composed of the board (10), the line (13), the ground layers (15) and the ground layers (14).

Description

インピーダンス変換器impedance converter
　本発明は、半導体高周波モジュールにおけるインピーダンス変換器に関するものである。 The present invention relates to an impedance converter in a semiconductor high frequency module.
　高周波用伝送線路基板上で高速信号を伝搬させるための代表的な構造として、マイクロストリップ線路、コプレナー線路、グランデッドコプレナー線路等が使用されている。例えば、マイクロストリップ線路は、誘電体基板の一方の面に平面的な導電体層のグランド面を形成し、誘電体基板の他方の面に帯状の線路を形成して伝送線路を構成している。これらの伝送線路の特性インピーダンスは、信号線路の幅と厚さ、誘電体基板の誘電率と厚さ、信号線路とグランドパターン間の距離によって決定される。 Microstrip lines, coplanar lines, grounded coplanar lines, etc. are used as typical structures for propagating high-speed signals on high-frequency transmission line substrates. For example, in a microstrip line, a transmission line is constructed by forming a ground plane of a planar conductive layer on one side of a dielectric substrate, and forming a strip-shaped line on the other side of the dielectric substrate. . The characteristic impedance of these transmission lines is determined by the width and thickness of the signal line, the dielectric constant and thickness of the dielectric substrate, and the distance between the signal line and the ground pattern.
　高周波回路に、例えば、ある一定のインピーダンスを有する負荷回路や信号源を接続する場合、接続部分で電力や信号を効率よく伝達させるために、高周波回路と負荷回路や信号源との特性インピーダンスを整合させる必要がある。このインピーダンス整合を行わせるため、線路の両端で特性インピーダンスが異なるように形成したインピーダンス変換器が用いられる（非特許文献１参照）。 For example, when connecting a load circuit or signal source with a certain impedance to a high-frequency circuit, the characteristic impedance of the high-frequency circuit and the load circuit or signal source must be matched in order to efficiently transmit power and signals at the connection part. It is necessary to do so. In order to perform this impedance matching, an impedance converter is used in which the characteristic impedance is different at both ends of the line (see Non-Patent Document 1).
　図１３Ａは従来のインピーダンス変換器の構造を示す平面図、図１３Ｂは図１３Ａのインピーダンス変換器のＡ－Ａ’線断面図、図１３Ｃは図１３Ａのインピーダンス変換器のＢ－Ｂ’線断面図である。伝送線路によるインピーダンス変換器は、高周波帯での急激なインピーダンス変化による伝送特性の劣化を防ぐため、図１３Ａ～図１３Ｃに示すように信号線路１０２の幅を徐々に変化させることにより、マイクロストリップ線路の特性インピーダンスを所望のインピーダンスに変換するようにしていた。図１３Ａ～図１３Ｃにおける１００は誘電体基板、１０１はグランド層である。 13A is a plan view showing the structure of a conventional impedance converter, FIG. 13B is a sectional view taken along line AA' of the impedance converter shown in FIG. 13A, and FIG. 13C is a sectional view taken along line BB' of the impedance converter shown in FIG. 13A. It is. An impedance converter using a transmission line is constructed using a microstrip line by gradually changing the width of the signal line 102 as shown in FIGS. The characteristic impedance of the impedance was converted into the desired impedance. In FIGS. 13A to 13C, 100 is a dielectric substrate, and 101 is a ground layer.
　近年、半導体高周波モジュールの信号数の増大、基板接続パッドの微細化が進んでいる。すなわち、半導体高周波モジュールの高機能化のために半導体高周波モジュールから入出力される信号が増加しているが、半導体高周波モジュールの高機能化、低コスト化のためにはモジュールの外形サイズを小さくする必要があるため、基板接続パッドとパッド間隔の微細化が進行している。その結果、半導体高周波モジュールと接続する配線基板において、高密度で多信号を引き回せる伝送線路や、伝送線路によって高周波特性を維持したままインピーダンス変換を行うインピーダンス変換器の実現が求められている。 In recent years, the number of signals in semiconductor high-frequency modules has increased and substrate connection pads have become smaller. In other words, in order to improve the functionality of semiconductor high-frequency modules, the number of signals input and output from semiconductor high-frequency modules is increasing, but in order to improve the functionality and reduce costs of semiconductor high-frequency modules, it is necessary to reduce the external size of the module. As a result, substrate connection pads and pad spacing are becoming increasingly finer. As a result, in wiring boards connected to semiconductor high-frequency modules, there is a need for transmission lines that can route multiple signals at high density, as well as impedance converters that perform impedance conversion while maintaining high-frequency characteristics using transmission lines.
　伝送線路によって高周波特性を維持したままインピーダンス変換を行う場合、従来技術では、線路幅をテーパー形状で徐々に変化させている。しかし、図１４Ａに示すように、信号線路１０２の間隔を十分に確保しようとすると、基板接続パッド１０３の間隔ｄ１も大きくなって、インピーダンス変換器のサイズが大きくなるという問題点があった。また、図１４Ｂに示すように、信号線路１０２の幅が大きくなって信号線路１０２の間隔ｄ２が小さくなると、信号線路１０２間のクロストークノイズが大きくなるという問題点があった。 When impedance conversion is performed using a transmission line while maintaining high frequency characteristics, in the conventional technology, the line width is gradually changed in a tapered shape. However, as shown in FIG. 14A, when attempting to ensure sufficient spacing between the signal lines 102, the spacing d1 between the substrate connection pads 103 also increases, resulting in a problem that the size of the impedance converter increases. Furthermore, as shown in FIG. 14B, when the width of the signal lines 102 increases and the interval d2 between the signal lines 102 decreases, there is a problem in that crosstalk noise between the signal lines 102 increases.
　信号線路１０２間のクロストークノイズは、一方の信号線路１０２によって信号パルスが伝送されたとき、他方の信号線路１０２の電子を変位させることにより生じるものである。このため、信号線路１０２の間隔が小さくなればなる程、他方の信号線路１０２の電子の変位量も大きくなり、クロストークノイズも大きくなっていく。以上のように、従来のインピーダンス変換器では、線路密度の向上と線路間のクロストークノイズの低減とを両立させることが難しく、高密度実装に適用することが困難であった。 Crosstalk noise between the signal lines 102 is caused by displacing electrons on the other signal line 102 when a signal pulse is transmitted by one signal line 102. Therefore, as the interval between the signal lines 102 becomes smaller, the amount of displacement of electrons on the other signal line 102 becomes larger, and the crosstalk noise also becomes larger. As described above, with conventional impedance converters, it is difficult to simultaneously improve line density and reduce crosstalk noise between lines, and it is difficult to apply them to high-density packaging.
　本発明は、上記課題を解決するためになされたもので、線路密度の向上と線路間のクロストークノイズの低減とを両立させることができるインピーダンス変換器を提供することを目的とする。 The present invention was made to solve the above problems, and an object of the present invention is to provide an impedance converter that can both improve line density and reduce crosstalk noise between lines.
　本発明のインピーダンス変換器は、第１のマイクロストリップ線路と、信号伝搬方向と交差する方向に沿って前記第１のマイクロストリップ線路と交互に配置された第２のマイクロストリップ線路と、前記第１のマイクロストリップ線路と接続された第１のグランデッドコプレナー線路と、前記信号伝搬方向と交差する方向に沿って前記第１のグランデッドコプレナー線路と交互に配置され、前記第２のマイクロストリップ線路と接続された第２のグランデッドコプレナー線路とを備え、前記第１のマイクロストリップ線路は、誘電体基板と、前記誘電体基板の裏面に形成された第１のグランド層と、前記誘電体基板の表面に形成された第１の信号線路とから構成され、前記第２のマイクロストリップ線路は、前記誘電体基板と、前記第１のグランド層と、前記信号伝搬方向と交差する方向に沿って前記第１の信号線路と交互に配置されるように前記誘電体基板中に形成された第２の信号線路とから構成され、前記第１のグランデッドコプレナー線路は、前記誘電体基板と、前記第１の信号線路と、前記第１の信号線路と同一層の、前記第１の信号線路の両側の位置に、前記第２の信号線路の上方に位置するように形成された第２のグランド層と、前記第２の信号線路と同一層の、前記第２の信号線路の両側の位置に、前記第１の信号線路の下方に位置するように形成された第３のグランド層とから構成され、前記第２のグランデッドコプレナー線路は、前記誘電体基板と、前記第２の信号線路と、前記第２の信号線路の両側に形成された前記第３のグランド層と、前記第２の信号線路の上方に位置するように形成された前記第２のグランド層とから構成されることを特徴とするものである。 The impedance converter of the present invention includes a first microstrip line, a second microstrip line alternately arranged with the first microstrip line along a direction intersecting a signal propagation direction, and a first a first grounded coplanar line connected to the microstrip line; and a second grounded coplanar line arranged alternately with the first grounded coplanar line along a direction intersecting the signal propagation direction, and the second microstrip line connected to the first grounded coplanar line. a second grounded coplanar line connected to the line, and the first microstrip line includes a dielectric substrate, a first ground layer formed on the back surface of the dielectric substrate, and a second grounded coplanar line connected to the dielectric line. and a first signal line formed on the surface of the dielectric substrate, and the second microstrip line connects the dielectric substrate, the first ground layer, and a signal line in a direction crossing the signal propagation direction. and second signal lines formed in the dielectric substrate so as to be arranged alternately with the first signal lines along the dielectric substrate, and the first grounded coplanar line is arranged along the dielectric substrate. and a second signal line formed on both sides of the first signal line, on the same layer as the first signal line, and above the second signal line. a third ground layer formed on both sides of the second signal line and below the first signal line, on the same layer as the second signal line; The second grounded coplanar line includes the dielectric substrate, the second signal line, and the third ground layer formed on both sides of the second signal line; and the second ground layer formed above the second signal line.
　本発明によれば、インピーダンス変換器の一方の側において第１のマイクロストリップ線路と第２のマイクロストリップ線路とを信号伝搬方向と交差する方向に沿って交互に配置し、インピーダンス変換器の他方の側において第１のグランデッドコプレナー線路と第２のグランデッドコプレナー線路とを信号伝搬方向と交差する方向に沿って交互に配置することにより、第１、第２の信号線路の幅や、第１の信号線路の中心線と第２のグランド層の中心線間の距離（第２の信号線路の中心線と第３のグランド層の中心線間の距離）を維持したまま、伝送線路の特性インピーダンスを調整することが可能になり、入力側の特性インピーダンスと出力側の特性インピーダンスとが異なるインピーダンス変換器を実現することができる。また、本発明では、線路間のクロストークノイズを低減することができる。その結果、本発明では、従来と同程度の量にクロストークノイズを抑えたまま、信号線路の間隔を微細化することができ、線路密度の向上と線路間のクロストークノイズの低減とを両立させることができるので、高密度実装に適用可能なインピーダンス変換器を実現することができる。 According to the present invention, the first microstrip line and the second microstrip line are arranged alternately along the direction intersecting the signal propagation direction on one side of the impedance converter, and the first microstrip line and the second microstrip line are arranged alternately along the direction crossing the signal propagation direction, and By alternately arranging the first grounded coplanar line and the second grounded coplanar line along the direction intersecting the signal propagation direction on the side, the width of the first and second signal lines can be adjusted. While maintaining the distance between the center line of the first signal line and the center line of the second ground layer (the distance between the center line of the second signal line and the center line of the third ground layer), It becomes possible to adjust the characteristic impedance, and it is possible to realize an impedance converter in which the characteristic impedance on the input side and the characteristic impedance on the output side are different. Further, in the present invention, crosstalk noise between lines can be reduced. As a result, in the present invention, it is possible to miniaturize the spacing between signal lines while suppressing crosstalk noise to the same level as in the past, thereby achieving both improvement in line density and reduction in crosstalk noise between lines. Therefore, an impedance converter applicable to high-density packaging can be realized.
図１Ａ－図１Ｃは、本発明のインピーダンス変換器の平面図および断面図である。1A-1C are a plan view and a cross-sectional view of an impedance converter of the present invention. 図２Ａ－図２Ｃは、本発明のインピーダンス変換器の断面図である。2A-2C are cross-sectional views of impedance converters of the present invention. 図３は、グランデットコプレナー線路の断面図である。FIG. 3 is a cross-sectional view of a granded coplanar line. 図４Ａ－図４Ｂは、従来のインピーダンス変換器のモデルを示す図である。4A and 4B are diagrams showing a model of a conventional impedance converter. 図５Ａ－図５Ｂは、本発明の第１の実施例に係るインピーダンス変換器のモデルを示す図である。5A and 5B are diagrams showing a model of an impedance converter according to a first embodiment of the present invention. 図６は、本発明の第１の実施例に係るインピーダンス変換器および従来のインピーダンス変換器のバックワード・クロストークのシミュレーション結果を示す図である。FIG. 6 is a diagram showing simulation results of backward crosstalk of the impedance converter according to the first embodiment of the present invention and the conventional impedance converter. 図７は、本発明の第１の実施例に係るインピーダンス変換器および従来のインピーダンス変換器のフォワード・クロストークのシミュレーション結果を示す図である。FIG. 7 is a diagram showing simulation results of forward crosstalk of the impedance converter according to the first embodiment of the present invention and the conventional impedance converter. 図８Ａ－図８Ｃは、本発明の第２の実施例に係るインピーダンス変換器の平面図および断面図である。8A to 8C are a plan view and a cross-sectional view of an impedance converter according to a second embodiment of the present invention. 図９Ａ－図９Ｃは、本発明の第２の実施例に係るインピーダンス変換器の断面図である。9A to 9C are cross-sectional views of an impedance converter according to a second embodiment of the present invention. 図１０Ａ－図１０Ｂは、本発明の第２の実施例に係るインピーダンス変換器のモデルを示す図である。10A and 10B are diagrams showing a model of an impedance converter according to a second embodiment of the present invention. 図１１は、本発明の第２の実施例に係るインピーダンス変換器および従来のインピーダンス変換器のバックワード・クロストークのシミュレーション結果を示す図である。FIG. 11 is a diagram showing simulation results of backward crosstalk of the impedance converter according to the second embodiment of the present invention and the conventional impedance converter. 図１２は、本発明の第２の実施例に係るインピーダンス変換器および従来のインピーダンス変換器のフォワード・クロストークのシミュレーション結果を示す図である。FIG. 12 is a diagram showing simulation results of forward crosstalk of the impedance converter according to the second embodiment of the present invention and the conventional impedance converter. 図１３Ａ－図１３Ｃは、従来のインピーダンス変換器の構造を示す平面図および断面図である。13A to 13C are a plan view and a cross-sectional view showing the structure of a conventional impedance converter. 図１４Ａ－図１４Ｂは、従来のインピーダンス変換器の問題点を説明する平面図である。14A and 14B are plan views illustrating problems with conventional impedance converters.
［発明の原理］
　図１Ａは本発明のインピーダンス変換器の平面図、図１Ｂは図１Ａのインピーダンス変換器のＡ－Ａ’線断面図、図１Ｃは図１Ａのインピーダンス変換器のＢ－Ｂ’線断面図である。図２Ａは図１Ａのインピーダンス変換器のＣ－Ｃ’線断面図、図２Ｂは図１Ａのインピーダンス変換器のＤ－Ｄ’線断面図、図２Ｃは図１Ａのインピーダンス変換器のＥ－Ｅ’線断面図である。 [Principle of the invention]
FIG. 1A is a plan view of the impedance converter of the present invention, FIG. 1B is a sectional view taken along line AA' of the impedance converter of FIG. 1A, and FIG. 1C is a sectional view taken along line BB' of the impedance converter of FIG. 1A. . 2A is a sectional view taken along line CC' of the impedance converter in FIG. 1A, FIG. 2B is a sectional view taken along line DD' of the impedance converter in FIG. 1A, and FIG. 2C is a sectional view taken along line EE' of the impedance converter in FIG. 1A. FIG.
　本発明のインピーダンス変換器は、入力側がマイクロストリップ線路の構造で、出力側がグランデッドコプレナー線路の構造をしている。第１のマイクロストリップ線路１は、誘電体基板１０と、誘電体基板１０の裏面に形成された導体からなるグランド層１１と、誘電体基板１０の表面に形成された帯状の導体からなる信号線路１２とから構成される。第２のマイクロストリップ線路２は、誘電体基板１０と、グランド層１１と、誘電体基板１０中に形成された帯状の導体からなる信号線路１３とから構成される。 The impedance converter of the present invention has a microstrip line structure on the input side and a grounded coplanar line structure on the output side. The first microstrip line 1 includes a dielectric substrate 10, a ground layer 11 made of a conductor formed on the back surface of the dielectric substrate 10, and a signal line made of a strip-shaped conductor formed on the surface of the dielectric substrate 10. It consists of 12. The second microstrip line 2 includes a dielectric substrate 10, a ground layer 11, and a signal line 13 made of a strip-shaped conductor formed in the dielectric substrate 10.
　このように、インピーダンス変換器の入力側では、信号線路１２が誘電体基板１０の表面に形成された第１のマイクロストリップ線路１と、信号線路１３が誘電体基板１０中に形成された第２のマイクロストリップ線路２とが、信号伝搬方向（図１Ａ～図１Ｃの左右方向）と直交する方向に沿って交互に配置されている。 As described above, on the input side of the impedance converter, the signal line 12 is formed on the surface of the dielectric substrate 10 in the first microstrip line 1, and the signal line 13 is formed in the dielectric substrate 10 in the second microstrip line 1. microstrip lines 2 are alternately arranged along a direction perpendicular to the signal propagation direction (left-right direction in FIGS. 1A to 1C).
　第１のグランデッドコプレナー線路３は、誘電体基板１０と、信号線路１２と、信号線路１２と同一層（誘電体基板１０の表面）の、信号線路１２の両側の位置に、信号線路１３の上方に位置するように形成された導体からなるグランド層１４と、信号線路１３と同一層（誘電体基板１０の内部）の、信号線路１３の両側の位置に、信号線路１２の下方に位置するように形成された導体からなるグランド層１５とから構成される。第２のグランデッドコプレナー線路４は、誘電体基板１０と、信号線路１３と、信号線路１３の両側に形成されたグランド層１５と、信号線路１３の上方に位置するように形成されたグランド層１４とから構成される。 The first grounded coplanar line 3 includes a dielectric substrate 10, a signal line 12, and a signal line 13 on both sides of the signal line 12 on the same layer as the signal line 12 (on the surface of the dielectric substrate 10). A ground layer 14 made of a conductor formed so as to be located above, a ground layer 14 formed on both sides of the signal line 13 on the same layer as the signal line 13 (inside the dielectric substrate 10), and a ground layer 14 formed below the signal line 12. and a ground layer 15 made of a conductor formed in such a manner. The second grounded coplanar line 4 includes a dielectric substrate 10, a signal line 13, a ground layer 15 formed on both sides of the signal line 13, and a ground layer 15 formed above the signal line 13. It is composed of a layer 14.
　このように、インピーダンス変換器の出力側では、信号線路１２が誘電体基板１０の表面に形成された第１のグランデッドコプレナー線路３と、信号線路１３が誘電体基板１０中に形成された第２のグランデッドコプレナー線路４とが、信号伝搬方向と直交する方向に沿って交互に配置されている。信号線路１２の両側のグランド層１４は、誘電体を間に挟んで信号線路１３と対向するように形成され、信号線路１３の両側のグランド層１５は、誘電体を間に挟んで信号線路１２と対向するように形成されている。信号伝搬方向と直交する方向（図２Ａ～図２Ｃの左右方向）において、グランド層１４の中心線の位置は信号線路１３の中心線の位置と一致し、グランド層１５の中心線の位置は信号線路１２の中心線の位置と一致する。 As described above, on the output side of the impedance converter, the signal line 12 is formed on the surface of the dielectric substrate 10 and the first grounded coplanar line 3 is formed, and the signal line 13 is formed in the dielectric substrate 10. Second grounded coplanar lines 4 are alternately arranged along the direction orthogonal to the signal propagation direction. The ground layers 14 on both sides of the signal line 12 are formed to face the signal line 13 with a dielectric in between, and the ground layers 15 on both sides of the signal line 13 are formed to face the signal line 12 with a dielectric in between. It is formed to face the In the direction perpendicular to the signal propagation direction (horizontal direction in FIGS. 2A to 2C), the position of the center line of the ground layer 14 coincides with the position of the center line of the signal line 13, and the position of the center line of the ground layer 15 coincides with the position of the center line of the signal line 13. This coincides with the position of the center line of the track 12.
　上記のとおり、信号線路１２と信号線路１３とは、信号伝搬方向と直交する方向に沿って交互に配置されている。信号線路１２の入力側の端部は基板接続パッド１６と接続され、信号線路１２の出力側の端部は基板接続パッド１７と接続されている。信号線路１３の入力側の端部はビア２０を介して基板接続パッド１８と接続され、信号線路１３の出力側の端部はビア２１を介して基板接続パッド１９と接続されている。 As described above, the signal lines 12 and the signal lines 13 are arranged alternately along the direction orthogonal to the signal propagation direction. The input side end of the signal line 12 is connected to the board connection pad 16, and the output side end of the signal line 12 is connected to the board connection pad 17. The input side end of the signal line 13 is connected to the substrate connection pad 18 via a via 20, and the output side end of the signal line 13 is connected to the substrate connection pad 19 via a via 21.
　信号線路１２の両側に形成されたグランド層１４は、ビア２２を介してグランド層１１と接続されている。信号線路１３の両側に形成されたグランド層１５は、ビア２３を介してグランド層１１と接続されている。また、グランド層１４とグランド層１５とは、ビア２４を介して互いに接続されている。 The ground layers 14 formed on both sides of the signal line 12 are connected to the ground layer 11 via vias 22. Ground layers 15 formed on both sides of the signal line 13 are connected to the ground layer 11 via vias 23. Further, the ground layer 14 and the ground layer 15 are connected to each other via a via 24.
　誘電体基板１０の誘電率はε１である。誘電体基板１０の厚さ方向の信号線路１２とグランド層１５の間隔、および信号線路１３とグランド層１４の間隔をｈ１とする。誘電体基板１０の厚さ方向の信号線路１３とグランド層１１の間隔、およびグランド層１４とグランド層１１の間隔をｈ２とする。信号線路１２の下面の長手方向の中心線から信号線路１３の上面の長手方向の中心線までの距離をｈ３とする。信号伝搬方向と直交する方向の信号線路１２とグランド層１４の間隔をｇ１、信号伝搬方向と直交する方向の信号線路１３とグランド層１５の間隔をｇ２とする。信号線路１２の長手方向の中心線とグランド層１４の長手方向の中心線間の距離、および信号線路１３の長手方向の中心線とグランド層１５の長手方向の中心線間の距離をｇ３とする。信号伝搬方向と直交する方向の信号線路１２の幅をｗ１、信号線路１３の幅をｗ２、グランド層１４の幅をｗｇ１、グランド層１５の幅をｗｇ２とする。信号線路１２とグランド層１４の厚さをａ１、信号線路１３とグランド層１５の厚さをａ２とする。 The dielectric constant of the dielectric substrate 10 is ε1. Let h1 be the distance between the signal line 12 and the ground layer 15 and the distance between the signal line 13 and the ground layer 14 in the thickness direction of the dielectric substrate 10. The distance between the signal line 13 and the ground layer 11 in the thickness direction of the dielectric substrate 10 and the distance between the ground layer 14 and the ground layer 11 are assumed to be h2. The distance from the longitudinal center line of the lower surface of the signal line 12 to the longitudinal center line of the upper surface of the signal line 13 is h3. The distance between the signal line 12 and the ground layer 14 in the direction perpendicular to the signal propagation direction is g1, and the distance between the signal line 13 and the ground layer 15 in the direction perpendicular to the signal propagation direction is g2. Let g3 be the distance between the longitudinal center line of the signal line 12 and the longitudinal center line of the ground layer 14, and the distance between the longitudinal center line of the signal line 13 and the longitudinal center line of the ground layer 15. . The width of the signal line 12 in the direction perpendicular to the signal propagation direction is w1, the width of the signal line 13 is w2, the width of the ground layer 14 is wg1, and the width of the ground layer 15 is wg2. The thickness of the signal line 12 and the ground layer 14 is a1, and the thickness of the signal line 13 and the ground layer 15 is a2.
　グランド層１４は、信号線路１２の信号伝搬方向に沿って幅が徐々に広くなるテーパー部１４０と、テーパー部１４０の幅の最太部と連なるように形成された幅が一定の矩形部１４１とを有する。本発明では、信号線路１２の両側にグランド層１４を設けることにより、信号線路１２とグランド層１４の間隔ｇ１が徐々に変化する構造を実現することができる。 The ground layer 14 includes a tapered portion 140 whose width gradually increases along the signal propagation direction of the signal line 12, and a rectangular portion 141 with a constant width formed so as to be continuous with the widest portion of the tapered portion 140. has. In the present invention, by providing the ground layers 14 on both sides of the signal line 12, it is possible to realize a structure in which the distance g1 between the signal line 12 and the ground layer 14 gradually changes.
　グランド層１５は、信号線路１３の信号伝搬方向に沿って幅が徐々に広くなるテーパー部１５０と、テーパー部１５０の幅の最太部と連なるように形成された幅が一定の矩形部１５１とを有する。本発明では、信号線路１３の両側にグランド層１５を設けることにより、信号線路１３とグランド層１５の間隔ｇ２が徐々に変化する構造を実現することができる。 The ground layer 15 includes a tapered portion 150 whose width gradually increases along the signal propagation direction of the signal line 13, and a rectangular portion 151 with a constant width formed so as to be continuous with the widest portion of the tapered portion 150. has. In the present invention, by providing the ground layers 15 on both sides of the signal line 13, it is possible to realize a structure in which the distance g2 between the signal line 13 and the ground layer 15 gradually changes.
　これにより、本発明では、高密度実装に適応可能であり、かつ信号線路１２，１３の幅ｗ１，ｗ２や、信号線路１２とグランド層１４の中心線間距離（信号線路１３とグランド層１５の中心線間距離）ｇ３を維持したまま、特性インピーダンスを連続的に変換することを可能にする。 As a result, the present invention is applicable to high-density packaging, and the widths w1 and w2 of the signal lines 12 and 13 and the distance between the center lines of the signal line 12 and the ground layer 14 (the distance between the signal line 13 and the ground layer 15) This makes it possible to continuously change the characteristic impedance while maintaining the centerline distance (distance between center lines) g3.
　伝送線路においては、信号線路とグランド層との距離が小さくなると、特性インピーダンスが小さくなる。従来の構成で特性インピーダンスを小さくするためには、信号線路幅を大きくする必要があり、高密度実装に適応することが困難であった。 In a transmission line, the smaller the distance between the signal line and the ground layer, the smaller the characteristic impedance. In order to reduce the characteristic impedance in the conventional configuration, it was necessary to increase the signal line width, making it difficult to adapt to high-density packaging.
　一方、本発明によれば、幅ｗ１，ｗ２や中心線間距離ｇ３を大きくすることなく、伝送線路の入力側の特性インピーダンスと出力側の特性インピーダンスが異なるインピーダンス変換器を実現することができる。 On the other hand, according to the present invention, it is possible to realize an impedance converter in which the characteristic impedance on the input side and the characteristic impedance on the output side of the transmission line are different, without increasing the widths w1, w2 or the center line distance g3.
　また、特性インピーダンスが等しい条件下で従来の構成と本発明を比較すると、従来の構成では信号線路とグランド層の中心線間距離が小さくなるのに対し、本発明では幅ｗ１，ｗ２や中心線間距離ｇ３を変えずに特性インイーダンスを調整することができ、信号線路１２，１３間のクロストークノイズを低減することができる。 Furthermore, when comparing the conventional configuration and the present invention under conditions of equal characteristic impedance, it is found that in the conventional configuration, the distance between the center lines of the signal line and the ground layer is small, whereas in the present invention, the distance between the widths w1, w2 and the center line The characteristic impedance can be adjusted without changing the distance g3, and crosstalk noise between the signal lines 12 and 13 can be reduced.
　さらに、インピーダンス変換器の出力側では、信号線路１２，１３とグランド層１４，１５の間隔が信号線路１２，１３間の距離より小さいことにより（ｇ１＜ｈ３、ｈ１＜ｈ３）、信号線路１２，１３とグランド層１４，１５との間に発生する電場が大きくなる。信号線路１２から発生する電場が、信号線路１２と同一層内の最も近くに存在するグランド層１４と、信号線路１２の直下に存在するグランド層１５とによって、これらグランド層１４，１５の存在する方向に偏向するため、信号線路１３の方向に伝わる電場が抑制される。同様に、信号線路１３から発生する電場がグランド層１４，１５の存在する方向に偏向するため、信号線路１２の方向に伝わる電場が抑制される。したがって、信号線路１２，１３間のクロストークノイズを低減することができる。このように、本発明では、クロストークノイズを低減する効果を、インピーダンス変換機能と同時に、また線路密度を低下させることなく得ることができる。 Furthermore, on the output side of the impedance converter, since the distance between the signal lines 12, 13 and the ground layers 14, 15 is smaller than the distance between the signal lines 12, 13 (g1<h3, h1<h3), the signal lines 12, 13, The electric field generated between 13 and ground layers 14 and 15 increases. The electric field generated from the signal line 12 is caused by the existence of these ground layers 14 and 15 due to the ground layer 14 that is closest to the signal line 12 and the ground layer 15 that is directly below the signal line 12. Since the electric field is deflected in the direction, the electric field propagating in the direction of the signal line 13 is suppressed. Similarly, since the electric field generated from the signal line 13 is deflected in the direction in which the ground layers 14 and 15 are present, the electric field transmitted in the direction of the signal line 12 is suppressed. Therefore, crosstalk noise between the signal lines 12 and 13 can be reduced. As described above, in the present invention, the effect of reducing crosstalk noise can be obtained simultaneously with the impedance conversion function and without reducing the line density.
　また、インピーダンス変換器の出力側では、信号線路１２と同一層内の最も近くに存在するグランド層１４と信号線路１２の直下に存在するグランド層１５とによって信号線路１２とグランド層１４，１５間の距離を調整し、信号線路１３と同一層内の最も近くに存在するグランド層１５と信号線路１３の直上に存在するグランド層１４とによって信号線路１３とグランド層１４，１５間の距離を調整することにより、通常のコプレナー線路あるいはマイクロストリップ線路よりも自在に線路の特性インピーダンスを設定することができる。 In addition, on the output side of the impedance converter, the signal line 12 is connected to the ground layers 14 and 15 by the ground layer 14 that is the closest to the signal line 12 and the ground layer 15 that is directly below the signal line 12. The distance between the signal line 13 and the ground layers 14 and 15 is adjusted by adjusting the distance between the signal line 13 and the ground layers 14 and 15, which is the closest ground layer 15 in the same layer as the signal line 13, and the ground layer 14 which is directly above the signal line 13. By doing so, the characteristic impedance of the line can be set more freely than in a normal coplanar line or microstrip line.
　インピーダンス変換器において、信号線路とグランド層との距離を変化させる技術としては、例えば特開２０１３－２５１８６３号公報に記載のものが知られている。しかし、特開２０１３－２５１８６３号公報に記載のインピーダンス変換器は、グランド層を傾斜させる三次元構造であるため、製造プロセスが現実には難しく、実用化が困難であった。
　本発明では、２次元的な構造の積層のみでインピーダンス変換を実現できることから、製造プロセスも簡単であり、低コストで実現が可能である。 As a technique for changing the distance between a signal line and a ground layer in an impedance converter, for example, the technique described in Japanese Patent Application Laid-open No. 2013-251863 is known. However, since the impedance converter described in JP-A-2013-251863 has a three-dimensional structure in which the ground layer is inclined, the manufacturing process is actually difficult, making it difficult to put it into practical use.
In the present invention, since impedance conversion can be realized only by laminating two-dimensional structures, the manufacturing process is simple and can be realized at low cost.
　したがって、本発明によれば、信号線路の幅を変化させずに特性インピーダンスを所望の値に設定可能であり、線路密度の向上と線路間のクロストークノイズの低減とを両立させることができるので、高密度実装に適用可能なインピーダンス変換器を実現することができる。 Therefore, according to the present invention, it is possible to set the characteristic impedance to a desired value without changing the width of the signal line, and it is possible to improve line density and reduce crosstalk noise between lines. , it is possible to realize an impedance converter applicable to high-density packaging.
［第１の実施例］
　次に、本発明の実施例について説明する。本実施例のインピーダンス変換器は、発明の原理で説明した構成の具体例なので、本実施例においても図１Ａ～図１Ｃ、図２Ａ～図２Ｃを用いて説明する。 [First example]
Next, examples of the present invention will be described. Since the impedance converter of this embodiment is a specific example of the configuration described in the principle of the invention, this embodiment will also be explained using FIGS. 1A to 1C and 2A to 2C.
　図１Ａ～図１Ｃ、図２Ａ～図２Ｃにおいて、誘電体基板１０は、ベンゾシクロブテン（ＢＣＢ）等の誘電体からなる。ＢＣＢの誘電率ε１は２．７である。誘電体基板１０の裏面には、Ａｕ等の導体からなるグランド層１１が形成されている。誘電体基板１０の表面にはＡｕ等の導体からなる帯状の信号線路１２が形成され、誘電体基板１０中にはＡｕ等の導体からなる帯状の信号線路１３が形成されている。インピーダンス変換器の出力側には、信号線路１２の両側にグランド層１４が形成され、信号線路１３の両側にグランド層１５が形成されている。 In FIGS. 1A to 1C and 2A to 2C, the dielectric substrate 10 is made of a dielectric material such as benzocyclobutene (BCB). The dielectric constant ε1 of BCB is 2.7. A ground layer 11 made of a conductor such as Au is formed on the back surface of the dielectric substrate 10. A band-shaped signal line 12 made of a conductor such as Au is formed on the surface of the dielectric substrate 10, and a band-shaped signal line 13 made of a conductor such as Au is formed in the dielectric substrate 10. On the output side of the impedance converter, ground layers 14 are formed on both sides of the signal line 12, and ground layers 15 are formed on both sides of the signal line 13.
　本実施例のインピーダンス変換器の一端（入力側）は入力インピーダンスＺｉを有し、他端（出力側）は出力インピーダンスＺｏを有するものとする（Ｚｉ＞Ｚｏ）。図１Ａ～図１Ｃの例では左端が入力側、右端が出力側となっている。信号線路１２と両側のグランド層１４間の距離、および信号線路１３と両側のグランド層１５間の距離は、入力側から出力側になるに従って徐々に小さくなっている。これにより、伝送線路の特性インピーダンスもＺｉからＺｏへと徐々に小さくなっていく。 It is assumed that one end (input side) of the impedance converter of this embodiment has an input impedance Z i and the other end (output side) has an output impedance Z o (Z i >Z o ). In the examples of FIGS. 1A to 1C, the left end is the input side and the right end is the output side. The distance between the signal line 12 and the ground layers 14 on both sides, and the distance between the signal line 13 and the ground layers 15 on both sides gradually become smaller from the input side to the output side. As a result, the characteristic impedance of the transmission line also gradually decreases from Z i to Z o .
　図３のようなグランデットコプレナー線路の場合、特性インピーダンスＺ０は下の式で求められる。 In the case of a granded coplanar line as shown in FIG. 3, the characteristic impedance Z 0 is obtained by the following formula.
Figure JPOXMLDOC01-appb-M000001
Figure JPOXMLDOC01-appb-M000001
　図３では、信号線路２０２の幅をａ、表面のグランド層２０３間の距離をｂ、信号線路２０２とグランド層２０１間の距離（誘電体基板２００の厚さ）をｈとしている。式（１）のη０は空間インピーダンス、εｒは誘電体基板２００の誘電率である。信号線路２０２とグランド層２０３間の距離（ｂ－ａ）／２が小さくなれば、式（１）より特性インピーダンスＺ０は小さくなる。したがって、本実施例によるマイクロストリップ線路は、入力側で特性インピーダンスが大きく、出力側で特性インピーダンスが小さくなるようなインピーダンス変換器を形成する。 In FIG. 3, the width of the signal line 202 is a, the distance between the ground layers 203 on the surface is b, and the distance between the signal line 202 and the ground layer 201 (thickness of the dielectric substrate 200) is h. In equation (1), η 0 is the spatial impedance, and ε r is the dielectric constant of the dielectric substrate 200. If the distance (ba)/2 between the signal line 202 and the ground layer 203 becomes smaller, the characteristic impedance Z 0 becomes smaller according to equation (1). Therefore, the microstrip line according to this embodiment forms an impedance converter having a large characteristic impedance on the input side and a small characteristic impedance on the output side.
　次に、従来のインピーダンス変換器と本実施例のインピーダンス変換器について、クロストーク量を比較してみる。図４Ａ、図４Ｂ、図５Ａ、図５Ｂは電磁界シミュレータＳｏｎｎｅｔ（登録商標）－ＥＭによるインピーダンス変換器のモデルを示す図である。図４Ａ、図４Ｂは従来のインピーダンス変換器のモデルの斜視図、図５Ａ、図５Ｂは本実施例のインピーダンス変換器のモデルの斜視図である。図４Ａ、図４Ｂにおける１００は誘電体基板、１０１はグランド層、１０２は信号線路である。 Next, the amount of crosstalk will be compared between the conventional impedance converter and the impedance converter of this embodiment. 4A, FIG. 4B, FIG. 5A, and FIG. 5B are diagrams showing models of impedance converters by the electromagnetic field simulator Sonnet (registered trademark)-EM. 4A and 4B are perspective views of a model of a conventional impedance converter, and FIGS. 5A and 5B are perspective views of a model of an impedance converter of this embodiment. 4A and 4B, 100 is a dielectric substrate, 101 is a ground layer, and 102 is a signal line.
　信号線路１２，１３，１０２とグランド層１４，１５，１０１の材料としてＡｕ（金）を使用し、誘電体基板１０，１００としてＢＣＢ基板（誘電率ε１＝２．７）を用いた。インピーダンス変換器の線路長は３００μｍとした。従来のインピーダンス変換器としては、コプレナー線路やマイクロストリップ線路があるが、ここではマイクロストリップ線路とする。シミュレーションでは、計算簡略化のため、インピーダンス変換器の出力側のみ計算している（ｈ３＞ｇ３）。また、特性インピーダンスを揃えて計算の簡略化を図るため、信号線路１２とグランド層１４の上にＢＣＢ層が形成されており、信号線路１３とグランド層１５のように周りをＢＣＢで覆われているものと仮定している。 Au (gold) was used as the material for the signal lines 12, 13, 102 and the ground layers 14, 15, 101, and a BCB substrate (dielectric constant ε1=2.7) was used as the dielectric substrate 10, 100. The line length of the impedance converter was 300 μm. Conventional impedance converters include coplanar lines and microstrip lines, but here the microstrip line is used. In the simulation, to simplify the calculation, only the output side of the impedance converter is calculated (h3>g3). In addition, in order to simplify the calculation by aligning the characteristic impedance, a BCB layer is formed on the signal line 12 and the ground layer 14, and the surroundings are covered with BCB like the signal line 13 and the ground layer 15. It is assumed that there is.
　クロストーク量を比較するため、信号線路１２とグランド層１４の中心線間距離（信号線路１３とグランド層１５の中心線間距離）ｇ３を１２μｍとし、従来の信号線路１０２の中心線間距離もｇ３＝１２μｍとする。また、信号線路１２とグランド層１４の厚さａ１、信号線路１３とグランド層１５の厚さａ２、信号線路１０２の厚さａを全て２μｍとし、従来と本実施例の特性インピーダンスを５０Ωに揃えた。 In order to compare the amount of crosstalk, the distance between the center lines of the signal line 12 and the ground layer 14 (the distance between the center lines of the signal line 13 and the ground layer 15) g3 is set to 12 μm, and the distance between the center lines of the conventional signal line 102 is also Let g3=12 μm. In addition, the thickness a1 of the signal line 12 and the ground layer 14, the thickness a2 of the signal line 13 and the ground layer 15, and the thickness a of the signal line 102 are all 2 μm, and the characteristic impedance of the conventional and this embodiment is equal to 50Ω. Ta.
　従来の信号線路１０２の幅Ｗを６μｍ、信号線路１０２の間隔Ｇを６μｍ、信号線路１０２とグランド層１０１間の距離ｈを４μｍとした。本実施例の信号線路１２，１３の幅ｗ１，ｗ２を６μｍ、信号線路１２とグランド層１４の間隔ｇ１を３μｍ、信号線路１３とグランド層１５の間隔ｇ２を３μｍ、信号線路１２とグランド層１５の間隔（信号線路１３とグランド層１４の間隔）ｈ１を８μｍとした。計算簡略化のため、従来の信号線路１０２の数、および本実施例の信号線路１２と１３の総数を共に５本としている。 The width W of the conventional signal line 102 was 6 μm, the interval G between the signal lines 102 was 6 μm, and the distance h between the signal line 102 and the ground layer 101 was 4 μm. In this embodiment, the widths w1 and w2 of the signal lines 12 and 13 are 6 μm, the distance g1 between the signal line 12 and the ground layer 14 is 3 μm, the distance g2 between the signal line 13 and the ground layer 15 is 3 μm, and the signal line 12 and the ground layer 15 are The interval h1 (the interval between the signal line 13 and the ground layer 14) was set to 8 μm. To simplify calculations, the number of conventional signal lines 102 and the total number of signal lines 12 and 13 in this embodiment are both five.
　図４Ａ、図４Ｂ、図５Ａ、図５Ｂのようにポート番号を設定したとき、Ｓパラメータの結果を調べることで、クロストーク量を直接評価できる。ｐ１は従来のインピーダンス変換器の隣接する２本の信号線路１０２のうち、一方の信号線路１０２の入力ポート、ｐ２は一方の信号線路１０２の出力ポートである。ポートｐ３は他方の信号線路１０２の入力ポート、ポートｐ４は他方の信号線路１０２の出力ポートである。本実施例の場合、ｐ１は信号線路１３の入力ポート、ｐ２は信号線路１３の出力ポートである。ｐ３は信号線路１２の入力ポート、ｐ４は信号線路１２の出力ポートである。 When port numbers are set as shown in FIGS. 4A, 4B, 5A, and 5B, the amount of crosstalk can be directly evaluated by examining the S-parameter results. p1 is the input port of one of the two adjacent signal lines 102 of a conventional impedance converter, and p2 is the output port of one of the signal lines 102. Port p3 is an input port of the other signal line 102, and port p4 is an output port of the other signal line 102. In this embodiment, p1 is the input port of the signal line 13, and p2 is the output port of the signal line 13. p3 is an input port of the signal line 12, and p4 is an output port of the signal line 12.
　Ｓ３１は、ポートｐ１に信号を与えたときのポートｐ１とポートｐ３の電圧比であり、バックワード（近端）・クロストークを表す。また、Ｓ４１は、ポートｐ１とポートｐ４の電圧比であり、フォワード（遠端）・クロストークを表す。図６、図７はそれぞれＳ３１、Ｓ４１のシミュレーション結果を示す図であり、差異を分かり易くするため、デシベル表示にしている。図６の６００は従来のインピーダンス変換器のバックワード・クロストークを示し、６０１は本実施例のインピーダンス変換器のバックワード・クロストークを示している。また、図７の７００は従来のインピーダンス変換器のフォワード・クロストークを示し、７０１は本実施例のインピーダンス変換器のフォワード・クロストークを示している。 S31 is the voltage ratio between port p1 and port p3 when a signal is applied to port p1, and represents backward (near end) crosstalk. Further, S41 is the voltage ratio between port p1 and port p4, and represents forward (far end) crosstalk. 6 and 7 are diagrams showing the simulation results of S31 and S41, respectively, and are expressed in decibels to make the differences easier to understand. 600 in FIG. 6 shows the backward crosstalk of the conventional impedance converter, and 601 shows the backward crosstalk of the impedance converter of this embodiment. Further, 700 in FIG. 7 shows the forward crosstalk of the conventional impedance converter, and 701 shows the forward crosstalk of the impedance converter of this embodiment.
　図６によれば、本実施例のインピーダンス変換器のバックワード・クロストークは、従来のインピーダンス変換器のバックワード・クロストークよりも小さく、特に０ＧＨｚ～１００ＧＨｚの広範囲において２０ｄＢ程度小さいことが分かる。また、図７によれば、本実施例のインピーダンス変換器のフォワード・クロストークは、従来のインピーダンス変換器のフォワード・クロストークよりも小さく、特に０ＧＨｚ～１００ＧＨｚの広範囲において２５～６０ｄＢ程度小さいことが分かる。 According to FIG. 6, it can be seen that the backward crosstalk of the impedance converter of this embodiment is smaller than the backward crosstalk of the conventional impedance converter, and is particularly smaller by about 20 dB over a wide range from 0 GHz to 100 GHz. Furthermore, according to FIG. 7, the forward crosstalk of the impedance converter of this embodiment is smaller than that of the conventional impedance converter, and is particularly smaller by about 25 to 60 dB over a wide range of 0 GHz to 100 GHz. I understand.
　以上のように、本実施例によれば、信号線路１２，１３の幅ｗ１，ｗ２や、信号線路１２とグランド層１４の中心線間距離（信号線路１３とグランド層１５の中心線間距離）ｇ３を維持したまま、特性インピーダンスを連続的に変換することができ、高密度実装に適応可能である。 As described above, according to this embodiment, the widths w1 and w2 of the signal lines 12 and 13, the distance between the center lines of the signal line 12 and the ground layer 14 (the distance between the center lines of the signal line 13 and the ground layer 15) The characteristic impedance can be continuously changed while maintaining g3, and it is applicable to high-density packaging.
　本実施例では、信号線路１２，１３間の距離ｈ３を、信号線路１２とグランド層１４の中心線間距離（信号線路１３とグランド層１５の中心線間距離）ｇ３よりも大きくすることにより、信号線路１２，１３間のクロストークノイズを低減することができる。また、信号線路１２とグランド層１４の間隔ｇ１および信号線路１２とグランド層１５の間隔（信号線路１３とグランド層１４の間隔）ｈ１を、信号線路１２，１３間の距離ｈ３よりも小さくすることにより、クロストークノイズをさらに低減することができる。 In this embodiment, by making the distance h3 between the signal lines 12 and 13 larger than the distance g3 between the center lines of the signal line 12 and the ground layer 14 (distance between the center lines of the signal line 13 and the ground layer 15), Crosstalk noise between the signal lines 12 and 13 can be reduced. Further, the distance g1 between the signal line 12 and the ground layer 14 and the distance h1 between the signal line 12 and the ground layer 15 (the distance between the signal line 13 and the ground layer 14) should be made smaller than the distance h3 between the signal lines 12 and 13. Accordingly, crosstalk noise can be further reduced.
　また、本実施例では、信号線路１２と同一層内の最も近くに存在するグランド層１４と信号線路１２の直下に存在するグランド層１５とによって信号線路１２とグランド層１４，１５間の距離を調整し、信号線路１３と同一層内の最も近くに存在するグランド層１５と信号線路１３の直上に存在するグランド層１４とによって信号線路１３とグランド層１４，１５間の距離を調整することにより、通常のコプレナー線路あるいはマイクロストリップ線路よりも自在に線路の特性インピーダンスを設定することができる。 In addition, in this embodiment, the distance between the signal line 12 and the ground layers 14 and 15 is determined by the ground layer 14 that is closest to the signal line 12 in the same layer and the ground layer 15 that is directly below the signal line 12. By adjusting the distance between the signal line 13 and the ground layers 14 and 15 by adjusting the ground layer 15 that is closest to the signal line 13 in the same layer and the ground layer 14 that is directly above the signal line 13. , the characteristic impedance of the line can be set more freely than with normal coplanar lines or microstrip lines.
　したがって、本実施例によれば、線路密度の向上と線路間のクロストークノイズの低減とを両立させることができるので、高密度実装に適用可能なインピーダンス変換器を実現することができる。 Therefore, according to this embodiment, it is possible to both improve line density and reduce crosstalk noise between lines, and thus it is possible to realize an impedance converter that is applicable to high-density packaging.
　なお、本実施例では、インピーダンス変換器の出力側の特性インピーダンスを小さくしているが、入力側の特性インピーダンスを小さくしたインピーダンス変換器を形成することもできる。入力側の特性インピーダンスを小さくするには、図１Ａ～図１Ｃにおいて右端を入力側、左端を出力側とすればよい。この場合、テーパー部１４０，１５０は、入力側から出力側に向かって徐々に狭くなる。 Note that in this embodiment, the characteristic impedance on the output side of the impedance converter is made small, but it is also possible to form an impedance converter with a small characteristic impedance on the input side. In order to reduce the characteristic impedance on the input side, the right end may be set as the input side and the left end may be set as the output side in FIGS. 1A to 1C. In this case, the tapered portions 140, 150 become gradually narrower from the input side to the output side.
　また、図１Ａ～図１Ｃ、図２Ａ～図２Ｃでは、平行に設ける信号線路１２，１３の総数が５本の場合について説明した。グランド層１４は信号線路１２の両側に設けられるので、グランド層１４の本数は信号線路１２の本数よりも１本多くなる。グランド層１５は信号線路１３の両側に設けられるので、グランド層１５の本数は信号線路１３の本数よりも１本多くなる。信号線路１２，１３の総数は５本に限るものではないことは言うまでもない。 Further, in FIGS. 1A to 1C and 2A to 2C, the case where the total number of signal lines 12 and 13 provided in parallel is five is described. Since the ground layers 14 are provided on both sides of the signal line 12, the number of ground layers 14 is one more than the number of signal lines 12. Since the ground layers 15 are provided on both sides of the signal line 13, the number of ground layers 15 is one more than the number of signal lines 13. It goes without saying that the total number of signal lines 12 and 13 is not limited to five.
［第２の実施例］
　次に、本発明の第２の実施例について説明する。図８Ａは本発明の第２の実施例に係るインピーダンス変換器の平面図、図８Ｂは図８Ａのインピーダンス変換器のＡ－Ａ’線断面図、図８Ｃは図８Ａのインピーダンス変換器のＢ－Ｂ’線断面図である。図９Ａは図８Ａのインピーダンス変換器のＣ－Ｃ’線断面図、図９Ｂは図８Ａのインピーダンス変換器のＤ－Ｄ’線断面図、図９Ｃは図８Ａのインピーダンス変換器のＥ－Ｅ’線断面図である。図８Ａ～図８Ｃ、図９Ａ～図９Ｃにおいて、図１Ａ～図１Ｃ、図２Ａ～図２Ｃと同一の構成には同一の符号を付してある。 [Second example]
Next, a second embodiment of the present invention will be described. 8A is a plan view of an impedance converter according to a second embodiment of the present invention, FIG. 8B is a sectional view taken along line AA' of the impedance converter of FIG. 8A, and FIG. 8C is a B-- It is a sectional view taken along the line B'. 9A is a sectional view taken along line CC' of the impedance converter in FIG. 8A, FIG. 9B is a sectional view taken along line DD' of the impedance converter in FIG. 8A, and FIG. 9C is a sectional view taken along line EE' of the impedance converter in FIG. 8A. FIG. In FIGS. 8A to 8C and 9A to 9C, the same components as those in FIGS. 1A to 1C and FIGS. 2A to 2C are denoted by the same reference numerals.
　本実施例のインピーダンス変換器は、入力側がマイクロストリップ線路の構造で、出力側がグランデッドコプレナー線路とマイクロストリップ線路を交互に配置した構造をしている。第１の実施例と同様に、第１のマイクロストリップ線路１は、誘電体基板１０と、グランド層１１と、信号線路１２とから構成される。第２のマイクロストリップ線路２は、誘電体基板１０と、グランド層１１と、信号線路１３とから構成される。このように、インピーダンス変換器の入力側では、第１のマイクロストリップ線路１と第２のマイクロストリップ線路２が、信号伝搬方向（図８Ａ～図８Ｃの左右方向）と直交する方向に沿って交互に配置されている。 The impedance converter of this embodiment has a microstrip line structure on the input side, and a structure in which grounded coplanar lines and microstrip lines are alternately arranged on the output side. Similar to the first embodiment, the first microstrip line 1 includes a dielectric substrate 10, a ground layer 11, and a signal line 12. The second microstrip line 2 includes a dielectric substrate 10, a ground layer 11, and a signal line 13. In this way, on the input side of the impedance converter, the first microstrip line 1 and the second microstrip line 2 are arranged alternately along the direction perpendicular to the signal propagation direction (the left-right direction in FIGS. 8A to 8C). It is located in
　グランデッドコプレナー線路３ａは、誘電体基板１０と、信号線路１２と、信号線路１２と同一層（誘電体基板１０の表面）の、信号線路１２の両側の位置に、信号線路１３の上方に位置するように形成された導体からなるグランド層１４と、誘電体基板１０の裏面に形成されたグランド層１１とから構成される。第３のマイクロストリップ線路５は、誘電体基板１０と、信号線路１３と、グランド層１４とから構成される。 The grounded coplanar line 3a is connected to the dielectric substrate 10, the signal line 12, on the same layer as the signal line 12 (on the surface of the dielectric substrate 10), on both sides of the signal line 12, and above the signal line 13. It is composed of a ground layer 14 made of a conductor and a ground layer 11 formed on the back surface of the dielectric substrate 10. The third microstrip line 5 includes a dielectric substrate 10, a signal line 13, and a ground layer 14.
　このように、インピーダンス変換器の出力側では、信号線路１２が誘電体基板１０の表面に形成されたグランデッドコプレナー線路３ａと、信号線路１３が誘電体基板１０中に形成された第３のマイクロストリップ線路５とが、信号伝搬方向と直交する方向に沿って交互に配置されている。信号線路１２の両側のグランド層１４は、誘電体を間に挟んで信号線路１３と対向するように形成されている。信号伝搬方向と直交する方向（図９Ａ～図９Ｃの左右方向）において、グランド層１４の中心線の位置は信号線路１３の中心線の位置と一致する。 In this way, on the output side of the impedance converter, the signal line 12 is formed on the surface of the dielectric substrate 10, which is the grounded coplanar line 3a, and the signal line 13 is formed on the surface of the dielectric substrate 10, which is the third grounded coplanar line 3a. Microstrip lines 5 are alternately arranged along a direction perpendicular to the signal propagation direction. The ground layers 14 on both sides of the signal line 12 are formed to face the signal line 13 with a dielectric material in between. The position of the center line of the ground layer 14 coincides with the position of the center line of the signal line 13 in the direction perpendicular to the signal propagation direction (horizontal direction in FIGS. 9A to 9C).
　グランド層１４は、ビア２２を介してグランド層１１と接続されている。第１の実施例と同様に、グランド層１４は、信号伝搬方向に沿って幅が徐々に広くなるテーパー部１４０と、テーパー部１４０の幅の最太部と連なるように形成された幅が一定の矩形部１４１とを有する。信号線路１２の両側にグランド層１４を設けることにより、信号線路１２とグランド層１４の間隔ｇ１が徐々に変化する構造を実現することができる。 The ground layer 14 is connected to the ground layer 11 via a via 22. Similar to the first embodiment, the ground layer 14 has a tapered portion 140 whose width gradually increases along the signal propagation direction, and a constant width formed so as to be continuous with the widest portion of the tapered portion 140. It has a rectangular portion 141. By providing the ground layers 14 on both sides of the signal line 12, it is possible to realize a structure in which the distance g1 between the signal line 12 and the ground layer 14 gradually changes.
　また、インピーダンス変換器の入力側では、誘電体基板１０と信号線路１３とグランド層１１とによって第２のマイクロストリップ線路２が構成され、出力側では、誘電体基板１０と信号線路１３とグランド層１４とによって第３のマイクロストリップ線路５が構成される。マイクロストリップ線路においては、信号線路とグランド層との距離が小さくなると、特性インピーダンスが小さくなる。したがって、本実施例によるマイクロストリップ線路２，５は、入力側で特性インピーダンスが大きく、出力側で特性インピーダンスが小さくなるようなインピーダンス変換器を形成する。 Further, on the input side of the impedance converter, a second microstrip line 2 is configured by the dielectric substrate 10, the signal line 13, and the ground layer 11, and on the output side, the dielectric substrate 10, the signal line 13, and the ground layer 11 are configured. 14 constitutes the third microstrip line 5. In a microstrip line, the smaller the distance between the signal line and the ground layer, the smaller the characteristic impedance. Therefore, the microstrip lines 2 and 5 according to this embodiment form an impedance converter having a large characteristic impedance on the input side and a small characteristic impedance on the output side.
　次に、従来のインピーダンス変換器と本実施例のインピーダンス変換器について、クロストーク量を比較してみる。図１０Ａ、図１０Ｂは電磁界シミュレータＳｏｎｎｅｔ－ＥＭによる本実施例のインピーダンス変換器のモデルを示す図である。従来のインピーダンス変換器のモデルは図４Ａ、図４Ｂに示したとおりである。 Next, the amount of crosstalk will be compared between the conventional impedance converter and the impedance converter of this embodiment. FIGS. 10A and 10B are diagrams showing a model of the impedance converter of this example by the electromagnetic field simulator Sonnet-EM. Models of conventional impedance converters are shown in FIGS. 4A and 4B.
　信号線路１２，１３，１０２とグランド層１４，１０１の材料としてＡｕ（金）を使用し、誘電体基板１０，１００としてＢＣＢ基板（誘電率ε１＝２．７）を用いた。インピーダンス変換器の線路長は３００μｍとした。シミュレーションでは、計算簡略化のため、インピーダンス変換器の出力側のみ計算している（ｈ３＞ｇ３）。また、特性インピーダンスを揃えて計算の簡略化を図るため、信号線路１２とグランド層１４の上にＢＣＢ層が形成されており、信号線路１３のように周りをＢＣＢで覆われているものと仮定している。 Au (gold) was used as the material for the signal lines 12, 13, 102 and the ground layers 14, 101, and a BCB substrate (dielectric constant ε1=2.7) was used as the dielectric substrate 10, 100. The line length of the impedance converter was 300 μm. In the simulation, to simplify the calculation, only the output side of the impedance converter is calculated (h3>g3). In addition, in order to simplify the calculation by aligning the characteristic impedance, it is assumed that a BCB layer is formed on the signal line 12 and the ground layer 14, and that the surrounding area is covered with BCB like the signal line 13. are doing.
　クロストーク量を比較するため、信号線路１２とグランド層１４の中心線間距離ｇ３を１２μｍとし、従来の信号線路１０２の中心線間距離もｇ３＝１２μｍとする。また、信号線路１２とグランド層１４の厚さａ１、信号線路１３の厚さａ２、信号線路１０２の厚さａを全て２μｍとし、従来と本実施例の特性インピーダンスを５０Ωに揃えた。 In order to compare the amount of crosstalk, the distance g3 between the center lines of the signal line 12 and the ground layer 14 is set to 12 μm, and the distance between the center lines of the conventional signal line 102 is also set to g3=12 μm. Further, the thickness a1 of the signal line 12 and the ground layer 14, the thickness a2 of the signal line 13, and the thickness a of the signal line 102 are all 2 μm, and the characteristic impedance of the conventional and this embodiment is made equal to 50Ω.
　従来の信号線路１０２の幅Ｗを６μｍ、信号線路１０２の間隔Ｇを６μｍ、信号線路１０２とグランド層１０１間の距離ｈを４μｍとした。本実施例の信号線路１２，１３の幅ｗ１，ｗ２を６μｍ、信号線路１２とグランド層１４の間隔ｇ１を２．５μｍ、信号線路１３とグランド層１４の間隔ｈ１を３μｍとした。計算簡略化のため、従来の信号線路１０２の数、および本実施例の信号線路１２と１３の総数を共に５本としている。 The width W of the conventional signal line 102 was 6 μm, the interval G between the signal lines 102 was 6 μm, and the distance h between the signal line 102 and the ground layer 101 was 4 μm. In this embodiment, the widths w1 and w2 of the signal lines 12 and 13 were 6 μm, the distance g1 between the signal line 12 and the ground layer 14 was 2.5 μm, and the distance h1 between the signal line 13 and the ground layer 14 was 3 μm. To simplify calculations, the number of conventional signal lines 102 and the total number of signal lines 12 and 13 in this embodiment are both five.
　図４Ａ、図４Ｂ、図１０Ａ、図１０Ｂのようにポート番号を設定したとき、Ｓパラメータの結果を調べることで、クロストーク量を直接評価できる。ｐ１は従来のインピーダンス変換器の隣接する２本の信号線路１０２のうち、一方の信号線路１０２の入力ポート、ｐ２は一方の信号線路１０２の出力ポートである。ポートｐ３は他方の信号線路１０２の入力ポート、ポートｐ４は他方の信号線路１０２の出力ポートである。本実施例の場合、ｐ１は信号線路１２の入力ポート、ｐ２は信号線路１２の出力ポートである。ｐ３は信号線路１３の入力ポート、ｐ４は信号線路１３の出力ポートである。 When port numbers are set as shown in FIGS. 4A, 4B, 10A, and 10B, the amount of crosstalk can be directly evaluated by examining the S-parameter results. p1 is the input port of one of the two adjacent signal lines 102 of a conventional impedance converter, and p2 is the output port of one of the signal lines 102. Port p3 is an input port of the other signal line 102, and port p4 is an output port of the other signal line 102. In this embodiment, p1 is the input port of the signal line 12, and p2 is the output port of the signal line 12. p3 is an input port of the signal line 13, and p4 is an output port of the signal line 13.
　図１１、図１２はそれぞれＳ３１、Ｓ４１のシミュレーション結果を示す図であり、差異を分かり易くするため、デシベル表示にしている。図１１の８００は従来のインピーダンス変換器のバックワード・クロストークを示し、８０１は本実施例のインピーダンス変換器のバックワード・クロストークを示している。また、図１２の９００は従来のインピーダンス変換器のフォワード・クロストークを示し、９０１は本実施例のインピーダンス変換器のフォワード・クロストークを示している。 FIGS. 11 and 12 are diagrams showing the simulation results of S31 and S41, respectively, and are expressed in decibels to make the differences easier to understand. 800 in FIG. 11 indicates backward crosstalk of the conventional impedance converter, and 801 indicates backward crosstalk of the impedance converter of this embodiment. Further, 900 in FIG. 12 indicates forward crosstalk of the conventional impedance converter, and 901 indicates forward crosstalk of the impedance converter of this embodiment.
　図１１によれば、本実施例のインピーダンス変換器のバックワード・クロストークは、従来のインピーダンス変換器のバックワード・クロストークよりも小さく、特に０ＧＨｚ～１００ＧＨｚの広範囲において０～８ｄＢ程度小さいことが分かる。また、図１２によれば、本実施例のインピーダンス変換器のフォワード・クロストークは、従来のインピーダンス変換器のフォワード・クロストークよりも小さく、特に０ＧＨｚ～１００ＧＨｚの広範囲において０～１５ｄＢ程度小さいことが分かる。 According to FIG. 11, the backward crosstalk of the impedance converter of this embodiment is smaller than the backward crosstalk of the conventional impedance converter, and is particularly smaller by about 0 to 8 dB over a wide range of 0 GHz to 100 GHz. I understand. Furthermore, according to FIG. 12, the forward crosstalk of the impedance converter of this embodiment is smaller than that of the conventional impedance converter, and is particularly smaller by about 0 to 15 dB over a wide range of 0 GHz to 100 GHz. I understand.
　以上のように、本実施例によれば、信号線路１２，１３の幅ｗ１，ｗ２や、信号線路１２とグランド層１４の中心線間距離ｇ３を維持したまま、特性インピーダンスを連続的に変換することができ、高密度実装に適応可能である。 As described above, according to this embodiment, the characteristic impedance is continuously changed while maintaining the widths w1 and w2 of the signal lines 12 and 13 and the distance g3 between the center lines of the signal line 12 and the ground layer 14. It is possible to adapt to high-density packaging.
　本実施例では、信号線路１２，１３間の距離ｈ３を、信号線路１２とグランド層１４の中心線間距離ｇ３よりも大きくすることにより、信号線路１２，１３間のクロストークノイズを低減することができる。また、信号線路１２とグランド層１４の間隔ｇ１および信号線路１３とグランド層１４の間隔ｈ１を、信号線路１２，１３間の距離ｈ３よりも小さくすることにより、クロストークノイズをさらに低減することができる。 In this embodiment, crosstalk noise between the signal lines 12 and 13 is reduced by making the distance h3 between the signal lines 12 and 13 larger than the distance g3 between the center lines of the signal line 12 and the ground layer 14. I can do it. Further, crosstalk noise can be further reduced by making the distance g1 between the signal line 12 and the ground layer 14 and the distance h1 between the signal line 13 and the ground layer 14 smaller than the distance h3 between the signal lines 12 and 13. can.
　また、本実施例では、信号線路１２と同一層内の最も近くに存在するグランド層１４によって信号線路１２とグランド層１４間の距離を調整し、信号線路１３の直上に存在するグランド層１４によって信号線路１３とグランド層１４間の距離を調整することにより、通常のコプレナー線路あるいはマイクロストリップ線路よりも自在に線路の特性インピーダンスを設定することができる。さらに、インピーダンス変換器の出力側ではマイクロストリップ線路とコプレナー線路を高密度に交互に配置することができる。 In addition, in this embodiment, the distance between the signal line 12 and the ground layer 14 is adjusted by the ground layer 14 existing closest to the signal line 12 in the same layer, and the distance between the signal line 12 and the ground layer 14 existing directly above the signal line 13 is adjusted. By adjusting the distance between the signal line 13 and the ground layer 14, the characteristic impedance of the line can be set more freely than in a normal coplanar line or microstrip line. Furthermore, microstrip lines and coplanar lines can be arranged alternately with high density on the output side of the impedance converter.
　したがって、本実施例によれば、線路密度の向上と線路間のクロストークノイズの低減とを両立させることができるので、高密度実装に適用可能なインピーダンス変換器を実現することができる。 Therefore, according to this embodiment, it is possible to both improve line density and reduce crosstalk noise between lines, and thus it is possible to realize an impedance converter that is applicable to high-density packaging.
　なお、本実施例では、インピーダンス変換器の出力側の特性インピーダンスを小さくしているが、入力側の特性インピーダンスを小さくしたインピーダンス変換器を形成することもできる。入力側の特性インピーダンスを小さくするには、図８Ａ～図８Ｃにおいて右端を入力側、左端を出力側とすればよい。この場合、テーパー部１４０は、入力側から出力側に向かって徐々に狭くなる。 Note that in this embodiment, the characteristic impedance on the output side of the impedance converter is made small, but it is also possible to form an impedance converter with a small characteristic impedance on the input side. In order to reduce the characteristic impedance on the input side, the right end may be set as the input side and the left end may be set as the output side in FIGS. 8A to 8C. In this case, the tapered portion 140 becomes gradually narrower from the input side to the output side.
　また、図８Ａ～図８Ｃ、図９Ａ～図９Ｃでは、平行に設ける信号線路１２，１３の総数が５本の場合について説明した。グランド層１４は信号線路１２の両側に設けられるので、グランド層１４の本数は信号線路１２の本数よりも１本多くなる。信号線路１２，１３の総数は５本に限るものではないことは言うまでもない。
　第１の実施例と本実施例を任意に組み合わせた構成においても、本発明の本質的な効果は損なわれることはない。 Furthermore, in FIGS. 8A to 8C and FIGS. 9A to 9C, the case where the total number of signal lines 12 and 13 provided in parallel is five has been described. Since the ground layers 14 are provided on both sides of the signal line 12, the number of ground layers 14 is one more than the number of signal lines 12. It goes without saying that the total number of signal lines 12 and 13 is not limited to five.
Even in a configuration in which the first embodiment and this embodiment are arbitrarily combined, the essential effects of the present invention are not impaired.
　上記の実施例の一部又は全部は、以下の付記のようにも記載されうるが、以下には限られない。 Part or all of the above embodiments may be described as in the following supplementary notes, but the embodiments are not limited to the following.
　（付記１）本発明のインピーダンス変換器は、第１のマイクロストリップ線路と、信号伝搬方向と交差する方向に沿って前記第１のマイクロストリップ線路と交互に配置された第２のマイクロストリップ線路と、前記第１のマイクロストリップ線路と接続された第１のグランデッドコプレナー線路と、前記信号伝搬方向と交差する方向に沿って前記第１のグランデッドコプレナー線路と交互に配置され、前記第２のマイクロストリップ線路と接続された第２のグランデッドコプレナー線路とを備え、前記第１のマイクロストリップ線路は、誘電体基板と、前記誘電体基板の裏面に形成された第１のグランド層と、前記誘電体基板の表面に形成された第１の信号線路とから構成され、前記第２のマイクロストリップ線路は、前記誘電体基板と、前記第１のグランド層と、前記信号伝搬方向と交差する方向に沿って前記第１の信号線路と交互に配置されるように前記誘電体基板中に形成された第２の信号線路とから構成され、前記第１のグランデッドコプレナー線路は、前記誘電体基板と、前記第１の信号線路と、前記第１の信号線路と同一層の、前記第１の信号線路の両側の位置に、前記第２の信号線路の上方に位置するように形成された第２のグランド層と、前記第２の信号線路と同一層の、前記第２の信号線路の両側の位置に、前記第１の信号線路の下方に位置するように形成された第３のグランド層とから構成され、前記第２のグランデッドコプレナー線路は、前記誘電体基板と、前記第２の信号線路と、前記第２の信号線路の両側に形成された前記第３のグランド層と、前記第２の信号線路の上方に位置するように形成された前記第２のグランド層とから構成される。 (Supplementary Note 1) The impedance converter of the present invention includes a first microstrip line and a second microstrip line alternately arranged with the first microstrip line along a direction intersecting the signal propagation direction. , a first grounded coplanar line connected to the first microstrip line and a grounded coplanar line alternately arranged with the first grounded coplanar line along a direction intersecting the signal propagation direction, and the first grounded coplanar line connected to the first microstrip line; a second grounded coplanar line connected to a second microstrip line, and the first microstrip line includes a dielectric substrate and a first ground layer formed on the back surface of the dielectric substrate. and a first signal line formed on the surface of the dielectric substrate, and the second microstrip line includes the dielectric substrate, the first ground layer, and the signal propagation direction. and second signal lines formed in the dielectric substrate so as to be arranged alternately with the first signal line along the intersecting direction, and the first grounded coplanar line includes: the dielectric substrate, the first signal line, the same layer as the first signal line, on both sides of the first signal line, and above the second signal line. A second ground layer is formed on the same layer as the second signal line, and a second ground layer is formed on both sides of the second signal line so as to be located below the first signal line. The second grounded coplanar line includes the dielectric substrate, the second signal line, and the third ground layer formed on both sides of the second signal line. It is composed of a ground layer and the second ground layer formed above the second signal line.
　（付記２）付記１記載のインピーダンス変換器において、前記第２のグランド層は、前記信号伝搬方向に沿って幅が徐々に変化する第１のテーパー部と、前記第１のテーパー部の最太部と連なるように形成された幅が一定の第１の矩形部とを有し、前記第３のグランド層は、前記信号伝搬方向に沿って幅が徐々に変化する第２のテーパー部と、前記第２のテーパー部の最太部と連なるように形成された幅が一定の第２の矩形部とを有する。 (Supplementary Note 2) In the impedance converter according to Supplementary Note 1, the second ground layer includes a first tapered portion whose width gradually changes along the signal propagation direction, and a thickest portion of the first tapered portion. a first rectangular part with a constant width formed so as to be continuous with the first rectangular part, and the third ground layer has a second tapered part with a width that gradually changes along the signal propagation direction; and a second rectangular portion having a constant width formed so as to be continuous with the thickest portion of the second tapered portion.
　（付記３）付記１または２記載のインピーダンス変換器において、前記第１の信号線路の中心線と前記第２のグランド層の中心線間の距離、前記第２の信号線路の中心線と前記第３のグランド層の中心線間の距離、前記第１の信号線路と前記第２のグランド層の間隔、前記第１の信号線路と前記第３のグランド層の間隔、および前記第２の信号線路と前記第２のグランド層の間隔は、前記第１の信号線路と前記第２の信号線路間の距離よりも小さい。 (Supplementary Note 3) In the impedance converter according to Supplementary Note 1 or 2, the distance between the center line of the first signal line and the center line of the second ground layer, the distance between the center line of the second signal line and the center line of the second ground layer, 3, the distance between the center lines of the ground layers, the distance between the first signal line and the second ground layer, the distance between the first signal line and the third ground layer, and the second signal line. and the second ground layer is smaller than the distance between the first signal line and the second signal line.
　（付記４）本発明のインピーダンス変換器は、第１のマイクロストリップ線路と、信号伝搬方向と交差する方向に沿って前記第１のマイクロストリップ線路と交互に配置された第２のマイクロストリップ線路と、前記第１のマイクロストリップ線路と接続されたグランデッドコプレナー線路と、前記信号伝搬方向と交差する方向に沿って前記グランデッドコプレナー線路と交互に配置され、前記第２のマイクロストリップ線路と接続された第３のマイクロストリップ線路とを備え、前記第１のマイクロストリップ線路は、誘電体基板と、前記誘電体基板の裏面に形成された第１のグランド層と、前記誘電体基板の表面に形成された第１の信号線路とから構成され、前記第２のマイクロストリップ線路は、前記誘電体基板と、前記第１のグランド層と、前記信号伝搬方向と交差する方向に沿って前記第１の信号線路と交互に配置されるように前記誘電体基板中に形成された第２の信号線路とから構成され、前記グランデッドコプレナー線路は、前記誘電体基板と、前記第１の信号線路と、前記第１の信号線路と同一層の前記第１の信号線路の両側の位置に、前記第２の信号線路の上方に位置するように形成された第２のグランド層と、前記第１の信号線路の下方に位置する前記第１のグランド層とから構成され、前記第３のマイクロストリップ線路は、前記誘電体基板と、前記第２の信号線路と、前記第２の信号線路の上方に位置するように形成された前記第２のグランド層とから構成される。 (Additional Note 4) The impedance converter of the present invention includes a first microstrip line and a second microstrip line alternately arranged with the first microstrip line along a direction intersecting the signal propagation direction. , a grounded coplanar line connected to the first microstrip line, and a grounded coplanar line alternately arranged with the grounded coplanar line along a direction intersecting the signal propagation direction, and connected to the second microstrip line. a third microstrip line connected thereto, the first microstrip line includes a dielectric substrate, a first ground layer formed on the back surface of the dielectric substrate, and a surface of the dielectric substrate. The second microstrip line includes the dielectric substrate, the first ground layer, and the first signal line formed along the direction intersecting the signal propagation direction. and second signal lines formed in the dielectric substrate so as to be arranged alternately with the first signal line, and the grounded coplanar line is configured to connect the dielectric substrate and the first signal line. a second ground layer formed above the second signal line on both sides of the first signal line on the same layer as the first signal line; the first ground layer located below the first signal line, and the third microstrip line includes the dielectric substrate, the second signal line, and the first ground layer located below the second signal line. and the second ground layer formed above.
　（付記５）付記４記載のインピーダンス変換器において、前記第２のグランド層は、前記信号伝搬方向に沿って幅が徐々に変化するテーパー部と、前記テーパー部の最太部と連なるように形成された幅が一定の矩形部とを有する。 (Supplementary Note 5) In the impedance converter according to Supplementary Note 4, the second ground layer is formed so as to be continuous with a tapered portion whose width gradually changes along the signal propagation direction and the thickest portion of the tapered portion. and a rectangular portion with a constant width.
　（付記６）付記４または５記載のインピーダンス変換器において、前記第１の信号線路の中心線と前記第２のグランド層の中心線間の距離、前記第１の信号線路と前記第２のグランド層の間隔、および前記第２の信号線路と前記第２のグランド層の間隔は、前記第１の信号線路と前記第２の信号線路間の距離よりも小さい。 (Supplementary Note 6) In the impedance converter according to Supplementary Note 4 or 5, the distance between the center line of the first signal line and the center line of the second ground layer, the distance between the first signal line and the second ground layer, The spacing between layers and the spacing between the second signal line and the second ground layer are smaller than the distance between the first signal line and the second signal line.
　本発明は、半導体高周波モジュールにおいてインピーダンスを変換する技術に適用することができる。 The present invention can be applied to a technique for converting impedance in a semiconductor high-frequency module.
　１，２，５…マイクロストリップ線路、３，３ａ，４…グランデッドコプレナー線路、１０…誘電体基板、１１，１４，１５…グランド層、１２，１３…信号線路、１６～１９…基板接続パッド、２０～２４…ビア、１４０，１５０…テーパー部、１４１，１５１…矩形部。 1, 2, 5... Microstrip line, 3, 3a, 4... Grounded coplanar line, 10... Dielectric substrate, 11, 14, 15... Ground layer, 12, 13... Signal line, 16-19... Board connection Pads, 20 to 24... Vias, 140, 150... Tapered portions, 141, 151... Rectangular portions.

Claims (6)

  1. 　第１のマイクロストリップ線路と、
    　信号伝搬方向と交差する方向に沿って前記第１のマイクロストリップ線路と交互に配置された第２のマイクロストリップ線路と、
    　前記第１のマイクロストリップ線路と接続された第１のグランデッドコプレナー線路と、
    　前記信号伝搬方向と交差する方向に沿って前記第１のグランデッドコプレナー線路と交互に配置され、前記第２のマイクロストリップ線路と接続された第２のグランデッドコプレナー線路とを備え、
    　前記第１のマイクロストリップ線路は、
    　誘電体基板と、
    　前記誘電体基板の裏面に形成された第１のグランド層と、
    　前記誘電体基板の表面に形成された第１の信号線路とから構成され、
    　前記第２のマイクロストリップ線路は、
    　前記誘電体基板と、
    　前記第１のグランド層と、
    　前記信号伝搬方向と交差する方向に沿って前記第１の信号線路と交互に配置されるように前記誘電体基板中に形成された第２の信号線路とから構成され、
    　前記第１のグランデッドコプレナー線路は、
    　前記誘電体基板と、
    　前記第１の信号線路と、
    　前記第１の信号線路と同一層の、前記第１の信号線路の両側の位置に、前記第２の信号線路の上方に位置するように形成された第２のグランド層と、
    　前記第２の信号線路と同一層の、前記第２の信号線路の両側の位置に、前記第１の信号線路の下方に位置するように形成された第３のグランド層とから構成され、
    　前記第２のグランデッドコプレナー線路は、
    　前記誘電体基板と、
    　前記第２の信号線路と、
    　前記第２の信号線路の両側に形成された前記第３のグランド層と、
    　前記第２の信号線路の上方に位置するように形成された前記第２のグランド層とから構成されることを特徴とするインピーダンス変換器。     a first microstrip line;
    second microstrip lines arranged alternately with the first microstrip lines along a direction intersecting the signal propagation direction;
    a first grounded coplanar line connected to the first microstrip line;
    a second grounded coplanar line arranged alternately with the first grounded coplanar line along a direction intersecting the signal propagation direction and connected to the second microstrip line;
    The first microstrip line is
    a dielectric substrate;
    a first ground layer formed on the back surface of the dielectric substrate;
    a first signal line formed on the surface of the dielectric substrate,
    The second microstrip line is
    the dielectric substrate;
    the first ground layer;
    and second signal lines formed in the dielectric substrate so as to be arranged alternately with the first signal lines along the direction intersecting the signal propagation direction,
    The first grounded coplanar line is
    the dielectric substrate;
    the first signal line;
    a second ground layer formed on both sides of the first signal line and above the second signal line on the same layer as the first signal line;
    a third ground layer formed on both sides of the second signal line on the same layer as the second signal line and below the first signal line;
    The second grounded coplanar line is
    the dielectric substrate;
    the second signal line;
    the third ground layer formed on both sides of the second signal line;
    and the second ground layer formed above the second signal line.
  2. 　請求項１記載のインピーダンス変換器において、
    　前記第２のグランド層は、前記信号伝搬方向に沿って幅が徐々に変化する第１のテーパー部と、前記第１のテーパー部の最太部と連なるように形成された幅が一定の第１の矩形部とを有し、
    　前記第３のグランド層は、前記信号伝搬方向に沿って幅が徐々に変化する第２のテーパー部と、前記第２のテーパー部の最太部と連なるように形成された幅が一定の第２の矩形部とを有することを特徴とするインピーダンス変換器。     The impedance converter according to claim 1,
    The second ground layer includes a first tapered portion whose width gradually changes along the signal propagation direction, and a first tapered portion whose width is constant and which is formed so as to be continuous with the thickest portion of the first tapered portion. 1 rectangular part,
    The third ground layer includes a second tapered portion whose width gradually changes along the signal propagation direction, and a second tapered portion whose width is constant and which is formed so as to be continuous with the thickest portion of the second tapered portion. An impedance converter having two rectangular parts.
  3. 　請求項１または２記載のインピーダンス変換器において、
    　前記第１の信号線路の中心線と前記第２のグランド層の中心線間の距離、前記第２の信号線路の中心線と前記第３のグランド層の中心線間の距離、前記第１の信号線路と前記第２のグランド層の間隔、前記第１の信号線路と前記第３のグランド層の間隔、および前記第２の信号線路と前記第２のグランド層の間隔は、前記第１の信号線路と前記第２の信号線路間の距離よりも小さいことを特徴とするインピーダンス変換器。     The impedance converter according to claim 1 or 2,
    the distance between the center line of the first signal line and the center line of the second ground layer; the distance between the center line of the second signal line and the center line of the third ground layer; The distance between the signal line and the second ground layer, the distance between the first signal line and the third ground layer, and the distance between the second signal line and the second ground layer are the same as the distance between the first signal line and the second ground layer. An impedance converter characterized in that the distance is smaller than the distance between the signal line and the second signal line.
  4. 　第１のマイクロストリップ線路と、
    　信号伝搬方向と交差する方向に沿って前記第１のマイクロストリップ線路と交互に配置された第２のマイクロストリップ線路と、
    　前記第１のマイクロストリップ線路と接続されたグランデッドコプレナー線路と、
    　前記信号伝搬方向と交差する方向に沿って前記グランデッドコプレナー線路と交互に配置され、前記第２のマイクロストリップ線路と接続された第３のマイクロストリップ線路とを備え、
    　前記第１のマイクロストリップ線路は、
    　誘電体基板と、
    　前記誘電体基板の裏面に形成された第１のグランド層と、
    　前記誘電体基板の表面に形成された第１の信号線路とから構成され、
    　前記第２のマイクロストリップ線路は、
    　前記誘電体基板と、
    　前記第１のグランド層と、
    　前記信号伝搬方向と交差する方向に沿って前記第１の信号線路と交互に配置されるように前記誘電体基板中に形成された第２の信号線路とから構成され、
    　前記グランデッドコプレナー線路は、
    　前記誘電体基板と、
    　前記第１の信号線路と、
    　前記第１の信号線路と同一層の前記第１の信号線路の両側の位置に、前記第２の信号線路の上方に位置するように形成された第２のグランド層と、
    　前記第１の信号線路の下方に位置する前記第１のグランド層とから構成され、
    　前記第３のマイクロストリップ線路は、
    　前記誘電体基板と、
    　前記第２の信号線路と、
    　前記第２の信号線路の上方に位置するように形成された前記第２のグランド層とから構成されることを特徴とするインピーダンス変換器。     a first microstrip line;
    second microstrip lines arranged alternately with the first microstrip lines along a direction intersecting the signal propagation direction;
    a grounded coplanar line connected to the first microstrip line;
    a third microstrip line arranged alternately with the grounded coplanar line along a direction intersecting the signal propagation direction and connected to the second microstrip line;
    The first microstrip line is
    a dielectric substrate;
    a first ground layer formed on the back surface of the dielectric substrate;
    a first signal line formed on the surface of the dielectric substrate,
    The second microstrip line is
    the dielectric substrate;
    the first ground layer;
    and second signal lines formed in the dielectric substrate so as to be arranged alternately with the first signal lines along the direction intersecting the signal propagation direction,
    The grounded coplanar line is
    the dielectric substrate;
    the first signal line;
    a second ground layer formed above the second signal line on both sides of the first signal line on the same layer as the first signal line;
    the first ground layer located below the first signal line,
    The third microstrip line is
    the dielectric substrate;
    the second signal line;
    and the second ground layer formed above the second signal line.
  5. 　請求項４記載のインピーダンス変換器において、
    　前記第２のグランド層は、前記信号伝搬方向に沿って幅が徐々に変化するテーパー部と、前記テーパー部の最太部と連なるように形成された幅が一定の矩形部とを有することを特徴とするインピーダンス変換器。     The impedance converter according to claim 4,
    The second ground layer has a tapered part whose width gradually changes along the signal propagation direction, and a rectangular part with a constant width formed so as to be continuous with the thickest part of the tapered part. Characteristic impedance converter.
  6. 　請求項４または５記載のインピーダンス変換器において、
    　前記第１の信号線路の中心線と前記第２のグランド層の中心線間の距離、前記第１の信号線路と前記第２のグランド層の間隔、および前記第２の信号線路と前記第２のグランド層の間隔は、前記第１の信号線路と前記第２の信号線路間の距離よりも小さいことを特徴とするインピーダンス変換器。     The impedance converter according to claim 4 or 5,
    The distance between the center line of the first signal line and the center line of the second ground layer, the distance between the first signal line and the second ground layer, and the distance between the second signal line and the second ground layer. An impedance converter characterized in that an interval between the ground layers is smaller than a distance between the first signal line and the second signal line.
PCT/JP2022/023436 2022-06-10 2022-06-10 Impedance converter WO2023238376A1 (en)

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Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH1041637A (en) * 1996-07-23 1998-02-13 Nec Corp High-density multilayer wiring board
JP2003069239A (en) * 2001-08-22 2003-03-07 Toppan Printing Co Ltd Multilayer interconnection board for high frequency circuit
JP2005101587A (en) * 2003-08-29 2005-04-14 Handotai Rikougaku Kenkyu Center:Kk Parallel wiring and integrated circuit
WO2021220460A1 (en) * 2020-04-30 2021-11-04 日本電信電話株式会社 Impedance converter

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JPH1041637A (en) * 1996-07-23 1998-02-13 Nec Corp High-density multilayer wiring board
JP2003069239A (en) * 2001-08-22 2003-03-07 Toppan Printing Co Ltd Multilayer interconnection board for high frequency circuit
JP2005101587A (en) * 2003-08-29 2005-04-14 Handotai Rikougaku Kenkyu Center:Kk Parallel wiring and integrated circuit
WO2021220460A1 (en) * 2020-04-30 2021-11-04 日本電信電話株式会社 Impedance converter

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