WO2023123741A1 - 一种高速永磁同步电机全速域飞启方法 - Google Patents

一种高速永磁同步电机全速域飞启方法 Download PDF

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WO2023123741A1
WO2023123741A1 PCT/CN2022/086967 CN2022086967W WO2023123741A1 WO 2023123741 A1 WO2023123741 A1 WO 2023123741A1 CN 2022086967 W CN2022086967 W CN 2022086967W WO 2023123741 A1 WO2023123741 A1 WO 2023123741A1
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speed
motor
short
initial
current
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French (fr)
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程明
李林
花为
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东南大学
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/13—Observer control, e.g. using Luenberger observers or Kalman filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/34—Arrangements for starting
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
    • H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
    • H02P25/022—Synchronous motors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/08—Arrangements for controlling the speed or torque of a single motor
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
    • H02P6/20—Arrangements for starting

Definitions

  • the invention relates to a motor belt-speed re-start control technology, specifically discloses a method for flying a high-speed permanent magnet synchronous motor at full speed, and belongs to the technical field of power generation, power transformation or power distribution.
  • High-Speed Permanent Magnet Synchronous Machine has the advantages of small size, high efficiency, high power density, and wide speed range.
  • HSPMSM High-Speed Permanent Magnet Synchronous Machine
  • the traditional vector control method is relatively mature and easy to implement, and the vector control strategy is mostly used in the control system of HSPMSM.
  • the position sensorless control strategy is mostly used in the HSPMSM vector control system.
  • HSPMSM position sensorless control system In the HSPMSM position sensorless control system, most motors are started from zero speed. However, in some special applications, such as the start-up of fans and water pumps with initial rotational speed, the idle restart of trams, the restart of large-inertia motors, the fault-tolerant control of position sensors, etc., the initial speed of the motor is not zero. In HSPMSM In position sensorless control, starting a motor with initial speed is called Flying-Start of HSPMSM.
  • the core problem of vector control is to obtain the motor speed and rotor position angle, so that the magnetic field vector generated by the stator current is perpendicular to the rotor magnetic field vector, so as to generate the maximum electromagnetic torque.
  • the inverter Before the HSPMSM position sensorless control system flies, the inverter is disconnected, and there is no current in the stator winding of the motor, so it is impossible to directly use the positionless algorithm to start the speed motor. At the same time, the back EMF of the speed motor exists and the inverter The initial output voltage does not match the problem, directly starting the speed motor leads to inrush current and motor out of step.
  • the traditional permanent magnet synchronous motor speed re-injection technology can be divided into two categories: the first category is to use the voltage sensor to measure the induced electromotive force in the stator winding of the rotating motor, and the speed and speed of the motor fly-off can be obtained by solving the induced electromotive force.
  • Rotor position adding additional voltage sensors, increasing the cost of the control system, and reducing the reliability of the non-position control system;
  • the second type is the method of high-frequency signal injection.
  • the initial speed and initial position of the motor can be estimated, and it has excellent control performance at zero and low speeds.
  • high-frequency signal injection can only be used in salient pole motors.
  • the design of filter and observer required to solve the high-frequency current response is complex, and the limitation of the output voltage of the inverter further limits the application of the high-frequency signal injection method in the fly-start of the motor with a high initial speed.
  • the purpose of the present invention is to propose a method for full-speed start-up of a high-speed permanent magnet synchronous motor, which can realize full-speed start-up without adding additional current and voltage sensors.
  • the purpose of the invention of the present invention is to address the deficiencies in the above-mentioned background technology, and provide a method for flying high-speed permanent magnet synchronous motors in the full-speed range.
  • the speed zone of the motor is judged according to the real-time short-circuit current vector amplitude generated by the short-circuit test pulse.
  • the positionless control based on the short-circuit current vector method is performed, and for the motor in the low-speed zone, it is The positionless control based on the back EMF decoupling method, the positionless control based on the constant current frequency ratio start-up is performed for the motor in the zero-low speed range, and the control strategy corresponding to each speed range is used to realize the full-speed range fly-off of the HSPMSM, which solves the problem of existing permanent magnets
  • the re-introduction method of synchronous motor with speed needs to add an additional voltage sensor, the high-frequency signal injection method is not suitable for the fly-off of the motor with a high initial speed, and the position control based on the two-time short-circuit current vector method has calculation errors and cannot be used immediately after the short-circuit current operation.
  • the present invention adopts the following technical solutions in order to realize the above-mentioned invention purpose: a high-speed permanent magnet synchronous motor full-speed domain flying method, which injects test short-circuit pulses into the stator winding of the motor, collects short-circuit current and calculates the amplitude of the short-circuit current vector, and according to the short-circuit current vector Determine the speed range of the initial speed of the motor with speed based on the magnitude of , and select the corresponding fly-start method according to the range of the initial speed.
  • the speed motor is controlled based on the model reference adaptive position sensor; if the amplitude of the short-circuit current vector is less than 40% of the rated current and greater than the rated current 3% of the current, it is determined that the initial speed of the speed motor is in the low-speed range, and the observed back EMF feed-forward decoupling is used to obtain the command voltage of vector control.
  • the inverter works under the action of vector control, and the stator winding of the speed motor After the current is generated, a positionless control algorithm based on model reference self-adaptation is used to realize the flying start of the motor in the low-speed range; if the magnitude of the short-circuit current vector is less than 3% of the rated current, it is determined that the initial speed of the motor with speed is in the zero-low speed range, The constant current frequency ratio control method is used to drag the speed motor to a given speed for stable operation, and then switch to the positionless control method based on model reference self-adaptation.
  • the specific method of estimating the initial speed of the speed motor and the initial position angle of the rotor using the short-circuit current vector method is as follows: first, all the upper bridge arms of the inverter are closed and The bridge arms are all opened, thereby generating a short-circuit current vector, repeating the above-mentioned short-circuit operation four times and sampling the short-circuit current by the current sensor.
  • the frequency of the current sensor is set to 10 times the operating frequency of the system, and the maximum value of the short-circuit current is collected; secondly, Since the short-circuit operation time is extremely short, generally several hundred microseconds, it is considered that the motor speed is approximately constant.
  • the angle difference between the two short-circuit current vectors in space is The electrical angle rotated in the separated short-circuit operation, the angle difference divided by the time of two short-circuit operations is the initial speed of the motor, four short-circuit operations to get three sets of motor speeds, the error between each set of speeds is less than 5% , take the average value of each speed as the initial speed of the motor; bring it into the initial speed of the motor, use Laplace transform to solve the value of the short-circuit current in the d-q coordinate system, and then solve the value of the last short-circuit current vector in the d-q coordinate system Lower angle; finally, the angle of the last short-circuit current vector in the d-q coordinate system obtained from the previous step minus the angle of the last short-circuit current vector in the ⁇ - ⁇ coordinate system, the angle difference is the d
  • the speed motor is controlled without a position sensor, specifically: the speed motor
  • the initial speed and initial rotor position angle are brought into the model reference adaptive positionless algorithm based on the stator current as the initial speed and initial rotor position of the positionless algorithm, so as to realize the flying start of the motor.
  • the voltage reference value of the vector control obtained by the back EMF decoupling method is as follows: the motor runs in the two-phase stationary coordinate system, the given current and the motor angle are set to 0, and the disturbance observer is used to calculate the extended back EMF of the motor and compare it to Feed-forward decoupling, vector control is performed after the decoupling operation is completed, the inverter starts to run, the stator winding of the motor generates current, and then the motor is directly started using the positionless control algorithm based on model reference self-adaptation, because the back EMF of the speed motor has already It is fed forward and decoupled to reduce the overcurrent caused by the mismatch between the back electromotive force and the initial output voltage of the inverter during the speed start process, so the motor will not have inrush current or out of step when starting question.
  • the initial speed of the speed motor is in the zero-low speed range, and the constant current frequency ratio control method is used to drag the speed motor to a given speed.
  • the potential decoupling method and the short-circuit current vector method are not available.
  • the zero-low-speed motor is regarded as a static state, and the constant current-frequency ratio is used to control the drag motor.
  • the given q-axis current amplitude is the rated current of the motor, and the given speed is a ramp
  • the input form and the slope of the slope are related to the motor's moment of inertia, viscosity coefficient, and load conditions.
  • both the speed loop controller and the current loop controller in the motor controller adopt PI control.
  • the single short-circuit time and the interval between short-circuit operations are one system operation cycle.
  • the current sensor samples the maximum value of the short-circuit current, that is, the short-circuit current is sampled and recorded at the falling delay of the short-circuit pulse, and the current sampling frequency is set to 10 times the system operating frequency to reduce the calculation error of the short-circuit current vector.
  • the present invention adopts the above-mentioned technical scheme, and has the following beneficial effects:
  • the present invention judges the interval of the initial rotational speed of the motor through the short-circuit current pulse test, and adopts position-free control based on the short-circuit current vector method and electromotive force feedback based on the Decoupling non-position control, non-position control based on constant current-frequency ratio startup, non-position control is realized by a closed-loop control system composed of a non-position controller based on current-type model reference self-adaptation, with good robustness and faster algorithm convergence speed Fast, can quickly estimate the motor speed and rotor position, realize the fast flight of the HSPMSM in the full speed range, without increasing hardware costs, simple design, and easy engineering implementation.
  • the present invention abandons the traditional high-frequency pulse vibration square wave injection method, and obtains the accurate initial rotational speed of the rotor through four short-circuit current operations, and uses the acquired initial rotational speed of the rotor And the initial position of the rotor is the initial data of the positionless control based on the current-type model reference self-adaptation.
  • the positionless algorithm can quickly converge to the current speed and rotor position of the speed motor, realize the field oriented control, and effectively restrain the fly-off caused by the angle difference.
  • the inrush current generated by matching can realize the rapid start-up of the motor after the short-circuit operation is completed.
  • multiple short-circuit operations can effectively reduce the calculation error caused by current sampling, enhance the fault tolerance of the system, and ensure reliable start-up.
  • the command voltage of vector control is obtained by using the back-emf decoupling method, and then the vector control is used to generate current, and then directly realize the motor fly-on through positionless control.
  • the present invention abandons the traditional high-frequency signal injection method, and adopts the strategy of I-F start to drag the motor to a given speed, and the given d-axis lags behind the real d-axis 90 degrees electrical angle, switch to the non-position control algorithm after generating enough back EMF, take the given speed and the corresponding rotor position angle as the initial value of the non-position control, the non-position algorithm quickly converges to the real value, and realizes from I-F Fast and smooth switching from start-up to no-position control, thereby realizing reliable fly-off of the motor with speed in the zero-low speed range.
  • Figure 1 is a control flow chart of the full-speed domain fly-on algorithm.
  • Figure 2 is a system block diagram of the HSPMSM flying-start strategy based on the short-circuit current vector method.
  • Fig. 3 is a schematic diagram of a short-circuit pulse and a short-circuit current vector.
  • Fig. 4 is a schematic diagram of the spatial position of the short-circuit current vector during four short-circuit operations.
  • Fig. 5 is a waveform diagram of the motor speed during the flight-start process based on the short-circuit current vector method.
  • Fig. 6 is a waveform diagram of the rotor position angle of the motor during the fly-up process based on the short-circuit current vector method.
  • Fig. 7 is a waveform diagram of the motor current during the flight-start process based on the short-circuit current vector method.
  • Fig. 8 is a waveform diagram of four groups of short-circuit currents during the flight-start process based on the short-circuit current vector method.
  • Fig. 9 is a control block diagram of the disturbance observation period device for observing the extended back EMF.
  • Figure 10 is a block diagram of the implementation of the back EMF decoupling operation.
  • Fig. 11 is a waveform diagram of the motor speed during the fly-off process based on the back EMF decoupling method.
  • Fig. 12 is a waveform diagram of the rotor position angle of the motor during the fly-off process based on the back EMF decoupling method.
  • Fig. 13 is a waveform diagram of the motor current during the fly-on process based on the back EMF decoupling method.
  • Figure 14 is a block diagram of the implementation of constant current frequency ratio control.
  • Fig. 15 is a waveform diagram of the motor speed during the flight-start process based on the constant current frequency ratio control.
  • Fig. 16 is a waveform diagram of the rotor position angle of the motor during the fly-on process based on the constant current frequency ratio control.
  • Fig. 17 is a waveform diagram of the motor current during the fly-start process based on the constant current frequency ratio control.
  • a short-circuit test pulse is firstly injected into the motor winding, and the short-circuit current is sampled at the falling edge of the pulse and the magnitude of the current vector is calculated. If the magnitude of the current vector is greater than 40% of the rated current, the initial speed of the motor is medium to high speed. If the current vector amplitude is less than 40% of the rated current and greater than 3% of the rated current, the motor speed is considered to be in the lower speed range. If the magnitude of the current vector is less than 3% of the rated current, the motor is considered to be in the zero-low speed range.
  • the short-circuit current vector method usually requires that the amplitude of the short-circuit current is greater than 40% of the rated current. According to the formula (1), the demarcation speed value ⁇ ec between the medium-high speed and the low speed can be obtained.
  • L q is the q-axis inductance
  • ⁇ f is the rotor flux linkage
  • the short-circuit current pulse is applied to the switching tube of the inverter so that the upper bridge arm of the three-phase inverter is turned off and the lower bridge arm is opened, four short-circuit operations are performed on the stator winding of the motor, and the short-circuit current is sampled and records, record the short-circuit current sampling values as i abc1 , i abc2 , i abc3 , and i abc4 .
  • Clark transformation the above-mentioned short-circuit current is transformed into the ⁇ - ⁇ coordinate system.
  • the block diagram of the system is shown in Figure 2. Calculate the space angle of the short-circuit current vector in the ⁇ - ⁇ coordinate system:
  • n the nth short-circuit operation.
  • the spatial angle difference of adjacent short-circuit current vectors is the angle at which the motor turns.
  • ⁇ scm is the spatial angle difference of adjacent short-circuit current vectors in the ⁇ - ⁇ coordinate system
  • T s is the short-circuit operation time
  • the interval between two adjacent short-circuit operations is also T s , as shown in Figure 3 . Since the first short-circuit current will affect the measurement of the second short-circuit current, it is necessary to ensure that the short-circuit current has decayed to zero before the second short-circuit operation, which requires that the interval between the two short-circuit operations should not be too short. At the same time, an excessively long interval time causes the rotation angle of the motor to exceed half an electric angle cycle in short-circuit operation, which brings deviations in speed calculation.
  • the motor rotor position angle is the angle between the d coordinate axis and the ⁇ coordinate axis.
  • the angle of the fourth short-circuit current vector in the dq coordinate system minus the angle of the fourth short-circuit current vector in the ⁇ - ⁇ coordinate system is the initial rotor position angle
  • the positionless control model of HSPMSM is established in MATLAB/Simulink.
  • Figure 5 and Figure 6 show the motor speed and motor rotor position angle waveforms during the fly-off process.
  • Figure 7 shows the motor current waveform during the fly-off process.
  • Figure 8 shows four groups of short-circuit currents during the flight-start process. It can be seen that in the middle and high speed range, the use of the flight-start strategy based on the short-circuit current vector method can reduce the inrush current during flight-start and reliably realize the flight-start of the HSPMSM with speed.
  • the back EMF identification value is approximately equal to the real motor back EMF.
  • Figure 9 shows the extended back EMF observer based on formula (10), and the structure diagram of the whole system is shown in Figure 10.
  • Figure 11 and Figure 12 show the motor speed and motor rotor position angle waveforms during the fly-on process based on the back EMF decoupling method at low speeds
  • Figure 13 shows the motor current waveform during the fly-off process. It can be seen that in the lower speed range, the use of the fly-start strategy based on the back EMF decoupling method can reduce the calculation error caused by the current sampling error, suppress the inrush current during the fly-start, and reliably realize the fly-start of the HSPMSM with speed.
  • FIG. 14 is a system block diagram of I-F control.
  • the amplitude design value of the given current vector is:
  • T l is the load torque
  • p is the number of pole pairs.
  • the given speed is accelerated in the form of a ramp signal, and the given slope is the rotational acceleration of the current vector. Excessive acceleration will make it difficult for the actual axis to synchronize with the rotational current, while too small acceleration will cause the motor to accelerate slowly.
  • the acceleration of the given motor speed is designed as:
  • J is the moment of inertia
  • the ramp acceleration time of the motor can be obtained according to the given speed.
  • Figure 15 and Figure 16 show the motor speed and motor rotor position angle waveforms during the fly-off process based on I-F control at zero and low speed
  • Figure 17 shows the motor current waveform during the fly-off process.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)
  • Control Of Motors That Do Not Use Commutators (AREA)

Abstract

提供一种高速永磁同步电机全速域飞启方法，属于发电、变电或配电的技术领域。针对基于模型参考自适应的HSPMSM无位置控制系统的带速启动。首先判断带速电机所处的转速区间，在中高速时，采用短路电流矢量法求解出电机的初始转速和转子位置角，将初始转速和转子位置角作为初始值带入无位置算法中，完成电机的飞启。在较低速时，采用反电势解耦法实现电机的低速飞启。而在极低转速甚至零速时，采用恒流频比控制，将电机拖至一定转速，再切换至无位置控制。提供的飞启方法能可靠地启动带速电机，有效抑制电机起动过程中出现的冲击电流，实现带速电机的全速域飞启，无需附加任何传感器且容易实现，具备良好的经济性与实用性。

Description

一种高速永磁同步电机全速域飞启方法 技术领域
本发明涉及电机带速重投控制技术，具体公开一种高速永磁同步电机全速域飞启方法，属于发电、变电或配电的技术领域。
背景技术
高速永磁同步电机(High-Speed Permanent Magnet Synchronous Machine，HSPMSM)具有体积小、效率高、功率密度高、调速范围宽等优点。在航空航天、新能源汽车、高档家电等领域，HSPMSM正逐渐成为研究和应用的热点。传统矢量控制方法较为成熟且容易实现，在HSPMSM的控制系统中多采用矢量控制的策略。进一步地，为了提高HSPMSM矢量控制系统的可靠性，减少控制系统的成本，并减小电机的体积，HSPMSM的矢量控制系统中多采用无位置传感器控制策略。
在HSPMSM无位置传感器控制系统中，电机的启动大多数都是从零转速开始启动。而在某些特殊的应用场合中，例如带初始转速的风机、抽水机的启动，电车惰行再启动，大惯量电机再启动、位置传感器的容错控制等，电机的初始速度都不为零，在HSPMSM无位置传感器控制中，启动带初始转速的电机称为HSPMSM的飞启(Flying-Start)。
矢量控制的核心问题在于获得电机的转速和转子位置角度，使得定子电流产生的磁场矢量垂直于转子磁场矢量，以此产生最大的电磁转矩。在HSPMSM的无位置传感器控制系统飞启前，逆变器处于断开状态，电机定子绕组不存在电流，无法直接使用无位置算法启动带速电机，同时带速电机的反电势存在与逆变器初始输出电压不匹配的问题，直接启动带速电机导致冲击电流及电机失步。
传统的永磁同步电机带速重投技术可以分为两大类：第一类为使用电压传感器测量旋转电机定子绕组中的感应电动势，解算感应动电势即可以得到实现电机飞启的转速和转子位置，增加额外的电压传感器，增加控制系统的成本，降低无位置控制系统的可靠性；
第二类为高频信号注入的方法，通过求解电流响应可以估算出电机的初始转速和初始位置，在零低速时具有优异的控制性能，然而高频信号注入只能在凸极电机中使用，解算高频电流响应所需要的滤波器和观测器设计复杂，同时逆变器 输出电压存在的限幅进一步限制了高频信号注入方法在初始转速较高的电机飞启中的应用。
为了克服需要额外机械传感器的永磁同步电机带速重投技术增加系统成本且降低无位置控制系统可靠性的缺陷，相关学者使用两次短路电流矢量法与基于有效磁链法的无位置控制实现电机的带速重投，然而为了解决有效磁链法的积分器重启问题，需要设置一段飞启过渡时期，也就是说，在完成短路操作后，无法立即启动带速电机，因此不适用于电机需要快速启动的场合。更重要地，两次短路电流矢量法十分依赖于电流采样和计算的准确性，电流采样存在误差时，电机转速和角度计算误差会很大，从而导致飞启过程中转速的波动，甚至飞启失败。为实现永磁同步电机全速域的飞启，相关学者提出基于复合控制策略的无位置传感器控制技术实现永磁同步电机全速域范围内的控制，例如，在静止时检测电机初始位置，在低速区采用高频脉振方波注入的方法辨识电机转速与转子位置；在零速和低速取采用高频脉振信号注入法，在中速过渡区采用高频脉振注入与模型参考自适应相结合的方法，在高速区采用模型参考自适应方法。然而，对于表贴式的永磁同步电机或者凸极率较小的永磁同步电机来说，高频脉振方波注入的方法并不适用，需要设计切换过程以使不同速域的控制方法平滑切换，虽然能够实现全速域内的控制但不能满足电机快速启动的应用需求。
本发明旨在提出一种高速永磁同步电机全速域飞启方法，在不增加额外电流与电压传感器的情况下实现全速域的飞启。
发明内容
本发明的发明目的是针对上述背景技术的不足，提供一种高速永磁同步电机全速域飞启方法，根据短路测试脉冲产生的所有短路电流矢量的幅值将电机全速域分为中高速区间、低速区间、零低速区，根据短路测试脉冲产生的实时短路电流矢量幅值判断电机所处的速度区间，对于中高速区间的电机进行基于短路电流矢量法的无位置控制，对于低速区间的电机进行基于反电势解耦法的无位置控制，对于零低速区间的电机进行基于恒流频比启动的无位置控制，采用各速度区间对应的控制策略实现HSPMSM的全速域飞启，解决现有永磁同步电机带速重投方法需要增加额外电压传感器、高频信号注入法不适于初始转速较高的电机飞启以及基于两次短路电流矢量法无位置控制存在计算误差且在短路电流操作后无法立即启动带速电机的技术问题。
本发明为实现上述发明目的采用如下技术方案：一种高速永磁同步电机全速域飞启方法，向电机定子绕组注入测试短路脉冲，采集短路电流并计算短路电流矢量的幅值，根据短路电流矢量的幅值判定带速电机的初始转速所处的转速区间，并根据初始转速所处的区间选择对应的飞启方法。
进一步地，根据初始转速所处的区间选择对应的飞启方法，具体为：若短路电流矢量的幅值大于额定电流的40％时，则判定带速电机的初始转速处于中高速区间，采用短路电流矢量法估计无位置算法的初始转速和转子初始位置角后，对带速电机进行基于模型参考自适应的无位置传感器的控制；若短路电流矢量的幅值小于额定电流的40％且大于额定电流的3％，则判定带速电机的初始转速处于低速区间，对观测得到的反电势前馈解耦得到矢量控制的指令电压，逆变器在矢量控制的作用下工作，带速电机定子绕组产生电流后，采用基于模型参考自适应的无位置控制算法实现低速区间电机的飞启；若短路电流矢量的幅值小于额定电流的3％，则判定带速电机的初始转速处于零低速区间，采用恒流频比控制方法将带速电机拖至给定转速稳定运行后再切换至基于模型参考自适应的无位置控制方法。
进一步地，中高速区间和低速区间的分界转速值ω ec为ω ec＝0.4I NL q/T testψ f。
进一步地，带速电机的初始转速处于中高速区间时，采用短路电流矢量法估计带速电机的初始转速和转子初始位置角的具体方法为：首先，将逆变器上桥臂全部关闭而下桥臂全部打开，由此产生短路电流矢量，重复四次上述短路操作并由电流传感器对短路电流进行采样，电流传感器频率设置为系统运行频率的10倍，且采集短路电流的最大值；其次，由于短路操作时间极短，一般为几百微秒，因此认为电机转速近似不变。在电机参数不发生改变的情况下，四次短路电流矢量幅值相同，而短路电流矢量的空间位置随着电机的旋转而改变，因此两个短路电流矢量在空间上的角度差即为电机在相隔的短路操作中旋转过的电角度，此角度差除以两次短路操作的时间即为电机的初始转速，四次短路操作得到三组电机转速，各组转速之间的误差均小于5％时，将各转速的平均值作为电机初始转速；带入电机初始转速，采用拉普拉斯变换求解出短路电流在d-q坐标系下的值，依此求解出最后一次短路电流矢量在d-q坐标系下角度；最后由上一步求解出的最后一次短路电流矢量在d-q坐标系下的角度减去最后一次短路电流矢量在α-β坐标系下的角度，此角度差即为d轴与α坐标轴的夹角，即转子位置角度。
带速电机的初始转速处于中高速区间时，采用短路电流矢量法估计带速电机的初始转速和转子初始位置角后，对带速电机进行无位置传感器的控制，具体为：将带速电机的初始转速和初始转子位置角带入基于定子电流的模型参考自适应无位置算法中，作为无位置算法的初始转速和初始转子位置，以此实现电机的飞启。
进一步地，带速电机的初始转速处于低速区间时，定子绕组中的电流较小，短路电流的检测误差很大，需要依赖高精度电流传感器的矢量控制法以及短路电流矢量法不再适用，采用反电势解耦法得到矢量控制的电压参考值的具体为：电机运行在两相静止坐标系下，给定电流与电机角度均设置为0，采用扰动观测器计算电机的拓展反电势并对其前馈解耦，解耦操作完成后进行矢量控制，逆变器开始运行，电机定子绕组产生电流，再采用基于模型参考自适应的无位置控制算法直接启动电机，因带速电机的反电势已经被前馈并解耦，以此来减少带速启动过程中因为反电势与逆变器初始输出电压不匹配而产生的过电流，因此启动时电机不会出现冲击电流，也不会出现失步问题。
进一步地，带速电机的初始转速处于零低速区间，采用恒流频比控制方法将带速电机拖至给定转速，具体为：由于电机转速过低而反电势十分微弱，基于反电势的反电势解耦法以及短路电流矢量法不可用，将零低速电机视作静止状态，采用恒流频比控制拖动电机，给定q轴电流幅值为电机的额定电流，而给定转速为斜坡输入的形式，斜坡的斜率与电机的转动惯量、粘滞系数、负载情况都有关系。待电机成功被拖动至给定转速时，此时电机的转子位置和转速都可以被计算出来，再切换至无位置算法即可。
进一步地，飞启策略中，电机控制器中的转速环控制器和电流环控制器均采用PI控制。
进一步地，单次短路时间以及短路操作的间隔时间为一个系统运行周期。
进一步地，电流传感器采样短路电流的最大值，即在短路脉冲的下降延处对短路电流进行采样并记录，电流采样频率设置为系统运行频率的10倍以便减少短路电流矢量的计算误差。
本发明采用上述技术方案，具有以下有益效果：
(1)本发明通过短路电流脉冲测试判断电机初始转速所处区间，并针对初始转速所处的中高速区间、低速区间、零低速区间分别采取基于短路电流矢量法 的无位置控制、基于电势反解耦的无位置控制、基于恒流频比启动的无位置控制，无位置控制通过基于电流型模型参考自适应的无位置控制器构成一个闭环控制系统实现，鲁棒性好，算法收敛速度更快，能快速估算出电机转速与转子位置，实现全速域内HSPMSM的快速飞启，不增加硬件成本，设计简单，易于工程实现。
(2)对于初始转速处于中高速区间的带速电机，本发明摒弃了传统的高频脉振方波注入法，通过四次短路电流操作获取准确的转子初始转速，并以获取的转子初始转速及转子初始位置为基于电流型模型参考自适应的无位置控制的初始数据，无位置算法能够迅速收敛至带速电机的当前转速与转子位置，实现磁场定向控制，有效抑制飞启时因角度不匹配而产生的冲击电流，在完成短路操作后即可实现电机的快速飞启，同时，多次短路操作能有效减少因为电流采样造成的计算误差，增强系统的容错能力，保证可靠飞启。
(3)对于初始转速处于低速区间的带速电机，为抑制飞启时的冲击电流及失步问题，采用反电势解耦法获得矢量控制的指令电压，再通过矢量控制使得电机电子绕组中产生电流，继而通过无位置控制直接实现电机飞启。
(4)对于初始转速处于零低速区间的带速电机，本发明摒弃了传统的高频信号注入法，采用I-F启动的策略将电机拖动至给定转速时，给定d轴滞后真实d轴90度电角度，产生足够的反电势后再切换至无位置控制算法，将给定转速及其对应的转子位置角作为无位置控制的初始值，无位置算法迅速收敛至真实值，实现从I-F启动到无位置控制的快速平滑切换，进而实现零低速区间带速电机的可靠飞启。
附图说明
图1为全速域飞启算法的控制流程图。
图2为基于短路电流矢量法的HSPMSM飞启策略的系统框图。
图3为短路脉冲与短路电流矢量的示意图。
图4为四次短路操作过程中的短路电流矢量空间位置的示意图。
图5为基于短路电流矢量法的飞启过程中电机转速的波形图。
图6为基于短路电流矢量法的飞启过程中电机转子位置角的波形图。
图7为基于短路电流矢量法的飞启过程中电机电流的波形图。
图8为基于短路电流矢量法的飞启过程中四组短路电流的波形图。
图9为用于观测拓展反电势的扰动观测期器的控制框图。
图10为反电势解耦操作的实施框图。
图11为基于反电势解耦法的飞启过程中电机转速的波形图。
图12为基于反电势解耦法的飞启过程中电机转子位置角的波形图。
图13为基于反电势解耦法的飞启过程中电机电流的波形图。
图14为恒流频比控制的实施框图。
图15为基于恒流频比控制的飞启过程中电机转速的波形图。
图16为基于恒流频比控制的飞启过程中电机转子位置角的波形图。
图17为基于恒流频比控制的飞启过程中电机电流的波形图。
具体实施方式
下面结合附图对发明的技术方案进行详细说明。
现以一个高速永磁同步电机控制为例，详细介绍本发明实施过程。飞启算法的控制逻辑如图1所示。
在发明所提飞启策略中，首先向电机绕组注入一个短路测试脉冲，在脉冲的下降沿对短路电流采样并计算电流矢量的幅值。若电流矢量的幅值大于额定电流的40％，则电机初始转速为中高速。若电流矢量幅值小于额定电流的40％且大于额定电流的3％，则认为电机转速位于较低转速区间。若电流矢量的幅值小于额定电流的3％，则认为电机处于零低速区间。
短路测试脉冲设定为T test＝100μs，而在此短路脉冲作用下的短路电流矢量需小于额定电流的80％，来避免冲击电流和过大的制动转矩。短路电流矢量法通常要求短路电流的幅值大于额定电流的40％，根据式子(1)可以得到中高转速与较低转速之间的分界转速值ω ec。
ω ec＝0.4I NL q/T testψ f       (1)
其中，L q为q轴电感、ψ f为转子磁链。
在中高速区间时，向逆变器的开关管施加短路电流脉冲使得三相逆变器的上桥臂关断而下桥臂开通，对电机定子绕组进行四次短路操作，并对短路电流采样和记录，记短路电流采样值为i abc1、i abc2、i abc3、i abc4。经过Clark变换，将上述短路电流转变到α-β坐标系下。系统的框图如图2所示。计算短路电流矢量在α-β坐标系下的空间角度：
Figure PCTCN2022086967-appb-000001
其中，下标n代表第n次短路操作。如图4所示，相邻短路电流矢量的空间角度差即为电机转过的角度。设估算的电机初始转速为
Figure PCTCN2022086967-appb-000002
则电机的初始转速：
Figure PCTCN2022086967-appb-000003
其中，Δθscm是相邻的短路电流矢量在α-β坐标系下的空间角度差，T s是短路操作时间，相邻两次短路操作之间的间隔时间也为T s，如图3所示。由于第一次短路电流会对第二次短路电流的测量造成影响，因此在进行第二次短路操作之前需保证短路电流已衰减到零，这就要求两次短路操作时间的间隔不能过短。同时，过长的间隔时间导致电机在短路操作中的旋转角度超过半个电角周期，带来速度计算的偏差。
由式子(3)可以得到三组估算转速，任意两组估算转速需满足误差小于5％，否则将重新进行短路操作，
Figure PCTCN2022086967-appb-000004
初始电机转速
Figure PCTCN2022086967-appb-000005
可以改写为：
Figure PCTCN2022086967-appb-000006
根据拉普拉斯变换，求解出第四次短路电流矢量在d-q坐标系下角度：
Figure PCTCN2022086967-appb-000007
根据图4所示，电机转子位置角即为d坐标轴与α坐标轴的夹角。第四次短路电流矢量在d-q坐标系下的角度减去第四次短路电流矢量在α-β坐标系下的角度即为初始转子位置角度
Figure PCTCN2022086967-appb-000008
Figure PCTCN2022086967-appb-000009
至此HSPMSM的初始转速和转子位置角全部被计算而出，将其作为MRAS无位置算法的初始值即可完成电机的飞启：
Figure PCTCN2022086967-appb-000010
在MATLAB/Simulink中建立了HSPMSM的无位置控制模型，图5和图6给出了飞启过程中的电机转速和电机转子位置角波形，图7给出了飞启过程中的电机电流波形，图8为飞启过程中的四组短路电流。可见，在中高速区间，采用基 于短路电流矢量法的飞启策略可以减小飞启时的涌流并可靠地实现带速HSPMSM的飞启。
在较低转速时，将两相静止坐标系下的电机电压方程写成：
Figure PCTCN2022086967-appb-000011
其中，L Δ＝L d-L q，E＝ω eψ f+(L d-L q)(i dω e-pi q)，e α＝-Esinθ e，e β＝Ecosθ e，等式(9)右边最后一项即为拓展反电动势。对于凸极率很小的电机来说ω eL Δ＜＜R s，因此ω eL Δ项可忽略不计。拓展反电势的辨识算法为：
Figure PCTCN2022086967-appb-000012
其中，其中
Figure PCTCN2022086967-appb-000013
和
Figure PCTCN2022086967-appb-000014
分别代表α-β坐标系下的指令电压。在凸极率小于等于2的情况下，反电势辨识值近似等于真实的电机反电势。
基于式(10)的拓展反电势观测器的图9所示，整个系统的结构图如图10所示。
图11和图12给出了在较低转速情况下基于反电势解耦法的飞启过程中的电机转速和电机转子位置角波形，图13给出了飞启过程中的电机电流波形。可见，在较低转速区间，采用基于反电势解耦法的飞启策略可以减少因电流采样误差所导致的计算误差、抑制飞启时的涌流并可靠地实现带速HSPMSM的飞启。
在零低速的情况下，采用I-F控制将零低速电机拖至给定转速。图14为I-F控制的系统框图。I-F启动中，给定电流矢量的幅值设计值为：
Figure PCTCN2022086967-appb-000015
其中，T l为负载转矩，p为极对数。飞启操作时无负载，因此i q设定值大于0即可，然而为了保证可靠启动，为了获得足够的电磁转矩，将i q设定为额定电流值大小。
其次，给定转速以斜坡信号的形式加速，给定的斜率即为电流矢量的旋转加速度，过大的加速度将导致实际轴难以与旋转电流同步，而过小的加速度导致电机的加速缓慢。在空载飞启时，电机给定转速的加速度设计为：
Figure PCTCN2022086967-appb-000016
其中，J为转动惯量，依据给定转速可以得到电机的斜坡加速时间。针对电机从加速区切换至稳定运行区的转速波动问题，采用基于有功波动的角度补偿。补偿算法如下：
Δω i＝-HPF(u αi α+u βi β)      (13)
图15和图16给出了在零低速情况下基于I-F控制的飞启过程中的电机转速和电机转子位置角波形，图17给出了飞启过程中的电机电流波形。可见，零低速区间，由于反电势十分微弱，不能采用基于反电势的短路电流矢量法以及反电势解耦法，因而在零低速时采用基于I-F控制的飞启策略可以减小飞启时的涌流并可靠地实现带速HSPMSM的飞启。

Claims (10)

  1. 一种高速永磁同步电机全速域飞启方法，其特征在于，
    控制逆变器向定子绕组侧注入测试短路脉冲；
    采集定子绕组短路电流，并根据定子绕组短路电流矢量幅值判定带速电机初始转速所处的转速区间；
    带速电机的初始转速处于中高速区间时，采用四次短路电流矢量法计算初始转速及转子初始位置角，依据计算得到的初始转速及转子初始位置角对电机进行无位置控制；
    带速电机的初始转速处于低速区间时，采用反电势解耦法获取矢量控制指令信号，依据矢量控制的指令信号控制定子绕组侧逆变器，在检测到定子绕组产生电流后对电机进行无位置控制；
    带速电机的初始转速处于零低速区间时，采用恒流频比控制法将带速电机拖动至给定转速，依据给定转速及其对应转子位置角对电机进行无位置控制。
  2. 根据权利要求1所述一种高速永磁同步电机全速域飞启方法，其特征在于，
    定子绕组短路电流矢量幅值大于额定电流的40％时，则判定带速电机初始转速处于中高速区间；
    定子绕组短路电流矢量幅值小于额定电流的40％且大于额定电流的3％时，则判定带速电机初始转速处于低速区间；
    定子绕组短路电流矢量幅值小于额定电流的3％时，则判定带速电机初始转速处于零低速区间。
  3. 根据权利要求1所述一种高速永磁同步电机全速域飞启方法，其特征在于，所述无位置控制通过基于电流型模型参考自适应算法实现。
  4. 根据权利要求1所述一种高速永磁同步电机全速域飞启方法，其特征在于，采用四次短路电流矢量法计算初始转速及转子初始位置角的具体方法为：对电机电子绕组进行四次短路操作，并采样每次短路操作获取的短路电流矢量，依据两个短路电流矢量在空间上的角度差以及两次短路操作的时间间隔确定带速电机的初始转速，根据四次短路操作获取的短路电流矢量求得三组初始转速，在各组初始转速满足阈值要求时，将各组初始转速的均值作为初始转速，依据初始转速求解出最后一个短路电流矢量在旋转坐标系下的角度，依据最后一个短路电流矢量在旋转坐标系下的角度与最后一个短路电流矢量在静止坐标系下的角度的差值求得转子初始位置角。
  5. 根据权利要求1所述一种高速永磁同步电机全速域飞启方法，其特征在于，采用反电势解耦法获取矢量控制指令信号的具体方法为：将两相静止坐标系下的给定电流以及电机角度均置为0，通过扰动观测法获取电机拓展反电势的观测值，对拓展反电势的观测值进行前馈解耦得到矢量控制的指令信号。
  6. 根据权利要求1所述一种高速永磁同步电机全速域飞启方法，其特征在于，所述给定转速为以电流矢量旋转加速度为斜率的斜坡信号，所述电流矢量旋转加速度满足如下约束：
    Figure PCTCN2022086967-appb-100001
    其中，a r为电流矢量旋转加速度，p为极对数，ψ f为转子磁链，i q为旋转坐标下的给定q轴电流，J为转动惯量。
  7. 根据权利要求5所述一种高速永磁同步电机全速域飞启方法，其特征在于，通过扰动观测器获取电机拓展反电势的观测值的表达式为：
    Figure PCTCN2022086967-appb-100002
    Figure PCTCN2022086967-appb-100003
    其中，
    Figure PCTCN2022086967-appb-100004
    为电机反电势观测值的α轴分量、β轴分量，
    Figure PCTCN2022086967-appb-100005
    为两相静止坐标系下的指令电压，
    Figure PCTCN2022086967-appb-100006
    p为极对数，i α、i β为两相静止坐标系下的电流。
  8. 根据权利要求6所述一种高速永磁同步电机全速域飞启方法，其特征在于，依据给定转速及其对应转子位置角对电机进行无位置控制的具体方法为：将旋转坐标下的给定q轴电流幅值设置为额定电流，并将旋转坐标系下的给定d轴电流置0，以给定转速的积分值作为对应的转子位置角，对定子绕组电流采样值进行CLARK变换以及PARK变换后得到两相旋转坐标系下的实际电流，依据旋转坐标系下的给定电流对实际电流进行基于矢量控制的闭环控制，用于坐标变换的转子位置角度为给定转速积分值。
  9. 根据权利要求6所述一种高速永磁同步电机全速域飞启方法，其特征在于，对给定转速进行基于有功波动的补偿，给定转速补偿值Δω i为Δω i＝-HPF(u αi α+u βi β)，其中，u α、u β为两相静止坐标系下的电压，i α、i β为两相静止坐标系下的电流。
  10. 根据权利要求2所述一种高速永磁同步电机全速域飞启方法，其特征在于，中高速区间与低速区间的分界转速值ω ec为ω ec＝0.4I NL q/T testψ f，其中，I N为额定电流，L q为q轴电感，T test为测试短路脉冲的时长，ψ f为转子磁链。
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