WO2022011833A1 - 一种三相逆变器及三相逆变器的控制方法 - Google Patents

一种三相逆变器及三相逆变器的控制方法 Download PDF

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Publication number
WO2022011833A1
WO2022011833A1 PCT/CN2020/117032 CN2020117032W WO2022011833A1 WO 2022011833 A1 WO2022011833 A1 WO 2022011833A1 CN 2020117032 W CN2020117032 W CN 2020117032W WO 2022011833 A1 WO2022011833 A1 WO 2022011833A1
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Prior art keywords
current
bridge arm
phase
output
switch tube
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PCT/CN2020/117032
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English (en)
French (fr)
Inventor
吴坚
祁飚杰
杨永春
罗宇浩
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浙江昱能科技有限公司
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Priority to BR112023000692A priority Critical patent/BR112023000692A2/pt
Priority to EP20945485.9A priority patent/EP4184773A1/en
Priority to US18/016,353 priority patent/US20230344362A1/en
Publication of WO2022011833A1 publication Critical patent/WO2022011833A1/zh

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/083—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the ignition at the zero crossing of the voltage or the current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815—Resonant converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53873—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current with digital control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/539—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency
    • H02M7/5395—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters with automatic control of output wave form or frequency by pulse-width modulation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/12—Arrangements for reducing harmonics from ac input or output
    • H02M1/126—Arrangements for reducing harmonics from ac input or output using passive filters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/44—Circuits or arrangements for compensating for electromagnetic interference in converters or inverters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present application relates to the technical field of industrial robots, and in particular, to a three-phase inverter and a control method of the three-phase inverter.
  • Inverter is a device that converts direct current into alternating current. It consists of inverter bridge, control logic and filter circuit, such as the common grid-connected inverter for photovoltaic power generation.
  • the realization of soft switching can reduce the switching loss, improve the conversion efficiency, reduce the stress of the switch tube, improve the reliability of the converter, and reduce the EMI noise.
  • Fig. 1 is the circuit diagram of the first three-phase inverter in the prior art
  • Fig. 2 is the circuit diagram of the second three-phase inverter in the prior art
  • Fig. 3 is the third three-phase inverter in the prior art Circuit diagram of a phase inverter.
  • Fig. 1 is the topology structure of the most common three-phase three-bridge-arm inverter at present, and each of the three bridge arms is provided with two switch tubes (switch tube Q' 1 , switch tube Q' 2 , switch tube Q' 2 in Fig. 1 , Switch tube Q' 3 , switch tube Q' 4 , switch tube Q' 5 , switch tube Q' 6 ), and one end of the output inductor (L' 1 , L' 2 , L' 3 in FIG. 1 ) are respectively connected to each The midpoint of the bridge arm, the other end of the output inductor is connected to the filter capacitor (C' 1 , C' 2 , C' 3 in Figure 1 ) correspondingly.
  • the inverter of this topology works in continuous mode and cannot directly achieve soft switching.
  • a filter loop can be formed only when the switch tube under the corresponding bridge arm is turned on, thereby filtering out the high-frequency switching harmonic current, while the power tube under the bridge arm is turned on.
  • the high-frequency switch harmonic current will be output together with the grid current, and the filter circuit cannot filter out the high-frequency switch harmonic current well. The proportion will be very large, resulting in low quality of the output grid current waveform.
  • the inverter of this topology can only work in the continuous mode, when the starting power is large, the inverter of this topology has high requirements on the output inductance, filter capacitor and switch tube, which is difficult to implement. , therefore, the common inverters of this type are inverters that work in continuous mode.
  • Figures 2 and 3 show the topology of the current three-phase soft-switching converter.
  • the high-frequency switching is filtered out by adding an auxiliary resonant network on the DC side (V dc ) or on the AC side. harmonic currents.
  • V dc DC side
  • harmonic currents harmonic currents.
  • the circuit of this topology structure is relatively complex, and an additional switch tube needs to be added to the auxiliary resonance network to control the resonance, which increases the hardware and software overhead of the entire system.
  • the purpose of the present invention is to provide a three-phase inverter and a control method of the three-phase inverter, which can realize the soft switching of the three-phase inverter more simply, and effectively filter out the harmonic current pollution of the high-frequency switch.
  • the present invention provides a three-phase inverter, comprising: three bridge arms, a resonance capacitor, an output inductor, a filter capacitor, and a controller;
  • each of the bridge arms is provided with two switch tubes, and the controller is respectively connected to the control end of each switch tube; the first end of one of the switch tubes of the bridge arm is connected to the positive pole of the DC bus. connection, the second end of the other switch tube of the bridge arm is connected to the negative pole of the DC bus, and the second end of the switch tube of the bridge arm is connected to the other end of the bridge arm.
  • the first end of the switch tube is connected as the midpoint of the bridge arm;
  • One of the bridge arms corresponds to two of the resonant capacitors, one of the output inductors and one of the filter capacitors, the first end of one of the resonant capacitors is connected to the positive pole of the DC bus, and the other of the resonant capacitors is connected to the positive pole of the DC bus.
  • the second end is connected to the negative pole of the DC bus, and the second end of one of the resonant capacitors and the first end of the other resonant capacitor are connected to the midpoint and the first end of the output inductor, so the second end of the output inductor is connected to the first end of the filter capacitor;
  • the second ends of the filter capacitors are connected to each other.
  • the second ends of each of the filter capacitors are grounded.
  • the controller is specifically a digital signal processor or a field programmable gate array.
  • the present invention also provides a control method for a three-phase inverter, based on the controller described in any one of the above, including:
  • the main control switch tube of the bridge arm corresponding to the current phase is controlled to be turned on;
  • the synchronous switch tube When the body diode of the synchronous switch tube starts to conduct freewheeling, the synchronous switch tube is controlled to be turned on to perform synchronous rectification, and when the current value of the output inductor of the current corresponding bridge arm drops to zero, controlling the synchronous switch to turn off;
  • the main control switch tube is the switch tube that is first turned on by the current corresponding bridge arm
  • the synchronous switch tube is another switch tube of the current corresponding bridge arm.
  • the fundamental wave component of the output inductance of the bridge arm corresponding to the current is specifically controlled to be a power frequency sine wave, so that all The output current of the current phase is a sine wave.
  • the fundamental wave component of the output inductance of the current corresponding bridge arm is a power frequency sine wave, so that the output current of the current phase is a sine wave, specifically:
  • the switching period is the period during which the current phase generates grid-connected current.
  • the fundamental wave component of the output inductance of the current corresponding bridge arm is a power frequency sine wave, so that the output current of the current phase is a sine wave, specifically:
  • the switching tube is controlled by frequency conversion, so that the output current of the current phase is a sine wave.
  • the preset reference current satisfies the following formula:
  • I ref is the preset reference current
  • T on is the turn-on time of the main control switch
  • T off is the turn-on time of the synchronous switch
  • T is the switching period
  • I out sin( ⁇ t+ ⁇ ) is the average equivalent current value of the current phase of the output inductance of the corresponding bridge arm in one switching cycle
  • ⁇ is the power frequency of the grid
  • ⁇ is the phase difference between the preset reference current and the grid voltage .
  • the power factor adjustment is performed by controlling the phase difference.
  • the adjusting the power factor by controlling the phase difference specifically includes:
  • phase difference is controlled to be zero
  • phase difference is controlled not to be zero, and reactive power is adjusted by controlling the phase difference.
  • the three-phase inverter provided by the present invention includes: three bridge arms, a resonant capacitor, an output inductor, a filter capacitor, and a controller, each bridge arm is provided with two switch tubes, and one end of the two switch tubes is connected to the DC The positive and negative poles of the bus are connected, and the other end is connected as the midpoint of the bridge arm.
  • the midpoint of each bridge arm is connected to two resonant capacitors and output inductors.
  • the other end of the resonant capacitor is connected to the positive and negative poles of the DC bus, and the other end of the output inductor is connected Filter capacitor, the other end of the filter capacitor is grounded.
  • the output inductor and the resonant capacitor are used to form a resonant unit without adding a switch tube, and a resonant network is established for the realization of soft switching.
  • circuit components are saved and the circuit components are simplified.
  • the control flow reduces the hardware and software overhead of the system.
  • the invention also provides a control method for a three-phase inverter.
  • the switching tube is turned on at zero voltage by using resonance, and synchronous rectification is realized at the same time.
  • the switching loss of the tube and the conduction loss of the freewheeling diode in the switching tube improve the conversion efficiency and reduce the electromagnetic noise caused by the switching tube.
  • FIG. 1 is a circuit diagram of a first three-phase inverter in the prior art
  • FIG. 2 is a circuit diagram of a second type of three-phase inverter in the prior art
  • FIG. 3 is a circuit diagram of a third three-phase inverter in the prior art
  • FIG. 4 is a circuit diagram of a three-phase inverter provided by an embodiment of the present application.
  • FIG. 5 is a flowchart of a control method of a three-phase inverter provided by an embodiment of the present application
  • FIG. 6 is a schematic diagram of six working modes of a three-phase inverter provided by an embodiment of the present application.
  • FIG. 7 is a schematic diagram of first to third working modes of a three-phase inverter provided by an embodiment of the present application.
  • FIG. 8 is a schematic diagram of the driving waveform of each bridge arm in the first to third working modes provided by an embodiment of the present application.
  • FIG. 9 is a waveform diagram of a bridge arm midpoint voltage and an output inductor current corresponding to a phase a bridge arm according to an embodiment of the present application.
  • FIG. 10(a) provides a schematic diagram of a first control state of a phase-a bridge arm according to an embodiment of the present application
  • FIG. 10(b) provides a schematic diagram of a second control state of the a-phase bridge arm according to an embodiment of the present application
  • FIG. 10(c) provides a schematic diagram of a third control state of the a-phase bridge arm according to an embodiment of the present application.
  • FIG. 10(d) provides a schematic diagram of a fourth control state of the a-phase bridge arm according to an embodiment of the present application.
  • FIG. 10(e) provides a schematic diagram of a fifth control state of the a-phase bridge arm according to an embodiment of the present application
  • FIG. 11 is a schematic diagram of a control effect of a switch tube provided by an embodiment of the present application.
  • FIG. 12 is a schematic diagram of a waveform during full active power output provided by an embodiment of the present application.
  • FIG. 13 is a schematic diagram of a waveform when an incomplete active power output is provided according to an embodiment of the present application.
  • the core of the present application is to provide a three-phase inverter and a control method of the three-phase inverter, which can realize the soft switching of the three-phase inverter more simply, and effectively filter out the harmonic current pollution of the high-frequency switching.
  • FIG. 4 is a circuit diagram of a three-phase inverter provided by an embodiment of the present application.
  • the three-phase inverter provided by the embodiment of the present application includes: three bridge arms, a resonance capacitor, an output inductor, a filter capacitor, and a controller;
  • each bridge arm is provided with two switch tubes, and the controller is connected to the control end of each switch tube respectively; the first end of one switch tube of the bridge arm is connected to the positive pole of the DC bus, and the other switch tube of the bridge arm is connected to the positive pole of the DC bus. The second end is connected to the negative pole of the DC bus, and the second end of one switch tube of the bridge arm is connected to the first end of the other switch tube of the bridge arm as the midpoint of the bridge arm;
  • One bridge arm corresponds to two resonant capacitors, one output inductor and one filter capacitor, the first end of one resonant capacitor is connected to the positive pole of the DC bus, the second end of the other resonant capacitor is connected to the negative pole of the DC bus, and one resonant capacitor The second end of the other resonant capacitor is connected to the midpoint and the first end of the output inductor, and the second end of the output inductor is connected to the first end of the filter capacitor;
  • the second ends of the filter capacitors are connected to each other.
  • the two switches of the bridge arm corresponding to a of the three-phase inverter as Q 1 and Q 2 , the corresponding resonance capacitors as C 1 and C 2 , the output inductance as L 1 , and the filter capacitor as C a ;
  • the two switches of the corresponding bridge arm are Q 3 and Q 4 respectively , the corresponding resonant capacitors are C 3 and C 4 , the output inductance is L 2 , and the filter capacitor is C b ;
  • the switch tubes are Q 5 and Q 6 respectively , the corresponding resonance capacitors are C 5 and C 6 , the output inductance is L 3 , and the filter capacitor is C c .
  • a first end of the switch Q, Q 3 of the switch 1 of the first end and the first end of the switching transistor Q 5 is connected to the positive electrode of the DC V dc bus, a second terminal of the switch Q 2 is , switch Q and second ends of the second switch transistor 4 and Q 6 is connected to the negative DC bus, a second end of the switch Q 1 and the Q switch 2 is connected to a first end of a corresponding arm midpoint, the second end of the switch Q 3 and the transistor Q b is connected to the midpoint of the corresponding end of the first arm 4, a second switch terminal of Q 5 and a first end of the switch Q 6 c is connected to the midpoint of the corresponding arm; an output terminal of the first inductor L 1, a second terminal of the resonant capacitor C, a midpoint of the first end of the resonant capacitor C 2 and a corresponding arm of the connector; a first end of the output inductor L 2, resonant capacitor C 3 to the second end, a first end midpoint of
  • Continuous mode and discontinuous mode can be determined according to whether the current flowing through the output inductor is zero during the high-frequency switching (on and off) time of the switch, that is, when the switch of the next bridge arm is turned on, if the previous The current of the output inductor in a bridge arm has been reduced to zero, and it is in discontinuous mode at this time, otherwise it is in continuous mode.
  • the three-phase converter in the prior art needs to add auxiliary resonant networks to the main circuit to realize soft switching. Whether it is added on the DC side or the AC side, these auxiliary resonant network circuits are complex and often have switch tubes. In order to achieve resonance, The working state has become numerous, increasing the difficulty of control. At the same time, these switch tubes require additional auxiliary power supply and drive circuit, which increases the cost of the entire system.
  • the embodiment of the present application does not add an additional switch tube to assist resonance, and uses the output inductance as a resonant inductance to establish a resonant network for the realization of soft switching, so that the output inductance and the resonant capacitor naturally resonate, creating conditions for the zero-voltage turn-on of the switch tube.
  • the switching loss can be reduced, the conversion efficiency can be improved, and the EMI noise can be reduced, and the stress of the switching tube can be reduced, and the system reliability can be improved. Since there is no need to set a larger output inductance, the multi-machine parallel system (the output terminals of multiple three-phase inverters are connected to the same interface of the power grid) can also work stably.
  • the controller may adopt a digital signal processor (Digital Signal Process, DSP) or a Field Programmable Gate Array (Field Programmable Gate Array, FPGA) or other controllers with similar functions.
  • DSP Digital Signal Process
  • FPGA Field Programmable Gate Array
  • the switch tube can be a MOS tube or an IGBT or a thyristor.
  • the first end of the filter capacitor at the output end of each bridge arm of the three-phase inverter is connected with the output inductor of the bridge arm, and the second end is connected together, as shown in FIG. 1 .
  • the high-frequency switching harmonic current on the inductance is fixed.
  • the proportion of high-frequency switching harmonic current to the total current gradually decreases. It is understandable that when the output power is low, the high-frequency switching harmonic current will account for a large proportion.
  • TDD Total Harmonic Distortion
  • the first end of the filter capacitor of each bridge arm is connected to the output inductance of the bridge arm, and the second end is directly connected (grounded) to the negative pole of the input DC bus.
  • the output filter is mainly a filter capacitor, and the output filter of a traditional three-phase inverter is an output inductor and a filter.
  • the filter capacitor compared with the traditional three-phase inverter, the three-phase inverter provided by the embodiment of the present application is not limited by the output inductance and does not need to increase the inductance.
  • a new filter circuit is formed for the high-frequency switching harmonic current through the filter capacitor. No matter whether the power tube under the bridge arm is turned off or on, the high-frequency switch harmonic current can completely flow back to the input DC from the filter capacitor through the filter circuit.
  • the busbar that is to say, the filter circuit can filter out the high-frequency switching harmonic current well, so as to output the grid current without the pollution of the high-frequency switching harmonic current, even when outputting low power, it can also output high-quality grid. current.
  • the switching frequency of the traditional three-phase inverter is generally below 20 kHz, and the high-frequency switching harmonic current is filtered out by the filter capacitor in the embodiment of the present application.
  • the frequency can reach more than 200kHz, and the quality of the grid current waveform can be guaranteed.
  • the present application also discloses a control method of a three-phase inverter corresponding to the above-mentioned three-phase inverter.
  • FIG. 5 is a flowchart of a control method of a three-phase inverter provided by an embodiment of the application
  • FIG. 6 is a schematic diagram of six working modes of a three-phase inverter provided by an embodiment of the application
  • FIG. 7 A schematic diagram of the first to third working modes of a three-phase inverter provided by an embodiment of the present application
  • FIG. 8 is a driving waveform of each bridge arm in the first to third working modes provided by an embodiment of the present application
  • Fig. 9 is a waveform diagram of the bridge arm midpoint voltage and output inductor current corresponding to a phase a bridge arm provided by an embodiment of the application
  • Fig. 10(a) provides a phase a bridge arm of an embodiment of the application A schematic diagram of the first control state
  • FIG. 10 provides a phase a bridge arm of an embodiment of the application A schematic diagram of the first control state
  • FIG. 10(b) is a schematic diagram of a second control state of the a-phase bridge arm according to an embodiment of the application;
  • FIG. 10(c) provides a third control state of the a-phase bridge arm according to an embodiment of the application A schematic diagram of a control state;
  • FIG. 10(d) provides a schematic diagram of a fourth control state of the a-phase bridge arm for this embodiment of the application;
  • FIG. 10(e) provides a fifth control state of the a-phase bridge arm for this embodiment of the application schematic diagram.
  • FIGS. 10( a ) to 10 ( e ) are schematic diagrams of various working states of a three-phase inverter implementing soft switching provided by the present application.
  • control method of the three-phase inverter includes:
  • the main control switch tube is the switch tube that is first turned on by the current corresponding bridge arm
  • the synchronous switch tube is another switch tube of the current corresponding bridge arm.
  • an embodiment of the present application provides a three-phase output of a three-phase inverter in a discontinuous mode.
  • the zero-crossing point of the three-phase output current is used as the working mode transition point.
  • the three-phase inverter provided by the embodiment of the present application can be divided into six working modes. As shown in Figure 6, taking the zero-crossing point of the phase a current ia as the start and end point as an example, the working modes 1, 2, and 3 are the positive current half cycles of ia, and the working modes 4, 5, and 6 are the negative currents of ia half cycle.
  • each phase has two switches working in each working mode, which are respectively defined as the main control switch and the synchronous switch.
  • the main control switch tube is responsible for controlling the current waveform of the corresponding output inductor, and the synchronous switch tube is turned on when the main control switch tube is turned off, instead of the body diode freewheeling to reduce losses.
  • step S501 since the output inductor works in the discontinuous mode, the switch tube satisfies zero current at any time, and the soft switching condition is that the voltage across the switch tube is zero.
  • the controller controls the main-control switch to turn on when the zero-voltage detection circuit detects that the voltage across the main-control switch is zero.
  • step S502 after the main control switch is turned on, the current of the corresponding output inductor increases, and the current of the output inductor of the current corresponding bridge arm is detected by setting a current detection circuit.
  • the main control switch tube is controlled to be turned off.
  • working modes of each switch tube in working modes 1, 2, and 3 are shown in Figure 7, and working modes 4, 5, and 6 can be And so on.
  • the three bridge arms of the switch tubes corresponding to the a-phase, b-phase and c-phase are controlled to work independently.
  • the switch tube Q 1 , the switch tube Q 3 , and the switch tube Q 5 are the main control switch tubes, and the switch tube Q 2 , the switch tube Q 4 , and the switch tube Q 6 are synchronous switch tubes.
  • the switch tube Q 2 , the switch tube Q 4 , and the switch tube Q 6 are the main control switch tubes, and the switch tube Q 1 , the switch tube Q 3 , and the switch tube Q 5 are synchronous switch tubes.
  • FIG. 10( a ) The control state in the period from t 0 to t 1 in FIG. 9 is shown in FIG. 10( a ).
  • the switch transistor Q 1 When the midpoint voltage U a of the bridge arm of phase a resonates to the peak at time t 0 , the switch transistor Q 1 is turned on. At this time, due to the resonance The potential at both ends of the capacitor C 1 is negative and positive and a back pressure appears, the body diode of the switch Q 1 is turned on, and the voltage difference between the DC bus voltage V dc at both ends of the switch Q 1 and the midpoint voltage U a of the a-phase bridge arm is about zero , to achieve zero-voltage turn-on.
  • FIG. 10( b ) The control state in the period from t 1 to t 2 in FIG. 9 is shown in FIG. 10( b ), when the peak value of the current i L1 of the output inductor L 1 reaches the preset reference current value or the turn-on time of the switch Q 1 reaches the preset time
  • the switch tube Q 1 is turned off, the output inductor L 1 freewheels, and the body diode of the switch tube Q 2 is turned on.
  • FIG 9 t control state period 3 ⁇ t 4 in FIG. 10 (d) when flowing through the switching transistor Q 2
  • the freewheeling current is reduced to zero, the controller of the Q switch drive signal set low 2 the switch Q 2 is turned off.
  • the power grid charges the output inductor L 1 and the resonant capacitor C 2 in the reverse direction, and the current i L1 reverses.
  • the output inductor L 1 releases energy in the reverse direction, and the voltage of the output inductor L 1 reverses, making the Q switch body diode conduction 1, so a phase leg is clamped midpoint voltage U a DC bus voltage V dc, the inductance L so as to output an energy back to the DC bus.
  • FIG. 10(e) The control state from t 4 to t 5 in FIG. 9 is shown in FIG. 10(e), when the energy of the output inductor L 1 is released, the output inductor L 1 , the resonant capacitor C 2 , the filter capacitor Ca and the output inductor L 1 , the resonant capacitor C 1 , the DC bus and the filter capacitor C a form two resonant circuits, and natural resonance occurs, wherein the filter capacitor C a is clamped by the grid and does not participate in the resonance.
  • the switching transistor Q 1 is turned on at the peak to enter the next switching cycle, or the switching transistor Q 1 is turned on at the trough to enter the working mode 4 .
  • control modes of the corresponding master control switch tubes and synchronous switch tubes can be set with reference to FIG. 9 and FIG. 10(a) to FIG. 10(e). Repeat.
  • the fundamental component of the output inductance of the current corresponding bridge arm is specifically controlled to be a power frequency sine wave, so that the output current of the current phase is a sine wave.
  • the average equivalent current value in each switching period can be controlled to change sinusoidally, so that the output current of the current phase is a sine wave; wherein the switching period is the period during which the current phase generates grid-connected current.
  • the switching tube can also be controlled by frequency conversion, so that the output current of the current phase is a sine wave.
  • FIG. 11 is a schematic diagram of the control effect of a switch tube provided by an embodiment of the present application
  • FIG. 12 is a schematic diagram of a waveform when a complete active power output is provided by an embodiment of the present application; Schematic diagram of the waveform during active output.
  • the current output by the output inductor is controlled by the main control switch tube, and the switch node of the main control switch tube needs to refer to the preset reference current.
  • the embodiment of the present application provides a specific implementation for controlling the main control switch tube with a preset reference current as a reference. As shown in FIG. 11 , if the reference current method is used to control the main control switch tube after the main control switch tube is turned on , the corresponding output inductor current i L increases. A current detector circuit for detecting an output current corresponding to the inductor current i L arm, at equal output inductor current i L of the peak current with a preset reference value I ref, the main control switch is turned off, this opening time Denoted as t on .
  • the switch tube After the main control switch tube is turned off, the freewheeling starts, the current i L decreases, and after the t off time drops to zero, the switch tube is turned on after a period of dead time.
  • the sum of t on , t off and dead time is one switching period T, and 1/T is the switching frequency.
  • the reference value of t on is calculated in advance. After the main control switch is turned on, after the timer counts t on, the peak value of the current i L of the output inductor is considered to be the same as the preset value. The values of the reference currents I ref are assumed to be equal.
  • the switching tube is controlled by frequency conversion, so that the output current of the current phase is a sine wave, that is, it is controlled in different phases, and the value of the switching period T is different, so that the output current of the current phase is sine wave.
  • the number of peaks or troughs after the main control switch tube is turned off can be specified to enter the next switching cycle, that is, the switching frequency is controlled by controlling the length of the dead time.
  • Figure 11 is a simplified drawing method.
  • the preset reference current satisfies the following formula:
  • I ref is the preset reference current
  • T on is the turn-on time of the main control switch
  • T off is the turn-on time of the synchronous switch
  • T is the switching period
  • I out sin( ⁇ t+ ⁇ ) is the current corresponding bridge
  • ⁇ is the power frequency of the grid
  • ⁇ is the phase difference between the preset reference current and the grid voltage.
  • the average equivalent current value in each switching cycle is controlled to be sinusoidal, so that the output current of the current phase is a sine wave, but the waveform of the preset reference current is not a sine wave, that is, the output inductance.
  • the average equivalent current value is different from the change of the preset reference current.
  • the power factor adjustment is performed by controlling the phase difference.
  • the control phase difference when the complete active power output control is performed, the control phase difference is zero; when the incomplete active power output control is performed, the control phase difference is not zero, and the reactive power is adjusted by controlling the phase difference.
  • the output current and the grid phase are exactly the same.
  • the phases of the output current I ref and the grid voltage U grid are different.
  • the phases of the preset reference current I ref and the grid voltage U grid are different, that is, ⁇ 0, and the zero-crossing points are staggered.
  • the phase of the output current i L and the preset reference current I ref is the same, then the phase of the output current i L and the grid voltage U grid is different, that is, ⁇ 0, so as to realize the output reactive power.

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Abstract

一种三相逆变器及三相逆变器的控制方法，通过在三个桥臂的中点与直流母线的正负极之间设置谐振电容，和输出电感组成谐振单元，无需增加开关管，为软开关的实现建立了谐振网络，相较于现有技术中三相逆变器的软开关实现方式，节约了电路器件，简化了控制流程，降低了系统的硬件和软件开销。通过控制该三相逆变器工作在断续模式下工作，利用谐振达到开关管零电压开通的状态，并且同时实现同步整流，降低了开关管的开关损耗和开关管内续流二极管的导通损耗，提高了转换效率的同时也减小了开关管所带来的电磁噪声，输出高质量的电网电流。

Description

一种三相逆变器及三相逆变器的控制方法
本申请要求于2020年7月15日提交中国专利局、申请号为202010681109.1、发明名称为“一种三相逆变器及三相逆变器的控制方法”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及工业机器人技术领域，特别是涉及一种三相逆变器及三相逆变器的控制方法。
背景技术
逆变器是将直流电转变成交流电的一种设备，它由逆变桥、控制逻辑和滤波电路组成，例如常见的用于光伏发电的并网逆变器。
在逆变器中，实现软开关能减小开关损耗，提高转换效率的同时减小开关管应力提高变流器可靠性、减小EMI噪声。
图1为现有技术中的第一种三相逆变器的电路图；图2为现有技术中的第二种三相逆变器的电路图；图3为现有技术中的第三种三相逆变器的电路图。
图1为目前最为常见的三相三桥臂的逆变器的拓扑结构，三个桥臂上各设有两个开关管(如图1中的开关管Q′ 1、开关管Q' 2、开关管Q′ 3、开关管Q' 4、开关管Q′ 5、开关管Q' 6)，输出电感(如图1中的L' 1、L' 2、L' 3)的一端分别连接各桥臂的中点，输出电感的另一端与滤波电容(如图1中的C′ 1、C' 2、C′ 3)对应连接。该拓扑结构的逆变器工作在连续模式下，无法直接实现软开关。这种拓扑结构的逆变器在进行输出滤波时，只有当对应的桥臂下方的开关管导通时，才能形成滤波回路，进而滤除高频开关谐波电流，而桥臂下方的功率管关断时，高频开关谐波电流就会与电网电流一起输出，滤波电路就不能很好地滤除高频开关谐波电流，当输出低功率时，高频开关谐波电流在总电流的占比就会很大，导致输出的电网电流波形质量低。此外，由于该拓扑结构的逆变器只能工作在连续模式下，在起步功率较大时，这种拓扑结构的逆变器对输出电感、滤波电容以及开关管的要求很高，实 现起来困难，因此，常见的这类逆变器都是工作在连续模式下的逆变器。
图2和图3为目前的三相软开关变流器的拓扑结构，在图1的基础上，具体通过在直流侧(V dc)或者在交流侧添加辅助谐振网络来实现滤除高频开关谐波电流。但这种拓扑结构电路较为复杂，需要在辅助谐振网络中增加额外的开关管控制谐振，增加了整个系统的硬件和软件开销。
提供一种更为简易地实现三相逆变器软开关的方案，是本领域技术人员需要解决的技术问题。
发明内容
本发明的目的是提供一种三相逆变器及三相逆变器的控制方法，可以更为简易地实现三相逆变器的软开关，有效滤除高频开关谐波电流污染。
为解决上述技术问题，本发明提供一种三相逆变器，包括：三个桥臂，谐振电容，输出电感，滤波电容，以及控制器；
其中，各所述桥臂均设有两个开关管，所述控制器分别与各所述开关管的控制端连接；所述桥臂的一个所述开关管的第一端与直流母线的正极连接，所述桥臂的另一个所述开关管的第二端与所述直流母线的负极连接，且所述桥臂的一个所述开关管的第二端与所述桥臂的另一个所述开关管的第一端连接为所述桥臂的中点；
一个所述桥臂对应两个所述谐振电容、一个所述输出电感和一个所述滤波电容，一个所述谐振电容的第一端与所述直流母线的正极连接，另一个所述谐振电容的第二端与所述直流母线的负极连接，且一个所述谐振电容的第二端与另一个所述谐振电容的第一端与所述中点及所述输出电感的第一端连接，所述输出电感的第二端与所述滤波电容的第一端连接；
各所述滤波电容的第二端相互连接。
可选的，各所述滤波电容的第二端均接地。
可选的，所述控制器具体为数字信号处理器或现场可编程门阵列。
为解决上述技术问题，本发明还提供一种三相逆变器的控制方法，基于上述任意一项所述的控制器，包括：
在当前相产生并网电流后，检测所述当前相达到软开关条件后，控制 与所述当前相对应的桥臂的主控开关管导通；
在所述当前相对应的桥臂的输出电感的电流峰值达到预设基准电流值时、或所述主控开关管的导通时间达到预设时间时，控制所述主控开关管关断；
当所述同步开关管的体二极管开始导通续流时，控制所述同步开关管导通以进行同步整流，并在所述当前相对应的桥臂的输出电感的电流值降为零时，控制所述同步开关管关断；
在所述当前相的死区时间内，所述当前相对应的桥臂的谐振电容、输出电感和滤波电容发生自然谐振时，当检测达到所述软开关条件后，返回所述控制与所述当前相对应的桥臂的主控开关管导通的步骤；
其中，所述主控开关管为所述当前相对应的桥臂首先开通的开关管，所述同步开关管为所述当前相对应的桥臂的另一个开关管。
可选的，在控制所述当前相对应的桥臂的开关管的开通与关断时，具体控制所述当前相对应的桥臂的输出电感的基波分量为工频正弦波，以使所述当前相的输出电流为正弦波。
可选的，所述控制所述当前相对应的桥臂的输出电感的基波分量为工频正弦波，以使所述当前相的输出电流为正弦波，具体为：
控制每个开关周期内所述平均等效电流值均呈正弦变化，以使所述当前相的输出电流为正弦波；
其中，所述开关周期为所述当前相产生并网电流的周期。
可选的，所述控制所述当前相对应的桥臂的输出电感的基波分量为工频正弦波，以使所述当前相的输出电流为正弦波，具体为：
对所述开关管进行变频控制，以使所述当前相的输出电流为正弦波。
可选的，所述预设基准电流满足下述公式：
Figure PCTCN2020117032-appb-000001
其中，I ref为所述预设基准电流，T on为所述主控开关管的开通时间，T off为所述同步开关管的开通时间，T为开关周期，I outsin(ωt+θ)为所述当前相对应的桥臂的输出电感的一个所述开关周期在所述当前相的平均等效电流 值，ω为电网工频，θ为所述预设基准电流和电网电压的相位差。
可选的，在控制所述当前相对应的桥臂的开关管的开通与关断时，通过控制所述相位差以进行功率因数调节。
可选的，所述通过控制所述相位差以进行功率因数调节，具体包括：
当进行完全有功输出控制时，控制所述相位差为零；
当进行非完全有功输出控制时，控制所述相位差不为零，并通过控制所述相位差以调节无功功率。
本发明所提供的三相逆变器，包括：三个桥臂，谐振电容，输出电感，滤波电容，以及控制器，各桥臂设有两个开关管，两个开关管的一端分别与直流母线的正、负极连接，另一端相连作为桥臂中点，各桥臂中点均连接两个谐振电容和输出电感，谐振电容的另一端分别连接直流母线的正、负极，输出电感另一端连接滤波电容，滤波电容另一端接地。通过输出电感和谐振电容组成谐振单元，无需增加开关管，为软开关的实现建立了谐振网络，相较于现有技术中三相逆变器的软开关实现方式，节约了电路器件，简化了控制流程，降低了系统的硬件和软件开销。
本发明还提供一种三相逆变器的控制方法，通过控制三相逆变器工作在断续模式下工作，利用谐振达到开关管零电压开通的状态，并且同时实现同步整流，降低了开关管的开关损耗和开关管内续流二极管的导通损耗，提高了转换效率的同时也减小了开关管所带来的电磁噪声。
附图说明
为了更清楚的说明本申请实施例或现有技术的技术方案，下面将对实施例或现有技术描述中所需要使用的附图作简单的介绍，显而易见地，下面描述中的附图仅仅是本申请的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动的前提下，还可以根据这些附图获得其他的附图。
图1为现有技术中的第一种三相逆变器的电路图；
图2为现有技术中的第二种三相逆变器的电路图；
图3为现有技术中的第三种三相逆变器的电路图；
图4为本申请实施例提供的一种三相逆变器的电路图；
图5为本申请实施例提供的一种三相逆变器的控制方法的流程图；
图6为本申请实施例提供的一种三相逆变器的六个工作模态的示意图；
图7为本本申请实施例提供的一种三相逆变器的第一至第三工作模态的示意图；
图8为本申请实施例提供的第一至第三工作模态下各桥臂的驱动波形的示意图；
图9为本申请实施例提供的一种a相桥臂对应的桥臂中点电压和输出电感电流的波形图；
图10(a)为本申请实施例提供一种a相桥臂第一控制状态的示意图；
图10(b)为本申请实施例提供一种a相桥臂第二控制状态的示意图；
图10(c)为本申请实施例提供一种a相桥臂第三控制状态的示意图；
图10(d)为本申请实施例提供一种a相桥臂第四控制状态的示意图；
图10(e)为本申请实施例提供一种a相桥臂第五控制状态的示意图；
图11是本申请实施例提供的一种开关管的控制效果示意图；
图12为本申请实施例提供的一种完全有功输出时的波形示意图；
图13为本申请实施例提供的一种非完全有功输出时的波形示意图。
具体实施方式
本申请的核心是提供一种三相逆变器及三相逆变器的控制方法，可以更为简易地实现三相逆变器的软开关，有效滤除高频开关谐波电流污染。
下面将结合本申请实施例中的附图，对本申请实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例仅仅是本申请一部分实施例，而不是全部的实施例。基于本申请中的实施例，本领域普通技术人员在没有做出创造性劳动前提下所获得的所有其他实施例，都属于本申请保护的范围。
图4为本申请实施例提供的一种三相逆变器的电路图。
本申请实施例提供的三相逆变器包括：三个桥臂，谐振电容，输出电感，滤波电容，以及控制器；
其中，各桥臂均设有两个开关管，控制器分别与各开关管的控制端连接；桥臂的一个开关管的第一端与直流母线的正极连接，桥臂的另一个开关管的第二端与直流母线的负极连接，且桥臂的一个开关管的第二端与桥臂的另一个开关管的第一端连接为桥臂的中点；
一个桥臂对应两个谐振电容、一个输出电感和一个滤波电容，一个谐振电容的第一端与直流母线的正极连接，另一个谐振电容的第二端与直流母线的负极连接，且一个谐振电容的第二端与另一个谐振电容的第一端与中点及输出电感的第一端连接，输出电感的第二端与滤波电容的第一端连接；
各滤波电容的第二端相互连接。
记三相逆变器的a相对应的桥臂的两个开关管分别为Q 1和Q 2，对应的谐振电容为C 1和C 2，输出电感为L 1，滤波电容为C a；b相对应的桥臂的两个开关管分别为Q 3和Q 4，对应的谐振电容为C 3和C 4，输出电感为L 2，滤波电容为C b；c相对应的桥臂的两个开关管分别为Q 5和Q 6，对应的谐振电容为C 5和C 6，输出电感为L 3，滤波电容为C c。
则如图4所示，开关管Q 1的第一端、开关管Q 3的第一端和开关管Q 5的第一端与直流母线V dc的正极连接，开关管Q 2的第二端、开关管Q 4的第二端和开关管Q 6的第二端与直流母线的负极连接，开关管Q 1的第二端和开关管Q 2的第一端连接为a相对应的桥臂的中点，开关管Q 3的第二端和开关管Q 4的第一端连接为b相对应的桥臂的中点，开关管Q 5的第二端和开关管Q 6的第一端连接为c相对应的桥臂的中点；输出电感L 1的第一端、谐振电容C 1的第二端、谐振电容C 2的第一端与a相对应的桥臂的中点连接；输出电感L 2的第一端、谐振电容C 3的第二端、谐振电容C 4的第一端与b相对应的桥臂的中点连接；输出电感L 3的第一端、谐振电容C 5的第二端、谐振电容C 5的第一端与从c相对应的桥臂的中点连接；输出电感L 1的第二端与滤波电容C a 的第一端连接，输出电感L 2的第二端与滤波电容C b的第一端连接，输出电感L 3的第二端与滤波电容C c的第一端连接；滤波电容C a的第二端、滤波电容C b的第二端和滤波电容C c的第二端相互连接。
连续模式和断续模式可以根据开关管在高频开关(开通和关断)的时间内，输出电感上流过的电流是否为零来确定，即在下一个桥臂的开关管导通时，若前一桥臂中的输出电感的电流已减小为零，此时为断续模式，否则为连续模式。
现有技术中的三相变流实现软开关需要在主电路中添加辅助谐振网络，不管是添加在直流侧或是交流侧，这些辅助谐振网络电路复杂，往往带有开关管，为了实现谐振，工作状态变得繁多，增加了控制难度。同时这些开关管需要额外增加辅助电源和驱动电路，升高了整个系统的成本。而本申请实施例没有添加额外的开关管辅助谐振，利用输出电感充当谐振电感，为软开关的实现建立了谐振网络，使输出电感和谐振电容自然谐振，为开关管零电压开通创造了条件，继而可以在降低开关损耗、提高转换效率、减小EMI噪声的同时减小开关管应力，提高系统可靠性。由于不需要再设置较大的输出电感，多机并联系统(多台三相逆变器的输出端子连接到电网的同一接口)也能够稳定工作。
在具体实施中，控制器可以采用数字信号处理器(Digital Signal Process，DSP)或现场可编程门阵列(Field Programmable Gate Array，FPGA)或其他功能类似的控制器。开关管可以采用MOS管或IGBT或晶闸管等。
现有技术中三相逆变器各个桥臂的输出端的滤波电容的第一端与该桥臂的输出电感连接，第二端连接在一起，即如图1所示。当输出低功率时，由于三相逆变器的母线电压和输出电网电压是固定的，当电感值确定后，无论输出的功率是多少，电感上的高频开关谐波电流就固定了。随着输出功率的增加，高频开关谐波电流占总电流的比例逐渐减小，可以理解的是，当输出低功率时，高频开关谐波电流占比就会很大，这个时候要保证并网电流的总谐波失真(Total Harmonic Distortion，THD)满足国家标准要求，必然需要增大输出电感以减小高频开关谐波电流，但是，增大电感又会带 来新的问题，例如多机并联系统容易发生谐振。
故如图4所示，在本申请实施例中，将各个桥臂的滤波电容的第一端与该桥臂的输出电感连接，第二端直接与输入直流母线的负极连接(接地)。可见本申请实施例改变了三相逆变器的输出滤波方式，提供了合适的滤波器以及滤波回路，则输出滤波器主要为滤波电容，传统三相逆变器的输出滤波器为输出电感和滤波电容，相对于传统三相逆变器，本申请实施例提供的三相逆变器就不在受限于输出电感，无需增大电感。通过滤波电容为高频开关谐波电流形成了新的滤波回路，无论桥臂下方的功率管是关断还是开通，高频开关谐波电流均可通过该滤波回路完全从滤波电容流回输入直流母线，也就是说，该滤波回路能够很好地滤除高频开关谐波电流，从而输出无高频开关谐波电流污染的电网电流，即使在输出低功率时，也可以输出高质量的电网电流。并且，相较于现有技术中的三相逆变器来说，也无需设置较大的滤波电容。
受拓扑结构的限制，传统三相逆变器开关频率一般在20kHz以下，而本申请实施例通过滤波电容滤除了高频开关谐波电流，则本申请实施例提供的三相逆变器的开关频率可以达到200kHz以上，且能够保证电网电流波形质量。
上文详述了三相逆变器对应的各个实施例，在此基础上，本申请还公开了与上述三相逆变器对应的三相逆变器的控制方法。
图5为本申请实施例提供的一种三相逆变器的控制方法的流程图；图6为本申请实施例提供的一种三相逆变器的六个工作模态的示意图；图7为本本申请实施例提供的一种三相逆变器的第一至第三工作模态的示意图；图8为本申请实施例提供的第一至第三工作模态下各桥臂的驱动波形的示意图；图9为本申请实施例提供的一种a相桥臂对应的桥臂中点电压和输出电感电流的波形图；图10(a)为本申请实施例提供一种a相桥臂第一控制状态的示意图；图10(b)为本申请实施例提供一种a相桥臂第二控制状态的示意图；图10(c)为本申请实施例提供一种a相桥臂第三控制状态的示意图；图10(d)为本申请实施例提供一种a相桥臂第四控 制状态的示意图；图10(e)为本申请实施例提供一种a相桥臂第五控制状态的示意图。
图10(a)～图10(e)为本申请所提供的一种实现软开关的三相逆变器的各个工作状态的示意图。
如图5所示，基于上述任意一项实施例中所述的控制器，本申请实施例提供的三相逆变器的控制方法包括：
S501：在当前相产生并网电流后，检测当前相达到软开关条件后，控制与当前相对应的桥臂的主控开关管导通。
S502：在当前相对应的桥臂的输出电感的电流峰值达到预设基准电流值时、或主控开关管的导通时间达到预设时间时，控制主控开关管关断。
S503：当同步开关管的体二极管开始导通续流时，控制同步开关管导通以进行同步整流，并在当前相对应的桥臂的输出电感的电流值降为零时，控制同步开关管关断。
S504：在当前相的死区时间内，当前相对应的桥臂的谐振电容、输出电感和滤波电容发生自然谐振时，返回步骤S501。
其中，主控开关管为当前相对应的桥臂首先开通的开关管，同步开关管为当前相对应的桥臂的另一个开关管。
在具体实施中，本申请实施例提供一种三相逆变器在断续模式下的三相输出。在本申请实施例中，无论是输出有功功率，还是输出无功功率，均是以三相输出电流过零点作为工作模态转换点。根据三相电流过零点，本申请实施例提供的三相逆变器可以分为六个工作模态。如图6所示，以a相电流ia的过零点为启动和终点为例，工作模态1、2、3为ia的正电流半周期，工作模态4、5、6为ia的负电流半周期。
本申请实施例中实现同步整流，每一相在每个工作模态有两个开关管工作，分别定义为主控开关管和同步开关管。其中，主控开关管负责控制对应的输出电感的电流波形，同步开关管在主控开关管关闭时开通，代替体二极管续流以减小损耗。
具体地，对于步骤S501来说，由于输出电感工作在断续模式下，任何时候开通开关管都满足零电流，则软开关条件为开关管两端电压为零。在 产生三相并网电流的其中一相并网电流时，控制器在通过零电压检测电路检测到主控开关管两端电压为零时控制主控开关管开通。
对于步骤S502来说，在主控开关管开通后，相应的输出电感的电流增加，通过设置电流检测电路的方式来检测当前相对应的桥臂的输出电感的电流，在当前相对应的桥臂的输出电感的电流峰值达到预设基准电流值时，控制主控开关管关断。
此外，还可以通过预先计算在主控开关管开通后，当前相对应的桥臂的输出电感的电流峰值达到预设基准电流值的时间记为预设时间，当主控开挂管的导通时间达到预设时间时，控制主控开关管关断。
考虑到交流电网正负半周的对称性及三相交流电网的对称性，在工作模态1、2、3下各开关管的工作模式如图7所示，工作模态4、5、6可以以此类推。
按照三相交流电压电流的关系，控制与a相、b相和c相对应的开关管其中三个桥臂各自独立工作。在输出电流为正时，开关管Q 1、开关管Q 3、开关管Q 5为主控开关管，开关管Q 2、开关管Q 4、开关管Q 6为同步开关管。在输出电流为负时，开关管Q 2、开关管Q 4、开关管Q 6为主控开关管，开关管Q 1、开关管Q 3、开关管Q 5为同步开关管。根据图6，可以推导出在工作模态1、2、3下各开关管的工作模式如图7所示，工作模态4、5、6可以以此类推。在工作模态1、2、3下各桥臂的驱动信号的波形如图8所示，较宽的高电平信号是主控开关管的开通信号，较窄的高电平信号是同步开关管的开通信号。
以工作模态1的工作区间a相为例，从图6可知，此时主控开关管是Q 1，同步开关管是Q 2，则a相桥臂对应的桥臂中点电压U a和输出电感L 1的电流i L1的波形如图9所示。
基于图9，对a相桥臂主要有五个控制状态。
图9的t 0～t 1时段的控制状态如图10(a)所示，在t 0时刻a相桥臂中点电压U a谐振至波峰时，开通开关管Q 1，此时，由于谐振电容C 1两端电位上负下正出现反压，开关管Q 1的体二极管开通，开关管Q 1两端的直流母线电压V dc和a相桥臂中点电压U a的压差约为零，实现零电压开通。
图9的t 1～t 2时段的控制状态如图10(b)所示，在输出电感L 1的电流i L1的峰值达到预设基准电流值或开关管Q 1的开通时间达到预设时间时，关断开关管Q 1，输出电感L 1续流，使开关管Q 2的体二极管导通。
图9的t 2～t 3时段的控制状态如图10(c)所示，当a相桥臂中点电压U a降至最低，控制器驱动开关管Q 2开通，开关管Q 2开通后实现同步整流。
图9的t 3～t 4时段的控制状态如图10(d)所示，当流经开关管Q 2的续流电流减小至零时，控制器对开关管Q 2的驱动信号置低，使开关管Q 2关断。电网对输出电感L 1、谐振电容C 2反向充电，电流i L1反向，在电流i L1到达反向峰值时，输出电感L 1反向释放能量，输出电感L 1的电压反向，使开关管Q 1的体二极管导通，从而a相桥臂中点电压U a被钳位为直流母线电压V dc，从而输出电感L 1的能量反馈到直流母线。
图9的t 4～t 5时段的控制状态如图10(e)所示，当输出电感L 1的能量释放完毕，输出电感L 1、谐振电容C 2和滤波电容C a以及输出电感L 1、谐振电容C 1、直流母线和滤波电容C a形成两个谐振回路，发生自然谐振，其中，滤波电容C a被电网钳位不参与谐振。在t 4～t 5时段中，在波峰开通开关管Q 1以进入下个开关周期，或在波谷开通开关管Q 1以进入工作模态4。
可以理解的是，在其他工作模态及其他工作区间下对相应的主控开关管和同步开关管的控制方式可以参考图9、图10(a)～图10(e)设置，在此不再赘述。
在上述实施例的基础上，对应电网工频，在控制当前相对应的桥臂的开关管的开通与关断时，具体控制当前相对应的桥臂的输出电感的基波分量为工频正弦波，以使当前相的输出电流为正弦波。
在具体实施中，可以通过控制每个开关周期内平均等效电流值均呈正弦变化，以使当前相的输出电流为正弦波；其中，开关周期为当前相产生并网电流的周期。
或者，还可以通过对对开关管进行变频控制，以使当前相的输出电流为正弦波。
图11是本申请实施例提供的一种开关管的控制效果示意图；图12为本申请实施例提供的一种完全有功输出时的波形示意图；图13为本申请实施例提供的一种非完全有功输出时的波形示意图。
在上述实施例中，通过主控开关管控制输出电感输出的电流，主控开关管的开关节点需参照预设基准电流。
本申请实施例提供一种以预设基准电流为参考来控制主控开关管的具体实施方式，如图11所示，若采用基准电流的方式来控制主控开关管在主控开关管开通后，相应的输出电感的电流i L增加。设置电流检测电路检测当前相对应的桥臂的输出电感的电流i L，在输出电感的电流i L峰值与预设基准电流I ref的值相等时，关断主控开关管，这段开通时间记为t on。
主控开关管关断后开始续流，电流i L下降，经过t off时间降到零后经过一段死区时间后再开通开关管。t on、t off和死区时间的总和为一个开关周期T，1/T为开关频率。
若采用预设时间的方式来控制主控开关管，则预先计算得到t on的基准值，在主控开关管开通后，通过计时器计时t on后，认为输出电感的电流i L峰值与预设基准电流I ref的值相等。
则上述实施例中所述的通过对开关管进行变频控制，以使当前相的输出电流为正弦波，即为控制在不同的相位，开关周期T的值不同，以使当前相的输出电流为正弦波。另外，可以指定在主控开关管关断后的第几个波峰或波谷进入下个开关周期，即通过控制死区时间的长度来控制开关的频率。
需要说明的是，预设基准电流的半个周期里，可能会控制开关管进行上千次启停，图11为简略画法。
在本申请实施例提供的三相逆变器的控制方法中，预设基准电流满足下述公式：
Figure PCTCN2020117032-appb-000002
其中，I ref为预设基准电流，T on为主控开关管的开通时间，T off为同步开关管的开通时间，T为开关周期，I outsin(ωt+θ)为当前相对应的桥臂的输 出电感的一个开关周期在当前相的平均等效电流值，ω为电网工频，θ为预设基准电流和电网电压的相位差。
需要说明的是，为了满足开关管零电压开通的时机，
Figure PCTCN2020117032-appb-000003
并非固定值。故虽然为了保证输出电流质量，通过控制每个开关周期内平均等效电流值均呈正弦变化，以使当前相的输出电流为正弦波，但预设基准电流的波形并非正弦波，即输出电感的平均等效电流值与预设基准电流的变化不同。
在此基础上，在控制当前相对应的桥臂的开关管的开通与关断时，通过控制相位差以进行功率因数调节。
在具体实施中，当进行完全有功输出控制时，控制相位差为零；当进行非完全有功输出控制时，控制相位差不为零，并通过控制相位差以调节无功功率。
在三相逆变器完全有功输出时，输出电流和电网的相位完全相同。以本申请实施例提供的图12所示，预设基准电流I ref和电网电压U grid的相位完全相同时，相位差为零即Δθ＝0，即过零点都相同，正负也相同。基于上文描述的控制方式，输出电流i L与预设基准电流I ref的相位相同，也就是和电网电压U grid的相位相同即Δθ＝0，实现完全有功输出
当要求输出无功或者是进行功率因数调节的时候，要求输出电流I ref和电网电压U grid的相位不同。以本申请实施例提供的图13所示，预设基准电流I ref和电网电压U grid的相位不同即Δθ≠0，过零点错开。基于上文描述的控制方式，输出电流i L与预设基准电流I ref的相位相同，则输出电流i L与电网电压U grid的相位不同即Δθ≠0，实现输出无功，通过设定，预设基准电流I ref和电网电压U grid之间的相位差Δθ，就可以调节无功功率的大小。
以上对本申请所提供的一种三相逆变器及三相逆变器的控制方法进行了详细介绍。说明书中各个实施例采用递进的方式描述，每个实施例重点说明的都是与其他实施例的不同之处，各个实施例之间相同相似部分互相 参见即可。应当指出，对于本技术领域的普通技术人员来说，在不脱离本申请原理的前提下，还可以对本申请进行若干改进和修饰，这些改进和修饰也落入本申请权利要求的保护范围内。
还需要说明的是，在本说明书中，诸如第一和第二等之类的关系术语仅仅用来将一个实体或者操作与另一个实体或操作区分开来，而不一定要求或者暗示这些实体或操作之间存在任何这种实际的关系或者顺序。而且，术语“包括”、“包含”或者其任何其他变体意在涵盖非排他性的包含，从而使得包括一系列要素的过程、方法、物品或者设备不仅包括那些要素，而且还包括没有明确列出的其他要素，或者是还包括为这种过程、方法、物品或者设备所固有的要素。在没有更多限制的情况下，由语句“包括一个……”限定的要素，并不排除在包括所述要素的过程、方法、物品或者设备中还存在另外的相同要素。

Claims (10)

  1. 一种三相逆变器，其特征在于，包括：三个桥臂，谐振电容，输出电感，滤波电容，以及控制器；
    其中，各所述桥臂均设有两个开关管，所述控制器分别与各所述开关管的控制端连接；所述桥臂的一个所述开关管的第一端与直流母线的正极连接，所述桥臂的另一个所述开关管的第二端与所述直流母线的负极连接，且所述桥臂的一个所述开关管的第二端与所述桥臂的另一个所述开关管的第一端连接为所述桥臂的中点；
    一个所述桥臂对应两个所述谐振电容、一个所述输出电感和一个所述滤波电容，一个所述谐振电容的第一端与所述直流母线的正极连接，另一个所述谐振电容的第二端与所述直流母线的负极连接，且一个所述谐振电容的第二端与另一个所述谐振电容的第一端与所述中点及所述输出电感的第一端连接，所述输出电感的第二端与所述滤波电容的第一端连接；
    各所述滤波电容的第二端相互连接。
  2. 根据权利要求1所述的三相逆变器，其特征在于，各所述滤波电容的第二端均接地。
  3. 根据权利要求1所述的三相逆变器，其特征在于，所述控制器具体为数字信号处理器或现场可编程门阵列。
  4. 一种三相逆变器的控制方法，其特征在于，基于权利要求1至3任意一项所述的控制器，包括：
    在当前相产生并网电流后，检测所述当前相达到软开关条件后，控制与所述当前相对应的桥臂的主控开关管导通；
    在所述当前相对应的桥臂的输出电感的电流峰值达到预设基准电流值时、或所述主控开关管的导通时间达到预设时间时，控制所述主控开关管关断；
    当所述同步开关管的体二极管开始导通续流时，控制所述同步开关管导通以进行同步整流，并在所述当前相对应的桥臂的输出电感的电流值降为零时，控制所述同步开关管关断；
    在所述当前相的死区时间内，所述当前相对应的桥臂的谐振电容、输 出电感和滤波电容发生自然谐振时，当检测达到所述软开关条件后，返回所述控制与所述当前相对应的桥臂的主控开关管导通的步骤；
    其中，所述主控开关管为所述当前相对应的桥臂首先开通的开关管，所述同步开关管为所述当前相对应的桥臂的另一个开关管。
  5. 根据权利要求4所述的控制方法，其特征在于，在控制所述当前相对应的桥臂的开关管的开通与关断时，具体控制所述当前相对应的桥臂的输出电感的基波分量为工频正弦波，以使所述当前相的输出电流为正弦波。
  6. 根据权利要求5所述的控制方法，其特征在于，所述控制所述当前相对应的桥臂的输出电感的基波分量为工频正弦波，以使所述当前相的输出电流为正弦波，具体为：
    控制每个开关周期内所述平均等效电流值均呈正弦变化，以使所述当前相的输出电流为正弦波；
    其中，所述开关周期为所述当前相产生并网电流的周期。
  7. 根据权利要求5所述的控制方法，其特征在于，所述控制所述当前相对应的桥臂的输出电感的基波分量为工频正弦波，以使所述当前相的输出电流为正弦波，具体为：
    对所述开关管进行变频控制，以使所述当前相的输出电流为正弦波。
  8. 根据权利要求4至7任意一项所述的控制方法，其特征在于，所述预设基准电流满足下述公式：
    Figure PCTCN2020117032-appb-100001
    其中，I ref为所述预设基准电流，T on为所述主控开关管的开通时间，T off为所述同步开关管的开通时间，T为开关周期，I outsin(ωt+θ)为所述当前相对应的桥臂的输出电感的一个所述开关周期在所述当前相的平均等效电流值，ω为电网工频，θ为所述预设基准电流和电网电压的相位差。
  9. 根据权利要求8所述的控制方法，其特征在于，在控制所述当前相对应的桥臂的开关管的开通与关断时，通过控制所述相位差以进行功率因数调节。
  10. 根据权利要求9所述的控制方法，其特征在于，所述通过控制所 述相位差以进行功率因数调节，具体包括：
    当进行完全有功输出控制时，控制所述相位差为零；
    当进行非完全有功输出控制时，控制所述相位差不为零，并通过控制所述相位差以调节无功功率。
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