WO2020172896A1 - 一种射频功率放大器及基站 - Google Patents

一种射频功率放大器及基站 Download PDF

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WO2020172896A1
WO2020172896A1 PCT/CN2019/076783 CN2019076783W WO2020172896A1 WO 2020172896 A1 WO2020172896 A1 WO 2020172896A1 CN 2019076783 W CN2019076783 W CN 2019076783W WO 2020172896 A1 WO2020172896 A1 WO 2020172896A1
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power amplifier
inverting
side power
input
output
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PCT/CN2019/076783
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English (en)
French (fr)
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陈晓凡
陈文华
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清华大学
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/42—Modifications of amplifiers to extend the bandwidth
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/189—High-frequency amplifiers, e.g. radio frequency amplifiers
    • H03F3/19—High-frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/451—Indexing scheme relating to amplifiers the amplifier being a radio frequency amplifier

Definitions

  • the present invention relates to the field of communication technology, in particular to a radio frequency power amplifier and a base station.
  • the memory effect of RF power amplifiers has many sources, the most important of which is the DC operating point drift effect introduced by the envelope components of the input and output currents of the power device.
  • the waveform of the drain current under the excitation of the modulating signal is a modulated half-sine wave, and its time-domain waveform and frequency spectrum can be obtained using numerical simulation methods, as shown in Figure 1.
  • the excitation signal uses a 200MHz bandwidth LTE-A signal
  • the baseband sampling rate is 1228.8MSPS
  • the carrier frequency is 3500MHz.
  • Figure 1(b) since an ideal Class-B power amplifier only intercepts the positive half of the carrier cycle, the drain current spectrum contains envelope components and harmonic components in addition to the fundamental frequency component.
  • the frequency of the envelope component is much lower than the fundamental frequency component, it is possible to introduce a memory effect across carrier periods.
  • the envelope component in the drain current is introduced by the even-order term in the Class-B transfer function, so its spectral width is much larger than the bandwidth of the modulation signal.
  • the bandwidth of the envelope component must be considered at least three to five times the modulation signal bandwidth, which is 1GHz in Figure 1.
  • the load impedance presented to the drain of the power amplifier in the envelope bandwidth is not zero, the envelope component in the drain current will produce a corresponding envelope voltage on the drain.
  • the waveform of this envelope voltage is not only related to the drain current, but also to the load impedance over the envelope bandwidth. Without loss of generality, for the typical inductive envelope impedance, due to the existence of the drain envelope voltage component, its time-domain waveform is superimposed on the drain DC voltage, which introduces the drift effect of the drain operating point.
  • the existing method is to design the matching network reasonably so that it can match the fundamental frequency and harmonics while giving the lowest possible value over the entire envelope bandwidth. Impedance to reduce the envelope voltage swing.
  • this memory effect elimination method works well on narrow-band power amplifiers. However, as the bandwidth increases, it is more and more difficult to achieve this design goal. For example, considering that the envelope component in the drain current can be extended to more than five times the signal bandwidth, this means that for a signal bandwidth of 200MHz, in order to completely eliminate the envelope voltage, zero impedance should be given in the frequency band near DC-1GHz. , And must not affect the fundamental frequency matching, which is difficult to achieve.
  • the difficulty of the above-mentioned traditional method is that only the different frequency characteristics between the envelope component and the fundamental frequency component are considered to distinguish between the two. That is, the frequency band of the envelope component is significantly lower than the fundamental frequency component, then this can be used.
  • the frequency discrimination design has a frequency selective matching network to perform different impedance matching on the envelope component and the fundamental frequency component. However, as the relative bandwidth of the modulated signal increases, the envelope component and the fundamental frequency component gradually approach in the frequency band. When there is no longer an order of magnitude difference between the high end of the envelope component and the low end of the fundamental frequency component, it becomes very difficult to distinguish the two only by frequency characteristics.
  • the present invention proposes a radio frequency power amplifier, which can not only eliminate the memory effect of the narrowband radio frequency power amplifier, but also eliminate the memory effect of the broadband radio frequency power amplifier.
  • a radio frequency power amplifier including: an input balun, a non-inverting side power amplifying unit, an inverting side power amplifying unit, and an output balun;
  • the input balanced-unbalanced converter is respectively connected to the input ends of the non-inverting side power amplifying unit and the reverse side power amplifying unit, and the output ends of the non-inverting side power amplifying unit and the reverse side power amplifying unit are respectively connected to The output balun;
  • the non-inverting power amplifying unit includes a non-inverting input matching circuit, a non-inverting power amplifying device, and a non-inverting output matching circuit that are sequentially connected.
  • the input ends of the non-inverting side power amplifier device and the inverting side power amplifier device are respectively connected to an input envelope elimination network, and the output ends of the non-inverting side power amplifier device and the reverse side power amplifier device are respectively connected to an output envelope elimination network.
  • the present invention further provides a radio frequency power amplifier, including: an input balun, a non-inverting side power amplifier, an inverting side power amplifier, and an output balun; the input baluns are connected respectively The input ends of the non-inverting side power amplifier and the inverting side power amplifier, and the output ends of the non-inverting side power amplifier and the inverting side power amplifier are respectively connected to the output balun;
  • the power amplifier on the same-phase side sequentially includes a power amplifier on the same-phase side, N power amplifier units on the same-phase side, and a combiner on the same-phase side.
  • the output ends of the same-phase side distributor are respectively connected to N power amplifier units on the same-phase side.
  • the anti-phase side power amplifier includes an anti-phase side distributor, N anti-phase side power amplifier units, and an anti-phase side combiner,
  • the output ends of the inverting-side distributor are respectively connected to N inverting-side power amplifying units, and the other ends of the N inverting-side power amplifying units are respectively connected to the input of the inverting-side combiner;
  • N is greater than or equal to 2;
  • the number of power amplifying units on the in-phase side and the power amplifying units on the inverting side are equal;
  • the non-inverting power amplifying unit includes a non-inverting input matching circuit, a non-inverting power amplifying device, and a non-inverting output matching circuit that are sequentially connected.
  • An input envelope elimination network is connected between the input end of each non-inverting power amplifier device and the input end of the corresponding inverting-side power amplifier device, and the output end of each non-inverting power amplifier device corresponds to An output envelope elimination network is connected between the output ends of the inverting-side power amplifier device.
  • the present invention also provides a base station, including any radio frequency power amplifier provided in the embodiments of the present invention.
  • the radio frequency power amplifier proposed in the present invention can effectively reduce the memory effect and realize a memoryless radio frequency power amplifier without significant memory effect.
  • This memoryless RF power amplifier can be linearized using simple memoryless DPD algorithms or other simple memoryless linearization methods, thereby greatly reducing the complexity of linearization, reducing the cost of the communication system, and improving the communication system The overall efficiency of the system has broad application prospects.
  • Fig. 1 is a schematic diagram of the drain current of an ideal Class-B power amplifier under the excitation of a broadband modulation signal in the prior art, where (a) is a time-domain waveform diagram, (b) is a spectrum decomposition diagram;
  • FIG. 2 is a schematic diagram of the phase characteristics of the envelope component and the fundamental frequency component in the drain current in the prior art
  • FIG. 3 is a functional block diagram of a radio frequency power amplifier according to an embodiment of the present invention.
  • FIG. 4 is a schematic diagram of the structure when the envelope elimination network is a coupled transmission line in the first embodiment of the present invention
  • Figure 5 is the effect diagram of the envelope elimination network based on the coupled transmission line, (a) is the envelope elimination network parameters, (b) is the even mode and odd mode impedance diagram of the envelope elimination network;
  • Fig. 6 is a simulation effect diagram of applying the envelope cancellation network structure of the embodiment of Fig. 4, (a) is the case of AM-AM distortion, (b) is the case of AM-PM distortion;
  • FIG. 7 shows several specific circuit structures of the envelope elimination network based on the coupled transmission line provided by the embodiment of FIG. 4;
  • FIG. 8 is a schematic structural diagram when the envelope elimination network in the radio frequency power amplifier provided by the second embodiment of the present invention is an inductance capacitor resonance circuit
  • FIG. 9 is an even mode equivalent circuit diagram of the inductance capacitor resonance circuit in the embodiment of FIG. 8;
  • FIG. 10 is an odd-mode equivalent circuit diagram of the inductor-capacitor resonance circuit in the embodiment of FIG. 8;
  • FIG. 11 is a schematic diagram of the odd-mode impedance and even-mode impedance of the envelope cancellation network in the embodiment of FIG. 8;
  • FIG. 12 is a schematic diagram of the structure when the envelope elimination network in the radio frequency power amplifier provided by the third embodiment of the present invention is a transformer;
  • FIG. 13 is a schematic diagram of an envelope elimination network implemented by a single-turn parallel double wire passing through a cylindrical magnetic core in the embodiment of FIG. 12;
  • FIG. 14 is a functional block diagram of a multi-channel radio frequency amplifier provided by Embodiment 4 of the present invention.
  • the matching network of the power amplifier is designed so that the fundamental frequency is matched with the harmonics, and the lowest possible impedance is provided over the entire envelope bandwidth to reduce the envelope voltage swing. But this method only considers the different frequency characteristics between the envelope component and the fundamental frequency component to distinguish the two. As the bandwidth of the modulation signal increases, the envelope component and the fundamental frequency component gradually approach in the frequency band, and it becomes very difficult to distinguish only by frequency characteristics.
  • the embodiment of the present invention provides such a radio frequency power amplifier.
  • the radio frequency power amplifier includes an input balun, an in-phase side power amplifier unit, and an inverted side power amplifier unit. And an output balanced-unbalanced converter; wherein the input balanced-unbalanced converter is connected to the input ends of the non-phase side power amplifier unit and the reverse side power amplifier unit, and the non-phase side power amplifier unit and the reverse side power amplifier unit The output terminals are respectively connected to the output balun.
  • the non-inverting side power amplifying unit includes a non-inverting side input matching circuit, a non-inverting side power amplifier device, and a non-inverting side output matching circuit connected in sequence
  • the inverting side power amplifying unit includes an inverting side input matching circuit and an inverting side power amplifier device connected in sequence And the inverting side output matching circuit; the input ends of the non-inverting side power amplifier device and the inverting side power amplifier device are respectively connected to an input envelope elimination network, and the output ends of the non-inverting side power amplifier device and the inverting side power amplifier device Connect the output envelope elimination network respectively.
  • Power amplifier devices refer to various devices with power amplifier functions, such as vacuum tubes and transistors with or without packaging, or modules composed of vacuum tubes or transistors plus necessary peripheral circuits.
  • the envelope elimination network is a special two-port network. Two modes can be defined on its two ports, namely, the odd mode of the two-port constant amplitude and inverted excitation and the even mode of the two-port constant amplitude and in-phase excitation. mold. Obviously, in order to eliminate the envelope component in the voltage and ensure a good match of the fundamental frequency component, the ideal envelope elimination network should be short-circuited in the even mode, and high resistance to the odd mode.
  • FIG. 4 is a circuit structure diagram of the input envelope elimination network and/or the output envelope elimination network in the radio frequency power amplifier according to the first embodiment of the present invention.
  • the envelope elimination network is a distributed parameter circuit based on coupled transmission lines.
  • TL1 and TL2 are a pair of coupled transmission lines
  • the even mode impedance is Ze
  • the odd mode impedance is Zo
  • the electrical length at the reference frequency f0 is L.
  • This structure has the function of a unique mode inverter: even mode excitation is applied to two ports, the odd mode with opposite propagation direction will be excited on the coupled transmission line; while the odd mode excitation is applied to the two ports on the coupled transmission line Will excite even modes that propagate in the same direction.
  • the envelope elimination network shown in Figure 5(a) is used to perform simulation verification under the excitation of broadband modulation signals.
  • the excitation signal used a 200MHz bandwidth LTE-A signal
  • the power tube used Wolfspeed's 10Watt GaN HEMT CGH40010.
  • the simulated amplitude-amplitude (AM-AM) and amplitude-phase (AM-PM) characteristics are shown in Figure 6.
  • Figure 6 also shows the AM-AM and AM-PM characteristics after removing the envelope cancellation network.
  • the narrower width in Figure 6 (a) and (b) is the case with the envelope cancellation network.
  • the wider width is the case of eliminating the network without envelope.
  • envelope elimination network based on coupled transmission lines can also have the following forms as shown in Figure 7:
  • Example 1 As shown in Figure 7(a), a DC blocking capacitor C1 and C2 are respectively provided at the signal input ends of the coupled transmission lines TL1 and TL2 to achieve ground isolation.
  • Example 2 As shown in Figure 7(b), a DC blocking capacitor C1 and C2 are respectively provided at the ground terminals of the coupled transmission lines TL1 and TL2 to also achieve ground isolation, which is physically easier to implement.
  • Example 3 due to the existence of the DC blocking capacitor, the junction point between the DC blocking capacitor C1 and the transmission line TL1, and between the DC blocking capacitor C2 and the transmission line TL2 is a point where a DC open circuit to ground and a short circuit to radio frequency. Considering that the power amplifier usually needs to insert a DC bias voltage at the input and output ends, this combination point is an ideal DC bias voltage insertion point. As shown in Figure 7(c), a DC bias power supply can be connected at this junction, which can greatly simplify the design of the feeder network, and is a very practical structure.
  • Example 4 On the basis of Example 3, in order to further improve the radio frequency isolation between the DC power supply and the power amplifier, a choke inductance can be added between the DC power supply and the bias voltage insertion point, as shown in Figure 7(d).
  • envelope elimination networks based on coupled transmission lines which are all deformations made on the basis of a pair of coupled transmission lines, such as introducing but not limited to inductance, capacitance, Devices such as resistors, diodes, and triodes to realize functions including but not limited to impedance matching, tuning, blocking, sampling, control, etc., all belong to the specific presentation of the inventive idea of the embodiments of the present invention and are included in the protection scope of the present invention.
  • the specific structure of the coupled transmission line can be flexibly selected according to the actual situation, such as microstrip line, strip line, coaxial line, parallel double line, twisted line, fin line, waveguide, coplanar waveguide, planar integrated waveguide, dielectric Transmission line structures such as waveguides and their variants can all be used to implement the envelope elimination network in this embodiment.
  • envelope elimination network composed of distributed parameter circuits can be realized by introducing structures such as resonators and transmission lines in addition to being realized by coupling transmission lines, which will not be described in detail here.
  • the envelope elimination network in the radio frequency power amplifier provided in the second embodiment of the present invention is composed of a centralized parameter circuit.
  • This embodiment provides an inductance capacitor resonance circuit, as shown in FIG. 8.
  • the circuit includes two inductors L1, L2 and a capacitor C1.
  • the capacitor is connected in series between the signal input ends of the two inductors.
  • the equivalent circuit of the envelope elimination network realized by the inductance-capacitance resonance circuit shown in Figure 8 under odd-mode excitation is shown in Figure 10. Since both ends of the capacitor C1 are reverse excitation under the even-mode excitation, the center line of the capacitor C1 is so on. The effect is a virtual ground. Therefore, in the odd-mode equivalent circuit, the two ports are decoupled, and the capacitors become ground capacitors with doubled capacitance.
  • the capacitance only appears in the odd-mode equivalent circuit, but not in the even-mode equivalent circuit, and the capacitance in the odd-mode equivalent circuit is equivalent to the capacitance to ground in parallel with the inductor. Therefore, as long as the capacitance value of the capacitor is reasonably selected to make it resonate in parallel with the inductor at a given frequency, the odd-mode impedance can be effectively increased without affecting the even-mode impedance.
  • inductor-capacitor resonant circuit shown in FIG. 8 can also be expanded similarly to FIG. 7 to obtain different variants to meet the needs of specific power amplifier design, which will not be repeated here.
  • circuit structure can be implemented in other ways.
  • a multi-level network composed of multiple capacitors and inductors can be used to obtain better performance
  • devices including but not limited to resistors, diodes, and triodes can also be introduced to implement impedance matching, tuning, blocking, sampling, and Functions such as control belong to the specific presentation of the inventive idea of the embodiment of the present invention and are included in the protection scope of the present invention.
  • the envelope elimination network in the radio frequency power amplifier provided in the third embodiment of the present invention is composed of a magnetic coupling circuit. At lower frequencies, the magnetic coupling circuit can effectively reduce the volume of the envelope elimination network and has better performance.
  • the transformer is also suitable for implementing the envelope elimination network in this embodiment.
  • An envelope elimination network implemented by a transformer is shown in Figure 12, which is composed of primary and secondary coils wound on a magnetic core. Obviously, under even-mode excitation, the magnetic fluxes generated by the primary and secondary coils cancel each other out in the core, and the even-mode impedance is 0; while under odd-mode excitation, the magnetic fluxes generated by the primary and secondary coils are in the core. The medium strengthens each other, and the odd mode impedance is high impedance.
  • the transformer in Figure 12 can be wound on an open-circuit magnetic core such as a magnetic bar, or on a closed-circuit magnetic core such as a magnetic ring; the primary and secondary coils of the transformer can use ordinary wires Winding, it can also use coaxial wire, twisted pair and other transmission wire winding; the number of turns of the transformer can be single turn or multiple turns.
  • an envelope elimination network structure that uses a single turn of parallel double wires to pass through a cylindrical core is shown in FIG. 13.
  • FIG. 14 is a schematic diagram of a multi-channel radio frequency amplifier provided by Embodiment 4 of the present invention.
  • the radio frequency amplifier includes: an input balun, a non-inverting side power amplifier, an inverting side power amplifier, and an output balun; the input balun is connected to the non-inverting side power amplifier and An input end of the inverting side power amplifier, and the output ends of the non-inverting side power amplifier and the inverting side power amplifier are respectively connected to the output balun;
  • the power amplifier on the same-phase side sequentially includes a power amplifier on the same-phase side, N power amplifier units on the same-phase side, and a combiner on the same-phase side.
  • the output ends of the same-phase side distributor are respectively connected to N power amplifier units on the same-phase side.
  • the anti-phase side power amplifier includes an anti-phase side distributor, N anti-phase side power amplifier units, and an anti-phase side combiner,
  • the output ends of the inverting-side distributor are respectively connected to N inverting-side power amplifying units, and the other ends of the N inverting-side power amplifying units are respectively connected to the input of the inverting-side combiner;
  • N is greater than or equal to 2;
  • the number of power amplifying units on the in-phase side and the power amplifying units on the inverting side are equal;
  • the non-inverting power amplifying unit includes a non-inverting input matching circuit, a non-inverting power amplifying device, and a non-inverting output matching circuit that are sequentially connected.
  • Side power amplifying device and inverting side output matching circuit in the same branch, the circuit structure of the non-inverting side power amplifying unit and the inverting side power amplifying unit are the same.
  • An input envelope elimination network is connected between the input end of each non-inverting power amplifier device and the input end of the corresponding inverting-side power amplifier device, and the output end of each non-inverting power amplifier device corresponds to An output envelope elimination network is connected between the output ends of the inverting-side power amplifier device.
  • the envelope elimination network in the multi-channel RF power amplifier is the same as that in the single-channel RF power amplifier. It can be implemented by any one of distributed parameter circuits, lumped parameter circuits, or magnetic coupling circuits.
  • the specific circuit configuration can be implemented in the first embodiment.
  • the envelope elimination networks implemented by different circuits can be mixed in the same multi-channel radio frequency power amplifier, and the specific structure and changes can be referred to other embodiments, which will not be described in detail here.
  • the non-inverting side power amplifier and the inverting side power amplifier are both Doherty power amplifiers.
  • Doherty power amplifiers are widely used.
  • a typical Doherty power amplifier contains a main power amplifier unit and at least one auxiliary power amplifier unit.
  • the number of auxiliary power amplifier units is N-1.
  • the most common is a dual-channel Doherty that contains a main amplifier unit and an auxiliary amplifier unit. Power amplifier, that is, when N is 2.
  • the structure of the power amplifying unit here is the same as the structure of the non-inverting side/inverting side power amplifying unit in Embodiment 1, and will not be described in detail.
  • the non-inverting side distributor, the inverting side distributor, the non-inverting side combiner, and the inverting side combiner in the Doherty power amplifier are all realized by a delay line.
  • This is a structure often used in Doherty power amplifiers.
  • Doherty power amplifiers There are other implementation forms, but this is not the focus of the embodiment of the present invention and will not be described in detail here.
  • the use of Doherty power amplifiers can achieve high efficiency under the excitation of the peak-to-average power ratio signal, and at the same time has a low memory effect, which has obvious practical value.
  • the embodiment of the present invention also provides a base station, including any radio frequency power amplifier provided in the embodiment of the present invention.

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Abstract

本发明公开一种射频功率放大器及基站，其中射频功率放大器包括：输入平衡-不平衡变换器、同相侧功率放大单元、反相侧功率放大单元和输出平衡-不平衡变换器；输入平衡-不平衡变换器分别连接同相侧功率放大单元和反相侧功率放大单元的输入端，同相侧功率放大单元和反相侧功率放大单元的输出端分别连接输出平衡-不平衡变换器；同相侧功率放大器件和反相侧功率放大器件的输入端分别连接输入包络消除网络，同相侧功率放大器件和反相侧功率放大器件的输出端分别连接输出包络消除网络。本发明提出的射频功率放大器可以有效的降低记忆效应，大幅度的降低线性化的复杂度，降低通信系统的成本，提高通信系统的整体效率，具有广阔的应用前景。

Description

一种射频功率放大器及基站 技术领域
本发明涉及通信技术领域，特别涉及一种射频功率放大器及基站。
背景技术
在现代通信系统中，随着调制信号带宽的增大和载波聚合技术的应用，由射频功率放大器的记忆效应引入的非线性问题越来越严重。为了应对这种记忆效应，在很多通信系统中不得不采用复杂的记忆数字预失真(DPD)算法对功率放大器进行线性化。然而，复杂的DPD技术需要引入额外的成本和功耗。尤其是随着大规模MIMO技术的应用，基站射频通道数量大大增加，复杂的DPD算法引入的资产成本和功耗逐渐上升到难以接受的程度。可以说，由宽带射频功率放大器的记忆效应引起的非线性问题，已经成为通信系统中亟待解决的关键技术问题。
射频功率放大器的记忆效应有多种来源，其中最重要的来源是功率器件输入端和输出端电流的包络分量引入的直流工作点漂移效应。对于理想的Class-B功率放大器，在调制信号激励下的漏极电流的波形为调制的半正弦波，其时域波形和频谱可以使用数值计算的仿真方法得到，如图1所示。在图1中，激励信号采用了200MHz带宽的LTE-A信号，基带采样率为1228.8MSPS，载波频率为3500MHz。由图1(b)可见，由于理想的Class-B功率放大器只截取了载波周期的正半周，漏极电流频谱中除了基频分量，还包含了包络分量和谐波分量。其中由于包络分量的频率远远低于基频分量，有可能引入跨载波周期的记忆效应。进一步的分析指出，漏极电流中的包络分量由Class-B传递函数中的偶次项引入，因此其频谱宽度要远远大于调制信号的带宽。对于记忆效应的研究而言，包络分量的带宽至少要考虑到三到五倍的调制信号带宽，在图1中为1GHz。显然，如果在包络带宽上呈现给功率放大器漏极的负载阻抗不为零，漏极电流中的包络成分会在漏极上产生相应的包络电压。这一包络电压的波形不但与漏极电流有关，还与包络带宽上的负载阻抗有关。不失一般性，对于典型的感性包络阻抗，由于漏极包络电压分量的存在，其时域波形叠加在漏极直流电压上，引入了漏极工作点的漂移效应。
在使用实际器件构建的功率放大器中，由于存在以膝电压(Vknee)效应为代表的漏极电压对漏极电流的反馈效应，漏极电压工作点的漂移不可避免的会产生漏极记忆效应。同样，在射频功率器件的栅极，由于FET器件中非线性输入电容的存在，也同样的存在由于包络电压引入的栅极工作点漂移效应，这一效应通过器件的跨导传输到输出信号中，引入了栅极记 忆效应。进一步的研究可以证明，这种栅极和漏极的电压工作点漂移效应，是宽带调制信号激励下功率放大器记忆效应的主要来源。
为了消除上述工作点漂移效应引入的记忆效应，现有的方法是对匹配网络进行合理的设计，使得其在进行基频和谐波匹配的同时，在整个包络带宽上给出尽可能低的阻抗，以降低包络电压摆幅。但这种记忆效应消除方法在窄带功率放大器上工作良好，然而随着带宽的增加，要实现这一设计目标越来越困难。例如，考虑到漏极电流中的包络分量可以扩展至五倍信号带宽以上，这意味着对于200MHz的信号带宽，为了完全消除包络电压，需要在近DC-1GHz的频段上给出零阻抗，同时不得影响基频匹配，这是很难实现的。更严重的是，当信号带宽增加到和载波频率相比拟的程度后，采用上述技术消除记忆效应在理论上就成为不可能。例如，对于载波频率2000MHz的功率放大器，当信号带宽增大到400MHz左右，其栅极和漏极电流中的包络频率就已经和基频频率发生了混叠，无法在频谱上对其进行区分了。
上述传统方法的困难在于仅仅考虑了包络分量与基频分量之间不同的频率特征以对二者进行区分，即包络分量所在的频段要明显低于基频分量，那么就可以利用这种频率区分度设计具有频率选择性的匹配网络，对包络分量和基频分量进行不同的阻抗匹配。然而，随着调制信号相对带宽的增加，包络分量与基频分量之间在频段上逐渐靠近。当包络分量的高端与基频分量的低端之间不再有数量级上的差距的时候，仅仅依靠频率特征来区分两者，就变得非常困难。
发明内容
针对上述问题，本发明提出一种射频功率放大器，不但可以消除窄带射频功率放大器的记忆效应，也可以消除宽带射频功率放大器的记忆效应。
一种射频功率放大器，包括：输入平衡-不平衡变换器、同相侧功率放大单元、反相侧功率放大单元和输出平衡-不平衡变换器；
所述输入平衡-不平衡变换器分别连接所述同相侧功率放大单元和反相侧功率放大单元的输入端，所述同相侧功率放大单元和所述反相侧功率放大单元的输出端分别连接所述输出平衡-不平衡变换器；
所述同相侧功率放大单元包括依次连接的同相侧输入匹配电路、同相侧功率放大器件和同相侧输出匹配电路，所述反相侧功率放大单元包括依次连接的反相侧输入匹配电路、反相侧功率放大器件和反相侧输出匹配电路；
所述同相侧功率放大器件和反相侧功率放大器件的输入端分别连接输入包络消除网络，所述同相侧功率放大器件和反相侧功率放大器件的输出端分别连接输出包络消除网络。
本发明又提出一种射频功率放大器，包括：输入平衡-不平衡变换器、同相侧功率放大器、 反相侧功率放大器和输出平衡-不平衡变换器；所述输入平衡-不平衡变换器分别连接所述同相侧功率放大器和反相侧功率放大器的输入端，所述同相侧功率放大器和反相侧功率放大器的输出端分别连接所述输出平衡-不平衡变换器；
所述同相侧功率放大器依次包括同相侧分配器、N个同相侧功率放大单元和同相侧合路器，所述同相侧分配器的输出端分别连接N个同相侧功率放大单元，所述N个同相侧功率放大单元另一端分别连接所述同相侧合路器的输入端；所述反相侧功率放大器包括反相侧分配器、N个反相侧功率放大单元和反相侧合路器，所述反相侧分配器的输出端分别连接N个反相侧功率放大单元，所述N个反相侧功率放大单元另一端分别连接所述反相侧合路器的输入端；N大于等于2；所述同相侧功率放大单元与所述反相侧功率放大单元数量相等；
所述同相侧功率放大单元包括依次连接的同相侧输入匹配电路、同相侧功率放大器件和同相侧输出匹配电路，所述反相侧功率放大单元包括依次连接的反相侧输入匹配电路、反相侧功率放大器件和反相侧输出匹配电路；
每个所述同相侧功率放大器件的输入端和对应的所述反相侧功率放大器件的输入端之间连接一输入包络消除网络，每个所述同相侧功率放大器件的输出端和对应的所述反相侧功率放大器件的输出端之间连接一输出包络消除网络。
本发明还提出一种基站，包括本发明实施例所提供的任意一种射频功率放大器。
本发明提出的射频功率放大器可以有效的降低记忆效应，实现没有显著记忆效应的无记忆射频功率放大器。这种无记忆射频功率放大器可以使用简单的无记忆DPD算法或其他简单的无记忆线性化手段对其进行线性化，从而大幅度的降低线性化的复杂度，降低通信系统的成本，提高通信系统的整体效率，具有广阔的应用前景。
附图说明
图1为现有技术中宽带调制信号激励下理想Class-B功率放大器漏极电流的示意图，其中(a)为时域波形图，(b)为频谱分解图；
图2为现有技术中漏极电流中包络分量与基频分量的相位特征示意图；
图3为本发明实施例一种射频功率放大器原理框图；
图4为本发明实施例一中包络消除网络为耦合传输线时的结构示意图；
图5为基于耦合传输线实现的包络消除网络效果图，(a)为包络消除网络参数，(b)为包络消除网络偶模和奇模阻抗示意图；
图6为应用图4实施例的包络消除网络结构的仿真效果图，(a)为AM-AM失真的情况，(b)为AM-PM失真的情况；
图7为图4实施例提供的基于耦合传输线的包络消除网络的几种具体电路结构；
图8为本发明实施例二提供的射频功率放大器中包络消除网络为电感电容谐振电路时的结构示意图；
图9为图8实施例中电感电容谐振电路的偶模等效电路图；
图10为图8实施例中电感电容谐振电路的奇模等效电路图；
图11为图8实施例中包络消除网络的奇模阻抗和偶模阻抗示意图；
图12为本发明实施例三提供的射频功率放大器中包络消除网络为变压器时的结构示意图；
图13为图12实施例中单匝平行双线穿过柱状磁芯实现的包络消除网络示意图；
[根据细则26改正18.03.2019]　
图14为本发明实施例四提供的多路射频放大器原理框图。
图15为本发明实施例四提供的N=2时同相侧功率放大器和反相侧功率放大器均为Doherty功率放大器的电路原理框图。
具体实施方式
为了使本发明的目的、技术方案及优点更加清楚明白，以下结合附图及实施例，对本发明进行进一步详细说明。应当理解，此处所描述的具体实施例仅仅用以解释本发明，并不用于限定本发明。
现有技术中为了消除记忆效应，对功率放大器的匹配网络进行设计，使得基频与谐波匹配的同时，在整个包络带宽上给出尽可能低的阻抗以降低包络电压摆幅。但这种方法仅仅考虑了包络分量与基频分量之间不同的频率特征以对二者进行区分。随着调制信号带宽的增加，包络分量与基频分量之间在频段上逐渐靠近，仅靠频率特征来区分变得非常困难。
然而，除了传统的频率区分度之外，功率放大器电流的包络分量和基频分量之间还存在由相位特征引入的区分度，利用这一区分度可以大大增大两者之间的差异。以理想Class-B功率放大器的漏极电流为例，由于漏极电流中的包络成分来源于功率放大器传递函数中的偶次项，而漏极电流中的基频分量则来自于传递函数中的奇次项。这就意味着，当激励信号相位旋转180°，漏极电流频谱中的基频分量相位也旋转180°，而包络分量的相位则会旋转360°。这一相位特征可以以双音激励信号为例直观的展示，如图2所示。由图2可见，当激励信号中的基频分量反相后，漏极电流中的基频分量也随之反相，而包络分量和谐波分量的相位则保持不变。同样的，对于栅极电流中的包络分量，也有类似的相位特征。
实施例一
基于这一相位特征，在调制信号激励下的宽带功率放大器的包络分量和基频分量之间，除了频率区分度之外，还可以引入模式区分度。本发明实施例根据此原理，提供这样一种射频功率放大器，如图3所示，该射频功率放大器包括一输入平衡-不平衡变换器、一同相侧功率放大单元、一反相侧功率放大单元和一输出平衡-不平衡变换器；其中输入平衡-不平衡变换器分别连接着同相侧功率放大单元和反相侧功率放大单元的输入端，而同相侧功率放大单元和反相侧功率放大单元的输出端分别连接到输出平衡-不平衡变换器。
同相侧功率放大单元包括依次连接的同相侧输入匹配电路、同相侧功率放大器件和同相侧输出匹配电路，反相侧功率放大单元包括依次连接的反相侧输入匹配电路、反相侧功率放大器件和反相侧输出匹配电路；所述同相侧功率放大器件和反相侧功率放大器件的输入端分别连接输入包络消除网络，所述同相侧功率放大器件和反相侧功率放大器件的输出端分别连接输出包络消除网络。
功率放大器件指各种具有功率放大器作用的器件，例如带封装或者不带封装的真空管和晶体管，或者由于真空管或者晶体管加上必要的外围电路组成的模块。
本发明实施例中包络消除网络是一种特殊的二端口网络，在其两个端口上可以定义两种模式，即两端口等幅反相激励的奇模和两端口等幅同相激励的偶模。显然，为了消除电压中包络分量，并保证基频分量的良好匹配，理想的包络消除网络应对偶模短路，而对奇模呈现高阻。
图4为本发明实施例一提供的射频功率放大器中输入包络消除网络和/或输出包络消除网络的一种电路结构图。如图4所示，该包络消除网络为基于耦合传输线的分布参数电路。其中TL1和TL2为一对耦合传输线，其偶模阻抗为Ze，奇模阻抗为Zo，在参考频率f0上的电长度为L。这一结构具有独特的模式反相器的作用：对两个端口施加偶模激励，在耦合传输线上会激励起传播方向相反的奇模；而对两个端口施加奇模激励，在耦合传输线上会激励起传播方向相同的偶模。这一特点决定了，只要对TL1和TL2进行合理的设计，使其Ze为高阻抗，Zo为低阻抗，并合理的选择L的长度，就可以实现包络消除网络所需的偶模低阻、奇模高阻的端口特性。
图5(a)中用给出了基于这一结构实现的包络消除网络实例，选择Ze＝200Ohm，Zo＝10Ohm，在3.5GHz参考频率下的电长度L＝15°。使用Keysight的ADS软件进行仿真，得到了其端口的偶模和奇模阻抗，如图5(b)。由图5(b)可见，这一包络消除网络在很宽的频率范围上给出了低的偶模阻抗：在DC-10GHz小于10Ohm，在DC-1GHz小于0.7Ohm；同时在很宽的频率上给出了高的奇模阻抗：在2.5-10GHz大于50Ohm，在3.5-10GHz大于70Ohm，实现了较为理想的包络消除网络。
为了达到记忆效应消除的效果，应用图5(a)所示的包络消除网络，进行了宽带调制信号激励下的仿真验证。在仿真中，激励信号使用了200MHz带宽的LTE-A信号，功率管选用了Wolfspeed公司的10Watt GaN HEMT CGH40010。仿真得到的幅度-幅度(AM-AM)和幅度-相位(AM-PM)特性如图6所示。作为对比，图6中也给出了去掉包络消除网络之后的AM-AM和AM-PM特性，图6(a)和(b)中幅宽较窄的为有包络消除网络的情况，幅宽较宽的为无包络消除网络的情况。由图6可以看到，基于本项目提出的包络消除技术，有效的消除了功率放 大器在宽带激励下的记忆效应，验证了上述原理和结构的可行性。
上述例子中所选数值仅为示例而非对具体值进行限制。在这一结构中，耦合线的Zo越大，L越接近四分之一波长，奇模高阻的效果越好；Ze越小，L越小，则偶模低阻的效果越好。因此，理论上希望Ze与Zo的差值越大越好，此时更容易同时得到好的奇模和偶模阻抗；但是在实际应用中，受传输线结构的限制，Ze与Zo应选择易于实现的值，这对于本领域专业人员来说属于公共知识，这里不再详细解释。
基于耦合传输线的包络消除网络除了图4所示的结构外，还可以有如图7所示的以下形式：
示例一、如图7(a)所示，在耦合传输线TL1和TL2的信号输入端分别设一隔直电容C1和C2，实现对地的隔离。
示例二、如图7(b)所示，在耦合传输线TL1和TL2的接地端分别设一隔直电容C1和C2，同样实现对地的隔离，该种方式在物理上更容易实现。
示例三、在示例二中，由于隔直电容的存在，隔直电容C1和传输线TL1之间、隔直电容C2和传输线TL2之间的结合点是一个对地直流开路而对射频短路的点。考虑到功率放大器通常需要在输入端和输出端介入直流偏置电压，因此这一个结合点是理想的直流偏置电压插入点。如图7(c)所示，可在此结合点处连接直流偏置电源，可以大大的简化馈电网络的设计，是一种非常实用的结构。
示例四、在示例三的基础上，为了进一步提高直流电源和功率放大器之间的射频隔离，可以在直流电源和偏置电压插入点之间增加扼流电感，如图7(d)所示。
以上示例为典型的应用举例，本领域技术人员应该理解到，所揭露的电路结构可以通过其它的方式实现。除了以上几个示例外，基于耦合传输线的包络消除网络还有其他形式，均是在一对耦合传输线的基础上作出的变形，比如在耦合传输线的基础上引入包括但不限于电感、电容、电阻、二极管、三极管等器件，以实现包括但不限于阻抗匹配、调谐、隔直、采样、控制等功能，都属于本发明实施例的发明思想的具体呈现，包括在本发明保护范围之内。
上述基于耦合传输线实现的包络消除网络的其他变形均属于本领域技术人员应当知晓的常用电路变形，其具体构造本发明实施例不再详细介绍。
在耦合传输线的具体结构上，可以根据实际情况灵活选择，例如微带线、带状线、同轴线、平行双线、扭绞线、鳍线、波导、共面波导、平面集成波导、介质波导等传输线结构及其变体均可用于实现本实施例中的包络消除网络。
此外，由分布参数电路组成的包络消除网络除了通过耦合传输线实现以外，还可以通过引入谐振器、传输线等结构来实现，这里不再详细描述。
实施例二
本发明实施例二提供的射频功率放大器中包络消除网络由集中参数电路组成。本实施例给出了一种电感电容谐振电路，如图8所示。该电路包括两个电感L1、L2和一个电容C1，电容串接在两个电感的信号输入端之间。
由图8所示电感电容谐振电路实现的包络消除网络，在偶模激励下的等效电路如图9所示，由于在偶模激励下电容C1两端为同相激励，其在偶模等效电路中等效为开路。图8所示电感电容谐振电路实现的包络消除网络在奇模激励下的等效电路如图10所示，由于在偶模激励下电容C1两端为反向激励，电容C1的中线处等效为虚地，因此在奇模等效电路中两个端口解耦，电容分别成为容值加倍的对地电容。
由上述分析可见，电容只出现于奇模等效电路，而不出现于偶模等效电路中，且在奇模等效电路中电容等效为与电感并联的对地电容。因此只要合理的选择电容的容值，使其在给定频率下与电感并联谐振，即可有效的提高奇模阻抗，同时不影响偶模阻抗。
作为示例，在图8中取L＝1nH，C＝10pF，此时的偶模阻抗与奇模阻抗如图11所示。由图11可见，在DC到1.4GHz的宽阔频段上，偶模阻抗都保持了较低的数值，同时在0.9-1.4GHz的工作频段内获得了较高的奇模阻抗，因此图8所示的电感电容谐振电路可以用作工作频段为0.9-1.4GHz宽带功率放大器的包络消除网络。
同样，图8所示的电感电容谐振电路，也可以进行类似于图7的扩展获得不同的变体，以适应具体功率放大器设计的需要，这里不再重复。
以上示例为典型的应用举例，本领域技术人员应该理解到，所揭露的电路结构可以通过其它的方式实现。例如可以使用多个电容电感构成的多级网络以获取更好的性能，还可以引入包括但不限于电阻、二极管、三极管等器件，以实现包括但不限于阻抗匹配、调谐、隔直、采样、控制等功能，都属于本发明实施例的发明思想的具体呈现，包括在本发明保护范围之内。
实施例三
本发明实施例三提供的射频功率放大器中包络消除网络由磁耦合电路组成。在较低的频率上，磁耦合电路可以有效的减小包络消除网络的体积，且具有更好的性能。变压器作为常见的磁耦合电路，也适用于实现本实施例中的包络消除网络。一种由变压器实现的包络消除网络如图12所示，由绕制于磁芯上的原边和副边线圈组成。显然，在偶模激励下，原边副边线圈产生的磁通在磁芯中互相抵消，偶模阻抗为0；而在奇模激励下，原边和副边线圈产生的磁通在磁芯中互相加强，奇模阻抗为高阻。
在具体电路结构上，图12中的变压器既可以在磁棒等开路磁芯上绕制，也可以在磁环等闭路磁芯上绕制；变压器的原边和副边线圈既可以使用普通导线绕制，也可以使用同轴线、双绞线等传输线绕制；变压器的匝数可以是单匝，也可以是多匝。作为示例，一种使用单匝平行双线穿过柱状磁芯的包络消除网络结构如图13所示。
实施例四
图14为本发明实施例四提供的多路射频放大器示意图。该射频放大器包括：输入平衡-不平衡变换器、同相侧功率放大器、反相侧功率放大器和输出平衡-不平衡变换器；所述输入平衡-不平衡变换器分别连接所述同相侧功率放大器和反相侧功率放大器的输入端，所述同相侧功率放大器和反相侧功率放大器的输出端分别连接所述输出平衡-不平衡变换器；
所述同相侧功率放大器依次包括同相侧分配器、N个同相侧功率放大单元和同相侧合路器，所述同相侧分配器的输出端分别连接N个同相侧功率放大单元，所述N个同相侧功率放大单元另一端分别连接所述同相侧合路器的输入端；所述反相侧功率放大器包括反相侧分配器、N个反相侧功率放大单元和反相侧合路器，所述反相侧分配器的输出端分别连接N个反相侧功率放大单元，所述N个反相侧功率放大单元另一端分别连接所述反相侧合路器的输入端；N大于等于2；所述同相侧功率放大单元与所述反相侧功率放大单元数量相等；
所述同相侧功率放大单元包括依次连接的同相侧输入匹配电路、同相侧功率放大器件和同相侧输出匹配电路，所述反相侧功率放大单元包括依次连接的反相侧输入匹配电路、反相侧功率放大器件和反相侧输出匹配电路；同一支路中，同相侧功率放大单元与反相侧功率放大单元的电路结构相同。
每个所述同相侧功率放大器件的输入端和对应的所述反相侧功率放大器件的输入端之间连接一输入包络消除网络，每个所述同相侧功率放大器件的输出端和对应的所述反相侧功率放大器件的输出端之间连接一输出包络消除网络。
多路射频功率放大器中的包络消除网络与单路射频功率放大器中的一样，可以由分布参数电路、集中参数电路或磁耦合电路的任意一种实现，具体电路构造可以是实施例一到实施例三中的任意一种，不同电路实现的包络消除网络可以混用在同一个多路射频功率放大器中，具体结构和变化可参见其他实施例，此处不再详细说明。
在一些实施方式中，同相侧功率放大器和反相侧功率放大器均为Doherty功率放大器。众所周知，在当前的通信基站等应用中，为了应对高的峰均功率比带来的效率下降问题，广泛采用了Doherty功率放大器。典型的Doherty功率放大器含有一个主路功率放大单元和至少一个辅路功率放大单元，辅路功率放大单元的数量为N-1，最常见的是含有一个主路放大单元和一个辅路放大单元的双路Doherty功率放大器，即N为2的情况。这里的功率放大单 元结构与实施例一中的同相侧/反相侧功率放大单元结构相同，不再详细描述。
图15示出了本实施例提供的N＝2时的射频功率放大器中同相侧功率放大器和反相侧功率放大器均为Doherty功率放大器的结构示意图。在本实施例中Doherty功率放大器中的同相侧分配器、反相侧分配器、同相侧合路器、反相侧合路器均通过延迟线实现，这是Doherty功率放大器中经常采用的结构，还有其他的实现形式，但这并非本发明实施例关注的内容，在此不做详述。采用Doherty功率放大器可以在高峰均功率比信号的激励下实现高效率，同时具有低的记忆效应，具有明显的实用价值。
本发明实施例还提供一种基站，包括本发明实施例所提供的任意一种射频功率放大器。
以上实施例仅用于说明本发明的技术方案，而非对其限制，尽管参照前述实施例对本发明进行了详细的说明，本领域的普通技术人员应当理解：其依然可以对前述各实施例所记载的技术方案进行修改，或者对其中部分技术特征进行等同替换；而这些修改或者替换，并不使相应技术方案的本质脱离本发明各实施例技术方案的精神和范围。

Claims (14)

  1. 一种射频功率放大器，其特征在于，包括：输入平衡-不平衡变换器、同相侧功率放大单元、反相侧功率放大单元和输出平衡-不平衡变换器；
    所述输入平衡-不平衡变换器分别连接所述同相侧功率放大单元和反相侧功率放大单元的输入端，所述同相侧功率放大单元和所述反相侧功率放大单元的输出端分别连接所述输出平衡-不平衡变换器；
    所述同相侧功率放大单元包括依次连接的同相侧输入匹配电路、同相侧功率放大器件和同相侧输出匹配电路，所述反相侧功率放大单元包括依次连接的反相侧输入匹配电路、反相侧功率放大器件和反相侧输出匹配电路；所述同相侧功率放大器件和反相侧功率放大器件的输入端分别连接输入包络消除网络，所述同相侧功率放大器件和反相侧功率放大器件的输出端分别连接输出包络消除网络。
  2. 根据权利要求1所述的射频功率放大器，其特征在于，所述输入包络消除网络和/或输出包络消除网络由分布参数电路组成。
  3. 根据权利要求2所述的射频功率放大器，其特征在于，所述分布参数电路包括一对耦合传输线。
  4. 根据权利要求3所述的射频功率放大器，其特征在于，每条传输线的信号输入端设有隔直电容；或者
    每条传输线的接地端设有隔直电容；或者
    每条传输线的接地端设置隔直电容，并且在所述隔直电容和所述传输线的结合点处施加偏置电压；或者
    每条传输线的接地端设置隔直电容，且在所述隔直电容和所述传输线的结合点与用于施加偏置电压的电源间加入扼流电感。
  5. 根据权利要求1所述的射频功率放大器，其特征在于，所述输入包络消除网络和/或输出包络消除网络由集中参数电路组成。
  6. 根据权利要求5所述的射频功率放大器，其特征在于，所述集中参数电路为电感电容谐振电路。
  7. 根据权利要求6所述的射频功率放大器，其特征在于，所述电感电容谐振电路至少包括两个电感和一个电容，所述电容串联在两个所述电感之间。
  8. 根据权利要求1所述的射频功率放大器，其特征在于，所述输入包络消除网络和/或输出包络消除网络由磁耦合电路组成。
  9. 根据权利要求8所述的射频功率放大器，其特征在于，所述磁耦合电路为变压器。
  10. 一种射频功率放大器，其特征在于，包括：输入平衡-不平衡变换器、同相侧功率放 大器、反相侧功率放大器和输出平衡-不平衡变换器；所述输入平衡-不平衡变换器分别连接所述同相侧功率放大器和反相侧功率放大器的输入端，所述同相侧功率放大器和反相侧功率放大器的输出端分别连接所述输出平衡-不平衡变换器；
    所述同相侧功率放大器依次包括同相侧分配器、N个同相侧功率放大单元和同相侧合路器，所述同相侧分配器的输出端分别连接N个同相侧功率放大单元，所述N个同相侧功率放大单元另一端分别连接所述同相侧合路器的输入端；所述反相侧功率放大器包括反相侧分配器、N个反相侧功率放大单元和反相侧合路器，所述反相侧分配器的输出端分别连接N个反相侧功率放大单元，所述N个反相侧功率放大单元另一端分别连接所述反相侧合路器的输入端；N大于等于2；所述同相侧功率放大单元与所述反相侧功率放大单元数量相等；
    所述同相侧功率放大单元包括依次连接的同相侧输入匹配电路、同相侧功率放大器件和同相侧输出匹配电路，所述反相侧功率放大单元包括依次连接的反相侧输入匹配电路、反相侧功率放大器件和反相侧输出匹配电路；
    每个所述同相侧功率放大器件的输入端和对应的所述反相侧功率放大器件的输入端之间连接一输入包络消除网络，每个所述同相侧功率放大器件的输出端和对应的所述反相侧功率放大器件的输出端之间连接一输出包络消除网络。
  11. 根据权利要求10所述的射频功率放大器，其特征在于，所述输入包络消除网络和/或输出包络消除网络由分布参数电路、集中参数电路或磁耦合电路的任意一种实现。
  12. 根据权利要求10所述的射频功率放大器，其特征在于，所述同相侧功率放大器和所述反相侧功率放大器均为Doherty功率放大器。
  13. 根据权利要求12所述的射频功率放大器，其特征在于，所述Doherty功率放大器包括一个主路功率放大单元和至少一个辅路功率放大单元，所述辅路功率放大单元的数量为N-1。
  14. 一种基站，其特征在于，包括权利要求1～13任一项所述的射频功率放大器。
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