WO2020001147A1 - 一种移动终端天线和移动终端 - Google Patents

一种移动终端天线和移动终端 Download PDF

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Publication number
WO2020001147A1
WO2020001147A1 PCT/CN2019/084145 CN2019084145W WO2020001147A1 WO 2020001147 A1 WO2020001147 A1 WO 2020001147A1 CN 2019084145 W CN2019084145 W CN 2019084145W WO 2020001147 A1 WO2020001147 A1 WO 2020001147A1
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WIPO (PCT)
Prior art keywords
unit
mobile terminal
dielectric substrate
segment
terminal antenna
Prior art date
Application number
PCT/CN2019/084145
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English (en)
French (fr)
Inventor
张鹏
胡伟
张飞飞
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中兴通讯股份有限公司
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Priority to US17/256,056 priority Critical patent/US11509041B2/en
Priority to EP19827121.5A priority patent/EP3817141B1/en
Publication of WO2020001147A1 publication Critical patent/WO2020001147A1/zh

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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/36—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith
    • H01Q1/38—Structural form of radiating elements, e.g. cone, spiral, umbrella; Particular materials used therewith formed by a conductive layer on an insulating support
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/12—Supports; Mounting means
    • H01Q1/22—Supports; Mounting means by structural association with other equipment or articles
    • H01Q1/24—Supports; Mounting means by structural association with other equipment or articles with receiving set
    • H01Q1/241—Supports; Mounting means by structural association with other equipment or articles with receiving set used in mobile communications, e.g. GSM
    • H01Q1/242—Supports; Mounting means by structural association with other equipment or articles with receiving set used in mobile communications, e.g. GSM specially adapted for hand-held use
    • H01Q1/243—Supports; Mounting means by structural association with other equipment or articles with receiving set used in mobile communications, e.g. GSM specially adapted for hand-held use with built-in antennas
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/48—Earthing means; Earth screens; Counterpoises
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q1/00—Details of, or arrangements associated with, antennas
    • H01Q1/50—Structural association of antennas with earthing switches, lead-in devices or lightning protectors
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
    • H01Q5/30—Arrangements for providing operation on different wavebands
    • H01Q5/378—Combination of fed elements with parasitic elements
    • H01Q5/385—Two or more parasitic elements
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/0407—Substantially flat resonant element parallel to ground plane, e.g. patch antenna
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
    • H01Q9/04—Resonant antennas
    • H01Q9/30—Resonant antennas with feed to end of elongated active element, e.g. unipole
    • H01Q9/42—Resonant antennas with feed to end of elongated active element, e.g. unipole with folded element, the folded parts being spaced apart a small fraction of the operating wavelength

Definitions

  • the present application relates to the field of antennas, but is not limited to the field of antennas, and in particular, to a mobile terminal antenna and a mobile terminal.
  • mobile terminal antennas are also constantly developing and innovating to meet actual needs.
  • mobile terminal antennas need to meet the communication requirements of 2G, 3G, and 4G, which must cover: LTE700 / GSM850 / GSM900DCS1800 / PCS1900 / UMTS / LTE2300 / LTE2600 and other multiple frequency bands. Therefore, mobile terminal antenna design needs to consider both multiband and broadband With characteristics.
  • more and more functions are integrated on mobile terminals, and the design space left for antennas is also becoming smaller.
  • PIFA Planar Inverted F-shaped Antenna
  • monopole antenna monopole antenna
  • loop antenna loop antenna.
  • the PIFA antenna also meets the advantages of small size, easy implementation, and good production consistency.
  • the monopole antenna is more compact and has a wider bandwidth.
  • the PIFA antenna has a narrow bandwidth, and the monopole antenna is easily affected by the surrounding environment. At this stage, it is difficult to make the ground clearance under the antenna completely.
  • Embodiments of the present application provide a mobile terminal antenna and a mobile terminal, so as to cover multiple frequency bands while satisfying the volume requirements of the mobile terminal antenna.
  • An embodiment of the present application provides a mobile terminal antenna, which includes a dielectric substrate and a floor located on one side of the dielectric substrate, and further includes a near-feed unit, a near-ground unit, and a coupling unit disposed on the other side of the dielectric substrate. among them
  • One end of the near-ground unit is connected to the coupling unit, and the other end is connected to the floor;
  • the coupling unit and the near-ground unit are equivalent to a left-handed inductor;
  • the near-feedback unit is equivalent to a right-handed inductor;
  • Coupling with the near-feed unit is equivalent to a left-handed capacitor;
  • the coupling unit is coupled with the floor and is equivalent to a right-handed capacitor;
  • the near-feed unit, the near-ground unit, the coupling unit and the floor form a composite left-handed transmission line structure.
  • An embodiment of the present application further provides a mobile terminal, including the foregoing mobile terminal antenna.
  • the embodiment of the present application is based on a mobile terminal antenna designed with a composite left and right hand transmission line, which can meet the needs of mobile communications, has a simple structure, a compact layout, and can greatly save antenna space.
  • FIG. 1 is a schematic diagram of a circuit model of an ideal composite left-right transmission line.
  • FIG. 2 is a schematic diagram of dispersion relationship of a composite left-right transmission line.
  • FIG. 3 is a schematic diagram of the overall structure of a mobile terminal antenna according to an embodiment of the present application.
  • FIG. 4 is a schematic structural diagram of a mobile terminal body antenna according to the embodiment of FIG. 3.
  • FIG. 5 is a top view of the mobile terminal antenna of the embodiment of FIG. 4.
  • FIG. 6 is a side view of the mobile terminal antenna of the embodiment of FIG. 4.
  • FIG. 7 is a schematic structural diagram of a mobile terminal antenna (removing a high-frequency resonance unit) according to an embodiment of the present application.
  • FIG. 8 is a schematic structural diagram of a mobile terminal antenna (removing a low-frequency resonance unit) according to an embodiment of the present application.
  • FIG. 9 is a schematic structural diagram of an antenna terminal antenna according to an embodiment of the present application (with metal components added).
  • FIG. 10 is a schematic structural diagram of an antenna terminal antenna according to another embodiment of the present application (the rectangular ring in the near-feed unit is replaced with an oval ring).
  • FIG. 11 is a schematic structural diagram (changed shape) of an antenna terminal antenna according to another embodiment of the present application.
  • FIG. 12 is a schematic diagram of simulation calculation of S11 parameters in the embodiments of FIGS. 3 to 6.
  • FIG. 13 is a schematic diagram of an input impedance of the embodiments in FIGS. 3 to 6.
  • FIG. 14 is a schematic diagram of radiation efficiency in the low-frequency working frequency band (690 MHz-960 MHz) of the embodiments of FIGS. 3 to 6.
  • FIG. 14 is a schematic diagram of radiation efficiency in the low-frequency working frequency band (690 MHz-960 MHz) of the embodiments of FIGS. 3 to 6.
  • FIG. 14 is a schematic diagram of radiation efficiency in the low-frequency working frequency band (690 MHz-960 MHz) of the embodiments of FIGS. 3 to 6.
  • FIG. 15 is a schematic diagram of radiation efficiency in the high-frequency working frequency band (1710MHz-2690MHz) of the embodiments of FIGS. 3 to 6.
  • FIG. 15 is a schematic diagram of radiation efficiency in the high-frequency working frequency band (1710MHz-2690MHz) of the embodiments of FIGS. 3 to 6.
  • FIG. 16 is an xoy-plane far-field radiation pattern of the embodiment of FIGS. 3 to 6 at 825 MHz.
  • FIG. 17 is an xoz-plane far-field radiation pattern of the embodiments of FIGS. 3 to 6 at 825 MHz.
  • FIGS. 3 to 6 is a yoz-plane far-field radiation pattern of the embodiments of FIGS. 3 to 6 at 825 MHz.
  • FIG. 19 is an xoy-plane far-field radiation pattern of the embodiment of FIGS. 3 to 6 at 2250 MHz.
  • FIG. 20 is an xoz-plane far-field radiation pattern of the embodiment of FIGS. 3 to 6 at 2250 MHz.
  • 21 is a yoz-plane far-field radiation pattern of the embodiment of FIGS. 3 to 6 at 2250 MHz.
  • FIG. 22 is an actual measurement chart of S11 parameters in the embodiments of FIGS. 3 to 6.
  • FIG. 22 is an actual measurement chart of S11 parameters in the embodiments of FIGS. 3 to 6.
  • FIG. 23 is a comparison diagram of the measured and simulated xoy-plane far-field radiation pattern of the embodiment of FIGS. 3 to 6 at 825 MHz.
  • FIG. 24 is a comparison chart of the measured and simulated xoz-plane far-field radiation pattern at 825 MHz in the embodiments of FIGS. 3 to 6.
  • FIG. 24 is a comparison chart of the measured and simulated xoz-plane far-field radiation pattern at 825 MHz in the embodiments of FIGS. 3 to 6.
  • FIG. 25 is a comparison diagram of the measured and simulated yoz-plane far-field radiation pattern of the embodiments of FIGS. 3 to 6 at 825 MHz.
  • FIG. 26 is a comparison diagram of the measured and simulated xoy-plane far-field radiation patterns at 2250 MHz in the embodiments of FIGS. 3 to 6.
  • FIG. 27 is a comparison diagram of the measured and simulated far field radiation pattern of the xoz-plane at 2250 MHz in the embodiments of FIGS. 3 to 6.
  • FIG. 28 is a comparison chart of the measured and simulated yoz-plane far-field radiation pattern at 2250 MHz in the embodiments of FIGS.
  • FIG. 29 is a schematic diagram of the low-frequency S11 parameter simulation of the embodiment of FIG. 7.
  • FIG. 30 is a schematic diagram of the high-frequency S11 parameter simulation of the embodiment of FIG. 8.
  • FIG. 31 is a schematic diagram of S11 parameter simulation in the embodiment of FIG. 9.
  • FIG. 32 is a schematic diagram of S11 parameter simulation in the embodiment of FIG. 10.
  • An embodiment of the present application provides a mobile terminal antenna, which uses a composite left and right-handed transmission line to cover multiple working frequency bands and meets a narrow terminal design space.
  • the maximum bandwidth that an electric small antenna can achieve is directly proportional to the space occupied by the antenna. To obtain a large bandwidth, you must ensure that sufficient space is reserved for the electric small antenna.
  • the direction of the Poynting vector S is the direction of electromagnetic wave propagation, that is, the direction of electromagnetic energy transmission. E, H, and S form a right-handed spiral relationship perpendicular to each other.
  • a unit length transmission line can be equivalent to a series distributed inductance and a parallel distributed capacitance.
  • phase propagation constant is negative, and the phase velocity and the group velocity are reversed.
  • the left-handed material is artificially constructed using the right-handed material existing in nature, so it is impossible to obtain a simple left-handed transmission line. Both exist simultaneously, that is, a composite left-handed transmission line.
  • both left-handed mode and left-handed mode are available.
  • the propagation constant is a pure real number, it is the transmission forbidden band.
  • This situation is an unbalanced state of the composite left-right transmission line, and the series resonance point and the parallel resonance point are different. If the series resonance and the parallel resonance are the same, an equilibrium state is obtained. At this time, there is no stopband between the left-hand characteristic frequency region and the right-hand characteristic frequency region. In this case, there is no necessary constraint relationship between the resonance frequency and the physical size.
  • the resonance center frequency of the zero-order resonance point can be changed. This can be used to achieve miniaturization of the antenna.
  • FIG. 1 it is an ideal composite left-handed transmission line circuit model, consisting of four parts: (a) right-hand inductor L ′ R , (b) right-hand capacitor C ′ R , (c) left-hand inductor L ′ L, and (d) left-hand Capacitance C ′ L.
  • (A) and (d) form the series part of the equivalent circuit, (b) and (c) form the parallel part of the equivalent circuit; (a) and (c) form the inductance of the equivalent circuit Part, (b) and (d) constitute the capacitive part in the equivalent circuit; (a) and (b) constitute the right-hand part in the equivalent circuit, and (b) and (d) constitute the Left hand part.
  • various electrical parameters in the equivalent circuit can be changed by adjusting the physical structure corresponding to the left-handed capacitor inductor and the right-handed capacitor inductor, so that the composite left-handed and right-handed transmission line works in a balanced state.
  • the composite left-handed transmission line has no stopband.
  • the embodiments of the present application implement a composite left and right-handed transmission line structure through the physical structure of the antenna, so as to meet the wide-band requirements of mobile terminal antennas.
  • LC networks can be formed by distributed components such as microstrip lines, strip lines, and coplanar waveguides.
  • the main form of the left-handed inductance L ′ L includes a spiral inductance and a short-circuited line inductance
  • the left-handed capacitance C ′ L is implemented by a cross-finger capacitor, a gap capacitance, and the like
  • the right-handed capacitance inductor is formed by With line, microstrip patch.
  • the mobile terminal antenna includes a dielectric substrate 1 and a floor 2 on one side of the dielectric substrate, and further includes a near-feed unit 7 disposed on the other side of the dielectric substrate.
  • the near-feed unit 7 is equivalent to a right-handed inductor;
  • the coupling unit 11 is coupled to the near-feed unit 7 and is equivalent to a left-hand capacitor;
  • the coupling unit is coupled to the floor and is equivalent to a right-hand capacitor;
  • the near-feed unit, the near-ground unit, the coupling unit and the floor constitute a composite left-right transmission line structure.
  • the embodiment of the present application is based on an antenna designed with a composite left and right hand transmission line, which can meet the needs of mobile communication, has a simple structure, a compact layout, and can greatly save antenna space.
  • the near-ground unit 5 is a short-circuit line.
  • the near-feed unit 7 includes a ring-shaped portion 71 and a feed line 72 connected to each other. One end of the feed line 72 is connected to the ring-shaped portion 71 and the other end is connected to the feeding point 8.
  • the annular portion 71 is parallel to the dielectric substrate 1, and may be, but not limited to, a rectangular or oval shape.
  • the feeder 72 may adopt, but is not limited to, an L-shaped structure or a linear structure.
  • the near-feedback unit 7 may not adopt a ring structure, but adopt a rectangular or oval structure.
  • a gap is provided between the coupling unit 11 and the near-feed unit 7, and a left-handed capacitive effect is formed between the coupling unit 11 and the near-feed unit 7 through the gap.
  • the coupling unit 11 may also adopt an interdigital structure to form a left-handed capacitor.
  • a dielectric substrate 1 is disposed between the coupling unit 11 and the floor 2 to form a right-handed capacitive effect.
  • the coupling unit 11 includes at least one of a low-frequency resonance unit 3 and a high-frequency resonance unit 4.
  • the low-frequency resonance unit 3 includes a first branch 31 and a second branch 32.
  • the first branch 31 has a U-shaped structure
  • the second branch 32 has a polyline structure
  • the first The branch 31 and the second branch 32 are connected by the near-earth unit 5.
  • the first branch 31 may include a first segment 311, a second segment 312, and a third segment 313 which are connected in sequence.
  • the first segment 311 is connected to the near-ground unit 5 and is located on the dielectric substrate. 1 surface, the second section 312 is perpendicular to the dielectric substrate 1, the third section 313 is far from the dielectric substrate 1, is located above the first section 311, and a part of the plane in the third section 313 is vertical A part of the plane of the dielectric substrate 1 is parallel to the dielectric substrate 1.
  • the second branch 32 may include a fourth segment 321, a fifth segment 322, and a sixth segment 323, which are connected in sequence.
  • the fourth segment 321 is connected to the near-ground unit 5 and is located on the surface of the dielectric substrate 1.
  • the fifth section 322 is perpendicular to the dielectric substrate 1
  • the sixth section 323 is far from the dielectric substrate 1 and extends in a direction away from the fourth section 321, and a part of the plane in the sixth section 323 is planar It is perpendicular to the dielectric substrate 1 and part of the plane is parallel to the dielectric substrate 1.
  • the high-frequency resonance unit 4 includes at least a first patch 41, and the first patch 41 is perpendicular to the dielectric substrate 1.
  • the first patch 41 may be, but is not limited to, a rectangle, and is located inside the first branch 31U of the low-frequency resonance unit.
  • the high-frequency resonance unit 4 further includes a second patch 42 that is perpendicular to the dielectric substrate 1.
  • the second patch 42 may be, but is not limited to, a rectangle, and is located between the fifth section 322 and the sixth section 323 in the second branch of the low-frequency resonance unit and the dielectric substrate 1.
  • the second patch 42 can be considered as a monopole patch.
  • the use of the second patch 42 can improve the antenna high-frequency resonance characteristics and increase the antenna impedance bandwidth.
  • the low-frequency resonance unit 3 and the near-feed unit 7 are coupled by a series left-handed capacitor, and the near-feed unit 7 itself is equivalent to a series right-handed inductor, which constitutes a series capacitor and a series inductor in a composite left-handed transmission circuit.
  • the size of the equivalent left-hand capacitor can be changed by changing the distance between the low-frequency resonance unit 3 and the near-feed unit 7.
  • the size of the corresponding right-hand inductor can be changed by changing the width and length of the near-feed unit 7.
  • the antenna's series resonance point can be adjusted by changing the physical size of the antenna.
  • the low-frequency resonance unit 3 also has a right-handed capacitor to the ground, and together with the near-earth unit 5 form a parallel capacitor and a parallel inductor in a composite left-handed transmission circuit.
  • a complete composite left-handed transmission line circuit capable of operating in a low frequency band is formed.
  • the size of the right-hand capacitor in the circuit can be changed by changing the area of the low-frequency resonance unit 3, and the size of the left-hand inductor can be changed by changing the size of the short-circuit line 5 and / or the low-frequency resonance unit 3, thereby changing the antenna correspondence by adjusting the physical size of the antenna Parallel resonance point.
  • the high-frequency resonance unit 4 and the near-feed unit 7 form a left-handed capacitance effect, and the near-feed unit 7 itself is equivalent to a series right-handed inductor, which forms a series capacitor in the composite left-handed transmission circuit.
  • the high-frequency resonance unit 4 also has a right-handed capacitor to the ground, and together with the near-earth unit 5 form a parallel capacitor and a parallel inductor in the composite left-handed transmission circuit.
  • a composite left-handed transmission circuit capable of operating in a high-frequency operating frequency band is formed.
  • Adjust the corresponding left-handed capacitance and right-handed inductance by changing the size of the first patch 41 and / or the second patch 42 in the high-frequency resonance unit 4 and the size of the near-feed unit 7, thereby adjusting the series resonance point of the equivalent circuit .
  • the parallel resonance point can be changed by changing the size of the first patch 41 and / or the second patch 42 in the high-frequency resonance unit 4 and the size of the short-circuit line 5.
  • a three-dimensional structure based on a composite left and right-handed transmission line is adopted, and a conventional rectangular monopole structure is introduced to achieve a wider frequency band requirement, which can respectively cover a plurality of low-frequency and high-frequency operating frequency bands and meet a narrow terminal design space.
  • the overall structure of the antenna is shown in FIGS. 3 to 6 and has a size of 65 mm ⁇ 10 mm ⁇ 5.8 mm.
  • the antenna floor is similar in size to the floor of a conventional smart phone device.
  • the dielectric substrate is a FR4 substrate with a size of 65mm ⁇ 120mm ⁇ 0.8mm.
  • the short line has a length of 5 to 7 mm and a width of 0.5 to 2 mm.
  • the ring portion 71 may be an outer ring: 64 mm ⁇ 4 mm, and an inner ring 63 mm ⁇ 2.6 mm.
  • the first segment 311 and the third segment 313 in the first branch have a length of 32 to 36 mm and a width of about 2 mm, and the second segment 312 has a length of about 5 mm and a width of about 1 mm; in the second branch, The length of the fourth segment 321 is 34 to 38 mm, the length of the fifth segment 322 is approximately 5 mm, and the width is approximately 1 mm. The length of the sixth segment 323 is 28 mm to 32 mm, and the width is approximately 2 mm. The gap is about 4mm.
  • the size of the first patch 41 in the high-frequency resonance unit 4 may be: 19.5 mm ⁇ 3 mm, and the size of the second patch 42 may be 17.5 mm ⁇ 3 mm.
  • Mobile terminal antennas can have a variety of sizes and can be combined with other sizes of floors and dielectric substrates of different materials.
  • another embodiment of the present application removes the high-frequency resonance unit based on the embodiments of FIGS. 3 to 6, and provides a low-band mobile terminal antenna based on a composite left-handed structure, which can be applied to mobile phones and the like.
  • Mobile terminal Its working principle is the same as that of the low-frequency operation in the embodiments of FIGS. 3 to 6.
  • this embodiment removes the low-frequency resonance unit 3 on the basis of the embodiments of FIGS. 3 to 6, and provides a high-frequency band mobile terminal antenna based on a composite left-handed structure, which can be applied to mobile terminals such as mobile phones. .
  • Its working principle is the same as the high-frequency working situation in the embodiments of FIGS. 3 to 6.
  • a rectangular monopole structure (second patch 42) is added to the antenna structure, which improves the antenna resonance characteristics and increases the impedance bandwidth.
  • a metal component 9 that may appear in practical applications is added below the antenna unit.
  • FIG. 10 it is another implementation form of a mobile terminal antenna, and its working principle is similar to the embodiments of FIGS. 3 to 6, where the rectangular ring in the near-feed unit is replaced with an oval ring.
  • FIG. 11 it is another implementation form of a mobile terminal antenna, in which the shape is greatly changed from that in FIGS. 3 to 6, but the principle is the same as that described above.
  • the coupling unit 11 is an integrated structure, and includes a first planar portion 111 and a second planar portion 112 connected to each other.
  • the first planar portion 111 is perpendicular to the dielectric substrate 1 and the second planar portion 112 is parallel to the dielectric substrate 1.
  • the near-ground unit 5 is a short-circuit line.
  • the near-feed unit 7 includes a patch portion 73 and a feeder line 72 connected to each other. One end of the feeder line 72 is connected to the patch portion 73 and the other end is connected to a feeding point.
  • the patch portion 73 is connected to the first portion.
  • a gap is provided between the two planar portions 112, and a left-handed capacitive effect is formed between the coupling unit 11 and the near-feed unit 7 through the gap.
  • the patch portion 73 may be, but is not limited to, rectangular, parallel to the dielectric substrate, and located on the same plane as the second planar portion.
  • FIG. 11 uses a reconfigurable manner to achieve a good working state in the working frequency band.
  • the S11 parameters of the embodiments in FIGS. 3 to 6 are simulated and calculated, and the results are shown in FIG. 12. Taking S11 as less than -6dB as a standard, the impedance bandwidth of the antenna in the embodiments of FIGS. 3 to 6 is 680MHz-1100MHz and 1690MHz-3000MHz. Note that it can directly cover multiple frequency bands such as LTE700, GSM850, GSM900, DCS1800, PCS1900, UMTS, LTE2300, and LTE2600, and has a wider operating frequency band.
  • the input impedance parameters of the embodiments in FIGS. 3 to 6 are simulated and calculated, and the results are shown in FIG. 13. It can be seen from FIG. 13 that the antenna has good resonance characteristics at low and high frequency portions.
  • the calculation of the radiation efficiency in the low frequency band (690-960MHz) in the embodiments of FIGS. 3 to 6 is performed by simulation, and the result is shown in FIG. 14. It can be seen that the radiation efficiency of the antenna in the low frequency band (690-960MHz) is greater than 48%.
  • the high-band radiation efficiency (1710MHz-2690MHz) in the embodiments of FIGS. 3 to 6 is simulated and calculated, and the result is shown in FIG. 15. It can be seen that the radiation efficiency of the antenna in the high frequency band (1710MHz-2690MHz) is greater than 62.5%.
  • Simulations are performed on the 825MHz xoy-plane far-field radiation pattern of the embodiments of FIGS. 3 to 6, and the results are shown in FIG. 16. Simulations are performed on the 825 MHz xoz-plane far-field radiation pattern of the embodiments of FIGS. 3 to 6, and the results are shown in FIG. 17. Simulations are performed on the 825 MHz yoz-plane far-field radiation pattern of the embodiments of FIGS. 3 to 6, and the results are shown in FIG. 18.
  • the above embodiments of FIGS. 3 to 6 are simulated at a 2250 MHz xoy-plane far-field radiation pattern, and the results are shown in FIG. 19. The above embodiments of FIGS.
  • FIGS. 3 to 6 are simulated at a 2250 MHz xoz-plane far-field radiation pattern, and the results are shown in FIG. 20.
  • the above embodiments of FIGS. 3 to 6 are simulated at a 2250 MHz yoz-plane far-field radiation pattern, and the results are shown in FIG. 21.
  • the above-mentioned FIGS. 16 to 21 show the directional patterns of the antenna frequency bands, which all show that the requirements for the directional patterns in the industry have been met.
  • the vector network analyzer was used to measure the return loss of the model object described in the embodiments of FIGS. 3 to 6, and the results are shown in FIG. 22.
  • the measured impedance bandwidth of the antenna in the embodiments of FIGS. 3 to 6 is 680MHz-1100MHz and 1480MHz-3000MHz. Note that it can cover multiple frequency bands such as LTE700, GSM850, GSM900, DCS1800, PCS1900, UMTS, LTE2300, LTE2600, etc., and has a wider operating frequency band.
  • Measurements are made on the model real object radiation pattern at 825 MHz and the x-y-plane far field described in the embodiments of FIGS. 3 to 6 above, and the results are shown in FIG. 23.
  • the measured far-field radiation pattern of the model described in the examples of FIGS. 3 to 6 is in good agreement with the simulation results at 825 MHz xoy-plane.
  • the far-field radiation pattern of the model object described in the embodiments of Figs. 3 to 6 above at 825 MHz is measured, and the results are shown in Fig. 24.
  • the measured far-field radiation pattern of the model described in the examples in Figures 3 to 6 is in good agreement with the simulation results at 825 MHz xoz-plane.
  • Measurements are made on the model real object radiation pattern at 825 MHz on the yoz-plane in the embodiments of Figs. 3 to 6, and the results are shown in Fig. 25.
  • the measured far-field radiation pattern of the model described in the examples of FIGS. 3 to 6 is in good agreement with the simulation results at 825 MHz.
  • Measurements are made on the model real object radiation pattern at 2250 MHz xoy-plane in the embodiments of Figs. 3 to 6, and the results are shown in Fig. 26.
  • the measured far-field radiation pattern of the model described in the examples in Figures 3 to 6 is in good agreement with the simulation results at the 2250 MHz xoy-plane.
  • Measurements are made on the model real object radiation pattern at 2250 MHz xoz-plane in the embodiments of Figs. 3 to 6, and the results are shown in Fig. 27.
  • the measured far-field radiation pattern of the model described in the examples of FIGS. 3 to 6 is in good agreement with the simulation results at the 2250 MHz xoz-plane.
  • the model real object at the 2250 MHz yoz-plane far-field radiation pattern of the model described in the embodiments of FIGS. 3 to 6 is measured, and the results are shown in FIG. 28.
  • the measured far-field radiation pattern of the model described in the examples of FIGS. 3 to 6 is in good agreement with the simulation results at 2250 MHz.
  • the low-band S11 parameters of the embodiment in FIG. 7 are simulated and calculated, and the result is shown in FIG. 29.
  • the impedance bandwidth of the antenna in the low frequency band is 730MHz-1100MHz. It is explained that the implementation manners provided in the embodiments of the present application can also be used alone to meet the needs of low frequencies, and the bandwidth is wider than when high frequencies are considered.
  • the high-band S11 parameters of the embodiment in FIG. 8 are simulated and calculated, and the results are shown in FIG. 30.
  • the impedance bandwidth of the antenna in the high frequency range is 1580MHz-2890MHz. It is explained that the implementation manners provided in the embodiments of the present application can also be used alone to meet the needs of high frequencies, and the bandwidth is wider than when low frequencies are considered.
  • the S11 parameter of the embodiment in FIG. 9 is simulated and calculated, and the result is shown in FIG. 31.
  • the impedance bandwidth of the antenna is 690MHz-1070MHz and 1630MHz-2940MHz. Therefore, it is proved that the antenna structure described in the embodiment of the present application can still maintain a good working state under a more complicated working environment.
  • the S11 parameter of the embodiment in FIG. 10 is simulated and calculated, and the result is shown in FIG. 32.
  • the impedance bandwidth of the antenna is 698MHz-1080MHz and 1680MHz-2920MHz.
  • the above embodiments are only examples of some antennas. If the size or material of the floor changes, the antenna unit only needs to be adjusted, and the antenna can continue to work. That is to say, for different kinds of working environments, the technical solution adopted in this civilization embodiment can be applied to construct a mobile terminal antenna based on a composite left and right hand transmission line.
  • the patch structure in the embodiment is not limited to regular geometric shapes such as a rectangle and a circle, and the shapes of LR, LR, CL, and LR are not limited to rectangular.
  • the embodiment of the present application designs a resonant unit with a high-frequency and wide-band on the basis of ensuring low-frequency operating characteristics by using a composite left-handed transmission line (CRLH-TL).
  • CRLH-TL composite left-handed transmission line
  • the CRLH-TL technology is used to design a resonance unit with a low frequency and broadband.
  • the impedance bandwidth of the high frequency band is improved, so that it can cover the high frequency operating frequency band.
  • These two structures together form a resonance unit that can cover multiple frequency bands such as LTE700 / GSM850 / GSM900 DCS1800 / PCS1900 / UMTS / LTE2300 / LTE2600.
  • An embodiment of the present application further provides a mobile terminal, where the mobile terminal includes the mobile terminal antenna described above.
  • the mobile terminal can be implemented in various forms.
  • the mobile terminals described in the embodiments of the present application may include mobile phones, smart phones, notebook computers, digital broadcast receivers, Personal Digital Assistants (PDAs), PADs (tablet computers), and Portable Media Players (PMPs). , Portable multimedia players), navigation devices, and more.
  • PDAs Personal Digital Assistants
  • PADs tablet computers
  • PMPs Portable Media Players
  • Portable multimedia players Portable multimedia players
  • navigation devices and more.

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Abstract

本申请实施例公开了一种移动终端天线和移动终端，所述移动终端天线包括介质基板和位于所述介质基板一侧的地板，还包括：设置于所述介质基板另一侧的近馈单元、近地单元和耦合单元，所述近地单元一端与所述耦合单元相连，另一端与所述地板相连；所述耦合单元和近地单元等效为左手电感；所述近馈单元等效为右手电感；所述耦合单元与近馈单元相耦合，等效为左手电容；所述耦合单元与所述地板相耦合，等效为右手电容；所述近馈单元、近地单元、耦合单元和地板构成复合左右手传输线结构。

Description

一种移动终端天线和移动终端
相关申请的交叉引用
本申请基于申请号为201810672340.7、申请日为2018年06月26日的中国专利申请提出，并要求该中国专利申请的优先权，该中国专利申请的全部内容在此引入本申请作为参考。
技术领域
本申请涉及天线领域但不限于天线领域，尤指一种移动终端天线和移动终端。
背景技术
随着时代的发展，移动通信系统经过几代革新，从最开始的1G、2G发展到如今的4G，相应于移动通信系统的发展，移动终端天线也在不断发展创新以适应实际需求。现阶段移动终端天线需要满足2G、3G和4G的通信需求，分别要覆盖：LTE700/GSM850/GSM900DCS1800/PCS1900/UMTS/LTE2300/LTE2600等多个频带，因此移动终端天线设计时需要兼顾多频带和宽频带特性。此外，为了满足消费者的智能化需要，移动终端上集成的功能越来越多，同时留给天线的设计空间也越来越小。
目前，手机天线设计方案中最常见的是采用平面倒F天线(Planar Inverted F-shaped Antenna，PIFA)、单极子天线和环形天线等。PIFA天线同时满足了外形小巧、易于实现、生产一致性好等优点。而单极子天线体积更加小巧，且具有更宽的带宽。但是PIFA天线带宽较窄，而单极子天线容易受到周围环境的影响，现阶段又很难使天线下方完全对地净空。
发明内容
本申请实施例提供了一种移动终端天线和移动终端，以在满足移动终端天线体积要求的情况下覆盖多个频带。
本申请实施例提供了一种移动终端天线，包括介质基板和位于所述介质基板一侧的地板，还包括：设置于所述介质基板另一侧的近馈单元、近地单元和耦合单元，其中
所述近地单元一端与所述耦合单元相连，另一端与所述地板相连；所述耦合单元和近地单元等效为左手电感；所述近馈单元等效为右手电感；所述耦合单元与近馈单元相耦合，等效为左手电容；所述耦合单元与所述地板相耦合，等效为右手电容；所述近馈单元、近地单元、耦合单元和地板构成复合左右手传输线结构。
本申请实施例还提供一种移动终端，包括上述移动终端天线。
本申请实施例基于复合左右手传输线设计的移动终端天线，能满足移动通信需要，且结构简单，布局紧凑，能大大节省天线空间。
本申请的其它特征和优点将在随后的说明书中阐述，并且，部分地从说明书中变得显而易见，或者通过实施本申请而了解。本申请的目的和其他优点可通过在说明书、权利要求书以及附图中所特别指出的结构来实现和获得。
附图说明
附图用来提供对本申请技术方案的进一步理解，并且构成说明书的一部分，与本申请的实施例一起用于解释本申请的技术方案，并不构成对本 申请技术方案的限制。
图1为理想复合左右手传输线电路模型示意图。
图2为复合左右手传输线色散关系示意图。
图3为本申请一实施例的移动终端天线整体结构示意图。
图4为图3实施例的移动终端体天线结构示意图。
图5为图4实施例的移动终端天线的俯视图。
图6为图4实施例的移动终端天线的侧视图。
图7为本申请一实施例的移动终端天线结构示意图(去掉高频谐振单元)。
图8为本申请一实施例的移动终端天线结构示意图(去掉低频谐振单元)。
图9为本申请一实施例的天线终端天线结构示意图(增加了金属元器件)。
图10为本申请另一实施例的天线终端天线结构示意图(近馈单元中的矩形环替换为椭圆形环)。
图11为本申请另一实施例的天线终端天线结构示意图(改变形状)。
图12为图3～6实施例的S11参数进行仿真计算示意图。
图13为图3～6实施例的输入阻抗示意图。
图14为图3～6实施例在低频工作频段(690MHz-960MHz)的辐射效率示意图。
图15为图3～6实施例在高频工作频段(1710MHz-2690MHz)的辐射效率示意图。
图16为图3～6实施例在825MHz时xoy-面远场辐射方向图。
图17为图3～6实施例在825MHz时xoz-面远场辐射方向图。
图18为图3～6实施例在825MHz时yoz-面远场辐射方向图。
图19为图3～6实施例在2250MHz时xoy-面远场辐射方向图。
图20为图3～6实施例在2250MHz时xoz-面远场辐射方向图。
图21为图3～6实施例在2250MHz时yoz-面远场辐射方向图。
图22为图3～6实施例的S11参数实测图。
图23为图3～6实施例在825MHz时xoy-面远场辐射方向图实测仿真对比图。
图24为图3～6实施例在825MHz时xoz-面远场辐射方向图实测仿真对比图。
图25为图3～6实施例在825MHz时yoz-面远场辐射方向图实测仿真对比图。
图26为图3～6实施例在2250MHz时xoy-面远场辐射方向图实测仿真对比图。
图27为图3～6实施例在2250MHz时xoz-面远场辐射方向图实测仿真对比图。
图28为图3～6实施例在2250MHz时yoz-面远场辐射方向图实测仿真对比图。
图29为图7实施例的低频S11参数仿真示意图。
图30为图8实施例的高频S11参数仿真示意图。
图31为图9实施例的S11参数仿真示意图。
图32为图10实施例的S11参数仿真示意图。
具体实施方式
为使本申请的目的、技术方案和优点更加清楚明白，下文中将结合附图对本申请的实施例进行详细说明。需要说明的是，在不冲突的情况下，本申请中的实施例及实施例中的特征可以相互任意组合。
本申请实施例提供一种移动终端天线，采用基于复合左右手传输线的 方式，以覆盖多个工作频带、满足终端设计空间狭小的情况。
下面简述一下复合左右手传输线的原理。
根据Chu定理，电小天线所能达到的最大带宽与天线占用的空间成正比，要获得大的带宽，必须保证为电小天线预留足够的空间。而Chu定理的建立是基于电磁波的右手定则，即电磁波在自然界的大部分介质中传播时(介电常数ε>0,磁导率μ>0)，该处电磁场的能量流密度S＝E×H，其中电场强度为E，磁场强度为H，玻印亭矢量S的方向是电磁波传播的方向，即电磁能传递的方向，E、H、S彼此垂直构成右手螺旋关系。
对于电磁波在一般介质中的传播，即右手材料，也可以用传输线理论进行分析，即单位长度的传输线可等效为串联分布电感和并联分布电容，色散关系，也就是相位常数与频率成正比。
如果存在一种材料，其ε<0、μ<0，那么电磁波在其中传播时电场强度、磁场强度和波矢量之间满足左手螺旋关系，谐振频率与物理尺寸之间不再存在必然的约束关系。
对于左手材料，可以等效为单位长度的串联分布电容和并联分布电感，相位传播常数为负，相速度和群速度反向。
实际中的左手材料都是利用自然界存在的右手材料人工构造的，所以不可能得到单纯的左手传输线，两者同时存在，即复合左右手传输线。
对于复合左右手传输线来说，兼具左手模式和左右手模式，当传播常数为纯实数时为传输禁带。这种情况是复合左右手传输线的不平衡状态，串联谐振点和并联谐振点不同。若串联谐振和并联谐振相同，则得到平衡态，此时左手特性频率区与右手特性频率区之间没有任何阻带。这种情况下谐振频率与物理尺寸之间就不存在必然的约束关系，只要通过改变物理结构来改变等效的电容和电感值，就能改变零阶谐振点的谐振中心频率。可以利用这一点实现天线的小型化。
如图1所示，为理想复合左右手传输线电路模型，由四部分组成：(a)右手电感L′ R、(b)右手电容C′ R、(c)左手电感L′ L和(d)左手电容C′ L。其中(a)和(d)构成了等效电路中的串联部分，(b)和(c)构成了等效电路中的并联部分；(a)和(c)构成了等效电路中的电感部分，(b)和(d)构成了等效电路中的电容部分；(a)和(b)构成了等效电路中的右手部分，(b)和(d)构成了等效电路中的左手部分。
复合左右手传输线的串联谐振点可用
Figure PCTCN2019084145-appb-000001
表征，并联谐振点可用
Figure PCTCN2019084145-appb-000002
来表征，其色散关系示意图如图2所示。通常情况下复合左右手传输线的串联谐振点和并联谐振点不同，这种情况就称为复合左右手传输线的不平衡状态，即ω se≠ω sh。当复合左右手传输线工作在非平衡状态时，在ω se和ω sh之间的工作频带内表现为阻带。为了获得较好的宽带特性，可以通过调整左手电容电感和右手电容电感对应的物理结构来改变等效电路中的各个电参数，从而使复合左右手传输线工作在平衡状态。当复合左右手传输线工作于平衡状态时，其串联谐振和并联谐振相等时，有ω se＝ω sh＝ω 0，即L′ RC′ L＝L′ LC′ R，此时复合左右手传输线达到平衡，在过度频率ω 0上相位常数β＝0，但是因为群速v g＝dω/dβ≠0，所以波还会传播，此时复合左右手传输线无阻带。
为了利用复合左右手传输线平衡状态下的这种宽频带特性，本申请实施例通过天线的物理结构来实现复合左右手传输线结构，从而满足移动终端天线的宽频带需要。通常情况下可以通过微带线、带状线、共面波导等等分布式组件来构成这种LC网络。例如利用微带线的实现方法，其左手电感L′ L的主要形式包括螺旋线电感和短路线电感，左手电容C′ L则由交指电 容、缝隙电容等形式实现，右手电容电感则由微带线、微带贴片实现。
如图3～6所示，本申请实施例的移动终端天线，包括介质基板1和位于所述介质基板一侧的地板2，还包括：设置于所述介质基板另一侧的近馈单元7、近地单元5和耦合单元11；所述近地单元5一端与所述耦合单元11相连，另一端与所述地板2相连；所述耦合单元11和近地单元5等效为左手电感；所述近馈单元7等效为右手电感；所述耦合单元11与近馈单元7相耦合，等效为左手电容；所述耦合单元与所述地板相耦合，等效为右手电容；所述近馈单元、近地单元、耦合单元和地板构成复合左右手传输线结构。
本申请实施例基于复合左右手传输线设计的天线，能满足移动通信需要，且结构简单，布局紧凑，能大大节省天线空间。
如图4所示，所述近地单元5为短路线。所述近馈单元7包括相连的环形部分71和馈线72，所述馈线72的一端与所述环形部分71相连，另一端与馈电点8相连。
其中，所述环形部分71与所述介质基板1平行，可采用但不限于矩形或椭圆型。所述馈线72可采用但不限于L型结构或直线型结构。
在其他实施例中，近馈单元7也可以不采用环形结构，而采用矩形、椭圆型等结构。
在本申请实施例中，所述耦合单元11与近馈单元7之间设置有缝隙，通过所述缝隙，所述耦合单元11与近馈单元7之间形成左手电容效应。
在其他实施例中，耦合单元11也可以采用交指结构，构成左手电容。
所述耦合单元11与所述地板2之间设置有介质基板1，形成右手电容效应。
在本申请实施例中，所述耦合单元11包括低频谐振单元3和高频谐振单元4中的至少之一。
其中，如图4所示，所述低频谐振单元3包括第一分支31和第二分支32，所述第一分支31为U型结构，所述第二分支32为折线结构，所述第一分支31和第二分支32通过所述近地单元5相连。
其中，所述第一分支31可包括依次相连的第一段311、第二段312和第三段313，其中，所述第一段311与所述近地单元5相连，位于所述介质基板1表面，所述第二段312垂直于所述介质基板1，所述第三段313远离所述介质基板1，位于所述第一段311上方，且所述第三段313中部分平面垂直于所述介质基板1，部分平面平行于所述介质基板1。
所述第二分支32可包括依次相连的第四段321、第五段322和第六段323，其中，所述第四段321与所述近地单元5相连，位于所述介质基板1表面，所述第五段322垂直于所述介质基板1，所述第六段323远离所述介质基板1，且向远离所述第四段321的方向延伸，所述第六段323中部分平面垂直于所述介质基板1，部分平面平行于所述介质基板1。
如图4所示，所述高频谐振单元4至少包括第一贴片41，所述第一贴片41垂直于所述介质基板1。
其中，所述第一贴片41可采用但不限于矩形，位于所述低频谐振单元中的第一分支31U型的内部。
在一实施例中，所述高频谐振单元4还包括第二贴片42，所述第二贴片42垂直于所述介质基板1。
其中，所述第二贴片42可采用但不限于矩形，位于所述低频谐振单元的第二分支中的第五段322和第六段323以及所述介质基板1之间。
所述第二贴片42可以认为是单极子贴片，采用第二贴片42可以改善天线高频谐振特性，增加天线阻抗带宽。
对于低频工作情况，低频谐振单元3与近馈单元7以串联左手电容相耦合，而近馈单元7本身即等效为串联右手电感，即组成了复合左右手传 输电路中的串联电容和串联电感。通过改变低频谐振单元3和近馈单元7之间的间距即可改变等效的左手电容的大小，同理通过改变近馈单元7的宽窄和长度即可改变对应的右手电感的大小。从而通过改变天线的物理尺寸来调整天线的串联谐振点。
低频谐振单元3又存在对地右手电容，与近地单元5共同形成了复合左右手传输电路中的并联电容和并联电感。从而形成了一个完整的可以在低频工作频带的复合左右手传输线电路。通过改变低频谐振单元3的面积可以对应改变电路中的右手电容大小，通过改变短路线5和/或低频谐振单元3的尺寸可以改变对应左手电感的大小，从而通过调整天线物理尺寸来改变天线对应的并联谐振点。
对于高频工作情况，与低频情况类似，高频谐振单元4与近馈单元7形成左手电容效应，而近馈单元7本身即等效为串联右手电感，组成了复合左右手传输电路中的串联电容和串联电感，高频谐振单元4又存在对地右手电容，与近地单元5共同形成了复合左右手传输电路中的并联电容和并联电感。从而形成了可以工作在高频工作频带的复合左右手传输电路。通过改变高频谐振单元4中第一贴片41和/或第二贴片42的大小和近馈单元7的尺寸来调整对应的左手电容和右手电感，从而调整对应等效电路的串联谐振点。通过改变高频谐振单元4中第一贴片41和/或第二贴片42的大小和短路线5的尺寸可以改变并联谐振点。
在本申请实施例中，采用了基于复合左右手传输线的立体结构，引入传统矩形单极子结构达到更宽频带需求，可以分别覆盖低频高频多个工作频带、满足终端设计空间狭小的情况。
在本申请一实施例中，天线整体结构参照图3～6所示，大小为65mm×10mm×5.8mm。天线地板与常规智能手机设备地板大小相仿，介质基板选用的为FR4基板，尺寸为65mm×120mm×0.8mm。其中，近地单元5 中，短线路的长度为5～7mm，宽度为0.5～2mm。近馈单元7中，环形部分71可以是外环：64mm×4mm，内环63mm×2.6mm。低频谐振单元3中，第一分支中第一分段311和第三分段313长度为32～36mm，宽度为2mm左右，第二分段312长度为5mm左右，宽度1mm左右；第二分支中第四分段321长度为34～38mm，第五分段322为5mm左右，宽度1mm左右，第六分段323长度28mm～32mm，宽度2mm左右；第一分支和第二分支与环形部分71的缝隙4mm左右。高频谐振单元4中第一贴片41的尺寸可以是：19.5mm×3mm，第二贴片42的尺寸可以是17.5mm×3mm。
需要说明的是，这只是列举了一种天线尺寸，如果地板或者介质基板发生改变，只需要对基于复合左右手传输线的移动终端天线进行适当调整即可正常工作，也就是说，基于复合左右手传输线的移动终端天线可以具有多种尺寸，可以与其他尺寸的地板和不同材质的介质基板相结合。
如图7所示，本申请另一实施例在图3～6实施例的基础上，去掉了高频谐振单元，提供了一种基于复合左右手结构的低频段移动终端天线，可应用于手机等移动终端。其工作原理与图3～6实施例中的低频工作情况相同。
如图8所示，本实施例在图3～6实施例的基础上，去掉了低频谐振单元3，提供了一种基于复合左右手结构的高频频段移动终端天线，可应用于手机等移动终端。其工作原理与图3～6实施例中的高频工作情况相同。同样的，天线结构中增加了一段矩形单极子结构(第二贴片42)，改善了天线谐振特性，增加了阻抗带宽。
如图9所示，在图3～6实施例的基础上，在天线单元的下方增加了在实际应用中可能出现的金属元器件9。
如图10所示，为另一种移动终端天线实现形式，其工作原理与图3～6实施例相似，其中，将近馈单元中的矩形环替换为椭圆形环。
如图11所示，为另一种移动终端天线实现形式，其中，其形状与图3～6改变较大，但其原理与上述描述相同。
其中，所述耦合单元11为一体结构，包括相连的第一平面部分111和第二平面部分112，所述第一平面部分111垂直于所述介质基板1，所述第二平面部分112平行于所述介质基板1。
所述近地单元5为短路线。所述近馈单元7包括相连的贴片部分73和馈线72，所述馈线72的一端与所述贴片部分73相连，另一端与馈电点相连；所述贴片部分73与所述第二平面部分112之间设置有缝隙，通过所述缝隙，所述耦合单元11与近馈单元7之间形成左手电容效应。
所述贴片部分73可以采用但不限于矩形，与所述介质基板平行，且与所述第二平面部分位于同一平面。
图11实施例利用可重构的方式，实现了在工作频带内良好的工作状态。
对上述图3～6实施例的S11参数进行仿真计算，结果如图12所示。以为S11小于-6dB为标准，图3～6实施例中天线的阻抗带宽为680MHz-1100MHz和1690MHz-3000MHz。说明可以直接覆盖LTE700，GSM850，GSM900，DCS1800，PCS1900，UMTS，LTE2300，LTE2600等多个频带，具有较宽的工作频带。
对上述图3～6实施例的输入阻抗参数进行仿真计算，结果如图13所示。从图13可以看出，天线在低频和高频部分具有很好的谐振特性。
对上述图3～6实施例的低频段(690-960MHz)辐射效率进行仿真计算，结果如图14所示。可以看出天线在低频段(690-960MHz)辐射效率大于48％。
对上述图3～6实施例的高频段辐射效率(1710MHz-2690MHz)进行仿真计算，结果如图15所示。可以看出天线在高频段(1710MHz-2690MHz)辐射效率大于62.5％。
对上述图3～6实施例在825MHz xoy-面远场辐射方向图进行仿真，结果如图16所示。对上述图3～6实施例在825MHz xoz-面远场辐射方向图进行仿真，结果如图17所示。对上述图3～6实施例在825MHz yoz-面远场辐射方向图进行仿真，结果如图18所示。对上述图3～6实施例在2250MHz xoy-面远场辐射方向图进行仿真，结果如图19所示。对上述图3～6实施例在2250MHz xoz-面远场辐射方向图进行仿真，结果如图20所示。对上述图3～6实施例在2250MHz yoz-面远场辐射方向图进行仿真，结果如图21所示。上述图16～21显示了天线各个频段的方向图，都说明达到了业内对方向图的要求。
利用矢量网络分析仪对图3～6实施例所述模型实物的回波损耗进行测量，结果如图22所示。以为S11小于-6dB为标准，图3～6实施例中天线的实测阻抗带宽为680MHz-1100MHz和1480MHz-3000MHz。说明可以覆盖LTE700，GSM850，GSM900，DCS1800，PCS1900，UMTS，LTE2300，LTE2600等多个频带，具有较宽的工作频带。
对上述图3～6实施例所述模型实物在825MHz xoy-面远场辐射方向图进行测量，结果如图23所示。图3～6实施例所述模型的实测远场辐射方向图在825MHz xoy-面与仿真结果一致性良好。
对上述图3～6实施例所述模型实物在825MHz xoz-面远场辐射方向图进行测量，结果如图24所示。图3～6实施例所述模型的实测远场辐射方向图在825MHz xoz-面与仿真结果一致性良好。
对上述图3～6实施例所述模型实物在825MHz yoz-面远场辐射方向图进行测量，结果如图25所示。图3～6实施例所述模型的实测远场辐射方向图在825MHz yoz-面与仿真结果一致性良好。
对上述图3～6实施例所述模型实物在2250MHz xoy-面远场辐射方向图进行测量，结果如图26所示。图3～6实施例所述模型的实测远场辐射 方向图在2250MHz xoy-面与仿真结果一致性良好。
对上述图3～6实施例所述模型实物在2250MHz xoz-面远场辐射方向图进行测量，结果如图27所示。图3～6实施例所述模型的实测远场辐射方向图在2250MHz xoz-面与仿真结果一致性良好。
对上述图3～6实施例所述模型实物在2250MHz yoz-面远场辐射方向图进行测量，结果如图28所示。图3～6实施例所述模型的实测远场辐射方向图在2250MHz yoz-面与仿真结果一致性良好。
对上述图7实施例的低频段S11参数进行仿真计算，结果如图29所示。天线在低频段的阻抗带宽为730MHz-1100MHz。说明本申请实施例提出的实现方式也可以单独用来满足低频需要，且带宽比考虑有高频时更宽。
对上述图8实施例的高频段S11参数进行仿真计算，结果如图30所示。天线在高频段的阻抗带宽为1580MHz-2890MHz。说明本申请实施例提出的实现方式也可以单独用来满足高频需要，且带宽比考虑有低频时更宽。
对上述图9实施例的S11参数进行仿真计算，结果如图31所示。天线的阻抗带宽为690MHz-1070MHz和1630MHz-2940MHz。从而证明本申请实施例所述的天线结构在较复杂的工作环境下仍可以保持良好的工作状态。
对上述图10实施例的S11参数进行仿真计算，结果如图32所示。天线的阻抗带宽为698MHz-1080MHz和1680MHz-2920MHz。从而证明本申请实施例在实现形式上的多样性，不仅局限于矩形，其他形式如椭圆形也可以获得较好的工作状态。
上述实施例只是列举了部分天线实例，如果地板尺寸或者材质发生改变，只需要对天线单元进行调整，天线仍可以继续工作。也就是说，对于不同种工作环境，均能应用本文明实施例所采用的技术方案，构建基于复合左右手传输线的移动终端天线。此外，实施例中的贴片结构不局限于矩 形、圆形等规则的几何形状，LR、LR、CL与LR的形状也不局限于矩形。
综上所述，本申请实施例通过复合左右手传输线(CRLH-TL)的形式在保证低频工作特性的基础上，设计具有高频宽带的谐振单元。其中，利用CRLH-TL技术，设计具有低频宽带的谐振单元。通过增加一段传统的矩形单极子结构，改善高频段的阻抗带宽，从而可以覆盖高频工作频段。这两种结构共同组成了可以覆盖LTE700/GSM850/GSM900 DCS1800/PCS1900/UMTS/LTE2300/LTE2600等多个频带的谐振单元。
本申请实施例还提供一种移动终端，所述移动终端包括上述的移动终端天线。
移动终端可以以各种形式来实施。例如，本申请实施例中描述的移动终端可以包括诸如移动电话、智能电话、笔记本电脑、数字广播接收器、PDA(Personal Digital Assistant，个人数字助理)、PAD(平板电脑)、PMP(Portable Media Player，便携式多媒体播放器)、导航装置等等的移动终端。
然而，本领域技术人员将理解的是，除了特别用于移动目的的元件之外，根据本申请的实施方式的构造也能够应用于固定类型的终端。以及诸如数字TV、台式计算机等等的固定终端。
虽然本申请所揭露的实施方式如上，但所述的内容仅为便于理解本申请而采用的实施方式，并非用以限定本申请。任何本申请所属领域内的技术人员，在不脱离本申请所揭露的精神和范围的前提下，可以在实施的形式及细节上进行任何的修改与变化，但本申请的专利保护范围，仍须以所附的权利要求书所界定的范围为准。

Claims (17)

  1. 一种移动终端天线，包括：介质基板、位于所述介质基板一侧的地板，设置于所述介质基板另一侧的近馈单元及近地单元和耦合单元，其中，
    所述近地单元一端与所述耦合单元相连，另一端与所述地板相连；所述耦合单元和近地单元等效为左手电感；所述近馈单元等效为右手电感；所述耦合单元与近馈单元相耦合，等效为左手电容；所述耦合单元与所述地板相耦合，等效为右手电容；所述近馈单元、近地单元、耦合单元和地板构成复合左右手传输线结构。
  2. 如权利要求1所述的移动终端天线，其中，
    所述耦合单元与近馈单元之间设置有缝隙。
  3. 如权利要求1或2所述的移动终端天线，其中，
    所述耦合单元包括低频谐振单元和高频谐振单元中的至少之一。
  4. 如权利要求3所述的移动终端天线，其中，所述低频谐振单元包括第一分支和第二分支，所述第一分支为U型结构，所述第二分支为折线结构，所述第一分支和第二分支通过所述近地单元相连。
  5. 如权利要求4所述的移动终端天线，其中，所述第一分支包括依次相连的第一段、第二段和第三段，其中，所述第一段与所述近地单元相连，位于所述介质基板表面，所述第二段垂直于所述介质基板，所述第三段远离所述介质基板，位于所述第一段上方，且所述第三段中部分平面垂直于所述介质基板，部分平面平行于所述介质基板。
  6. 如权利要求4所述的移动终端天线，其中，所述第二分支包括依次相连的第四段、第五段和第六段，其中，所述第四段与所述近地单元相连，位于所述介质基板表面，所述第五段垂直于所述介质基板，所述第六段远离所述介质基板，且向远离所述第四段的方向延伸，所述第六段中部分平面垂直于所述介质基板，部分平面平行于所述介质基板。
  7. 如权利要求3所述的移动终端天线，其中，所述高频谐振单元包括第一贴片，所述第一贴片垂直于所述介质基板。
  8. 如权利要求7所述的移动终端天线，其中，所述第一贴片为矩形，位于所述低频谐振单元中的第一分支U型的内部。
  9. 如权利要求7所述的移动终端天线，其中，所述高频谐振单元还包括第二贴片，所述第二贴片垂直于所述介质基板。
  10. 如权利要求9所述的移动终端天线，其中，所述第二贴片为矩形，位于所述低频谐振单元的第二分支中的第五段和第六段以及所述介质基板之间。
  11. 如权利要求1或2所述的移动终端天线，其中，所述近馈单元包括相连的环形部分和馈线，所述馈线的一端与所述环形部分相连，另一端与馈电点相连。
  12. 如权利要求11所述的移动终端天线，其中，
    所述环形部分与所述介质基板平行，为矩形或椭圆型；
    所述馈线为L型结构或直线型结构。
  13. 如权利要求1或2所述的移动终端天线，其中，所述耦合单元包括相连的第一平面部分和第二平面部分，所述第一平面部分垂直于所述介质基板，所述第二平面部分平行于所述介质基板。
  14. 如权利要求13所述的移动终端天线，其中，所述近馈单元包括相连的贴片部分和馈线，所述馈线的一端与所述贴片部分相连，另一端与馈电点相连；所述贴片部分与所述第二平面部分之间设置有缝隙。
  15. 如权利要求14所述的移动终端天线，其中，
    所述贴片部分为矩形，与所述介质基板平行，且与所述第二平面部分位于同一平面。
  16. 如权利要求1或2所述的移动终端天线，其中，
    所述近地单元为短路线。
  17. 一种移动终端，包括如权利要求1～16中任意一项所述的移动终端天线。
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US20210226321A1 (en) 2021-07-22
CN110649375A (zh) 2020-01-03
US11509041B2 (en) 2022-11-22

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