WO2018102872A1 - Contrôleur de moteur en ca sans capteurs - Google Patents

Contrôleur de moteur en ca sans capteurs Download PDF

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Publication number
WO2018102872A1
WO2018102872A1 PCT/AU2017/051343 AU2017051343W WO2018102872A1 WO 2018102872 A1 WO2018102872 A1 WO 2018102872A1 AU 2017051343 W AU2017051343 W AU 2017051343W WO 2018102872 A1 WO2018102872 A1 WO 2018102872A1
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WIPO (PCT)
Prior art keywords
controller
speed
motor
value
torque
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PCT/AU2017/051343
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English (en)
Inventor
Gregory Peter Hunter
Original Assignee
University Of Technology Sydney
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Priority claimed from AU2016905064A external-priority patent/AU2016905064A0/en
Application filed by University Of Technology Sydney filed Critical University Of Technology Sydney
Publication of WO2018102872A1 publication Critical patent/WO2018102872A1/fr

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/0085—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed
    • H02P21/0089—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation specially adapted for high speeds, e.g. above nominal speed using field weakening
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/26—Rotor flux based control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
    • H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
    • H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters
    • H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using dc to ac converters or inverters with pulse width modulation

Definitions

  • the present disclosure relates generally to the field of AC motor speed control and in particular to Feed Forward Torque Control (FFTC) for controlling the torque and speed of an electric motor without the use of sensors for measuring the rotor angular position or speed of rotation.
  • FFTC Feed Forward Torque Control
  • a feature of the traditional control structure is the need to measure or estimate the rotor position and in particular the electrical phase angle of the rotor flux direction or rotor angle.
  • the rotor position must be estimated from the motor voltages and currents.
  • PMSM permanent magnet synchronous motors
  • Described herein is a sensorless motor controller where fast torque response together with low torque ripple is achieved using low bandwidth current sensing whilst requiring low processor speed. Also described is a speed controller to complement the torque controller, and a position controller that has an inbuilt speed profiler to bring rotor position to its final value, obviating the need for an external movement profiler on the position input. Instead of measuring the rotor position, the position is determined indirectly by providing the conditions to force the rotor to lock into the required rotor position.
  • a controller for an AC electric motor comprising: a power converter driving the AC electric motor; a feed forward converter that derives voltage values provided to the power converter; and a load model unit configured to derive an applied rotor speed used for an input to the feed forward converter, wherein the applied rotor speed is derived using a stabilising speed correction value with a substantially zero average value.
  • the load model unit may be configured to derive the applied rotor speed by adding the stabilising speed correction value.
  • the load model unit may comprise a load model and a load model compensator.
  • the feed forward converter may derive the voltage values from a d-axis current value, a q-axis current value, and an angle of rotation value.
  • the controller may further comprise an integrator, and the integrator may derive the angle of rotation value by integration of the applied rotor speed and may provide the angle of rotation to the feed forward converter.
  • a controller for an AC electric motor comprising: a power converter driving the AC electric motor; a feed forward converter that derives voltage values provided to the power converter; and a load model unit configured to provide an angle of rotation value to the feed forward converter, the load model unit comprising: a load model; and a load model compensator comprising: a
  • the feedback proportional controller may operate to limit a DC gain of the proportional plus integral controller.
  • the motor speed may be an estimated motor speed.
  • the load model unit may further comprise a stability controller that adds a stabilising speed correction value to the estimated motor speed resulting in a corrected estimated motor speed; and the controller may further comprise an integrator that integrates the corrected estimated motor speed to produce the angle of rotation value used to determine the voltage values.
  • a controller for an AC electric motor comprising: a power converter driving the AC electric motor; a feed forward converter that derives voltage values provided to the power converter; and a flux weakening unit for operating the AC electric motor above base speed, the flux weakening unit comprising a flux weakening controller that determines and sets a d-axis current value based on a calculated flux linkage reference so that a peak output voltage of the power converter remains below a predetermined value during flux weakening.
  • the d-axis current value, i d ' may be determined from a relationship with the calculated flux linkage reference, l r , the applied flux linkage, d j , and the estimated stator phase inductance, L , as described by
  • the applied flux linkage may be described as follows: sign(w') 3 ⁇ 4 - v d ' 2 - v R ' q
  • the flux weakening controller may comprise: a comparator that compares an applied d-axis flux linkage value and a threshold d-axis flux linkage; and an integral controller configured to stabilise the d-axis current value used to generate the d-axis flux linkage value.
  • the controller may further comprise a z-domain integrator with an output limiter, and a flux gain operator.
  • the integral controller may cause a feedback loop gain of about 0.5 at a sampling frequency.
  • the controller may further comprise a load model unit configured to derive an applied rotor speed used for an input to the feed forward converter, the load model unit comprising: a load model; and a load model compensator.
  • the feed forward converter may generate the voltage signals based on d and q axis flux linkage values.
  • a controller for an AC electric motor comprising: a power converter driving the AC electric motor; a torque controller that provides control signals to the power converter; and a speed controller that controls a motor torque, the speed controller comprising: a comparator that compares a reference speed with an estimated rotor speed to produce a difference value; a proportional compensator that modifies the difference value to produce a first torque reference value; and a summing device that adds the first torque reference value and an estimated load torque to produce a torque command.
  • the estimated rotor speed may be fed back so that the first torque reference value is substantially an inertia-only torque reference.
  • the proportional compensator limits a compensator output based on a maximum acceleration setting.
  • the controller may further comprise a position controller that comprises a proportional gain compensator with a proportional gain that is a function of a position difference signal.
  • the position controller may further comprise a comparator that compares a reference position signal with an estimated rotor position signal to produce the position difference signal.
  • the position controller may provide a speed reference signal for use by the speed controller to control a speed of a rotor of the AC motor.
  • the speed controller after receiving the speed reference signal, may do one or both of the following: limit a rotor acceleration, and limit the rotor speed.
  • the controller may further comprise a load model unit configured to derive an applied rotor speed used for an input to the feed forward converter, the load model unit comprising: a load model; and a load model compensator.
  • a controller for an AC electric motor comprising: a speed controller for controlling a speed of a rotor of the AC motor; and a position controller provides a speed reference signal for use by the speed controller to control the speed of the rotor, wherein the position controller comprises a proportional gain compensator with a proportional gain that is a function of a position difference signal.
  • the position controller may further comprise a comparator that compares a reference position signal with an estimated rotor position signal to produce the position difference signal.
  • the controller may further comprise a resistive impedance correction that enables low speed damping.
  • a method of controlling an AC motor comprising: determining an applied rotor speed and an angle of rotation; providing the angle of rotation to a feed forward converter; and based on the angle of rotation, deriving and providing voltage values to a power converter for driving the AC motor.
  • Determining the applied rotor speed may comprise adding a stabilising speed correction value with a substantially zero average value.
  • Determining the applied rotor speed may comprise using feedback proportional control that is based on a function of an estimated motor speed.
  • the method may further comprise operating the AC motor in flux weakening by setting a d-axis current value based on a calculated flux linkage reference so that a peak output voltage of a power converter of the AC motor remains below a predetermined value during flux weakening.
  • the method may further comprise controlling a motor speed based on the applied rotor speed.
  • the controlling may comprise: comparing a reference speed with an estimated rotor speed to produce a difference value; modifying the difference value to produce a first torque reference value; and adding the first torque reference value and an estimated load torque to produce a torque command.
  • FIG. 1 schematically illustrates an embodiment of a torque controller
  • FIG. 2 is a diagrammatic representation of the fixed and rotating axes of a two-phase motor
  • FIG. 3 schematically illustrates an embodiment of a feed-forward converter
  • Fig. 4 schematically illustrates an embodiment of a pulse clipping and lengthening unit
  • FIG. 5 schematically illustrates an embodiment of a vector saturation unit
  • FIG. 6 is a diagram of an embodiment of a bridge PWM inverter
  • Fig. 7 graphically illustrates PWM output signals for the a phase of an embodiment of a motor
  • FIG. 8 schematically illustrates an embodiment of a load model with error compensation and damping control
  • FIG. 9 schematically illustrates the use of an embodiment of a flux weakening unit
  • Fig. 10 is a phasor diagram for a rotor retarded by an angle ⁇ ;
  • FIG. 11 schematically illustrates a high speed quasi-steady-state mechanical equivalent circuit
  • Fig. 12 schematically illustrates a high speed equivalent circuit of a controller and a motor with a load torque disturbance
  • FIG. 13 schematically illustrates a system equivalent circuit for a compliantly coupled load
  • FIG. 14 schematically illustrates a quasi-stationary open circuit q axis equivalent circuit with fixed d axis current
  • FIG. 15 schematically illustrates a complete q axis quasi steady state equivalent circuit of the motor at zero speed
  • FIG. 16 schematically illustrates a high speed quasi-steady state equivalent circuit
  • FIG. 17 schematically illustrates a zero speed equivalent circuit for a torque disturbance
  • Fig. 18 graphically illustrates a rotor angle offset response to a step torque disturbance at zero speed
  • Fig. 19 schematically illustrates a speed control loop
  • Fig. 20 schematically illustrates a position control loop
  • Fig. 21 graphically illustrates a speed and position trapezoidal profile
  • Fig. 22 is a block diagram of an embodiment of a motor controller. [65] In the drawings, like reference numerals designate similar parts.
  • a star superscript (*) is used to indicate command values
  • a tick O is used to indicate applied values
  • a tilde ( ⁇ ) is used to indicate estimated values.
  • Feed Forward Torque Control also makes use of the lack of need for rotor position observability but uses a different mechanism.
  • FFTC uses a load model to replace the rotor position estimator to provide rotor position and speed. A FFTC is described in
  • the load model calculates the rotor speed and position from the torque command.
  • Another modification to the traditional torque controller is the replacement of the feedback current controllers with feed forward controllers to generate the motor voltages. This frees up the q axis current feedback allowing it to be used to provide corrections to the load model.
  • the q axis current feedback signal is now used first to ensure the speed and position calculated from the load model track the actual values and second to provide rotor damping.
  • FIG. 1 of the drawings The basic structure of an embodiment of a torque controller 100 is shown in Fig. 1 of the drawings, as applied to a 2-phase motor 102 such as a hybrid stepper motor. It can equally be applied to a 3-phase motor by using a 3-phase output pulse width modulation (PWM) modulator instead of the 2-phase PWM modulator 104 used here, and then converting the three phase current measurements to two phase before further processing.
  • PWM pulse width modulation
  • the q axis current command ⁇ q together with the applied d axis current i'd and the applied rotor angle ⁇ ' are the inputs to a feed forward converter 106 which uses the machine equations to generate two phase voltages 108 for the PWM modulator 104.
  • the PWM modulator 104 uses a power inverter 2203 (see Fig. 22) to interface with the motor 102, with the PWM modulator and inverter 2203 forming a power converter 2205 to generate the motor voltages from the two phase voltages 108.
  • Other types of power converters that may be used include a matrix converter, or a cycloconverter (CCV).
  • the applied d axis current i'd is the command value i d with a correction applied by subtracting, via a difference operator 110, a proportion of the integral of the d axis current error Ai d generated by an integral controller 7 112. This correction operates slowly and is included to reduce the d axis current's sensitivity to errors in the estimated stator resistance and errors in the PWM.
  • a load model 120 includes additional compensation and stability control (described elsewhere herein).
  • the load model 120 is fed by the input q axis current command signal i * q and generates the applied rotor speed ⁇ ' which in turn is integrated at integrator 122 to generate the applied rotor angle ⁇ '.
  • q axis current error feedback ⁇ i q used to compensate for extra load torque and errors in the estimate for the inertia.
  • This ⁇ i q feedback signal is calculated via a difference operator 124, and is also used to dampen any motor instability by modulating the applied rotor speed ⁇ '.
  • the load model 120 and a following speed integrator 122 for generating the applied rotor angle replace the rotor position estimator required in conventional torque controllers, overcoming the problem of estimating the rotor position at zero speed.
  • the load model 120 and speed integrator 122 provide accurate and immediate values of load torque, speed and position for use in the outer speed and position control loops (described elsewhere herein). Details of an embodiment of a load model with compensator and stability correction are given elsewhere herein.
  • the feedback torque current error ⁇ i q does not respond to static rotor position errors and so cannot be used to correct the static applied rotor angle.
  • Static rotor position errors are addressed by locking the rotor position to the correct angle by applying a positive d axis current. This prevents the applied and actual rotor angles drifting apart at zero speed without affecting the dynamics of the controller. This d axis current can be reduced to zero to reduce power consumption at higher speeds where it is not needed.
  • the feed forward converter 106 generates the motor voltages required for the motor currents to track the input dq axes currents: applied i'd and command current q for an applied rotor angle ⁇ '.
  • the outputs of the feed forward converter 106 are the two-phase applied output motor voltages 108, v' a and ⁇ ' ⁇ , for use by the PWM modulator 104. These output voltages 108 are generated using the motor equations linking v' a and ⁇ ' ⁇ to i'd and i' q which are derived as described below with reference to Fig. 2 of the drawings.
  • FIG. 2 of the drawings illustrates fixed or stationary ⁇ axes 202, and rotating dq axes 204 of a two-phase motor 200 with rotor magnet 206.
  • the motor currents and voltages are related by the equation:
  • R and L are the stator resistance and inductance
  • ⁇ ⁇ and ⁇ ⁇ are the flux linkages in the a and ⁇ axes from the rotor magnet 206 and p is a differential operator. It is assumed the motor 200 has no saliency and thus the stator inductances do not vary with the rotor angle ⁇ .
  • equation (3) can be expanded further to the following familiar motor voltage equation in the dq frame:
  • Fig. 3 of the drawings shows a block diagram of an embodiment of a feed-forward converter 106 implementing equation (3) and using estimated values of the motor parameters.
  • the inputs to the feed-forward converter are the applied d axis current i'd, the applied q axis current i' q , and the applied angle ⁇ '.
  • the flux linkages X' d and X' q are obtained using an estimate of the motor inductance L 302, and are provided to a dq to ⁇ conversion block 304 that outputs the applied a and ⁇ axes flux linkages ⁇ ' ⁇ and ⁇ ' ⁇ .
  • the flux linkages are converted to the applied voltages, v'x a and v3 ⁇ 4, by a differentiation block 306.
  • Variables ⁇ ' ⁇ , v' « ? , ⁇ ' « ⁇ and v3 ⁇ 4 are the calculated IR voltage drops in the dq and ⁇ axes, and are added to voltages, v'x a and ⁇ ' ⁇ , at a summing operators 308, to provide the applied ⁇ axes voltages v' a and v'p.
  • the differentiation block 306 subtracts the previous sample value of the flux in the stationary frame from the present sample value and then scales by the inverse of the sample period x s .
  • the sample period for calculations is normally set to the PWM carrier period.
  • the dq to ⁇ conversion block 304 is usually implemented using sine and cosine look-up tables.
  • FIG. 4 A simple method that can be used to implement the required pulse lengthening is shown in Fig. 4 of the drawings.
  • This pulse lengthening and clipping unit 400 modifies the applied output stationary axes voltages v' a and ⁇ ' by clipping their vector magnitude using a Vector Saturation unit 402 and adding the resultant preceding errors 404 to the inputs at the following sample.
  • the Vector Saturation unit 402 clips the [v'o, v'p] vector to a fixed magnitude so it is limited to a circle in the space vector plane.
  • An embodiment of a method 500 of implementing clipping to a circle which avoids calculating the vector angle is shown in Fig. 5.
  • the vector components are normalised by dividing by a vector magnitude 502 then rescaled using the clipped value of the magnitude 504. This should be clipped to the maximum voltage output, V max , of the PWM modulator 104.
  • Another method of solving the clipping problem is to limit the rate of change of the q axis torque command current i q * to keep the peak output voltage to below the clipping level. This has the added advantage of reducing transients in the controller from sudden changes in i q * making for a smoother motor control.
  • the maximum rate of change, di q * /dt, should be set to +V DC /L where L is the estimate of the motor inductance.
  • This method may not always limit the output sufficiently to prevent clipping, therefore in some embodiments the controller also employs the pulse lengthening method.
  • a bridge PWM modulator 600 consisting of half bridge A 602 and half bridge B 604 on each output phase may be used for a two phase motor.
  • PWM modulator there are many possible modulation schemes.
  • double edge modulation of each half bridge 602, 604 using triangular carrier comparison may be used.
  • a triangular carrier wave 702 of half the PWM frequency is compared with a desired output 704, v' a , to generate the A half bridge output 706.
  • the triangular carrier wave 702 is also compared with minus the desired output 108, -v' a , for a B half bridge output 710. This scheme minimizes switching losses as only one half bridge switches at each change in output state and the switching losses are shared equally amongst the four output transistors.
  • FIG. 8 A block diagram of an embodiment of a load model unit 800, that includes error compensation and damping control, is shown in Fig. 8. The transfer functions are shown in the Laplace domain.
  • the load model unit 800 includes a load model 802 (modelled as l r / J s, where / is an estimate of the inertia of the load), and a load compensator 820.
  • the load model unit 800 receives inputs ⁇ q and ⁇ i q , and the output of the load model 802 is the speed co'/ which can be considered as a filtered version of the applied output speed ⁇ '. The difference is the stabilising speed correction term ⁇ on the output of stability controller P 0 804.
  • the stabilising speed correction value is a proportion of the torque error ⁇ , which is the q axis current error ⁇ i q multiplied by the rotor flux linkage ⁇ ⁇ , to stabilize the drive to prevent hunting.
  • the required stabilising speed correction value ⁇ is given by:
  • ⁇ is the damping factor, typically set to 1 for critical damping.
  • the required damping is provided by stability controller Po 804 with gain set by equation (6) with constant Ko set to the damping factor and using estimates of motor inductance L and inertia /.
  • the command torque current i * q is corrected by subtracting the applied load torque current i' q L at difference operator 806, thereby providing an estimated inertial torque component i * q - i' q L- Ignoring for now block P2 808 which takes effect at low speed, the applied load torque current i' q L is found from a proportional-integral (PI) controller 810 that includes proportion controller block 812, and integral controller block 814, with the torque current error ⁇ i q as its input.
  • PI proportional-integral
  • Constants Ko, K ⁇ and K 2 may be selected to be between 0 and 1 to achieve the damping and response characteristics required for the specific implementation.
  • constant Ko in stability control block 804 (the damping factor) may be set to 1.0, while constants K ⁇ and K 2 may be set to 0.25 and 0.5 respectively to achieve desired response characteristics.
  • the integrator 814 is turned into a low pass filter by adding a feedback path 816 around it that includes block P 2 808.
  • This path 816 is needed at or near zero speed so its gain is reduced as speed is increased by including the function t o(co'/).
  • This function is set to 1 at zero speed reducing to zero as speed is increased.
  • the input parameter used for this function is the filtered version co'/ of the applied speed obtained before the proportional term is added.
  • this function is a ramp function staying at 1 until co'/ nears the natural frequency estimate ⁇ ⁇ then ramping down to zero at a speed above ⁇ ⁇ .
  • This choice of the function is the same as that described in equation (7) for the reduction in the d axis current with speed. An example of such a function is shown in the implementation shown in Fig. 22.
  • This new connection of the P 2 block 808 output is used for correct operation of the new speed controller described later. Removing or limiting the DC component of the signal in this way is an advantage provided by the configuration of the load compensator 820 that forms part of the load model unit 800 presented herein.
  • the DC gain of the integrator 814 at zero speed is the inverse of the gain constant K3.
  • K3 depends on the application.
  • the value stored on the output of the integrator h 814 is the stored load torque from when the motor speed was high enough for the load torque to be measurable, which is when the motor back EMF is high enough to drive a large change in ⁇ i q when the motor flux phase angle differs from the applied phase angle ⁇ '.
  • this value slowly returns to zero at a rate depending on the value of K3.
  • the rotor angle settles to an offset value from the applied angle depending on the standstill load torque and the level of d axis current i d that has been applied to hold the rotor in position.
  • a low value for K 3 is preferred, perhaps 0.1 or less.
  • a high value for K3 is called for, perhaps as high as 1.0.
  • a compromise value which works well for most applications is 0.5.
  • the load compensator 820 therefore includes the P 2 block 808, as well as the proportional plus integral controller that includes both the proportional controller block Pi 812 and the integrator block 814.
  • the d axis current controller 130 uses feed forward control (in the feed forward converter 106) with added feedback compensation 132.
  • the feedback corrects for any steady state errors in the d axis current i d caused by errors in the estimation of parameters in the feed forward converter 106.
  • the feedback compensator uses a simple integral controller 112 where the integral of the current error is added to the reference current at the difference operator 110.
  • the integral gain is set to ⁇ ⁇ to match the effective gain of the integral controller 814 in the q axis load model compensator as shown in Fig8 .
  • the d axis reference current i d * must be set high enough to provide a holding torque greater than the difference between the load model torque (which for the load model used here only corrects for the inertial torque) and the actual torque. For example, if there is a start up friction torque 7> then the d axis reference current i d * at start up must be set to greater than T F / r . As the motor speed rises and the rising back EMF allows the q axis current feedback loop 140 to react to load torque errors, this reference current i d * can be reduced. In one embodiment, i d * is reduced as speed rises by the simple two stage ramp function d ( ⁇ u ) of equation (7). Other functions could be used.
  • Ke and ⁇ are set to 0.5 and 1.5 respectively but other values could be better for a given application, and Ke and ⁇ may fall within a range from about 0.1 to about 2.0.
  • FIG. 9 of the drawings shows the addition of a flux weakening unit 900 (shown in dotted lines) to the FFTC. Flux weakening is used to operate the motor at a speed above base speed.
  • This feedback loop 900 effectively negates the effect of the d axis current error integral controller 112 whilst keeping the integral controller 112 operating, thereby ensuring smooth transition into and out of flux weakening.
  • the feedback loop also sets the value of the applied d axis current i' d , thereby avoiding the need to back calculate it.
  • the feedback compensator of this local feedback loop 900 is a z domain integrator 902 with the output limited to negative values only as shown in Fig. 9. This ensures flux weakening comes in and out of operation smoothly.
  • the gain Kf may be set to about 0.5 for a fast stable response time.
  • the reference value X* d f is the calculated value of the d axis flux required to keep the output motor voltage below the maximum available voltage. V mm as shown in Fig. 22 at 2242.
  • the output of the z domain integrator 902 is added to the d axis reference current input at summing device 906, now designated i d * l to create the new reference current i d * .
  • V q ⁇ ' ⁇ ' ⁇ + v R ' q (9)
  • Voltage constant VM should be set slightly lower than the maximum available voltage V max to allow some voltage overhead to handle transient components.
  • V max maximum available voltage
  • VM V m a
  • the pulse lengthening of Fig. 4 may be disabled during flux weakening.
  • voltages may be normalized to the modulation index for exact control of the output voltage headroom as a proportion of the maximum available output voltage.
  • Equation (12) An inspection of equation (12) shows that v'd must be kept less than VM-
  • the applied d axis voltage v'd mainly depends on the applied torque so it can be limited in magnitude by controlling the torque current limits, as indicated at 2201 in Fig. 22.
  • VDM - ⁇ yd' - v R' a)
  • Vd negative. It must be limited to
  • Vd is positive. It must be limited to
  • iq' max (VDM + sign( ⁇ u a d ) (19)
  • iq' min ⁇ ( " 3 ⁇ 4M + Sign( ⁇ u')l3 ⁇ 4 d ) (20)
  • the resonant frequency of this circuit is the natural frequency coumble.
  • the load compensator 820 of Fig. 8 controls the damping of this circuit via control of ⁇ '. It should have its gain constants chosen to obtain adequate damping at the highest possible gain. Ignoring for now the second order integral block 814 and its feedback block Pi which have only a secondary effect on the response characteristics, the effect of the load compensator 820 can be incorporated into the high speed equivalent circuit. This will aid in determining suitable values of constants Ko and Ki.
  • This equivalent circuit 1200 allows easy selection of gain constants Ko and K ⁇ .
  • the gain constant K 2 of the second order integral controller 814 in Fig. 8 is set to as high a value as possible without interfering too much with the response characteristic provided by the load compensator 820.
  • a value of about K 2 0.5 may be used.
  • a load connected with a high compliance coupling may cause a low frequency mechanical resonance.
  • the command torque signal must be filtered to avoid exciting this resonance.
  • Fig. 13 of the drawings the effect of adding a two-mass coupled load on the controller can be seen by adding the load's equivalent mechanical circuit 1302 to the high speed equivalent circuit.
  • Capacitances CM and CL are the mechanical equivalent of the motor and load inertias and inductance Lc is the equivalent coupling compliance.
  • This circuit 1300 shows that by adjustment of the controller gains it may be possible to provide at least partial damping of the load coupling resonance, which is usually sufficient to stop oscillations.
  • the rotor At zero speed, the rotor is kept in position by the application of a positive d axis current but the dynamics are determined by the q axis external impedance imposed by the inverter. To optimize this, the q axis equivalent circuit of the motor needs to be derived, for example by using the motor resonances, or by using the motor equations.
  • the tuned circuit component values can be found by matching terminal voltage, energy and resonant frequency.
  • the q axis terminal voltage is X r d ⁇ /dt , which in the equivalent circuit is the voltage on the capacitor, giving the energy stored in the capacitor as 0.5Cp (A r d ⁇ /dt) 2 . This is the kinetic energy 0.5/(d ⁇ /dt) 2 , thereby giving:
  • a special case to be considered for the dynamic response is the effect of a step torque disturbance applied at zero speed.
  • a load torque at zero speed results in a rotor angle offset, the amount of which depends on the torque and the level of d axis holding current. Ideally when the torque is changed, the rotor offset angle should settle to its new value as quickly as possible with minimal overshoot.
  • the resulting sudden movement of the rotor produces a back EMF which in turn causes a pulse of current in the q axis, the amplitude of which depends on the combined impedance of the stator and the inverter. The resistance component of this impedance provides damping.
  • An added voltage source Vp 1702 shows the effect of the change in applied speed on the feed forward converter 106 due to the change in q axis current.
  • the voltage across Cp is the back EMF generated by the rotor movement from the torque disturbance.
  • the voltage source Vp is the countering applied back EMF generated by the applied speed change ⁇ ' acting on the feed forward converter 106. Because this also rotates the d axis current vector, affecting the resultant torque generated by the rotor offset, it appears inside the LpCp tuned circuit.
  • the controller is modified to artificially add a resistive component RE 1704 to the inverter output impedance.
  • the changes to the controller required to create this resistive component on the output can be seen at 2204 in the full controller embodiment illustrated in Fig. 22. This changes the effective motor stator resistance from R to KRR bookmark.
  • the feed forward converter 106 is also modified to use this new value of stator resistance.
  • Constant KR sets the damping for a zero speed torque disturbance. Simulations show that a suitable value is about 1 for most motor types and about any settings of the d axis holding current.
  • FIG. 19 A block diagram of an embodiment of a speed control loop 1900 is shown in Fig. 19. Advantage is taken of the availability of the applied load torque current i' Q L at 1902 when using a FFTC controller to eliminate the integrator normally used to correct for an offset torque. As well as removing integrator wind-up problems, this results in stable operation without re-tuning for wide changes in inertia.
  • Other types of torque controllers could be used provided they have a load torque current signal available. For example, a torque controller using proportional- integral current controllers could generate the load torque current signal with an observer.
  • the speed command input co * is compared to the filtered applied speed co'/ at comparator 1904.
  • the error signal passes through a proportional compensator 1906 with a gain Gap to create the command inertial torque current ⁇ q i. This is limited to plus and minus the maximum acceleration setting AM (in radians/s 2 times //l r to convert to the equivalent q axis current) at the limiting operator 1908.
  • the applied load torque current i' Q L at 1902 is then added at summing device 1910 to create the command torque current i * q after clipping to the torque current limit IM at limiting operator 2201 , as shown in Fig. 22 and mentioned earlier
  • the speed loop compensator gain can be normalized to:
  • G OJP ⁇ ⁇ ⁇ / (35) [173]
  • the normalized gain ⁇ ⁇ 0 then sets the response characteristic independent of the motor and load parameters.
  • a value for ⁇ ⁇ 0 of about 0.4 may be suitable for most situations.
  • ⁇ ⁇ 0 may be replaced with 2K 4 K 5 .
  • FIG. 20 A block diagram of a position control loop 2000 is shown in Fig. 20.
  • the command position angle ⁇ * is compared to the applied position angle ⁇ ' available from the torque controller.
  • the error is amplified by a proportional gain block 2004 of gain
  • a trajectory profile is pre-calculated then applied to ⁇ * .
  • the most common profile used is a trapezoidal speed profile where the motor accelerates at a constant rate until a maximum speed is reached. The speed is then maintained up to a pre- calculated point when the motor decelerates to standstill which corresponds to the final position. This method has many problems. Changing the trajectory mid-flight is difficult and a temporary speed drop due to overload cannot be accommodated.
  • the position cascaded control loops can be modified to provide the required tracking profile. To do this, use is made of the available speed and acceleration limits ⁇ ⁇ and A M and a non-linear gain
  • the initial acceleration rate and the maximum speed are controlled by A M and ⁇ ⁇ .
  • Control of the point when deceleration is started and the deceleration rate is achieved by a suitable choice of the non-linear function ⁇ ⁇ ( ⁇ ).
  • the required function ⁇ ⁇ ( ⁇ ) can be derived as follows, with At being the remaining time to reach the set point position:
  • ⁇ ⁇ ( ⁇ ) may be calculated directly rather than calculating Kg P (A9) separately first. Note that Kg P (A9) will increase indefinitely as ⁇ approaches zero. To maintain stability, the maximum value of ⁇ ⁇ may be limited to that given by equation (39).
  • Kg P A9 can be adjusted to:
  • ⁇ ⁇ ( ⁇ ) may be set as follows:
  • KgpW £ME ⁇ m > 2 e s
  • jerk limiting which is limiting the rate of change of acceleration, is often required. This can be implemented by making the speed proportional gain ⁇ ⁇ a function of the speed error in the same way that ⁇ ⁇ is made a function of ⁇ for position velocity profile control. A ramp limit on the acceleration would also need to be added to the acceleration clamp, shown as limiting operator 1908 in Figs. 19 and 20.
  • the position controller 2000 has been described to work with the speed controller 1900 described previously for a Feed Forward Torque Control system, but it can also be used with other types of speed controllers.
  • Other speed controllers such as a traditional proportional integral controller, do not have the ability to limit acceleration within their control loop, but this can be replaced with a ramp or rate of change limit to the speed command input to the controller instead.
  • FIG. 22 One embodiment of a complete controller in block diagram form suitable for practical implementation is shown in Fig. 22.
  • the controller is implemented in software on a
  • microcontroller for example on a digital signal processor, DSP. Operations shown in each block are calculated once per sample period except for the blocks in the Position Controller 2000 and the Speed Controller 1900, which may be calculated at a reduced sample rate depending on the response time required to position and speed changes. The breaks between the end of calculations for one sample and the start of calculations for the next sample are indicated by blocks marked z 1 . At these breaks the results of calculations are passed to the next sample. [197] Typical values of the constants used in the calculations, Ko to K and KR, KL and KG are shown in the diagram.
  • the applied stator flux linkages and the applied stator IR voltages are normalized (at 2210) to the modulation index, by dividing by the maximum PWM output voltage before saturation, V mm , which could be updated continuously from a reading of the DC link voltage in an actual system.
  • Variables normalized in this way are indicated in Fig. 22 by a hat ( ⁇ ) symbol.
  • Equalising delays 2214, 2216, 2218 are shown added to the d and q reference currents and to the sine and cosine values of the applied rotor angle to compensate for the delay in the PWM modulator 104 and any hardware filters in the current sensors.
  • Noise filters 2220 are shown for the error signals Ai d and Ai q . The nature of these filters will depend on the nature of the noise in the current sensors but typically would be a moving window filter averaging over the previous two sample measurements.
  • the applied speed value ⁇ ' is not used directly in calculations but is passed through a low pass filter 2222 first to remove noise to create an averaged speed co' flV . which is used instead.
  • the low pass filter 2222 is a single pole filter with a roll off frequency set high enough to not affect normal operation of the controller. In this case, the roll off frequency fo is set to four times the estimate of the natural frequency 4 ⁇ 3 ⁇ 4.
  • the applied load torque current is first added to the applied q axis current (after a Rate Limit operator 2226) then subtracted from it (at difference operator 2224) after it passes through the current limits 2201 , 2202, and before it is applied to the load model 802.
  • a Rate Limit operator 2226 When not in current limit, in some embodiments it may be better to just bypass the current limit blocks 2201 , 2202 and connect the output of the Rate Limit operator 2226 directly to the input of the load model 802.
  • This allows for a much higher resolution of the applied q axis current to be used for the load model 802 when not in current limit, when using fixed point integer variables as used in this implementation.
  • Low resolution is required on the input of Torque Current Limit at 2201 , to allow headroom for possible large overshoots in the value of the applied load torque current i q ' L .
  • switch 5A 2230 can be closed to force the rotor flux linkage value l re to match the actual rotor flux linkage.
  • the integrator I A 112 adjusts l re until the applied d axis current value i d ' becomes zero, resulting in l re matching the actual rotor flux linkage.
  • the rotor flux linkage estimate l r can be adjusted to the new measured value.
  • the gain of the integrator I A 112 is set to a value low enough to guarantee stability.
  • an Overload Limit 2232 to prevent excessive q axis torque current under severe transient overloads.
  • the output of the KG gain block 2234 is thus given by:
  • the error current ⁇ i q is approximately equal to sinA9 X r / L where ⁇ is the error in the controller applied flux angle ⁇ '.
  • the error limit hmit is set to K L X R /L where the constant KL corresponding to sinA9 is set typically to about 0.4.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Control Of Ac Motors In General (AREA)

Abstract

L'invention concerne un contrôleur pour un moteur électrique en CA, le contrôleur incluant : un convertisseur de puissance entraînant le moteur électrique en CA; un convertisseur avant qui déduit des valeurs de tensions appliquées au convertisseur de puissance; et une unité de modèle de charge servant à déduire une vitesse de rotor appliquée servant d'entrée au convertisseur avant, la vitesse de rotor appliquée étant déduite au moyen d'une valeur de correction de vitesse de stabilisation avec une valeur moyenne sensiblement nulle.
PCT/AU2017/051343 2016-12-08 2017-12-07 Contrôleur de moteur en ca sans capteurs WO2018102872A1 (fr)

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Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN109039206A (zh) * 2018-08-23 2018-12-18 江苏经纬轨道交通设备有限公司 牵引电机控制系统、控制方法及计算机可读存储介质
CN113395033A (zh) * 2021-07-06 2021-09-14 东方日立(成都)电控设备有限公司 一种用于高压变频器低电压穿越的控制方法及系统
US20210408951A1 (en) * 2020-06-30 2021-12-30 Dmytro KHACHATUROV Vector control method and system of a permanent magnet motor operation
CN113965129A (zh) * 2021-11-17 2022-01-21 大连海事大学 一种永磁同步电机控制系统电流测量偏移误差的补偿方法
CN114337467A (zh) * 2021-12-23 2022-04-12 昂宝电子（上海）有限公司 用于调整电机转速的方法及计算机存储介质

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1758240A2 (fr) * 2005-08-26 2007-02-28 Sanyo Electric Co., Ltd. Dispositif de commande de moteur
US20130221885A1 (en) * 2009-11-06 2013-08-29 University Of Technology, Sydney Sensorless ac motor controller

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1758240A2 (fr) * 2005-08-26 2007-02-28 Sanyo Electric Co., Ltd. Dispositif de commande de moteur
US20130221885A1 (en) * 2009-11-06 2013-08-29 University Of Technology, Sydney Sensorless ac motor controller

Non-Patent Citations (2)

* Cited by examiner, † Cited by third party
Title
HUNTER, G. P. ET AL.: "A Sensorless PMSM Fundamental Mode Controller with High Dynamic Full Range Speed Control", 37TH ANNUAL CONFERENCE ON IEEE INDUSTRIAL ELECTRONICS SOCIETY , IECON 2011, 7 November 2011 (2011-11-07), Melbourne, VIC, Australia, XP032104741 *
HUNTER, G., A SENSORLESS PM SYNCHRONOUS MOTOR DRIVE WITH FULL RANGE SPEED AND POSITION CONTROL USING FEED FORWARD TORQUE CONTROL, 16 January 2016 (2016-01-16), Retrieved from the Internet <URL:https://web.archive.org/web/20160804160750/http://ghunter.net/FFTC%20Deseription.pdf> [retrieved on 20160215] *

Cited By (10)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN109039206A (zh) * 2018-08-23 2018-12-18 江苏经纬轨道交通设备有限公司 牵引电机控制系统、控制方法及计算机可读存储介质
CN109039206B (zh) * 2018-08-23 2020-10-02 江苏经纬轨道交通设备有限公司 牵引电机控制系统、控制方法及计算机可读存储介质
US20210408951A1 (en) * 2020-06-30 2021-12-30 Dmytro KHACHATUROV Vector control method and system of a permanent magnet motor operation
US11817802B2 (en) * 2020-06-30 2023-11-14 Dmytro KHACHATUROV Vector control method and system of a permanent magnet motor operation
CN113395033A (zh) * 2021-07-06 2021-09-14 东方日立(成都)电控设备有限公司 一种用于高压变频器低电压穿越的控制方法及系统
CN113395033B (zh) * 2021-07-06 2022-07-15 东方日立(成都)电控设备有限公司 一种用于高压变频器低电压穿越的控制方法及系统
CN113965129A (zh) * 2021-11-17 2022-01-21 大连海事大学 一种永磁同步电机控制系统电流测量偏移误差的补偿方法
CN113965129B (zh) * 2021-11-17 2023-07-04 大连海事大学 一种永磁同步电机控制系统电流测量偏移误差的补偿方法
CN114337467A (zh) * 2021-12-23 2022-04-12 昂宝电子（上海）有限公司 用于调整电机转速的方法及计算机存储介质
CN114337467B (zh) * 2021-12-23 2024-01-12 昂宝电子（上海）有限公司 用于调整电机转速的方法及计算机存储介质

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