WO2017219389A1 - 大规模mimo系统中实现完美全向预编码的同步信号和信号的发送与接收方法 - Google Patents

大规模mimo系统中实现完美全向预编码的同步信号和信号的发送与接收方法 Download PDF

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WO2017219389A1
WO2017219389A1 PCT/CN2016/088335 CN2016088335W WO2017219389A1 WO 2017219389 A1 WO2017219389 A1 WO 2017219389A1 CN 2016088335 W CN2016088335 W CN 2016088335W WO 2017219389 A1 WO2017219389 A1 WO 2017219389A1
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perfect
signal
perfect omnidirectional
precoding matrix
precoding
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French (fr)
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高西奇
孟鑫
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东南大学
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Priority to US16/066,051 priority Critical patent/US10998968B2/en
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/24—Radio transmission systems, i.e. using radiation field for communication between two or more posts
    • H04B7/26—Radio transmission systems, i.e. using radiation field for communication between two or more posts at least one of which is mobile
    • H04B7/2662—Arrangements for Wireless System Synchronisation
    • H04B7/2671—Arrangements for Wireless Time-Division Multiple Access [TDMA] System Synchronisation
    • H04B7/2678—Time synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/0413—MIMO systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/0413—MIMO systems
    • H04B7/0426—Power distribution
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/02—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas
    • H04B7/04—Diversity systems; Multi-antenna system, i.e. transmission or reception using multiple antennas using two or more spaced independent antennas
    • H04B7/0413—MIMO systems
    • H04B7/0456—Selection of precoding matrices or codebooks, e.g. using matrices antenna weighting
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B7/00—Radio transmission systems, i.e. using radiation field
    • H04B7/14—Relay systems
    • H04B7/15—Active relay systems
    • H04B7/155—Ground-based stations
    • H04B7/15528—Control of operation parameters of a relay station to exploit the physical medium
    • H04B7/1555—Selecting relay station antenna mode, e.g. selecting omnidirectional -, directional beams, selecting polarizations
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04W—WIRELESS COMMUNICATION NETWORKS
    • H04W56/00—Synchronisation arrangements

Definitions

  • the present invention relates to a wireless communication method, and more particularly to a method for transmitting and receiving a synchronization signal and a signal for implementing perfect omni-directional precoding in a massive MIMO system.
  • a MIMO (Large Scale MIMO) wireless communication system in which a large number of antennas are arranged at a base station side has received extensive attention in academia and industry.
  • the base station side is generally configured with dozens or more antenna units (such as 128 or 256) and simultaneously serves dozens of users (such as 40).
  • a large number of antenna units at the base station can greatly increase the freedom of wireless communication space, greatly improve transmission rate, spectrum efficiency and power efficiency, and eliminate inter-cell interference to a considerable extent.
  • the increase in the number of antennas allows each antenna unit to be made smaller and less expensive.
  • the base station of each cell can simultaneously communicate with many users in the cell on the same time-frequency resource, thereby greatly improving the spectrum efficiency.
  • a large number of antenna units at the base station also enable better uplink directivity for each user uplink and downlink transmission, thereby significantly reducing the transmission power of the base station and the mobile terminal, and greatly improving power efficiency.
  • the random channels between each user and the base station can be orthogonal, which can eliminate the interference between cells and users and the influence of noise.
  • Common channels play a very important role in cellular systems, and much of the information at the base station needs to be communicated to users via common channels, such as synchronization signals, cell reference signals, control signaling, and Multimedia Broadcast Multicast Service (MBMS).
  • MBMS Multimedia Broadcast Multicast Service
  • a basic requirement for common channel design is that the transmitted signal has an omnidirectional characteristic to ensure reliable coverage.
  • the existing omnidirectional transmission schemes (such as single antenna transmission, cyclic delay diversity (CDD) and space time block code (STBC)) are only applicable to MIMO systems with fewer transmitters (no more than eight). The solution cannot be directly applied to large-scale MIMO systems configured with large-scale array antennas.
  • a conventional method is to use a single omnidirectional antenna to broadcast signals (which can be selected from one of a plurality of transmit antennas), but the selected single antenna must have a larger and larger configuration than the other antennas.
  • Expensive amplifiers are used to achieve the same power coverage as all antennas are used.
  • a key advantage of large-scale MIMO is the significantly improved power efficiency, that is, the power amplifier configured for each antenna unit can be significantly reduced as the number of antennas increases. It can be seen that if single-antenna transmission is used directly in massive MIMO. The expected power efficiency advantages will be lost.
  • CDD Space-Time Coded Transmission and Cyclic Delay Diversity
  • a synchronous signal transmitting method for perfect omni-directional precoding is realized, and a base station or a transmitting device intermittently generates a K ⁇ L synchronization signal, and then performs perfect omnidirectional precoding processing on the synchronization signal, and synchronizes
  • the signal is multiplied by M ⁇ K perfect omnidirectional precoding matrix W to obtain a M ⁇ L transmission signal, which is used as a digital baseband signal transmitted by the antenna array, where K is the dimension of the synchronization signal, 2 ⁇ K ⁇ M, M is the base station The number of antennas, L is the length of the synchronization signal in one transmission period.
  • the perfect omnidirectional precoding matrix W satisfies the following design criteria:
  • the power of the transmitted signal is the same in each spatial direction, ensuring perfect omnidirectional coverage
  • the perfect omnidirectional precoding matrix W satisfies the following conditions:
  • each row vector of the perfect omnidirectional precoding matrix W has the same 2 norm
  • the perfect omnidirectional precoding matrix W is composed of K sequences in a polyphase complementary orthogonal sequence set, and the multiphase complementary orthogonal sequence set includes K sequences of length M.
  • the two column sequences in the multi-phase complementary orthogonal sequence set are Golay sequence pairs.
  • the perfect omnidirectional precoding matrix W is obtained by: first constructing a dimension as a perfect omnidirectional precoding matrix W', and then performing K-times interpolation processing on each column of the perfect omnidirectional precoding matrix W' to obtain a perfect omnidirectional precoding matrix W, and the perfect omnidirectional precoding matrix W Each row has only one non-zero element.
  • the perfect omnidirectional precoding matrix W is obtained by multiplying the perfect omnidirectional precoding matrix W 0 by a K ⁇ K order ⁇ matrix U to obtain a M ⁇ K-order perfect omnidirectional precoding matrix W.
  • the synchronous signal receiving method for implementing perfect omni-directional precoding in the massive MIMO system according to the present invention, after the perfect omni-directional pre-coding processed synchronization signal is received by the mobile terminal or the receiving device, the mobile terminal or receiving The device performs reception synchronization processing using the received synchronization signal and a local copy of the synchronization signal.
  • the process for the mobile terminal or the receiving device to perform the receiving synchronization process by using the received synchronization signal and the local copy of the synchronization signal includes the following steps:
  • S1 the mobile terminal or the receiving device performs cross-correlation operation on the synchronization signal received at the positioning position ⁇ and the local copy of the synchronization signal, and obtains energy;
  • the base station or the transmitting device generates a K-dimensional vector signal through low-dimensional space-time coding, and then performs a perfect omnidirectional precoding process on the K-dimensional vector signal.
  • a perfect omnidirectional precoding matrix W multiplied by M ⁇ K for each vector signal is obtained, and an M-dimensional vector transmission signal is obtained, which is used as a digital baseband signal transmitted by the antenna array, where K ⁇ M, M is the number of base station antennas.
  • the transmission signal includes a pilot signal and a data signal
  • the base station or the transmitting device intermittently inserts the pilot signal while transmitting the data signal, and the data signal and the pilot signal are performed using the same perfect omnidirectional precoding matrix.
  • the same perfect omnidirectional precoding processed signal includes a pilot signal and a data signal, and the base station or the transmitting device intermittently inserts the pilot signal while transmitting the data signal, and the data signal and the pilot signal are performed using the same perfect omnidirectional precoding matrix.
  • the perfect omnidirectional precoding matrix W satisfies the following design criteria:
  • the power of the transmitted signal is the same in each spatial direction, ensuring perfect omnidirectional coverage
  • the perfect omnidirectional precoding matrix W satisfies the following design criteria: the degree of diversity of the high-dimensional space-time signal after perfect omni-directional precoding is the same as the degree of diversity of the low-dimensional space-time signal without perfect omni-directional precoding .
  • the perfect omnidirectional precoding matrix W satisfies the following conditions:
  • each row vector of the perfect omnidirectional precoding matrix W has the same 2 norm
  • the perfect omnidirectional precoding matrix W is composed of K sequences in a polyphase complementary orthogonal sequence set, and the multiphase complementary orthogonal sequence set includes K sequences of length M.
  • the two column sequences in the multi-phase complementary orthogonal sequence set are Golay sequence pairs.
  • the perfect omnidirectional precoding matrix W is obtained by: first constructing a dimension as a perfect omnidirectional precoding matrix W', and then performing K-times interpolation processing on each column of the perfect omnidirectional precoding matrix W' to obtain a perfect omnidirectional precoding matrix W, and the perfect omnidirectional precoding matrix W Each row has only one non-zero element.
  • the perfect omnidirectional precoding matrix W is obtained by multiplying the perfect omnidirectional precoding matrix W 0 by a K ⁇ K order ⁇ matrix U to obtain a M ⁇ K-order perfect omnidirectional precoding matrix W.
  • a signal receiving method for implementing perfect omni-directional precoding in a massive MIMO system wherein the transmitted signal after the perfect omni-directional precoding process is received by a mobile terminal or a receiving device after the transmission channel, the mobile terminal or The receiving device performs reception signal processing using the received transmission signal.
  • channel estimation and data signal detection of the mobile terminal or the receiving device are performed on an equivalent channel of the perfect omnidirectional precoding domain dimension reduction; using the received pilot signal, obtaining precoding through channel estimation The estimated value of the domain equivalent channel parameter; using the received data signal and the channel parameter estimation value, performing space-time decoding on the equivalent channel of the precoding domain to obtain the recovered data signal; the equivalent channel in the precoding domain is the actual high-dimensional channel Multiply by the perfect omnidirectional precoding matrix W.
  • the signals transmitted by the base station have the same power in each spatial direction and have perfect omnidirectional coverage;
  • the diversity gain can be obtained to improve the reliability of omnidirectional signal transmission.
  • FIG. 1 is a schematic flow chart of a synchronous signal transmitting method and a receiving method for implementing perfect omnidirectional precoding in a massive MIMO system according to the present invention
  • FIG. 2 is a signal transmission method for implementing perfect omnidirectional precoding in a massive MIMO system according to the present invention
  • the large-scale antenna array configured by the base station has multiple sectors, each sector is composed of a large number of antenna units, and when each antenna unit adopts an omnidirectional antenna or a 120-degree sector antenna,
  • the spacing of adjacent antenna elements can be designed as ⁇ /2 and Where ⁇ is the carrier wavelength.
  • Large-scale antenna arrays can also use circular arrays or other array structures that are easy to install.
  • Each antenna unit in the large-scale antenna array is connected to the digital baseband processing unit through a respective transceiver RF unit, an analog/digital conversion unit, a digital optical module, and a fiber transmission channel.
  • the invention discloses a synchronization signal transmission method for realizing perfect omnidirectional precoding in a massive MIMO system.
  • the base station or the transmitting device intermittently generates a K ⁇ L synchronization signal, and then performs a perfect synchronization signal.
  • the perfect omnidirectional precoding matrix W of the M ⁇ K is multiplied by the synchronization signal to obtain a M ⁇ L transmission signal, which is used as a digital baseband signal transmitted by the antenna array, where K is the dimension of the synchronization signal, 2 ⁇ K ⁇ M, M is the number of base station antennas, and L is the length of the synchronization signal in one transmission period.
  • the invention also discloses a synchronous signal receiving method for realizing perfect omnidirectional precoding in a massive MIMO system, as shown in FIG. 1 , at the receiving end, with y(n+ ⁇ ), Representing a continuous L-point discrete-time digital baseband signal obtained by a single receiving antenna of a mobile terminal or receiving device at a positioning position of ⁇ .
  • Cross-correlation is performed between y(n+ ⁇ ) and the local copy of the synchronization signal s(n), and the energy of the correlation value is obtained, and then the energy of the plurality of delay path positions is combined, and the total energy obtained is compared with the threshold. If it is greater than the threshold, it is determined that the positioning position ⁇ is aligned.
  • the transformation position ⁇ is repeated to repeat the above cross-correlation operation and the threshold comparison process until the positioning position ⁇ is aligned to obtain the positioning. Synchronize location information.
  • the first line to the right of the equal sign Represents a Zadoff-Chu sequence of length L
  • the second line is equivalent to A cyclic shift of L/2 was performed.
  • the digital baseband signal y(n) received by the receiving end at the positioning position ⁇ can be expressed as:
  • Equation (2) P is the total number of delay paths that can be resolved between the base station and the user, h p is the M-dimensional channel vector corresponding to the p-th delay path, and ⁇ p is the delay corresponding to the p-th delay path, z ( n) indicates additive white Gaussian noise.
  • the invention also discloses a signal transmission method for realizing perfect omnidirectional precoding in a massive MIMO system.
  • the base station or the transmitting device generates a K-dimensional vector signal through low-dimensional space-time coding, and then generates a K-dimensional vector.
  • the signal performs perfect omni-directional precoding processing, and each vector signal is multiplied by M ⁇ K perfect omnidirectional precoding matrix W to obtain an M-dimensional vector transmission signal, which is used as a digital baseband signal transmitted by the antenna array.
  • the transmitted signals are pilot signals and data signals.
  • the base station transmission and the downlink received by the mobile terminal When performing data signal transmission, only the discrete time domain narrowband channel is considered.
  • the narrowband channel under consideration there is only a single composite path, and the considered narrowband channel can be regarded as a subcarrier channel in a conventional wideband OFDM system, and accordingly,
  • the digital baseband transmit and receive signals involved are signals on time-frequency resources of a wideband OFDM system.
  • the number of antenna units equipped with the base station is M, and the mobile terminal is equipped with a single antenna.
  • the transmitted data signal d(l) is first subjected to low-dimensional space-time coding to generate a K-dimensional vector signal (K is smaller than M) s d (m); then, a K-dimensional vector pilot signal of length P is periodically inserted.
  • K is smaller than M
  • s d (m) a K-dimensional vector pilot signal of length P is periodically inserted.
  • the perfect omnidirectional precoding matrix the resulting signal x(n) is used as a digital baseband signal transmitted by a large-scale antenna array.
  • the invention also discloses a signal receiving method for realizing perfect omnidirectional precoding in a massive MIMO system, as shown in FIG. 2, at the receiving end, y(n) represents a single receiving antenna of the mobile terminal or the receiving device.
  • a digital baseband signal, y(n) including a received pilot signal y p (m') and a received data signal y d (m); using the received pilot signal y p (m'), obtaining channel parameters by channel estimation Estimated value; space-time decoding using the received data signal y d (m) and channel parameter estimates to obtain a recovered data signal
  • the low-dimensional space-time coding may be various space-time transmission methods used in a conventional low-order MIMO transmission system, such as space-time block code transmission, or cyclic delay diversity transmission, or a transmission method of multiplexing and diversity compromise.
  • the receiver channel parameter estimation is an estimate of the equivalent channel parameters of the precoding domain, and space time decoding is also implemented on this equivalent channel.
  • pilot insertion period there are P pilot vector signals and D data vector signals. Assuming that the channel is approximately constant, the digital baseband signal y(n) received at the receiving end can be expressed as:
  • h is the M-dimensional channel vector between the base station and the user
  • z(n) represents additive white Gaussian noise.
  • the precoding domain equivalent channel vector has a dimension of K. Separating the received pilot signal y p (m') and the received data signal y d (m) from the received signal can be expressed as:
  • z p (m') and z d (m) respectively represent corresponding noise terms.
  • the channel estimation and space-time decoding at the receiving end are based on the signal relationships given by the above two equations, respectively.
  • the estimated value of the pilot length P needs to be greater than or equal to K, but may be less than M. This is different from the conventional non-precoding omnidirectional diversity transmission scheme.
  • the non-precoding transmission scheme in order to perform coherent detection on the mobile terminal, it is necessary to know the instantaneous channel information h, and the base station needs to transmit the appropriate downlink pilot, so that The mobile terminal performs channel estimation, and the pilot length cannot be less than M.
  • the number of antenna elements M is large, a very large pilot overhead is caused, which seriously degrades system performance.
  • the omnidirectional precoding diversity transmission scheme provided by the present invention can increase the pilot overhead without increasing with M, and only relates to the spatial dimension K of the selected low-dimensional space-time coding, and the pilot overhead can be reduced by M/K times. .
  • the recovered data signal can be obtained.
  • the received data signal vector of the bth block can be expressed as:
  • the transmitted signal data signal d(1) is a signal obtained by channel coding, interleaving, and modulation symbol mapping of the information bit sequence.
  • demapping, deinterleaving, and channel decoding are required, thereby restoring the information bit sequence.
  • the omnidirectional transmission problem in a massive MIMO system is transformed into a perfect omnidirectional precoding matrix W and a low dimensional space-time signal (including a synchronization signal and a data signal).
  • Low-dimensional space-time signal transmission can borrow various space-time transmission methods used in conventional low-order MIMO transmission systems, including space-time block code transmission, cyclic delay diversity transmission, and transmission methods of multiplexing and diversity compromise.
  • the design of the perfect omnidirectional precoding matrix W becomes the key to affecting the transmission performance.
  • the perfect omnidirectional precoding matrix W should meet the following design criteria. then:
  • the power transmitted by the base station is the same in each spatial direction to ensure perfect omnidirectional coverage
  • the degree of diversity can reach the degree of diversity of the low-dimensional space-time signals used.
  • the perfect omnidirectional precoding matrix W should satisfy the following three design criteria:
  • diag() represents a vector consisting of the main diagonal elements of the matrix
  • 1 M represents an M-dimensional vector with each element being 1
  • a discrete time Fourier transform vector representing a sequence of M points of frequency ⁇ .
  • a perfect omnidirectional precoding matrix W of dimension M ⁇ K can be obtained by one of the following methods:
  • the sequence set is a Golay sequence pair;
  • the columns of the perfect omnidirectional precoding matrix W may be composed of sequences in a set of polyphase complementary orthogonal sequences.
  • a polyphase complementary orthogonal sequence set Each of these sequences Are polyphase sequences of length M, multiphase means All K sequences in this set satisfy both complementary and orthogonal characteristics.
  • the complementary characteristic means that the sum of the aperiodic autocorrelation functions of the respective K sequences is a ⁇ function, ie
  • a typical polyphase complementary orthogonal sequence set is a Golay sequence pair.
  • Example 2 You can construct a dimension by instance 1 first. Perfect omnidirectional precoding matrix W', which is recorded as
  • each column of the perfect omnidirectional precoding matrix W having a dimension of M ⁇ K can be obtained by performing K-time interpolation on each column of W′, that is,
  • the interpolated dimension is a 64 ⁇ 2 perfect omnidirectional precoding matrix W.

Abstract

本发明公开了大规模MIMO系统中实现完美全向预编码的同步信号发送方法和接收方法，基站或者发送装置间歇地生成同步信号，然后对同步信号进行完美全向预编码处理；经过完美全向预编码处理后的同步信号经过传输信道后由移动终端或者接收装置进行接收，移动终端或者接收装置利用接收到的同步信号和同步信号的本地副本进行接收同步处理。本发明还公开了大规模MIMO系统中实现完美全向预编码的信号发送方法和接收方法，基站或者发送装置经过低维空时编码生成矢量信号，然后对矢量信号进行完美全向预编码处理；经过完美全向预编码处理后的发送信号经过传输信道后由移动终端或者接收装置进行接收，移动终端或者接收装置利用接收到的发送信号进行接收信号处理。

Description

大规模MIMO系统中实现完美全向预编码的同步信号和信号的发送与接收方法 技术领域
本发明涉及无线通信方法，特别是涉及大规模MIMO系统中实现完美全向预编码的同步信号和信号的发送与接收方法。
背景技术
近年来，一种在基站端配置大量天线的MIMO(大规模MIMO)无线通信系统获得了学术界与工业界的广泛关注。在大规模MIMO系统中，基站端一般配置数十个以上天线单元(如128或256个)，且同时服务数十个用户(如40个)。基站端的大量天线单元可以较大幅度地提升无线通信空间自由度，大幅提升传输速率、频谱效率和功率效率，并在相当程度上消除小区间干扰。天线在数量上的增加使得每个天线单元可以被做得尺寸更小、成本更低。利用大规模天线单元提供的空间自由度，每个小区的基站可以同时与小区内众多用户在同一时频资源上进行通信，大幅提升频谱效率。基站端大量的天线单元也使得各用户上下行传输具有更好的空间指向性，由此显著降低基站和移动终端的发射功率，大幅提升功率效率。当基站天线数量足够多时，各个用户与基站间的随机信道可以趋近正交，能够消除小区间和用户间的干扰以及噪声的影响。以上所述的多个优势使得大规模MIMO具有极佳的应用前景。
公共信道在蜂窝系统中具有非常重要的作用，基站端的许多信息都需经由公共信道传递给用户，诸如同步信号、小区参考信号、控制信令及多媒体广播多播业务(MBMS)等。公共信道设计的一个基本要求是所发送的信号具有全向特性，以保证可靠覆盖。目前已有的全向发射方案(如单天线发射、循环延迟分集(CDD)及空时分组码(STBC))只适用于发射端天线数较少(不多于八根)的MIMO系统，这些方案无法直接应用于配置大规模阵列天线的大规模MIMO系统。例如，一种传统的方法是使用单根全向天线来广播信号(可以从多根发射天线中选定一根天线来发射)，然而所选的单根天线必须具有比其它天线配置更大更昂贵的功放，这样才能达到与所有天线都被使用情况下相同的功率覆盖范围。由于大规模MIMO的一个关键优势是显著提高的功率效率，即每个天线单元配置的功放可以随着天线数的增多而显著减小，可以看出，如果直接在大规模MIMO中采用单天线发射方案，所期望的功率效率优势将会丢失。空时编码传输和循环时延分集(CDD)是被LTE等系统广泛采用的全向分集传输技术，如果将其直接扩展应用于大规模MIMO系 统，不仅系统设计存在着困难，而且存在着导频开销随着基站端天线数线性增加的应用瓶颈问题。近年来，针对大规模MIMO系统中全向信号传输问题，已有空域信号扩展或预编码方法被提出，但是都不能保证完美的全向传输，各空间方向的发送信号功率不相同，在一定程度上会影响同步性能和全向信号传输的性能。
发明内容
发明目的：本发明的目的是提供一种能够克服现有技术中存在的问题的大规模MIMO系统中实现完美全向预编码同步信号和信号的发送与接收方法。
技术方案：为达到此目的，本发明采用以下技术方案：
本发明所述的大规模MIMO系统中实现完美全向预编码的同步信号发送方法，基站或者发送装置间歇地生成K×L的同步信号，然后对同步信号进行完美全向预编码处理，对同步信号左乘M×K的完美全向预编码矩阵W，得到M×L的发送信号，用作天线阵列发送的数字基带信号，其中K为同步信号的维度，2≤K＜M，M为基站天线数，L为一个发送周期内同步信号的长度。
进一步，所述完美全向预编码矩阵W满足如下设计准则：
(1)发送信号在各空间方向的功率相同，保证完美的全向覆盖；
(2)各天线单元上的发射信号功率相同，最大化各射频通道和天线阵列的功率效率。
进一步，所述完美全向预编码矩阵W满足如下条件：
(1)完美全向预编码矩阵W各列的M点序列的离散时间傅里叶变换的模平方之和为常数；
(2)完美全向预编码矩阵W的各行向量具有相同的2范数；
(3)完美全向预编码矩阵W的各列向量具有相同的2范数且相互正交。
进一步，所述完美全向预编码矩阵W由一个多相互补正交序列集合中的K个序列构成，且所述多相互补正交序列集合包括K个长度为M的序列。
进一步，当K＝2时，所述多相互补正交序列集合中的两个列序列为Golay序列对。
进一步，所述完美全向预编码矩阵W通过如下方法得到：首先构造出一个维度为
Figure PCTCN2016088335-appb-000001
的完美全向预编码矩阵W′，然后对完美全向预编码矩阵W′的每列进行K倍插零处理，得到完美全向预编码矩阵W，且所述完美全向预编码矩阵W的各行只有一个非零元素。
进一步，所述完美全向预编码矩阵W通过如下方法得到：对完美全向预编码矩阵W0右乘以一个K×K阶酉矩阵U，得到M×K阶完美全向预编码矩阵W。
本发明所述的大规模MIMO系统中实现完美全向预编码的同步信号接收方法，经过完美全向预编码处理后的同步信号经过传输信道后由移动终端或者接收装置进行接收，移动终端或者接收装置利用接收到的同步信号和同步信号的本地副本进行接收同步处理。
进一步，所述移动终端或者接收装置利用接收到的同步信号和同步信号的本地副本进行接收同步处理的过程包括以下步骤：
S1：移动终端或者接收装置对定位位置θ处接收到的同步信号与同步信号的本地副本进行互相关运算并求得能量；
S2：合并多个延迟径对应的的能量，得到总能量；
S3：将总能量与阈值进行比较：如果总能量大于阈值，则判定定位位置θ已对准；否则，则判定定位位置θ未对准，并变换定位位置θ，返回步骤S1。
本发明所述的大规模MIMO系统中实现完美全向预编码的信号发送方法，基站或者发送装置经过低维空时编码生成K维矢量信号，然后对K维矢量信号进行完美全向预编码处理，对每个矢量信号左乘M×K的完美全向预编码矩阵W，得到M维矢量发送信号，用作天线阵列发送的数字基带信号，其中K＜M，M为基站天线数。
进一步，所述发送信号包括导频信号和数据信号，基站或者发送装置中发送数据信号的同时间歇地插入导频信号，且数据信号和导频信号是使用了相同的完美全向预编码矩阵进行了相同的完美全向预编码处理后的信号。
进一步，所述完美全向预编码矩阵W满足如下设计准则：
(1)发送信号在各空间方向的功率相同，保证完美的全向覆盖；
(2)各天线单元上的发射信号功率相同，最大化各射频通道和天线阵列的功率效率。
进一步，所述完美全向预编码矩阵W满足如下设计准则：经过完美全向预编码后的高维空时信号的分集度与未经完美全向预编码的低维空时信号的分集度相同。
进一步，所述完美全向预编码矩阵W满足如下条件：
(1)完美全向预编码矩阵W各列的M点序列的离散时间傅里叶变换的模平方之和为常数；
(2)完美全向预编码矩阵W的各行向量具有相同的2范数；
(3)完美全向预编码矩阵W的各列向量具有相同的2范数且相互正交。
进一步，所述完美全向预编码矩阵W由一个多相互补正交序列集合中的K个序列构成，且所述多相互补正交序列集合包括K个长度为M的序列。
进一步，当K＝2时，所述多相互补正交序列集合中的两个列序列为Golay序列对。
进一步，所述完美全向预编码矩阵W通过如下方法得到：首先构造出一个维度为
Figure PCTCN2016088335-appb-000002
的完美全向预编码矩阵W′，然后对完美全向预编码矩阵W′的每列进行K倍插零处理，得到完美全向预编码矩阵W，且所述完美全向预编码矩阵W的各行只有一个非零元素。
进一步，所述完美全向预编码矩阵W通过如下方法得到：对完美全向预编码矩阵W0右乘以一个K×K阶酉矩阵U，得到M×K阶完美全向预编码矩阵W。
本发明所述的大规模MIMO系统中实现完美全向预编码的信号接收方法，所述经过完美全向预编码处理后的发送信号经过传输信道后由移动终端或者接收装置进行接收，移动终端或者接收装置利用接收到的发送信号进行接收信号处理。
进一步，所述接收信号处理中，移动终端或接收装置的信道估计和数据信号检测在完美全向预编码域降维的等效信道上实施；利用接收导频信号，通过信道估计，获得预编码域等效信道参数的估计值；利用接收数据信号和信道参数估计值，在预编码域等效信道上进行空时解码，得到恢复的数据信号；预编码域等效信道为实际的高维信道乘以完美全向预编码矩阵W。
有益效果：与现有技术相比，本发明具有如下的有益效果：
(1)基站发送的信号在各个空间方向上的功率相同，具有完美的全向覆盖效果；
(2)基站发送的信号在各个天线上的功率相同，能够最大化功率效率；
(3)通过完美全向预编码方法实现降维传输，可以大幅降低数据传输时所需的导频开销，大幅降低空时编码和解码的复杂度，大幅提高全向信号传输的整体效率；
(4)能够获得分集增益，提高全向信号传输的可靠性。
附图说明
图1为本发明的大规模MIMO系统中实现完美全向预编码的同步信号发送方法和接收方法流程示意图；
图2为本发明的大规模MIMO系统中实现完美全向预编码的信号发送方法和 接收方法流程示意图。
具体实施方式
下面结合具体实施方式和附图对本发明的技术方案作进一步的介绍。
对于大规模MIMO系统，考虑单小区情况，基站所配置的大规模天线阵列有多个扇面，每个扇面由大量的天线单元构成，当各天线单元采用全向天线或120度扇区天线时，相邻天线单元的间距可设计为λ/2和
Figure PCTCN2016088335-appb-000003
其中λ为载波波长。大规模天线阵列也可以采用圆阵列或其它方便安装的阵列结构。大规模天线阵列中各天线单元通过各自的收发射频单元、模数/数模转换单元、数字光模块及光纤传输通道与数字基带处理单元相连。
本发明公开了一种大规模MIMO系统中实现完美全向预编码的同步信号发送方法，如图1所示，基站或者发送装置间歇地生成K×L的同步信号，然后对同步信号进行完美全向预编码处理，对同步信号左乘M×K的完美全向预编码矩阵W，得到M×L的发送信号，用作天线阵列发送的数字基带信号，其中K为同步信号的维度，2≤K＜M，M为基站天线数，L为一个发送周期内同步信号的长度。
考虑基站发送和移动终端接收的下行链路。在进行同步信号传输时，考虑离散时间域宽带多径信道。此外，不失一般性，仅考虑单个移动终端的接收，并设基站端配备的天线单元数为M，移动终端配备单根天线。
在发送端，首先生成长度为L的K维(K大于等于2、且小于M)同步信号s(n)，
Figure PCTCN2016088335-appb-000004
s(n)中的所有K路分量的能量应相同，并互相正交，即
Figure PCTCN2016088335-appb-000005
其中c为一常数，而IK为K×K的恒等矩阵；然后，对s(n)进行完美全向预编码信号扩展处理，得到M维矢量发送信号x(n)＝Ws(n)，其中W为M×K的完美预编码矩阵，所得到信号x(n)用作大规模天线阵列发送的离散时间数字基带信号。
本发明还公开了一种大规模MIMO系统中实现完美全向预编码的同步信号接收方法，如图1所示，在接收端，以y(n+θ)，
Figure PCTCN2016088335-appb-000006
表示移动终端或接收装置单个接收天线在定位位置为θ处所得到连续L点离散时间数字基带信号。将y(n+θ)与同步信号s(n)的本地副本做互相关运算，并求得相关值的能量，然后合并多个延迟径位置的能量，将得到的总能量与阈值进行比较，如果大于阈值则判决为定位位置θ已对准，如果小于阈值则判决为定位位置θ未对准，并变换定位位置θ重复以上互相关运算及阈值比较过程，直至定位位置θ对准，得到定位同步位置信息。
其中，长度为L的K维同步信号s(n)，
Figure PCTCN2016088335-appb-000007
可以根据已有的经典序列产生，如Zadoff-Chu序列、Golay序列等。以Zadoff-Chu序列且K＝2的情况为例，s(n)可以通过对同一个Zadoff-Chu序列进行不同的循环移位得到，如
Figure PCTCN2016088335-appb-000008
式(1)中，等号右边的第一行
Figure PCTCN2016088335-appb-000009
表示一个长度为L的Zadoff-Chu序列，第二行相当于对
Figure PCTCN2016088335-appb-000010
进行了L/2的循环移位。假设信道在每次传输同步信号时是近似不变的，接收端在定位位置θ处接收到的数字基带信号y(n)可以表示为：
Figure PCTCN2016088335-appb-000011
式(2)中，P为基站与用户间信道可分辨的总延迟径数，hp为第p个延迟径对应的M维信道向量，τp为第p个延迟径对应的延迟，z(n)表示加性高斯白噪声。
将定位位置θ处的连续L点接收信号y(n+θ)，
Figure PCTCN2016088335-appb-000012
与同步信号的本地副本进行互相关运算，求得相关值的能量，并合并多个延迟径的能量，即
Figure PCTCN2016088335-appb-000013
其中||·||2表示矢量的2范数。将Λ(θ)与阈值进行比较，如果大于阈值则判决为定位位置θ已对准，如果小于阈值则判决为定位位置θ未对准，并变换定位位置θ重复以上互相关运算及阈值比较过程，直至定位位置θ对准。
本发明还公开了一种大规模MIMO系统中实现完美全向预编码的信号发送方法，如图2所示，基站或者发送装置经过低维空时编码生成K维矢量信号，然后对K维矢量信号进行完美全向预编码处理，对每个矢量信号左乘M×K的完美全向预编码矩阵W，得到M维矢量发送信号，用作天线阵列发送的数字基带信号。其中的发送信号是导频信号和数据信号。
考虑基站发送和移动终端接收的下行链路。在进行数据信号传输时，仅考虑离散时间域窄带信道，在所考虑的窄带信道中只有单个复合径，所考虑的窄带信道可以看做是常规宽带OFDM系统中的子载波信道，相应地，所涉及的数字基带发送和接收信号为宽带OFDM系统时频资源上的信号。此外，不失一般性，仅考虑单个移动终端的接收，并设基站端配备的天线单元数为M，移动终端配备单根天线。
在发送端，发送数据信号d(l)首先经过低维空时编码生成K维矢量信号(K小 于M)sd(m)；然后，周期地插入长度为P的K维矢量导频信号sp(m')，
Figure PCTCN2016088335-appb-000014
生成K维矢量发送信号s(n)；接着，对s(n)进行完美全向预编码信号扩展处理，得到M维矢量发送信号x(n)＝Ws(n)，其中W为M×K的完美全向预编码矩阵，所得到信号x(n)用作大规模天线阵列发送的数字基带信号。
本发明还公开了一种大规模MIMO系统中实现完美全向预编码的信号接收方法，如图2所示，在接收端，以y(n)表示由移动终端或接收装置单个接收天线所得到的数字基带信号，y(n)中包含接收导频信号yp(m′)和接收数据信号yd(m)；利用接收导频信号yp(m′)，通过信道估计，获得信道参数的估计值；利用接收数据信号yd(m)和信道参数估计值进行空时解码，得到恢复的数据信号
Figure PCTCN2016088335-appb-000015
其中，低维空时编码可以是常规低阶MIMO传输系统中采用的各种空时传输方法，如空时分组码传输，或循环时延分集传输，或复用与分集折中的传输方法。接收端信道参数估计是对预编码域等效信道参数的估计，空时解码也是在此等效信道上实施。在一个导频插入周期内，有P个导频矢量信号，D个数据矢量信号，假设信道是近似不变的，接收端接收到的数字基带信号y(n)可以表示为：
Figure PCTCN2016088335-appb-000016
式(4)中，h为基站与用户间的M维信道向量，z(n)表示加性高斯白噪声，
Figure PCTCN2016088335-appb-000017
为预编码域等效信道矢量，其维度为K。将接收导频信号yp(m′)和接收数据信号yd(m)从接收信号中分离出来，可以分别表示为：
Figure PCTCN2016088335-appb-000018
Figure PCTCN2016088335-appb-000019
式(5)中zp(m')和zd(m)分别表示相应的噪声项。接收端的信道估计和空时解码分别依据上面两个式子给出的信号关系式。
设
Figure PCTCN2016088335-appb-000020
Figure PCTCN2016088335-appb-000021
并设Sp满足Sp(Sp)H＝IK，其中上标H表示共轭翻转，IK为K×K的恒等矩阵，则
Figure PCTCN2016088335-appb-000022
的最小二乘估计可以由下式计算：
Figure PCTCN2016088335-appb-000023
为在接收端获得
Figure PCTCN2016088335-appb-000024
的估计值，导频长度P需要大于或等于K，但可以小于M。这不同于常规的无预编码的全向分集传输方案，在无预编码传输方案中，为在移动终端 进行相干检测，需要知道瞬时信道信息h，基站端需要发射相适应的下行导频，以便移动终端进行信道估计，而此时导频长度不能小于M，在天线单元数M很大的情况下会带来非常大的导频开销，严重降低系统性能。本发明提供的全向预编码分集传输方案，可以使得导频开销不随M的增加而增加，只与所选择的低维空时编码的空间维度K有关，导频开销可以降低达M/K倍。
利用由此得到的
Figure PCTCN2016088335-appb-000025
的估计值、(6)式给出的数据信号关系式、以及空时编码的特定信号生成方式，可以得到恢复的数据信号
Figure PCTCN2016088335-appb-000026
以下给出K＝2且采用Alamouti空时编码时信号检测的实例。设D＝KB，即一个导频周期内有B个空时码块，第b个块的发送数据信号矩阵可以表示为：
Figure PCTCN2016088335-appb-000027
式(8)中，
Figure PCTCN2016088335-appb-000028
上标*表示取共轭。第b个块的接收数据信号矢量可以表示为：
Figure PCTCN2016088335-appb-000029
式(9)中，zd(b)＝[zd(2b) zd(2b+1)]。在接收端，发送数据信号d(l)的估计值
Figure PCTCN2016088335-appb-000030
可以由下式计算：
Figure PCTCN2016088335-appb-000031
由此具体实施方式可见：在本发明提出的全向预编码分集传输方案中，空时解码在等效信道上实施，等效信道的空间维度可以显著降低，空时解码的复杂度亦可得以显著降低，因此，在显著降低导频开销的同时，也显著降低空时编码和解码的复杂度。在存在信道编码的无线传输系统中，发送信号数据信号d(l)是信息比特序列经过信道编码、交织和调制符号映射后得到的信号。在接收端，当获得d(l)的估计值
Figure PCTCN2016088335-appb-000032
后，需要进行解映射、解交织及信道解码，由此恢复出信息比特序列。
在前述基于预编码信号扩展处理的全向传输方案中，大规模MIMO系统中全向传输问题被转化为完美全向预编码矩阵W与低维空时信号(包括同步信号与数据信号)传输的设计问题。低维空时信号传输可以借用常规低阶MIMO传输系统中采用的各种空时传输方法，包括空时分组码传输、循环时延分集传输、以及复用与分集折中的传输方法等。在此情况下，完美全向预编码矩阵W的设计成为影响传输性能的关键。为保证传输性能，完美全向预编码矩阵W应当满足如下设计准 则：
(1)基站端发送信号在各空间方向的功率相同，以保证完美全向覆盖；
(2)基站端各天线单元上的发射信号功率相同，以最大化各射频通道和大规模天线阵列的功率效率；
(3)分集度可以达到所采用的低维空时信号的分集度。
假设发送信号矢量s(n)的各元素期望值为零、方差为
Figure PCTCN2016088335-appb-000033
且满足独立同分布的条件，即：
Figure PCTCN2016088335-appb-000034
其中E{·}表示求期望，0表示各元素均为零的矢量。在此情况下，为保证传输性能，完美全向预编码矩阵W应满足如下三个设计准则：
(1)W各列的M点序列的离散时间傅里叶变换的模平方之和为常数；
(2)W的所有M行具有相同的2范数；
(3)W的所有列具有相同的2范数且彼此正交。
不失一般性，上述三个设计准则可以具体描述为如下三个条件：
Figure PCTCN2016088335-appb-000035
Figure PCTCN2016088335-appb-000036
WHW＝IK        (13)
其中，diag()表示取矩阵的主对角线元素构成的矢量，1M表示各元素均为1的M维矢量，
Figure PCTCN2016088335-appb-000037
表示频率为ω的M点序列的离散时间傅里叶变换矢量。
维度为M×K的完美全向预编码矩阵W可以通过如下方法之一得到：
1)完美全向预编码矩阵W的K列可以由一个多相互补正交序列集合(其中包括K个长度为M的序列)中的K个序列构成，K＝2时的一个典型多相互补正交序列集合是Golay序列对；
2)可以先通过方法1)构造出一个维度为
Figure PCTCN2016088335-appb-000038
的完美全向预编码矩阵W′，而维度为M×K的完美全向预编码矩阵W的每列可以分别通过对W′的每列进行K倍插零得到；
3)若已知一满足上述式(11)、(12)和(13)的完美全向预编码矩阵，通过右乘以任一K×K的酉矩阵U得到新的完美全向预编码矩阵。
下面给出两个满足式(11)、(12)和(13)的完美全向预编码矩阵W的设计实施例：
实施例1
完美全向预编码矩阵W的各列可以由一个多相互补正交序列集合中的各序列构成。考虑一个多相互补正交序列集合
Figure PCTCN2016088335-appb-000039
其中每个序列
Figure PCTCN2016088335-appb-000040
都是长度为M的多相序列，多相意味着
Figure PCTCN2016088335-appb-000041
这个集合中的所有K个序列同时满足互补特性与正交特性。互补特性意味着这K个序列各自的非周期自相关函数的和为δ函数，即
Figure PCTCN2016088335-appb-000042
而正交特性意味着
Figure PCTCN2016088335-appb-000043
则完美全向预编码矩阵W被构造为
Figure PCTCN2016088335-appb-000044
当K＝2时，一个典型的多相互补正交序列集合是Golay序列对。
实例2：可以先通过实例1构造出一个维度为
Figure PCTCN2016088335-appb-000045
的完美全向预编码矩阵W′，它被记为
Figure PCTCN2016088335-appb-000046
则维度为M×K的完美全向预编码矩阵W的每列可以分别通过对W′的每列进行K倍插零得到，即
Figure PCTCN2016088335-appb-000047
式(18)中，
Figure PCTCN2016088335-appb-000048
表示维度为
Figure PCTCN2016088335-appb-000049
的K倍插零矩阵，
Figure PCTCN2016088335-appb-000050
表示取M×M的恒等矩阵的第k列，根据式(18)进行插值可以保证得到的完美全向预编码矩阵W的每行只有一个非零元素。以K＝2、M＝64为例，将维度为32×2的完美全向预编码矩阵W′记为
Figure PCTCN2016088335-appb-000051
则插值后的维度为64×2的完美全向预编码矩阵W为
Figure PCTCN2016088335-appb-000052
可以看出其中每行只有一个非零元素。
在本申请所提供的实施例中，应该理解到，所揭露的方法，在没有超过本申请的精神和范围内，可以通过其他的方式实现。当前的实施例只是一种示范性的例子，不应该作为限制，所给出的具体内容不应该限制本申请的目的。例如，多个单元或组件可以结合或者可以集成到另一个系统，或一些特征可以忽略，或不执行。所披露的完美全向预编码可以在离散时间域数字实现，也可以在模拟域实现，或在模拟与数字域混合实现。
以上所述，仅为本发明的具体实施方式，但本发明的保护范围并不局限于此，任何熟悉本技术领域的技术人员在本发明揭露的技术范围内，可轻易想到变化或替换，都应涵盖在本发明的保护范围之内。因此，本发明的保护范围应以所述权利要求的保护范围为准。

Claims (20)

  1. 大规模MIMO系统中实现完美全向预编码的同步信号发送方法，其特征在于：基站或者发送装置间歇地生成K×L的同步信号，然后对同步信号进行完美全向预编码处理，对同步信号左乘M×K的完美全向预编码矩阵W，得到M×L的发送信号，用作天线阵列发送的数字基带信号，其中K为同步信号的维度，2≤K＜M，M为基站天线数，L为一个发送周期内同步信号的长度。
  2. 根据权利要求1所述的大规模MIMO系统中实现完美全向预编码的同步信号发送方法，其特征在于：所述完美全向预编码矩阵W满足如下设计准则：
    (1)发送信号在各空间方向的功率相同，保证完美的全向覆盖；
    (2)各天线单元上的发射信号功率相同，最大化各射频通道和天线阵列的功率效率。
  3. 根据权利要求1所述的大规模MIMO系统中实现完美全向预编码的同步信号发送方法，其特征在于：所述完美全向预编码矩阵W满足如下条件：
    (1)完美全向预编码矩阵W各列的M点序列的离散时间傅里叶变换的模平方之和为常数；
    (2)完美全向预编码矩阵W的各行向量具有相同的2范数；
    (3)完美全向预编码矩阵W的各列向量具有相同的2范数且相互正交。
  4. 根据权利要求1所述的大规模MIMO系统中实现完美全向预编码的同步信号发送方法，其特征在于：所述完美全向预编码矩阵W由一个多相互补正交序列集合中的K个序列构成，且所述多相互补正交序列集合包括K个长度为M的序列。
  5. 根据权利要求4所述的大规模MIMO系统中实现完美全向预编码的同步信号发送方法，其特征在于：当K＝2时，所述多相互补正交序列集合中的两个列序列为Golay序列对。
  6. 根据权利要求1所述的大规模MIMO系统中实现完美全向预编码的同步信号发送方法，其特征在于：所述完美全向预编码矩阵W通过如下方法得到：首先构造出一个维度为
    Figure PCTCN2016088335-appb-100001
    的完美全向预编码矩阵W′，然后对完美全向预编码矩阵W′的每列进行K倍插零处理，得到完美全向预编码矩阵W，且所述完美全向预编码矩阵W的各行只有一个非零元素。
  7. 根据权利要求1所述的大规模MIMO系统中实现完美全向预编码的同步信号发送方法，其特征在于：所述完美全向预编码矩阵W通过如下方法得到：对完美全向预编码矩阵W0右乘以一个K×K阶酉矩阵U，得到M×K阶完美全向预编码矩阵 W。
  8. 大规模MIMO系统中实现完美全向预编码的同步信号接收方法，其特征在于：经过完美全向预编码处理后的同步信号经过传输信道后由移动终端或者接收装置进行接收，移动终端或者接收装置利用接收到的同步信号和同步信号的本地副本进行接收同步处理。
  9. 根据权利要求8所述的大规模MIMO系统中实现完美全向预编码的同步信号接收方法，其特征在于：所述移动终端或者接收装置利用接收到的同步信号和同步信号的本地副本进行接收同步处理的过程包括以下步骤：
    S1：移动终端或者接收装置对定位位置θ处接收到的同步信号与同步信号的本地副本进行互相关运算并求得能量；
    S2：合并多个延迟径对应的的能量，得到总能量；
    S3：将总能量与阈值进行比较：如果总能量大于阈值，则判定定位位置θ已对准；否则，则判定定位位置θ未对准，并变换定位位置θ，返回步骤S1。
  10. 大规模MIMO系统中实现完美全向预编码的信号发送方法，其特征在于：基站或者发送装置经过低维空时编码生成K维矢量信号，然后对K维矢量信号进行完美全向预编码处理，对每个矢量信号左乘M×K的完美全向预编码矩阵W，得到M维矢量发送信号，用作天线阵列发送的数字基带信号，其中K＜M，M为基站天线数。
  11. 根据权利要求10所述的大规模MIMO系统中实现完美全向预编码的信号发送方法，其特征在于：所述发送信号包括导频信号和数据信号，基站或者发送装置中发送数据信号的同时间歇地插入导频信号，且数据信号和导频信号是使用了相同的完美全向预编码矩阵进行了相同的完美全向预编码处理后的信号。
  12. 根据权利要求10所述的大规模MIMO系统中实现完美全向预编码的信号发送方法，其特征在于：所述完美全向预编码矩阵W满足如下设计准则：
    (1)发送信号在各空间方向的功率相同，保证完美的全向覆盖；
    (2)各天线单元上的发射信号功率相同，最大化各射频通道和天线阵列的功率效率。
  13. 根据权利要求10所述的大规模MIMO系统中实现完美全向预编码的信号发送方法，其特征在于：所述完美全向预编码矩阵W满足如下设计准则：经过完美全向预编码后的高维空时信号的分集度与未经完美全向预编码的低维空时信号的分集度相同。
  14. 根据权利要求10所述的大规模MIMO系统中实现完美全向预编码的信号发送方法，其特征在于：所述完美全向预编码矩阵W满足如下条件：
    (1)完美全向预编码矩阵W各列的M点序列的离散时间傅里叶变换的模平方之和为常数；
    (2)完美全向预编码矩阵W的各行向量具有相同的2范数；
    (3)完美全向预编码矩阵W的各列向量具有相同的2范数且相互正交。
  15. 根据权利要求10所述的大规模MIMO系统中实现完美全向预编码的信号发送方法，其特征在于：所述完美全向预编码矩阵W由一个多相互补正交序列集合中的K个序列构成，且所述多相互补正交序列集合包括K个长度为M的序列。
  16. 根据权利要求15所述的大规模MIMO系统中实现完美全向预编码的信号发送方法，其特征在于：当K＝2时，所述多相互补正交序列集合中的两个列序列为Golay序列对。
  17. 根据权利要求10所述的大规模MIMO系统中实现完美全向预编码的信号发送方法，其特征在于：所述完美全向预编码矩阵W通过如下方法得到：首先构造出一个维度为
    Figure PCTCN2016088335-appb-100002
    的完美全向预编码矩阵W′，然后对完美全向预编码矩阵W′的每列进行K倍插零处理，得到完美全向预编码矩阵W，且所述完美全向预编码矩阵W的各行只有一个非零元素。
  18. 根据权利要求10所述的大规模MIMO系统中实现完美全向预编码的信号发送方法，其特征在于：所述完美全向预编码矩阵W通过如下方法得到：对完美全向预编码矩阵W0右乘以一个K×K阶酉矩阵U，得到M×K阶完美全向预编码矩阵W。
  19. 大规模MIMO系统中实现完美全向预编码的信号接收方法，其特征在于：所述经过完美全向预编码处理后的发送信号经过传输信道后由移动终端或者接收装置进行接收，移动终端或者接收装置利用接收到的发送信号进行接收信号处理。
  20. 根据权利要求19所述的大规模MIMO系统中实现完美全向预编码的信号接收方法，其特征在于：所述接收信号处理中，移动终端或接收装置的信道估计和数据信号检测在完美全向预编码域降维的等效信道上实施；利用接收导频信号，通过信道估计，获得预编码域等效信道参数的估计值；利用接收数据信号和信道参数估计值，在预编码域等效信道上进行空时解码，得到恢复的数据信号；预编码域等效信道为实际的高维信道乘以完美全向预编码矩阵W。
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