WO2017113753A1 - 一种同频噪声处理方法、装置及系统 - Google Patents

一种同频噪声处理方法、装置及系统 Download PDF

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WO2017113753A1
WO2017113753A1 PCT/CN2016/090100 CN2016090100W WO2017113753A1 WO 2017113753 A1 WO2017113753 A1 WO 2017113753A1 CN 2016090100 W CN2016090100 W CN 2016090100W WO 2017113753 A1 WO2017113753 A1 WO 2017113753A1
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noise
frequency
signal
value
electrode
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PCT/CN2016/090100
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English (en)
French (fr)
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魏海军
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深圳市汇顶科技股份有限公司
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    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06F—ELECTRIC DIGITAL DATA PROCESSING
    • G06F3/00—Input arrangements for transferring data to be processed into a form capable of being handled by the computer; Output arrangements for transferring data from processing unit to output unit, e.g. interface arrangements
    • G06F3/01—Input arrangements or combined input and output arrangements for interaction between user and computer
    • G06F3/03—Arrangements for converting the position or the displacement of a member into a coded form
    • G06F3/041—Digitisers, e.g. for touch screens or touch pads, characterised by the transducing means
    • G06F3/0416—Control or interface arrangements specially adapted for digitisers
    • G06F3/0418—Control or interface arrangements specially adapted for digitisers for error correction or compensation, e.g. based on parallax, calibration or alignment
    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06F—ELECTRIC DIGITAL DATA PROCESSING
    • G06F3/00—Input arrangements for transferring data to be processed into a form capable of being handled by the computer; Output arrangements for transferring data from processing unit to output unit, e.g. interface arrangements
    • G06F3/01—Input arrangements or combined input and output arrangements for interaction between user and computer
    • G06F3/03—Arrangements for converting the position or the displacement of a member into a coded form
    • G06F3/041—Digitisers, e.g. for touch screens or touch pads, characterised by the transducing means
    • G—PHYSICS
    • G06—COMPUTING; CALCULATING OR COUNTING
    • G06F—ELECTRIC DIGITAL DATA PROCESSING
    • G06F3/00—Input arrangements for transferring data to be processed into a form capable of being handled by the computer; Output arrangements for transferring data from processing unit to output unit, e.g. interface arrangements
    • G06F3/01—Input arrangements or combined input and output arrangements for interaction between user and computer
    • G06F3/03—Arrangements for converting the position or the displacement of a member into a coded form
    • G06F3/041—Digitisers, e.g. for touch screens or touch pads, characterised by the transducing means
    • G06F3/044—Digitisers, e.g. for touch screens or touch pads, characterised by the transducing means by capacitive means

Definitions

  • the invention relates to the technical field of co-frequency noise processing, in particular to a method, device and system for processing same-frequency noise.
  • the position detection system is a key technology in the field of human-computer interaction, which greatly improves the human-computer interaction experience and is widely used in the fields of smart phones, notebook computers, personal consumer electronics, ATMs and ordering platforms.
  • the position detection system is required to accurately respond to the user's input intention.
  • the general position detection system realizes the user's touch position detection through the microcapacitance detection technology. That is, the micro-electric signal (ie, the driving signal) is loaded in the touch sensing area, the signal loop is formed by capacitive coupling, and the signal (ie, the sensing signal) is detected at the back end of the loop. When a conductor approaches the sensing area, a capacitive electrical effect is generated, which is equivalent to changing the signal loop impedance value, thereby affecting the detected value of the sensing signal, and finally determining the actual touch position by the amount of change in the detected value.
  • the micro-electric signal ie, the driving signal
  • the signal loop is formed by capacitive coupling
  • the signal ie, the sensing signal
  • the interference of the position detection system mainly comes from wireless communication equipment, LCD and charger.
  • the main frequency components of these interferences are randomly distributed in the [1KHZ--10MHZ] interval, while the drive signal frequency of the position detection system is usually at [50KHZ— Within 500KHZ]. Therefore, designing a reasonable low-pass or high-pass filter on the inductive signal loop helps to suppress some noise.
  • frequency reuse technology For the same-frequency noise of the signal frequency of the proximity detection system or other noise source system (in mobile communication systems, in order to improve the frequency utilization and increase the capacity of the system, frequency reuse technology is often used, and frequency reuse refers to being separated. After a certain distance, there are many cells using the same set of frequencies in a given coverage area. These cells are called co-frequency cells. The interference between co-frequency cells is called co-channel interference, and the same-frequency interference is the same frequency. Noise), the filter is obviously not an effective method. Mixed in position detection system or other noise source system The same-frequency noise in the system is terrible, and once the same-frequency noise and the induced signal are mixed, it is difficult to separate them again.
  • the noise frequency demodulation Under the premise of not generating the driving signal, the theoretical position detection system only has the noise signal, demodulates the noise frequency component, and selects the frequency point with the smallest frequency component as the next driving-sensing signal frequency. .
  • the method can effectively avoid the noise with slow change of the frequency component, but does not evade the noise whose frequency component changes rapidly. Because the noise detection and the sensing signal detection do not occur at the same time, the noise-free frequency point f obtained by the noise detection and demodulation in the T1 period, when the sensing signal is detected in the T2 period, the f-frequency point may have a large noise, therefore,
  • the noise processing of the noise frequency demodulation method is not high in real time.
  • the noise source clock synchronization the noise generated by some special noise sources occurs intermittently in the time domain.
  • the clock synchronization relationship can be established with the noise source, so that the detection of the induced signal is only performed during the intermittent time of the noise, so It can effectively avoid such noise.
  • it is difficult to establish the synchronization relationship and the application of the method is limited due to the short interval time.
  • the embodiment of the invention provides a co-frequency noise processing method, a co-frequency noise processing device and a co-frequency noise processing system, aiming at real-time detection and elimination of the same-frequency noise, suppressing the same-frequency noise in the position detection system or other noise sources. Negative effects in the system.
  • a co-channel noise processing method includes the following steps:
  • Step a loading a driving signal on the first electrode, the driving signal passing through the first electrode and the second a coupling capacitance between the electrodes is coupled to the second electrode to generate an inductive signal;
  • Step b Demodulating the sensing signal on the second electrode, calculating a system phase value of the noise source system according to the demodulated value of the sensing signal, and determining whether the current sensing signal exists by using the calculated system phase value of the noise source system Same frequency noise, if there is co-frequency noise, perform step c;
  • Step c calculating a noise quantization value of the driving signal frequency and the frequency noise of the nearby frequency by the mutual influence relationship between the demodulation value of the arbitrary frequency point and the demodulation value of the adjacent frequency point, and obtaining the corrected driving signal according to the noise quantization value
  • the noise-free demodulation value of the frequency and its nearby frequency noise eliminates the same-frequency noise in the drive signal frequency.
  • the technical solution adopted by the embodiment of the present invention further includes: in the step a, the driving signal loaded on the first electrode is:
  • the generated sensing signal is:
  • TX is the drive signal
  • RX is the sensing signal
  • A is the signal attenuation coefficient
  • ⁇ is the signal phase, which is the system phase value of the noise source system.
  • the demodulating manner for demodulating the sensing signal on the second electrode specifically includes: expanding the sensing signal RX, and performing the expanded sensing signal RX performs quadrature demodulation to obtain a cosine component and a sine component of the induced signal RX, and calculates an amplitude R of the induced signal RX at the driving frequency f according to the cosine component and the sine component; the expansion signal of the sensing signal RX is:
  • B and ⁇ are respectively a noise attenuation coefficient and a noise phase
  • the specific formula for calculating the amplitude R of the sensing signal RX at the driving frequency f according to the cosine component and the sine component is obtained by acquiring the cosine component and the sine component:
  • I is a cosine component and Q is a sinusoidal component.
  • the technical solution adopted by the embodiment of the present invention further includes: in the step b, determining, by using the calculated system phase value of the noise source system, whether the co-frequency noise exists in the current sensing signal: determining the step Whether the system phase value of the noise source system calculated in b is equal to the true system phase value of the noise source system, and if the system phase value of the noise source system calculated in the step b is equal to the true system phase value of the noise source system, then It is determined that the current sensing signal does not have the same-frequency noise; if the system phase value of the noise source system calculated in the step c is not equal to the true system phase value of the noise source system, it is determined that the current sensing signal has the same-frequency noise.
  • the technical solution adopted by the embodiment of the present invention further includes: in the step c, the calculating manner of obtaining the corrected driving signal frequency and the noise-free demodulation value of the frequency noise in the vicinity according to the noise quantization value is specifically:
  • the influence coefficients of the demodulated values at the five frequencies of f-2*bitfreq, f-bitfreq, f, f+bitfreq, and f+2*bitfreq are [0, 0.5, 1, 0.5, 0], respectively.
  • the relationship between the frequency point demodulation value and the demodulation value of its adjacent frequency point is:
  • [R -2 , R -1 , R 0 , R +1 , R +2 ] are f-2*bitfreq, f-bitfreq, f, f+bitfreq, and f+2*bitfreq, respectively.
  • Demodulation values of frequency points, The signal for driving the signal frequency f is corrected for the processed noise-free demodulation value at the corresponding frequency point.
  • an intra-frequency noise processing apparatus including a CPU controller, a modulator, a first electrode, a second electrode, and a demodulator; and the CPU controller is configured to control the modulator in a driving signal is loaded on the first electrode, and the driving signal is coupled to the second electrode through a coupling capacitance between the first electrode and the second electrode to generate an inductive signal;
  • the CPU controller controls the demodulator to demodulate the sensing signal on the second electrode
  • the CPU controller calculates a system phase value of the noise source system according to the demodulated value of the sensing signal, and determines whether the same-frequency noise exists in the current sensing signal by using the calculated system phase value of the noise source system; if there is co-frequency noise, Calculating a noise quantization value of a frequency of the driving signal and a frequency noise thereof in the vicinity of the demodulation value of the arbitrary frequency point and a demodulation value of the adjacent frequency point, and obtaining the corrected driving signal frequency according to the noise quantization value and Noise-free demodulation values of nearby frequency noise and eliminate co-channel noise in the drive signal frequency.
  • the technical solution adopted by the embodiment of the present invention further includes: demodulating the demodulation signal of the sensing signal on the second electrode by the demodulator, specifically: expanding the sensing signal RX, and orthogonalizing the expanded sensing signal RX Demodulating, obtaining a cosine component and a sine component of the sensing signal RX, and calculating an amplitude R of the sensing signal RX at the driving frequency f according to the cosine component and the sine component; the expansion signal of the sensing signal RX is:
  • a and ⁇ are the signal attenuation coefficient and the signal phase, respectively, B and ⁇ are the noise attenuation coefficient and the noise phase, respectively;
  • the specific formula for calculating the amplitude R of the sensing signal RX at the driving frequency f according to the cosine component and the sine component is obtained by acquiring the cosine component and the sine component:
  • I is a cosine component and Q is a sinusoidal component.
  • the technical solution adopted by the embodiment of the present invention further includes: determining, by the CPU controller, whether there is co-frequency noise in the current sensing signal: determining whether the calculated system phase value of the noise source system is equal to the real system of the noise source system Phase value, if the calculated system phase value of the noise source system is equal to the true system phase value of the noise source system, it is determined that there is no co-channel noise in the current sensing signal; if the calculated system phase value of the noise source system is not equal to the noise source The true system phase value of the system determines that there is co-channel noise in the current sensing signal.
  • the technical solution adopted by the embodiment of the present invention further includes: the CPU controller calculates a noise quantization value of the driving signal frequency and the frequency noise in the vicinity thereof, and obtains the corrected driving signal frequency and the noiseless solution of the nearby frequency noise according to the noise quantization value.
  • the minimum change unit; the influence coefficient of the demodulation value at the five frequency points of f-2*bitfreq, f-bitfreq, f, f+bitfreq, and f+2*bitfreq is [0, 0.5] , 1, 0.5, 0], the relationship between the demodulation value of any frequency point and the demodulation value of its adjacent frequency point is:
  • [R -2 , R -1 , R 0 , R +1 , R +2 ] are f-2*bitfreq, f-bitfreq, f, f+bitfreq, and f+2*bitfreq, respectively.
  • Demodulation values of frequency points, The signal for driving the signal frequency f is corrected for the processed noise-free demodulation value at the corresponding frequency point.
  • an intra-frequency noise processing system including a noise source system and an equal-frequency noise processing device, wherein the noise source system and the same-frequency noise processing device are connected by signals;
  • the noise processing device eliminates co-channel noise in the noise source system.
  • the beneficial effects of the embodiments of the present invention are: the same-frequency noise processing method, the same-frequency noise processing device, and the same-frequency noise processing system of the embodiment of the present invention adopt a solution by loading a driving signal on the first electrode.
  • the modulation mode obtains the demodulation value of the induced signal on the second electrode, calculates the phase of the system according to the demodulated value, and determines whether the current sensing signal contains the same-frequency noise through the calculated value of the system phase, thereby solving the real-time problem of the same-frequency noise processing; And through the mutual influence relationship between the demodulation value of the arbitrary frequency point and the demodulation value of its adjacent frequency point, the noise-free demodulation value of the modified driving signal and its nearby frequency noise is obtained, and finally the noise elimination processing is realized, and the same-frequency noise is effectively suppressed.
  • the negative effects in the position detection system greatly improve the user experience.
  • FIG. 1 is a schematic structural diagram of a same-frequency noise processing method according to an embodiment of the present invention.
  • Figure 2 is a spectrum graph of a sampled signal
  • FIG. 3 is a schematic structural diagram of a co-frequency noise processing system according to an embodiment of the present invention.
  • FIG. 4 is a circuit diagram of a noise processing system according to an embodiment of the present invention.
  • FIG. 5 is a schematic structural diagram of an application system of a co-frequency noise processing system according to an embodiment of the present invention.
  • the same-frequency noise processing of the position detecting system is taken as an example, but it is not limited thereto, and the present invention is also applicable to the same-frequency noise processing of other noise source systems, for example, a cellular system, a cluster. System or satellite communication system, etc.
  • FIG. 1 is a flowchart of a method for processing the same frequency noise according to an embodiment of the present invention.
  • the same-frequency noise processing method of the embodiment of the present invention includes the following steps:
  • Step S100 The CPU controller controls the modulator to load a driving signal on the first electrode, and the driving signal is coupled to the second electrode through a coupling capacitance between the first electrode and the second electrode to generate an sensing signal;
  • step S100 the driving signal is loaded on the first electrode, and the driving signal is coupled to the second electrode through a coupling capacitance between the first electrode and the second electrode.
  • the generating the sensing signal specifically includes: the CPU controller controls the modulator.
  • the driving signal loaded on the first electrode is TX, then:
  • the TX signal is coupled to the second electrode through a coupling capacitance between the first electrode and the second electrode, the process equivalent to the TX signal passing through the capacitive impedance and then connected to the second electrode, then the second
  • the induced signal detected by the electrode is RX:
  • a and ⁇ are the signal attenuation coefficient and the signal phase, respectively, the system phase of the position detection system; in general, the system phase ⁇ of the position detection system is fixed and can be regarded as a known amount.
  • B and ⁇ are respectively a noise attenuation coefficient and a noise phase, and the noise is randomly changed, that is, B and ⁇ are represented as random values; in the embodiment of the invention, the driving signal is loaded on the first electrode. It is a sine wave drive signal, but it is not limited to this. It is also applicable to load other types of drive signals, such as square wave signals.
  • Step S200 The CPU controller controls the demodulator to demodulate the sensing signal on the second electrode to obtain a demodulated value of the sensing signal.
  • step S200 the sensing signal coupled by the second electrode can be transmitted to the demodulator through ADC analog-to-digital conversion, and the demodulator is controlled by the CPU controller for demodulation, and the ADC can be passed through the CPU controller, the modulator, or the demodulation.
  • the device or the like is implemented in any module; the method for demodulating the induced signal on the second electrode by the demodulator specifically includes: expanding the sensing signal RX, and expanding the formula of the sensing signal RX as:
  • Step S300 calculating a system phase value arctan(Q/I) of the position detecting system according to the demodulated value of the sensing signal, and determining whether there is co-channel noise in the current sensing signal by using the calculated system phase value of the position detecting system; if present In the same frequency noise, step S400 is performed; if there is no co-channel noise, step S500 is performed;
  • step 300 the system phase values of the position detection system are:
  • the determining, by the calculated system phase value of the position detecting system, whether the co-frequency noise exists in the sensing signal is: determining whether the calculated system phase value of the position detecting system is equal to the real system phase value of the position detecting system, if The calculated system phase value of the position detecting system is equal to the real system phase value of the position detecting system, and it is determined that the current sensing signal does not have co-frequency noise; if the calculated system phase value of the position detecting system is not equal to the real system of the position detecting system The phase value determines that there is co-channel noise in the current sensing signal.
  • Step S400 calculating a noise quantization value of the frequency of the driving signal and the frequency noise of the nearby frequency by the mutual influence relationship between the demodulation value of the arbitrary frequency point and the demodulation value of the adjacent frequency point, according to the noise quantization value Obtaining the corrected noise-free demodulation value of the driving signal frequency and the frequency noise in the vicinity thereof, and eliminating the same-frequency noise in the driving signal frequency;
  • step S400 the corrected noise-free mediation value of the driving signal frequency and the frequency noise in the vicinity thereof has eliminated the influence of the same-frequency noise;
  • the calculation method of the noise-free mediation value of the driving signal frequency and the frequency noise in the vicinity thereof is specifically as follows:
  • the first electrode is coupled to the driving signal on the second electrode, and through the Hanning window, the sampling time is T, then the spectral curve of the sampling signal is as shown in FIG. 2, which is a spectrum curve of the sampling signal (the technology is in the industry) Known techniques, the present invention will not be described again).
  • f is the drive signal frequency
  • bitfreq 1/T
  • bitfreq is considered as the smallest unit of variation of the drive signal frequency f in the position detection system.
  • [R -2 , R -1 , R 0 , R +1 , R +2 ] are f-2*bitfreq, f-bitfreq, f, f+bitfreq, respectively.
  • the demodulated values of the five frequency points f+2*bitfreq can be obtained directly from step 200. and
  • the noise-free demodulation value after the correction is processed at the corresponding frequency point for the signal of the driving signal frequency f, and is obtained by the formula (9) and the formula (10).
  • Step S500 Calculating the user touch position by using the induced signal demodulated value or the corrected noiseless demodulated value Set
  • the first electrode and the second electrode are multi-electrode arrays, and the plurality of electrode arrays are vertically wired to each other, forming dense intersection nodes throughout the touch panel, and each of the intersection nodes is regarded as a coupling capacitor.
  • the size of the coupling capacitor near the touch position is changed. Therefore, it is only necessary to detect the amount of change of each capacitor to calculate the user's touch position.
  • the first electrode array of the first electrode is loaded with the driving signal, and the sensing signals on all the electrode arrays of the second electrode are detected and demodulated, and the demodulated value of the sensing signal can be regarded as the intersection of the two electrode arrays. Coupling capacitor value.
  • the driving signal loading of all the electrode arrays on the first electrode is sequentially performed in the same manner, and all the cross-coupling capacitance values of the touch panel plane can be obtained.
  • the capacitance value when the user has no operation is set as the reference value.
  • the real-time value of the capacitance value and the reference value can be compared, and the accurate touch position of the user can be quickly calculated; when the present invention is applied to the noise source
  • other types of signal metrics can also be calculated using the induced signal demodulated values or the corrected noiseless demodulated values.
  • FIG. 3 is a schematic structural diagram of an equal-frequency noise processing apparatus according to an embodiment of the present invention
  • FIG. 4 is a circuit diagram of a noise processing apparatus according to an embodiment of the present invention.
  • the same-frequency noise processing apparatus of the embodiment of the present invention includes a CPU controller, a modulator, a first electrode, a second electrode, and a demodulator; wherein the CPU controller is respectively connected to a modulator and a demodulator, and the modulator and the modulator a first electrode is connected, and the demodulator is connected to the second electrode;
  • the CPU controller is configured to control the modulator to load a driving signal on the first electrode, and the driving signal is coupled to the second electrode through a coupling capacitance between the first electrode and the second electrode to generate an sensing signal; wherein the CPU controller controls The loading of the driving signal by the modulator on the first electrode specifically includes: the driving signal loaded on the first electrode is TX, then:
  • the driving signal TX is coupled to the second electrode through a coupling capacitance between the first electrode and the second electrode, the process equivalent to the driving signal TX passing through the capacitive impedance and then connected to the second electrode,
  • the induced signal detected by the second electrode is RX:
  • a and ⁇ are the signal attenuation coefficient and the signal phase, respectively, the system phase of the position detection system; in general, the system phase ⁇ of the position detection system is fixed and can be regarded as a known amount.
  • B and ⁇ are the noise attenuation coefficient and the noise phase, respectively, and the noise is randomly changed, that is, B and ⁇ appear as random values.
  • the CPU controller controls the demodulator to demodulate the sensing signal on the second electrode to obtain a demodulated value of the sensing signal.
  • the manner in which the demodulator demodulates the sensing signal on the second electrode specifically includes: sensing
  • the signal RX is expanded and the induced signal RX expansion formula is:
  • the CPU controller specifically includes a noise judging module, a noise canceling module, and a position calculating module, and the noise judging module, the noise canceling module, and the position calculating module are sequentially connected;
  • the noise judging module is configured to calculate a system phase value arctan(Q/I) of the position detecting system according to the demodulated value of the sensing signal, and determine whether the same-frequency noise exists in the current sensing signal by using the calculated system phase value of the position detecting system; If there is co-channel noise, the corrected noise-free demodulation value is obtained by the noise cancellation module; if there is no co-channel noise, the user touch position is calculated by the position calculation module; wherein the system phase value of the position detection system is:
  • the noise judging module judges whether the co-channel noise is present in the sensing signal by determining whether the calculated system phase value of the position detecting system is equal to the real system phase value of the position detecting system, and if the calculated position detecting system is If the phase value is equal to the true system phase value of the position detecting system, it is determined that the current sensing signal does not have the same-frequency noise; if the calculated system phase value of the position detecting system is not equal to the true system phase value of the position detecting system, the current sensing signal is determined. There is co-frequency noise.
  • the noise cancellation module is configured to calculate a noise quantization value of the driving signal frequency and the frequency noise of the nearby frequency frequency by using an interaction relationship between the demodulation value of the arbitrary frequency point and the demodulation value of the adjacent frequency point, and obtain the corrected quantization value according to the noise quantization value.
  • the noise-free demodulation value of the driving signal frequency and the frequency noise in the vicinity thereof eliminates the same-frequency noise in the frequency of the driving signal; wherein the calculation method of the noise-free mediation value of the driving signal frequency and the frequency noise in the vicinity thereof is specifically: by the first electrode
  • the sinusoidal signal coupled to the second electrode is passed through the Hanning window, and the sampling time is T.
  • the spectral curve of the sampled signal is as shown in FIG.
  • [R -2 , R -1 , R 0 , R +1 , R +2 ] are f-2*bitfreq, f-bitfreq, f, f+bitfreq, respectively.
  • the demodulation values of the five frequency points of f+2*bitfreq can be obtained directly by demodulator demodulation.
  • the noise-free demodulation value after the correction is processed at the corresponding frequency point for the signal of the driving signal frequency f, and is obtained by the formula (9) and the formula (10).
  • the position calculation module is configured to calculate a user touch position by using the induced signal demodulation value or the modified noiseless demodulation value; wherein, as shown in FIG. 4, the first electrode and the second electrode are multi-electrode arrays, and the plurality of electrode arrays are mutually Vertical routing forms dense intersections throughout the touchpad, and each cross node is considered a coupling capacitor.
  • the size of the coupling capacitor near the touch position is changed. Therefore, it is only necessary to detect the amount of change of each capacitor to calculate the user's touch position.
  • the first electrode array of the first electrode is loaded with the driving signal, and the sensing signals on all the electrode arrays of the second electrode are detected and demodulated, and the demodulated value can be regarded as the two electrode cross-coupling capacitance values.
  • the driving signal loading of all the electrode arrays on the first electrode is sequentially performed in the same manner, and all the cross-coupling capacitance values of the touch panel plane can be obtained.
  • the capacitance value when the user has no operation is set as a reference value, when a user performs a touch operation, Simply compare the real-time value of the capacitor value with the reference value to quickly calculate the user's accurate touch position.
  • FIG. 5 is a schematic structural diagram of a co-frequency noise processing system according to an embodiment of the present invention.
  • the same-frequency noise processing system of the embodiment of the invention comprises a position detection system and an equal-frequency noise processing device, wherein the position detection system and the co-frequency noise processing device are connected to each other;
  • the co-frequency noise processing device comprises a CPU controller, a modulator, a first electrode, a second electrode and a demodulator; wherein the CPU controller is respectively connected to the modulator and the demodulator, the modulator is connected to the first electrode, and the demodulator is connected to the second electrode;
  • the CPU controller is configured to control the modulator to load a driving signal on the first electrode, and the driving signal is coupled to the second electrode through a coupling capacitance between the first electrode and the second electrode to generate an sensing signal; wherein the CPU controller controls The loading of the driving signal by the modulator on the first electrode specifically includes: the driving signal loaded on the first electrode is TX, then:
  • the driving signal TX is coupled to the second electrode through a coupling capacitance between the first electrode and the second electrode, the process equivalent to the driving signal TX passing through the capacitive impedance and then connected to the second electrode,
  • the induced signal detected by the second electrode is RX:
  • a and ⁇ are the signal attenuation coefficient and the signal phase, respectively, the system phase of the position detection system; in general, the system phase ⁇ of the position detection system is fixed and can be regarded as a known amount.
  • B and ⁇ are the noise attenuation coefficient and the noise phase, respectively, and the noise is randomly changed, that is, B and ⁇ appear as random values.
  • the CPU controller controls the demodulator to demodulate the induced signal on the second electrode to obtain the sensing signal.
  • a demodulation value wherein the demodulator demodulates the induced signal on the second electrode by specifically: deriving the sensing signal RX, and expanding the analog signal RX into:
  • the CPU controller specifically includes a noise judging module, a noise canceling module, and a position calculating module, and the noise judging module, the noise canceling module, and the position calculating module are sequentially connected;
  • the noise judging module is configured to calculate a system phase value arctan(Q/I) of the position detecting system according to the demodulated value of the sensing signal, and determine whether there is co-channel noise in the sampling sensing signal by using the calculated system phase value of the position detecting system; If there is co-channel noise, the corrected noise-free demodulation value is obtained by the noise cancellation module; if there is no co-channel noise, the user touch position is calculated by the position calculation module; wherein the system phase value of the position detection system is:
  • the noise judging module judges whether the co-frequency noise exists in the current sensing signal by acquiring the real system phase value of the position detecting system, and determining whether the calculated system phase value of the position detecting system is equal to the real system of the position detecting system.
  • the phase value if the calculated system phase value of the position detecting system is equal to the true system phase value of the position detecting system, it is determined that the current sensing signal does not have the same-frequency noise; if the calculated system phase value of the position detecting system is not equal to the position detecting
  • the true system phase value of the system determines that there is co-channel noise in the current sensing signal.
  • the noise cancellation module is configured to calculate a noise quantization value of the driving signal frequency and the frequency noise of the nearby frequency frequency by using an interaction relationship between the demodulation value of the arbitrary frequency point and the demodulation value of the adjacent frequency point, and obtain the corrected quantization value according to the noise quantization value.
  • the noise-free demodulation value of the driving signal frequency and the frequency noise in the vicinity thereof eliminates the same-frequency noise in the frequency of the driving signal; wherein the calculation method of the noise-free mediation value of the driving signal frequency and the frequency noise in the vicinity thereof is specifically: by the first electrode
  • the sinusoidal signal coupled to the second electrode is passed through the Hanning window, and the sampling time is T.
  • the spectral curve of the sampled signal is as shown in FIG.
  • [R -2 , R -1 , R 0 , R +1 , R +2 ] are f-2*bitfreq, f-bitfreq, f, f+bitfreq, respectively.
  • the demodulation values of the five frequency points of f+2*bitfreq can be obtained directly by demodulator demodulation.
  • the noise-free demodulation value after the correction is processed at the corresponding frequency point for the signal of the driving signal frequency f, and is obtained by the formula (9) and the formula (10).
  • the position calculation module is configured to calculate the user touch position using the sampled induced signal demodulated value or the corrected noiseless demodulated value.
  • the same-frequency noise processing method, the same-frequency noise processing device and the same-frequency noise processing system of the embodiment of the present invention acquire the demodulation value of the second-electrode sine wave signal by orthogonal demodulation by loading the driving signal on the first electrode.

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Abstract

本发明实施例涉及一种同频噪声处理方法、装置及系统。所述方法包括：在第一电极上加载驱动信号，所述驱动信号通过第一电极与第二电极之间的耦合电容耦合到第二电极上，生成感应信号；对第二电极上的感应信号进行解调,根据感应信号的解调值计算噪声源系统的系统相位值，并通过计算出的噪声源系统的系统相位值判断当前感应信号中是否存在同频噪声，如果存在同频噪声，通过任意频率点解调值与其相邻频率点解调值的相互影响关系式，计算驱动信号频率及其附近频率噪声的噪声量化值，根据所述噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值，消除驱动信号频率中的同频噪声。本发明实施例提供的技术方案可极大改善用户体验。

Description

一种同频噪声处理方法、装置及系统 【技术领域】
本发明涉及同频噪声处理技术领域，尤其涉及一种同频噪声处理方法、装置及系统。
【背景技术】
位置检测系统是人机交互领域的一项关键技术，极大地改善人机交互体验，广泛用于智能手机、笔记本电脑、个人消费电子、ATM及点餐平台等领域。当用户在触摸板或者其他交互界面移动手指或电子笔头等导体进行点击或划线操作，位置检测系统要求能够准确响应用户的输入意愿。
一般位置检测系统是通过微电容检测技术实现用户触摸位置检测的。即在触摸感应区域加载微电信号(即驱动信号)，通过电容耦合形成信号回路，在回路后端检测信号(即感应信号)。当有导体接近感应区域时，会产生电容电效应，等同于改变信号回路阻抗值，从而影响感应信号的检测值，最后以检测值的变化量判定实际的触摸位置。
尽管位置检测系统日趋成熟，但其应用环境也越来越复杂多变，设计工程师面临着诸多挑战，其中提升抗干扰性能是最为迫切的一项，同时也是整个微电容检测行业的一个难点。位置检测系统的干扰主要源自无线通信设备、LCD及充电器等设备，这些干扰的主要频率成份随机分布于[1KHZ--10MHZ]区间内，而位置检测系统的驱动信号频率通常处于[50KHZ—500KHZ]区间内。因此在感应信号回路上设计合理的低通或高通滤波器有助于抑制部分噪声。而对于接近位置检测系统或其他噪声源系统驱动信号频率的同频噪声(在移动通信系统中，为了提高频率利用率，增加系统的容量，常常采用频率复用技术，频率复用是指在相隔一定距离后，在给定的覆盖区域内，存在着许多使用同一组频率的小区，这些小区称为同频小区，同频小区之间的干扰称为同频干扰，同频干扰即为同频噪声)，滤波器显然不是一个有效的方法。混在位置检测系统或其他噪声源系 统中的同频噪声是可怕的，同频噪声与感应信号一旦混杂起来，便难以再次将它们分离。这样带来最为直接的后果就是引起后端电路中感应信号检测值产生变化量，这与用户触摸引起的变化量在数据端的效应是等同的，严重时甚至影响位置检测系统或其他噪声源系统的正常使用。
为了解决上述问题，现有技术中针对同频噪声处理所采取的技术方案为：
一、噪声频率解调：在不产生驱动信号的前提下，理论上位置检测系统中只有噪声信号，解调出噪声频率成份，选择频率成份最小的频率点，作为下一次的驱动-感应信号频率。该方法能够有效规避频率成份变化缓慢的噪声，却无法规避频率成份变化较快的噪声。因为噪声检测与感应信号检测不是同一时刻发生的，T1时段内噪声检测解调得到的无噪声频率点f，在T2时段内进行感应信号检测时，f频率点可能存在较大的噪声，因此，噪声频率解调方法的噪声处理实时性不高。
二、噪声源时钟同步：有些特殊噪声源产生的噪声在时域上是间歇性出现的，理论上可以与噪声源建立时钟同步关系，使得感应信号检测只在噪声的间歇时间内进行，如此便可实现有效规避该类噪声。但由于同步关系难以建立，且由于间歇时间太短限制了该方法的推广应用。
因此，有必要提供一种量化同频噪声的实时计算方法，从而抑制同频噪声在位置检测系统或其他噪声源系统中的负面效应。
【发明内容】
本发明实施例提供了一种同频噪声处理方法、同频噪声处理装置及同频噪声处理系统，旨在实现同频噪声的实时检测及消除，抑制同频噪声在位置检测系统或其他噪声源系统中的负面效应。
为了解决以上提出的问题，本发明实施例采用的技术方案为：
一种同频噪声处理方法，包括以下步骤：
步骤a：在第一电极上加载驱动信号，所述驱动信号通过第一电极与第二 电极之间的耦合电容耦合到第二电极上，生成感应信号；
步骤b：对第二电极上的感应信号进行解调,根据感应信号的解调值计算噪声源系统的系统相位值，并通过计算出的噪声源系统的系统相位值判断当前感应信号中是否存在同频噪声，如果存在同频噪声，执行步骤c；
步骤c：通过任意频率点解调值与其相邻频率点解调值的相互影响关系式，计算驱动信号频率及其附近频率噪声的噪声量化值，根据所述噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值，消除驱动信号频率中的同频噪声。
本发明实施例采取的技术方案还包括：在所述步骤a中，所述加载在第一电极的驱动信号为：
TX＝sin(ωt)
所述生成的感应信号为：
Rx＝Asin(ωt+φ)+noise
在上述公式中，TX为驱动信号，
Figure PCTCN2016090100-appb-000001
为驱动信号频率f，RX为感应信号，A为信号衰减系数，φ为信号相位，即噪声源系统的系统相位值。
本发明实施例采取的技术方案还包括：在所述步骤b中，所述对第二电极上的感应信号进行解调的解调方式具体包括：将感应信号RX展开，对展开后的感应信号RX进行正交解调，获取感应信号RX的余弦分量和正弦分量，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R；所述感应信号RX展开公式为：
Rx＝Asin(ωt+φ)+Bsin(ωt+θ)
＝(Acos(φ)+Bcos(θ))sin(ωt)+(Asin(φ)+Bsin(θ))cos(ωt)
在上述公式中，B、θ分别为噪声衰减系数和噪声相位；
所述获取余弦分量和正弦分量，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R的具体公式为：
I＝Acos(φ)+Bcos(θ)
Q＝Asin(φ)+Bsin(θ)
Figure PCTCN2016090100-appb-000002
上述公式中，I为余弦分量，Q为正弦分量。
本发明实施例采取的技术方案还包括：在所述步骤b中，所述通过计算出的噪声源系统的系统相位值判断当前感应信号中是否存在同频噪声的判断方式为：判断所述步骤b中计算出的噪声源系统的系统相位值是否等于噪声源系统的真实系统相位值，如果所述步骤b中计算出的噪声源系统的系统相位值等于噪声源系统的真实系统相位值，则判定当前感应信号不存在同频噪声；如果所述步骤c中计算出的噪声源系统的系统相位值不等于噪声源系统的真实系统相位值，则判定当前感应信号存在同频噪声。
本发明实施例采取的技术方案还包括：在所述步骤c中，所述根据噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值的计算方式具体为：由第一电极耦合到第二电极上的驱动信号，通过汉宁窗，采样时间为T，f为驱动信号频率，bitfreq＝1/T，bitfreq为驱动信号频率f的最小变化单位；驱动信号频率f的信号在f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq五个频点处解调值的影响系数分别为[0,0.5,1,0.5,0]，则任意频率点解调值与其相邻频率点解调值的相互影响关系式为：
Figure PCTCN2016090100-appb-000003
Figure PCTCN2016090100-appb-000004
在上述公式中，[R-2,R-1,R0,R+1,R+2]分别为f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq这五个频率点的解调值，
Figure PCTCN2016090100-appb-000005
为驱动信号频率f的信号在对应频率点上修正处理后的无噪声解调值。
本发明实施例采取的另一技术方案为：一种同频噪声处理装置，包括CPU控制器、调制器、第一电极、第二电极和解调器；所述CPU控制器用于控制调制器在第一电极上加载驱动信号，所述驱动信号通过第一电极与第二电极之间的耦合电容耦合到第二电极上，生成感应信号；
所述CPU控制器控制解调器对第二电极上的感应信号进行解调；
所述CPU控制器根据感应信号的解调值计算噪声源系统的系统相位值，并通过计算出的噪声源系统的系统相位值判断当前感应信号中是否存在同频噪声；如果存在同频噪声，通过任意频率点解调值与其相邻频率点解调值的相互影响关系式，计算驱动信号频率及其附近频率噪声的噪声量化值，根据所述噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值，并消除驱动信号频率中的同频噪声。
本发明实施例采取的技术方案还包括：所述解调器对第二电极上的感应信号进行解调的解调方式具体为：将感应信号RX展开，对展开后的感应信号RX进行正交解调，获取感应信号RX的余弦分量和正弦分量，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R；所述感应信号RX展开公式为：
Rx＝Asin(ωt+φ)+Bsin(ωt+θ)
＝(Acos(φ)+Bcos(θ))sin(ωt)+(Asin(φ)+Bsin(θ))cos(ωt)
在上述公式中，
Figure PCTCN2016090100-appb-000006
为驱动信号频率f，A、φ分别为信号衰减系数和信号相 位，B、θ分别为噪声衰减系数和噪声相位；
所述获取余弦分量和正弦分量，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R的具体公式为：
I＝Acos(φ)+Bcos(θ)
Q＝Asin(φ)+Bsin(θ)
Figure PCTCN2016090100-appb-000007
上述公式中，I为余弦分量，Q为正弦分量。
本发明实施例采取的技术方案还包括：所述CPU控制器判断当前感应信号中是否存在同频噪声的判断方式为：判断计算出的噪声源系统的系统相位值是否等于噪声源系统的真实系统相位值，如果计算出的噪声源系统的系统相位值等于噪声源系统的真实系统相位值，则判定当前感应信号不存在同频噪声；如果计算出的噪声源系统的系统相位值不等于噪声源系统的真实系统相位值，则判定当前感应信号存在同频噪声。
本发明实施例采取的技术方案还包括：所述CPU控制器计算驱动信号频率及其附近频率噪声的噪声量化值，根据噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值的计算方式具体为：由第一电极耦合到第二电极上的驱动信号，通过汉宁窗，采样时间为T，f为驱动信号频率，bitfreq＝1/T，bitfreq为驱动信号频率f的最小变化单位；驱动信号频率f的信号在f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq五个频点处解调值的影响系数分别为[0,0.5,1,0.5,0]，则任意频率点解调值与其相邻频率点解调值的相互影响关系式为：
Figure PCTCN2016090100-appb-000008
Figure PCTCN2016090100-appb-000009
在上述公式中，[R-2,R-1,R0,R+1,R+2]分别为f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq这五个频率点的解调值，
Figure PCTCN2016090100-appb-000010
为驱动信号频率f的信号在对应频率点上修正处理后的无噪声解调值。
本发明实施例采取的又一技术方案为：一种同频噪声处理系统，包括噪声源系统和同频噪声处理装置，所述噪声源系统及同频噪声处理装置信号连接；通过所述同频噪声处理装置消除噪声源系统中的同频噪声。
与现有技术相比，本发明实施例的有益效果在于：本发明实施例的同频噪声处理方法、同频噪声处理装置及同频噪声处理系统通过在第一电极上加载驱动信号，采用解调方式获取第二电极上感应信号的解调值，根据解调值计算系统相位，并通过系统相位的计算值判定当前感应信号是否包含同频噪声，解决了同频噪声处理的实时性问题；并通过任意频率点解调值与其相邻频率点解调值的相互影响关系，获取修正后的驱动信号及其附近频率噪声的无噪声解调值，最终实现噪声消除处理，有效抑制同频噪声在位置检测系统中的负面效应，极大改善用户体验。
【附图说明】
图1为本发明实施例的同频噪声处理方法结构示意图；
图2为采样信号的频谱曲线图；
图3是本发明实施例的同频噪声处理系统的结构示意图；
图4是本发明实施例的噪声处理系统的电路图；
图5是本发明实施例的同频噪声处理系统的应用系统的结构示意图。
【具体实施方式】
为了便于理解本发明，下面将参照相关附图对本发明进行更全面的描述。附图中给出了本发明的较佳实施例。但是，本发明可以以许多不同的形式来实现，并不限于本文所描述的实施例。相反地，提供这些实施例的目的是使对本发明的公开内容的理解更加透彻全面。
除非另有定义，本文所使用的所有的技术和科学术语与属于本发明的技术领域的技术人员通常理解的含义相同。本文中在本发明的说明书中所使用的术语只是为了描述具体的实施例的目的，不是旨在于限制本发明。
在本发明以下实施例中，仅以位置检测系统的同频噪声处理为例进行说明，但并不仅限于此，本发明同样适用于其他噪声源系统的同频噪声处理，例如，蜂窝系统、集群系统或卫星通话系统等。
请一并参阅图1，是本发明实施例的同频噪声处理方法的流程图。本发明实施例的同频噪声处理方法包括以下步骤：
步骤S100：通过CPU控制器控制调制器在第一电极上加载驱动信号，驱动信号通过第一电极与第二电极之间的耦合电容耦合到第二电极上，生成感应信号；
在步骤S100中，所述在第一电极上加载驱动信号，驱动信号通过第一电极与第二电极之间的耦合电容耦合到第二电极上，生成感应信号具体包括：CPU控制器控制调制器加载在第一电极的驱动信号为TX，则：
TX＝sin(ωt)  (1)
在公式(1)中，
Figure PCTCN2016090100-appb-000011
为驱动信号频率f；TX信号通过第一电极与第二电极之间的耦合电容，耦合到第二电极上，该过程可等效为TX信号通过电容阻抗之后连通到第二电极，则第二电极检测的感应信号为RX：
Rx＝Asin(ωt+φ)+noise  (2)
在公式(2)中，A、φ分别为信号衰减系数和信号相位，即位置检测系统的系统相位；一般情况下，位置检测系统的系统相位φ是固定不变的，可视为已知量，则
noise＝Bsin(ωt+θ)  (3)
在公式(3)中，B、θ分别为噪声衰减系数和噪声相位，噪声是随机变化的，即B、θ表现为随机值；在本发明实施例中，加载在第一电极上的驱动信号为弦波驱动信号，但并不仅限于此，加载其他类型的驱动信号同样适用，例如方波信号等。
步骤S200：通过CPU控制器控制解调器对第二电极上的感应信号进行解调，获取感应信号的解调值；
在步骤S200中，第二电极耦合出的感应信号可通过ADC模数转换后传输给解调器，通过CPU控制器控制解调器进行解调，ADC可以通过CPU控制器、调制器或解调器等任一模块里实现；解调器对第二电极上的感应信号进行解调的方式具体包括：将感应信号RX展开，感应信号RX展开公式为：
Rx＝Asin(ωt+φ)+Bsin(ωt+θ)
＝(Acos(φ)+Bcos(θ))sin(ωt)+(Asin(φ)+Bsin(θ))cos(ωt)  (4)
对展开后的感应信号RX进行正交解调，获取感应信号RX的余弦分量(I值)和正弦分量(Q值)，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R，具体公式为：
I＝Acos(φ)+Bcos(θ)  (5)
Q＝Asin(φ)+Bsin(θ)  (6)
Figure PCTCN2016090100-appb-000012
步骤S300：根据感应信号的解调值计算位置检测系统的系统相位值arctan(Q/I)，并通过计算出的位置检测系统的系统相位值判断当前感应信号中是否存在同频噪声；如果存在同频噪声，执行步骤S400；如果不存在同频噪声，执行步骤S500；
在步骤300中，位置检测系统的系统相位值为：
Figure PCTCN2016090100-appb-000013
当无同频噪声时，即B＝0，则
Figure PCTCN2016090100-appb-000014
计算出的位置检测系统的系统相位值等于位置检测系统的真实系统相位值；
当有同频噪声时：即B≠0且φ≠θ，
Figure PCTCN2016090100-appb-000015
所述通过计算出的位置检测系统的系统相位值判断感应信号中是否存在同频噪声的判断方式为：判断计算出的位置检测系统的系统相位值是否等于位置检测系统的真实系统相位值，如果计算出的位置检测系统的系统相位值等于位置检测系统的真实系统相位值，则判定当前感应信号不存在同频噪声；如果计算出的位置检测系统的系统相位值不等于位置检测系统的真实系统相位值，则判定当前感应信号存在同频噪声。
步骤S400：通过任意频率点解调值与其相邻频率点解调值的相互影响关系式，计算驱动信号频率及其附近频率噪声的噪声量化值，根据所述噪声量化值 获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值，消除驱动信号频率中的同频噪声；
在步骤S400中，修正后的驱动信号频率及其附近频率噪声的无噪声调解值已经消除了同频噪声的影响；驱动信号频率及其附近频率噪声的无噪声调解值的计算方式具体为：由第一电极耦合到第二电极上的驱动信号，且通过汉宁窗，采样时间为T，则该采样信号的频谱曲线如图2所示，为采样信号的频谱曲线图(该技术为行业内公知技术，本发明将不再赘述)。其中，f为驱动信号频率，bitfreq＝1/T，bitfreq视为位置检测系统中驱动信号频率f的最小变化单位。从上述曲线理论推导可知，驱动信号频率f的信号在f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq五个频点处解调值的影响系数分别为[0,0.5,1,0.5,0]，则任意频率点解调值与其相邻频率点解调值的相互影响关系式为：
Figure PCTCN2016090100-appb-000016
Figure PCTCN2016090100-appb-000017
在公式(9)和公式(10)中，[R-2,R-1,R0,R+1,R+2]分别为f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq这五个频率点的解调值，可直接从步骤200中获得。而
Figure PCTCN2016090100-appb-000018
为驱动信号频率f的信号在对应频率点上修正处理后的无噪声解调值，由公式(9)和公式(10)计算获取。
步骤S500：利用感应信号解调值或修正后的无噪声解调值计算用户触摸位 置；
在步骤S500中，第一电极与第二电极为多电极阵列，多个电极阵列相互垂直布线，在整个触摸板形成密集的交叉节点，每一个交叉节点视为一个耦合电容。用户触摸时，会改变触摸位置附近耦合电容的大小，因此，只需检测每个电容的变化量，即可计算出用户触摸位置。具体操作而言，将第一电极的第一根电极阵列加载驱动信号，同时检测第二电极的所有电极阵列上的感应信号并解调，感应信号的解调值可视为两个电极阵列交叉耦合电容值。以同样的方式顺序完成第一电极上所有电极阵列的驱动信号加载，即可获得触摸板平面所有交叉耦合电容值。将用户无操作时的电容值设为基准值，当有用户进行触摸操作时，只需对比电容值实时值与基准值，便可快速计算出用户准确的触摸位置；当本发明应用于噪声源系统的其他领域时，也可利用感应信号解调值或修正后的无噪声解调值进行其他类型的信号指标的计算。
请一并参阅图3和图4，图3是本发明实施例的同频噪声处理装置的结构示意图；图4是本发明实施例的噪声处理装置的电路图。本发明实施例的同频噪声处理装置包括CPU控制器、调制器、第一电极、第二电极和解调器；其中，CPU控制器分别与调制器和解调器连接，所述调制器与第一电极连接，所述解调器与第二电极连接；
CPU控制器用于控制调制器在第一电极上加载驱动信号，驱动信号通过第一电极与第二电极之间的耦合电容耦合到第二电极上，生成感应信号；其中，所述CPU控制器控制调制器在第一电极上加载驱动信号具体包括：加载在第一电极的驱动信号为TX，则：
TX＝sin(ωt)  (1)
在公式(1)中，
Figure PCTCN2016090100-appb-000019
为驱动信号频率f；驱动信号TX通过第一电极与第二电极之间的耦合电容，耦合到第二电极上，该过程可等效为驱动信号TX通过电 容阻抗之后连通到第二电极，则第二电极检测的感应信号为RX：
Rx＝Asin(ωt+φ)+noise  (2)
在公式(2)中，A、φ分别为信号衰减系数和信号相位，即位置检测系统的系统相位；一般情况下，位置检测系统的系统相位φ是固定不变的，可视为已知量，则
noise＝Bsin(ωt+θ)  (3)
在公式(3)中，B、θ分别为噪声衰减系数和噪声相位，噪声是随机变化的，即B、θ表现为随机值。
CPU控制器控制解调器对第二电极上的感应信号进行解调，获取感应信号的解调值；其中，解调器对第二电极上的感应信号进行解调的方式具体包括：将感应信号RX展开，感应信号RX展开公式为：
Rx＝Asin(ωt+φ)+Bsin(ωt+θ)
＝(Acos(φ)+Bcos(θ))sin(ωt)+(Asin(φ)+Bsin(θ))cos(ωt)  (4)
对展开后的感应信号RX进行正交解调，获取感应信号RX的余弦分量(I值)和正弦分量(Q值)，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R，具体公式为：
I＝Acos(φ)+Bcos(θ)  (5)
Q＝Asin(φ)+Bsin(θ)  (6)
Figure PCTCN2016090100-appb-000020
CPU控制器具体还包括噪声判断模块、噪声消除模块和位置计算模块，噪声判断模块、噪声消除模块和位置计算模块依次连接；
噪声判断模块用于根据感应信号的解调值计算位置检测系统的系统相位值arctan(Q/I)，并通过计算出的位置检测系统的系统相位值判断当前感应信号中是否存在同频噪声；如果存在同频噪声，通过噪声消除模块获取修正后的无噪声解调值；如果不存在同频噪声，通过位置计算模块计算用户触摸位置；其中，位置检测系统的系统相位值为：
Figure PCTCN2016090100-appb-000021
当无同频噪声时，即B＝0，则
Figure PCTCN2016090100-appb-000022
计算出的位置检测系统的系统相位值等于位置检测系统的真实系统相位值；
当有同频噪声时：即B≠0且φ≠θ，
Figure PCTCN2016090100-appb-000023
所述噪声判断模块判断感应信号中是否存在同频噪声的判断方式为：判断计算出的位置检测系统的系统相位值是否等于位置检测系统的真实系统相位值，如果计算出的位置检测系统的系统相位值等于位置检测系统的真实系统相位值，则判定当前感应信号不存在同频噪声；如果计算出的位置检测系统的系统相位值不等于位置检测系统的真实系统相位值，则判定当前感应信号存在同频噪声。
噪声消除模块用于通过任意频率点解调值与其相邻频率点解调值的相互影响关系式，计算驱动信号频率及其附近频率噪声的噪声量化值，根据所述噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值，消除驱动信号频率中的同频噪声；其中，驱动信号频率及其附近频率噪声的无噪声调解值的计算方式具体为：由第一电极耦合到第二电极上的弦波信号，且通过汉宁窗，采样时间为T，则该采样信号的频谱曲线如图2所示，为采样信号的频谱曲线图。其中，f为驱动信号频率，bitfreq＝1/T，视为位置检测系统中驱动信 号频率f的最小变化单位。从上述曲线理论推导可知，驱动信号频率f的信号在f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq五个频点处解调值的影响系数分别为[0,0.5,1,0.5,0]，则任意频率点解调值与其相邻频率点解调值的相互影响关系式为：
Figure PCTCN2016090100-appb-000024
Figure PCTCN2016090100-appb-000025
在公式(9)和公式(10)中，[R-2,R-1,R0,R+1,R+2]分别为f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq这五个频率点的解调值，可直接通过解调器解调获得。而
Figure PCTCN2016090100-appb-000026
为驱动信号频率f的信号在对应频率点上修正处理后的无噪声解调值，由公式(9)和公式(10)计算获取。
位置计算模块用于利用感应信号解调值或修正后的无噪声解调值计算用户触摸位置；其中，如图4所示，第一电极与第二电极为多电极阵列，多个电极阵列相互垂直布线，在整个触摸板形成密集的交叉节点，每一个交叉节点视为一个耦合电容。用户触摸时，会改变触摸位置附近耦合电容的大小，因此，只需检测每个电容的变化量，即可计算出用户触摸位置。具体操作而言，将第一电极的第一根电极阵列加载驱动信号，同时检测第二电极所有电极阵列上的感应信号并解调，解调值可视为两个电极交叉耦合电容值。以同样的方式顺序完成第一电极上所有电极阵列的驱动信号加载，即可获得触摸板平面所有交叉耦合电容值。将用户无操作时的电容值设为基准值，当有用户进行触摸操作时， 只需对比电容值实时值与基准值，便可快速计算出用户准确的触摸位置。
请图5，是本发明实施例的同频噪声处理系统的结构示意图。本发明实施例的同频噪声处理系统包括位置检测系统和同频噪声处理装置，位置检测系统和同频噪声处理装置相互连接；同频噪声处理装置包括CPU控制器、调制器、第一电极、第二电极和解调器；其中，CPU控制器分别与调制器和解调器连接，所述调制器与第一电极连接，所述解调器与第二电极连接；
CPU控制器用于控制调制器在第一电极上加载驱动信号，驱动信号通过第一电极与第二电极之间的耦合电容耦合到第二电极上，生成感应信号；其中，所述CPU控制器控制调制器在第一电极上加载驱动信号具体包括：加载在第一电极的驱动信号为TX，则：
TX＝sin(ωt)  (1)
在公式(1)中，
Figure PCTCN2016090100-appb-000027
为驱动信号频率f；驱动信号TX通过第一电极与第二电极之间的耦合电容，耦合到第二电极上，该过程可等效为驱动信号TX通过电容阻抗之后连通到第二电极，则第二电极检测的感应信号为RX：
Rx＝Asin(ωt+φ)+noise  (2)
在公式(2)中，A、φ分别为信号衰减系数和信号相位，即位置检测系统的系统相位；一般情况下，位置检测系统的系统相位φ是固定不变的，可视为已知量，则
noise＝Bsin(ωt+θ)  (3)
在公式(3)中，B、θ分别为噪声衰减系数和噪声相位，噪声是随机变化的，即B、θ表现为随机值。
CPU控制器控制解调器对第二电极上的感应信号进行解调，获取感应信号的 解调值；其中，解调器对第二电极上的感应信号进行解调的解调方式具体包括：将感应信号RX展开，感应信号RX展开公式为：
Rx＝Asin(ωt+φ)+Bsin(ωt+θ)
＝(Acos(φ)+Bcos(θ))sin(ωt)+(Asin(φ)+Bsin(θ))cos(ωt)  (4)
对展开后的感应信号RX进行正交解调，获取感应信号RX的余弦分量(I值)和正弦分量(Q值)，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R，具体公式为：
I＝Acos(φ)+Bcos(θ)  (5)
Q＝Asin(φ)+Bsin(θ)  (6)
Figure PCTCN2016090100-appb-000028
CPU控制器具体还包括噪声判断模块、噪声消除模块和位置计算模块，噪声判断模块、噪声消除模块和位置计算模块依次连接；
噪声判断模块用于根据感应信号的解调值计算位置检测系统的系统相位值arctan(Q/I)，并通过计算出的位置检测系统的系统相位值判断采样感应信号中是否存在同频噪声；如果存在同频噪声，通过噪声消除模块获取修正后的无噪声解调值；如果不存在同频噪声，通过位置计算模块计算用户触摸位置；其中，位置检测系统的系统相位值为：
Figure PCTCN2016090100-appb-000029
当无同频噪声时，即B＝0，则
Figure PCTCN2016090100-appb-000030
计算出的位置检测系统的系统相位值等于位置检测系统的真实系统相位值；
当有同频噪声时：即B≠0且φ≠θ，
Figure PCTCN2016090100-appb-000031
所述噪声判断模块判断当前感应信号中是否存在同频噪声的判断方式为：获取位置检测系统的真实系统相位值，并判断计算出的位置检测系统的系统相位值是否等于位置检测系统的真实系统相位值，如果计算出的位置检测系统的系统相位值等于位置检测系统的真实系统相位值，则判定当前感应信号不存在同频噪声；如果计算出的位置检测系统的系统相位值不等于位置检测系统的真实系统相位值，则判定当前感应信号存在同频噪声。
噪声消除模块用于通过任意频率点解调值与其相邻频率点解调值的相互影响关系式，计算驱动信号频率及其附近频率噪声的噪声量化值，根据所述噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值，消除驱动信号频率中的同频噪声；其中，驱动信号频率及其附近频率噪声的无噪声调解值的计算方式具体为：由第一电极耦合到第二电极上的弦波信号，且通过汉宁窗，采样时间为T，则该采样信号的频谱曲线如图2所示，为采样信号的频谱曲线图。其中，f为驱动信号频率，bitfreq＝1/T，视为位置检测系统中驱动信号频率f的最小变化单位。从上述曲线理论推导可知，驱动信号频率f的信号在f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq五个频点处解调值的影响系数分别为[0,0.5,1,0.5,0]，则任意频率点解调值与其相邻频率点解调值的相互影响关系式为：
Figure PCTCN2016090100-appb-000032
Figure PCTCN2016090100-appb-000033
在公式(9)和公式(10)中，[R-2,R-1,R0,R+1,R+2]分别为f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq这五个频率点的解调值，可直接通过解调器解调获得。而
Figure PCTCN2016090100-appb-000034
为驱动信号频率f的信号在对应频率点上修正处理后的无噪声解调值，由公式(9)和公式(10)计算获取。
位置计算模块用于利用采样的感应信号解调值或修正后的无噪声解调值计算用户触摸位置。
本发明实施例的同频噪声处理方法、同频噪声处理装置及同频噪声处理系统通过在第一电极上加载驱动信号，采用正交解调方式获取第二电极上弦波信号的解调值，根据解调值计算系统相位，并通过系统相位的计算值判定当前感应信号是否包含同频噪声，解决了同频噪声处理的实时性问题；并通过出任意频率点解调值与其相邻频率点解调值的相互影响关系，获取修正后的驱动信号及其附近频率噪声的无噪声解调值，最终实现噪声消除处理，有效抑制同频噪声在位置检测系统中的负面效应，极大改善用户体验。
上述实施例为本发明较佳的实施方式，但本发明的实施方式并不受上述实施例的限制，其他的任何未背离本发明的精神实质与原理下所作的改变、修饰、替代、组合、简化，均应为等效的置换方式，都包含在本发明的保护范围之内。

Claims (10)

  1. 一种同频噪声处理方法，其特征在于，包括以下步骤：
    步骤a：在第一电极上加载驱动信号，所述驱动信号通过第一电极与第二电极之间的耦合电容耦合到第二电极上，生成感应信号；
    步骤b：对第二电极上的感应信号进行解调,根据感应信号的解调值计算噪声源系统的系统相位值，并通过计算出的噪声源系统的系统相位值判断当前感应信号中是否存在同频噪声，如果存在同频噪声，执行步骤c；
    步骤c：通过任意频率点解调值与其相邻频率点解调值的相互影响关系式，计算驱动信号频率及其附近频率噪声的噪声量化值，根据所述噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值，消除驱动信号频率中的同频噪声。
  2. 根据权利要求1所述的同频噪声处理方法，其特征在于：在所述步骤a中，所述加载在第一电极的驱动信号为：
    TX＝sin(ωt)
    所述生成的感应信号为：
    Rx＝A sin(ωt+φ)+noise
    在上述公式中，TX为驱动信号，
    Figure PCTCN2016090100-appb-100001
    为驱动信号频率f，RX为感应信号，A为信号衰减系数，φ为信号相位，即噪声源系统的系统相位值。
  3. 根据权利要求2所述的同频噪声处理方法，其特征在于：在所述步骤b中，所述对第二电极上的感应信号进行解调的解调方式具体包括：将感应信号RX展开，对展开后的感应信号RX进行正交解调，获取感应信号RX的余弦分量和正弦分量，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R；所述感应信号RX展开公式为：
    Rx＝A sin(ωt+φ)+B sin(ωt+θ)
    ＝(A cos(φ)+B cos(θ))sin(ωt)+(A sin(φ)+B sin(θ))cos(ωt)
    在上述公式中，B、θ分别为噪声衰减系数和噪声相位；
    所述获取余弦分量和正弦分量，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R的具体公式为：
    I＝A cos(φ)+B cos(θ)
    Q＝A sin(φ)+B sin(θ)
    Figure PCTCN2016090100-appb-100002
    上述公式中，I为余弦分量，Q为正弦分量。
  4. 根据权利要求3所述的同频噪声处理方法，其特征在于：在所述步骤b中，所述通过计算出的噪声源系统的系统相位值判断当前感应信号中是否存在同频噪声的判断方式为：判断所述步骤b中计算出的噪声源系统的系统相位值是否等于噪声源系统的真实系统相位值，如果所述步骤b中计算出的噪声源系统的系统相位值等于噪声源系统的真实系统相位值，则判定当前感应信号不存在同频噪声；如果所述步骤c中计算出的噪声源系统的系统相位值不等于噪声源系统的真实系统相位值，则判定当前感应信号存在同频噪声。
  5. 根据权利要求4所述的同频噪声处理方法，其特征在于：在所述步骤c中，所述根据噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值的计算方式具体为：由第一电极耦合到第二电极上的驱动信号，通过汉宁窗，采样时间为T，f为驱动信号频率，bitfreq＝1/T，bitfreq为驱动信号频率f的最小变化单位；驱动信号频率f的信号在f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq五个频点处解调值的影响系数分别为[0,0.5,1,0.5,0]，则任意频率点解调值与其相邻频率点解调值的相互影响关系式为：
    Figure PCTCN2016090100-appb-100003
    Figure PCTCN2016090100-appb-100004
    在上述公式中，[R-2,R-1,R0,R+1,R+2]分别为f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq这五个频率点的解调值，
    Figure PCTCN2016090100-appb-100005
    为驱动信号频率f的信号在对应频率点上修正处理后的无噪声解调值。
  6. 一种同频噪声处理装置，其特征在于：包括CPU控制器、调制器、第一电极、第二电极和解调器；所述CPU控制器用于控制调制器在第一电极上加载驱动信号，所述驱动信号通过第一电极与第二电极之间的耦合电容耦合到第二电极上，生成感应信号；
    所述CPU控制器控制解调器对第二电极上的感应信号进行解调；
    所述CPU控制器根据感应信号的解调值计算噪声源系统的系统相位值，并通过计算出的噪声源系统的系统相位值判断当前感应信号中是否存在同频噪声；如果存在同频噪声，通过任意频率点解调值与其相邻频率点解调值的相互影响关系式，计算驱动信号频率及其附近频率噪声的噪声量化值，根据所述噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值，并消除驱动信号频率中的同频噪声。
  7. 根据权利要求6所述的同频噪声处理装置，其特征在于：所述解调器对第二电极上的感应信号进行解调的解调方式具体为：将感应信号RX展开，对展开后的感应信号RX进行正交解调，获取感应信号RX的余弦分量和正弦分量，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R；所述感 应信号RX展开公式为：
    Rx＝A sin(ωt+φ)+B sin(ωt+θ)
    ＝(A cos(φ)+B cos(θ))sin(ωt)+(A sin(φ)+B sin(θ))cos(ωt)
    在上述公式中，
    Figure PCTCN2016090100-appb-100006
    为驱动信号频率f，A、φ分别为信号衰减系数和信号相位，B、θ分别为噪声衰减系数和噪声相位；
    所述获取余弦分量和正弦分量，根据余弦分量和正弦分量计算出在驱动频率f处感应信号RX的幅值R的具体公式为：
    I＝A cos(φ)+B cos(θ)
    Q＝A sin(φ)+B sin(θ)
    Figure PCTCN2016090100-appb-100007
    上述公式中，I为余弦分量，Q为正弦分量。
  8. 根据权利要求7所述的同频噪声处理装置，其特征在于：所述CPU控制器判断当前感应信号中是否存在同频噪声的判断方式为：判断计算出的噪声源系统的系统相位值是否等于噪声源系统的真实系统相位值，如果计算出的噪声源系统的系统相位值等于噪声源系统的真实系统相位值，则判定当前感应信号不存在同频噪声；如果计算出的噪声源系统的系统相位值不等于噪声源系统的真实系统相位值，则判定当前感应信号存在同频噪声。
  9. 根据权利要求8所述的同频噪声处理装置，其特征在于：所述CPU控制器计算驱动信号频率及其附近频率噪声的噪声量化值，根据噪声量化值获取修正后的驱动信号频率及其附近频率噪声的无噪声解调值的计算方式具体为：由第一电极耦合到第二电极上的驱动信号，通过汉宁窗，采样时间为T，f为驱动信号频率，bitfreq＝1/T，bitfreq为驱动信号频率f的最小变化单位；驱动 信号频率f的信号在f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq五个频点处解调值的影响系数分别为[0,0.5,1,0.5,0]，则任意频率点解调值与其相邻频率点解调值的相互影响关系式为：
    Figure PCTCN2016090100-appb-100008
    Figure PCTCN2016090100-appb-100009
    在上述公式中，[R-2,R-1,R0,R+1,R+2]分别为f-2*bitfreq、f-bitfreq、f、f+bitfreq、f+2*bitfreq这五个频率点的解调值，
    Figure PCTCN2016090100-appb-100010
    为驱动信号频率f的信号在对应频率点上修正处理后的无噪声解调值。
  10. 一种同频噪声处理系统，包括噪声源系统，其特征在于：还包括权利要求6至9任一项所述的同频噪声处理装置，所述噪声源系统及同频噪声处理装置信号连接；通过所述同频噪声处理装置消除噪声源系统中的同频噪声。
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