WO2017107938A1 - 一种重叠复用调制方法、装置和系统 - Google Patents

一种重叠复用调制方法、装置和系统 Download PDF

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Publication number
WO2017107938A1
WO2017107938A1 PCT/CN2016/111405 CN2016111405W WO2017107938A1 WO 2017107938 A1 WO2017107938 A1 WO 2017107938A1 CN 2016111405 W CN2016111405 W CN 2016111405W WO 2017107938 A1 WO2017107938 A1 WO 2017107938A1
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Prior art keywords
envelope waveform
waveform
modulation
envelope
sequence
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PCT/CN2016/111405
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English (en)
French (fr)
Inventor
刘若鹏
季春霖
张莎莎
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深圳超级数据链技术有限公司
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Priority claimed from CN201510976895.7A external-priority patent/CN106911611A/zh
Priority claimed from CN201510981466.9A external-priority patent/CN106911460A/zh
Priority claimed from CN201510977027.0A external-priority patent/CN106911615A/zh
Priority claimed from CN201510979680.0A external-priority patent/CN106911457A/zh
Priority claimed from CN201510977213.4A external-priority patent/CN106911396A/zh
Priority claimed from CN201510977028.5A external-priority patent/CN106911449A/zh
Priority claimed from CN201510979093.1A external-priority patent/CN106911453A/zh
Priority claimed from CN201510979188.3A external-priority patent/CN106911455A/zh
Priority claimed from CN201510977187.5A external-priority patent/CN106911616A/zh
Priority claimed from CN201510981427.9A external-priority patent/CN106911619A/zh
Priority claimed from CN201510977202.6A external-priority patent/CN106911452A/zh
Priority claimed from CN201510977201.1A external-priority patent/CN106911451A/zh
Priority claimed from CN201510977154.0A external-priority patent/CN106911450A/zh
Priority claimed from CN201510979091.2A external-priority patent/CN106911618A/zh
Priority claimed from CN201510976723.XA external-priority patent/CN106911608A/zh
Priority claimed from CN201510981428.3A external-priority patent/CN106911620A/zh
Priority claimed from CN201510981430.0A external-priority patent/CN106911459A/zh
Priority claimed from CN201510976982.2A external-priority patent/CN106911613A/zh
Priority claimed from CN201510979707.6A external-priority patent/CN106911458A/zh
Priority claimed from CN201510976950.2A external-priority patent/CN106911448A/zh
Priority claimed from CN201510977078.3A external-priority patent/CN106911413A/zh
Priority claimed from CN201510976738.6A external-priority patent/CN106911609A/zh
Priority claimed from CN201510976985.6A external-priority patent/CN106911614A/zh
Priority claimed from CN201510976691.3A external-priority patent/CN106911412A/zh
Priority claimed from CN201510976810.5A external-priority patent/CN106911446A/zh
Priority claimed from CN201510979525.9A external-priority patent/CN106911456A/zh
Priority claimed from CN201510976808.8A external-priority patent/CN106911440A/zh
Priority claimed from CN201510976894.2A external-priority patent/CN106911610A/zh
Priority claimed from CN201510977211.5A external-priority patent/CN106911441A/zh
Priority claimed from CN201510979187.9A external-priority patent/CN106911454A/zh
Priority claimed from CN201510977212.XA external-priority patent/CN106911617A/zh
Priority claimed from CN201510976914.6A external-priority patent/CN106911447A/zh
Priority to JP2018552107A priority Critical patent/JP6704470B2/ja
Application filed by 深圳超级数据链技术有限公司 filed Critical 深圳超级数据链技术有限公司
Priority to KR1020187019865A priority patent/KR102277047B1/ko
Priority to EP16877748.0A priority patent/EP3396891A4/en
Publication of WO2017107938A1 publication Critical patent/WO2017107938A1/zh
Priority to US16/017,012 priority patent/US10630408B2/en

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J4/00—Combined time-division and frequency-division multiplex systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J1/00—Frequency-division multiplex systems
    • H04J1/02—Details
    • H04J1/06—Arrangements for supplying the carrier waves ; Arrangements for supplying synchronisation signals
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J14/00—Optical multiplex systems
    • H04J14/08—Time-division multiplex systems
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J3/00—Time-division multiplex systems
    • H04J3/02—Details
    • H04J3/04—Distributors combined with modulators or demodulators
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J3/00—Time-division multiplex systems
    • H04J3/16—Time-division multiplex systems in which the time allocation to individual channels within a transmission cycle is variable, e.g. to accommodate varying complexity of signals, to vary number of channels transmitted
    • H04J3/1676—Time-division multiplex with pulse-position, pulse-interval, or pulse-width modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J99/00—Subject matter not provided for in other groups of this subclass
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00—Arrangements affording multiple use of the transmission path
    • H04L5/22—Arrangements affording multiple use of the transmission path using time-division multiplexing
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0006—Assessment of spectral gaps suitable for allocating digitally modulated signals, e.g. for carrier allocation in cognitive radio
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
    • H04L27/24—Half-wave signalling systems

Definitions

  • the present invention relates to the field of communications, and in particular, to an overlapping multiplexing modulation method, apparatus, and overlapping multiplexing system.
  • Time Division Multiplexing is a technique for sharing a plurality of signal symbols occupying a narrow time duration in a digital communication for a wide time duration.
  • Frequency Division Multiplexing is a technology that allows multiple signals occupying a narrow bandwidth to share a wider bandwidth.
  • the used signal bandwidths are B1, B2, B3, B4, ...., of course, they can also occupy the same bandwidth, ⁇ B is the minimum protection bandwidth, and the actual protection bandwidth can be a little more.
  • ⁇ B should be greater than the transition bandwidth of the demultiplexing filter used plus the maximum frequency drift of the system and the maximum frequency spread of the channel. This is the most common frequency division multiplexing technology, and most of the existing broadcasting systems, communication systems, and radar systems use this technology. The biggest feature of this technology is that the used signal spectrums are isolated from each other without mutual interference.
  • FIG. 1A is a schematic diagram of a conventional time division multiplexing technique; the time duration of each multiplexed signal symbol in FIG. 1A (called time slot width in engineering) is T1, T2, T3, T4, .. In engineering, they are usually allowed to occupy the same time slot width, ⁇ T is the minimum protection time slot, and the actual protection time slot width should be a bit more. ⁇ T should be greater than the transition time width of the demultiplexed gate used plus the maximum amount of time jitter of the system. This is the most common time division multiplexing technique. Most of the existing multi-channel digital broadcasting systems and multi-channel digital communication systems use this technology.
  • FIG. 1B corresponds to a schematic diagram of a frequency division multiplexing technique.
  • the used signal bandwidths are B1, B2, B3, B4, ...., of course, they can also occupy the same bandwidth, ⁇ B is the minimum protection bandwidth, and the actual protection bandwidth can be a little more.
  • ⁇ B should be greater than the transition bandwidth of the demultiplexing filter used plus the maximum frequency drift of the system and the maximum frequency spread of the channel. This is the most common frequency division multiplexing technology, and most of the existing broadcasting systems, communication systems, and radar systems use this technology. The biggest feature of this technology is that the used signal spectrums are isolated from each other without mutual interference.
  • the most important feature of this technology when applied to digital communication is that the multiplexed signal symbols are completely isolated from each other in time, and there is never mutual interference. There is no restriction on the multiplexed signal symbols, and the symbols of the respective signals.
  • the duration (slot width) can have different widths, and can also be applied to different communication systems, as long as their time slots do not overlap each other, and thus are most widely used. But with this multiplexing, multiplexing itself has no effect on improving the spectral efficiency of the system.
  • the conventional view is that adjacent channels do not overlap in the time domain to avoid interference between adjacent channels, but this technique restricts the improvement of spectral efficiency.
  • the prior art time division multiplexing technology has the view that the channels do not need to be isolated from each other, and can have strong mutual overlap, as shown in FIG. 2A.
  • the prior art considers the overlap between channels as a new coding constraint relationship, and proposes a corresponding modulation and demodulation technique according to the constraint relationship, so it is called Overlapped Time Division Multiplexing (OvTDM).
  • This technique causes the spectral efficiency to increase proportionally with the number of overlaps K; and in the frequency domain corresponds to Overlapped Frequency Division Multiplexing, which corresponds to FIG. 2B.
  • the number of overlaps K can be increased indefinitely, so the spectral efficiency can be increased indefinitely, but in the laboratory research stage, it is found that with the number of overlaps.
  • the increase of K although the spectral efficiency is increased, but the transmission power also increases, and the increase of transmission power in turn limits the increase of the number of overlaps K to a certain extent, thereby limiting the increase of spectral efficiency.
  • the present application provides an overlapping time division multiplexing modulation method, including:
  • the initial envelope waveform is shifted in a time domain according to a predetermined shift interval according to the number of overlapping multiplexing to obtain an offset envelope waveform of the transmitted signal at each moment;
  • the modulation envelope waveforms at various times are superimposed in the time domain to obtain a complex modulation envelope waveform carrying the output signal sequence.
  • the present application further provides an overlapping time division multiplexing modulation apparatus, including:
  • a waveform generating module configured to generate an initial envelope waveform of the waveform smoothing in the time domain according to the design parameter
  • a shifting module configured to shift the initial envelope waveform by a predetermined shift interval in the time domain according to the number of overlapping multiplexing, to obtain an offset envelope waveform of the transmitted signal at each moment;
  • a modulation module for converting the input digital signal sequence into a sequence of positive and negative symbols
  • a multiplication module configured to multiply the input positive and negative symbol sequence by an offset envelope waveform of the signal transmitted at each time after the offset to obtain a modulation envelope waveform at each moment;
  • the superposition module is configured to superimpose the modulation envelope waveforms at each moment in the time domain to obtain a complex modulation envelope waveform carrying the output signal sequence.
  • an overlapping frequency division multiplexing modulation method comprising the steps of:
  • the initial envelope waveform is shifted in a frequency domain according to a predetermined spectral interval according to the number of overlapping multiplexing, to obtain an envelope waveform of each subcarrier;
  • the complex modulation envelope waveform in the frequency domain is transformed to obtain a complex modulation envelope waveform in the time domain.
  • a fourth aspect of the present invention provides an overlapping frequency division multiplexing modulation apparatus, including:
  • a waveform generating module configured to generate an initial envelope waveform of a waveform smoothing in a frequency domain
  • a shifting module configured to shift the initial envelope waveform in a frequency domain according to a predetermined spectral interval according to the number of overlapping multiplexing, to obtain an envelope waveform of each subcarrier
  • a conversion module for converting the input digital signal sequence into a sequence of positive and negative symbols
  • a multiplication module configured to multiply the symbols in the sequence of positive and negative symbols by respective corresponding subcarrier envelope waveforms to obtain a modulation envelope waveform of each subcarrier
  • a superimposing module configured to superimpose a modulation envelope waveform of each subcarrier in a frequency domain to obtain a complex modulation envelope waveform in a frequency domain;
  • a transform module configured to transform the complex modulation envelope waveform in the frequency domain to obtain a complex modulation envelope waveform in the time domain.
  • the overlapping time division multiplexing modulation method, device and system since the time domain waveform of the initial envelope waveform is relatively smooth, the frequency domain bandwidth is narrow, and the superposed waveform is smooth and limited to a narrow bandwidth, thereby improving The spectrum utilization rate and transmission rate of the system reduce the bit error rate of the system; and the overlapping frequency division multiplexing modulation method, device and system, because the generated initial envelope waveform is smooth in the frequency domain, correspondingly, in the time domain
  • the energy is concentrated and the duration is short, so the polymodulation envelope waveform formed by its modulation is concentrated in the time domain and has a short duration, so its spectrum utilization is high, the signal transmission rate is also high, and only a low transmission is required. Power, which has a lower bit error rate when demodulated.
  • 1A is a schematic diagram of a conventional time division multiplexing technique
  • 1B is a schematic diagram of a conventional frequency division multiplexing technique
  • 2A is a schematic diagram of the principle of overlapping time division multiplexing
  • 2B is a schematic diagram of the principle of overlapping frequency division multiplexing
  • 3A is a schematic structural diagram of an overlapping time division multiplexing system according to an embodiment of the present invention.
  • FIG. 3B is a schematic structural diagram of an overlapping frequency division multiplexing system according to an embodiment of the present invention
  • FIG. 4A is a schematic structural diagram of an overlapping time division multiplexing modulation apparatus according to an embodiment of the present invention
  • 4B is a schematic structural diagram of an overlapping frequency division multiplexing modulation apparatus according to an embodiment of the present invention.
  • 5A is a schematic diagram showing the hardware structure of an overlapping time division multiplexing modulation apparatus according to an embodiment of the present invention
  • 5B is a schematic structural diagram of hardware of an overlapping frequency division multiplexing modulation apparatus according to an embodiment of the present invention.
  • FIG. 6 is a schematic structural diagram of a receiver preprocessing apparatus according to an embodiment of the present invention.
  • FIG. 7 is a schematic structural diagram of a receiver sequence detecting apparatus according to an embodiment of the present invention.
  • FIG. 8 is a time domain waveform and a frequency domain waveform diagram of a Chebyshev envelope waveform according to an embodiment of the present invention.
  • FIG. 10 is a schematic diagram of superposition of waveforms to be transmitted when a Chebyshev envelope waveform is used in an embodiment of the present invention
  • FIG. 11 is a schematic diagram of the principle of K-way waveform multiplexing
  • FIG. 12 is a schematic diagram showing the principle of a symbol superposition process of a K-path waveform
  • Figure 14 is a node state transition diagram
  • Figure 15 is a time-domain and frequency-domain waveform diagram of a rectangular wave
  • Figure 16 is a diagram showing each of the signals generated when the rectangular wave envelope waveform is selected by the envelope waveform and the superimposed waveform;
  • 17 is a time-domain waveform and a frequency domain waveform diagram of a Blackman first-order derivative envelope waveform according to an embodiment of the present invention.
  • FIG. 19 is a schematic diagram showing superposition of waveforms to be transmitted when a Blackman first-order derivative envelope waveform is used in an embodiment of the present invention.
  • FIG. 20 is a schematic diagram of superposition of waveforms to be transmitted when a Blackman-Harris first-order derivative envelope waveform is used in an embodiment of the present invention.
  • 21 is a time-domain waveform and a frequency domain waveform diagram of a Bartlett envelope waveform in an embodiment of the present invention.
  • 22 is an envelope waveform diagram of each time after the Bartlett window is shifted in an embodiment of the present invention.
  • FIG. 23 is a schematic diagram showing superposition of waveforms to be transmitted when a Bartlett envelope waveform is used in an embodiment of the present invention.
  • 24 is a time domain waveform and a frequency domain waveform diagram of a Gaussian envelope waveform in an embodiment of the present invention.
  • 25 is an envelope waveform diagram of each time after a Gaussian window is shifted according to an embodiment of the present invention.
  • 26 is a schematic diagram of superposition of waveforms to be transmitted when a Gaussian envelope waveform is used in an embodiment of the present invention
  • 27 is a time-domain waveform and a frequency domain waveform diagram of a Hanning envelope waveform in an embodiment of the present invention
  • 29 is a schematic diagram showing superposition of waveforms to be transmitted when a Hanning envelope waveform is used in an embodiment of the present invention.
  • FIG. 30 is a time-domain waveform and a frequency domain waveform diagram of a Kaiser envelope waveform in an embodiment of the present invention
  • FIG. 31 is an envelope waveform diagram of each time after a Kaiser window is shifted according to an embodiment of the present invention.
  • 33 is a time-domain waveform and a frequency domain waveform diagram of a Hamming envelope waveform in an embodiment of the present invention.
  • FIG. 34 is a diagram showing envelope waveforms of respective moments after a Hamming window is shifted according to an embodiment of the present invention.
  • FIG. 35 is a schematic diagram of superposition of waveforms to be transmitted when a Hamming envelope waveform is used in an embodiment of the present invention.
  • 36 is a time-domain waveform and a frequency domain waveform diagram of a Bartlett-Hanning envelope waveform in an embodiment of the present invention
  • FIG. 38 is a schematic diagram showing superposition of waveforms to be transmitted when a Bartlett-Hanning envelope waveform is used in an embodiment of the present invention
  • 39 is a time domain waveform and a frequency domain waveform diagram of a Blackman envelope waveform in an embodiment of the present invention.
  • 40 is an envelope waveform diagram of each time after a Blackman window is shifted according to an embodiment of the present invention.
  • FIG. 41 is a schematic diagram showing superposition of waveforms to be transmitted when a Blackman envelope waveform is used in an embodiment of the present invention
  • 43 is an envelope waveform diagram of each time after a Berman window is shifted according to an embodiment of the present invention.
  • FIG. 44 is a schematic diagram showing superposition of waveforms to be transmitted when a Berman envelope waveform is used in an embodiment of the present invention.
  • FIG. 46 is an envelope waveform diagram of each time after the flat top window is shifted according to an embodiment of the present invention.
  • 47 is a schematic diagram showing superposition of waveforms to be transmitted when a flat top envelope waveform is used in an embodiment of the present invention.
  • 49 is an envelope waveform diagram of each time after the Natal window is shifted according to an embodiment of the present invention.
  • FIG. 50 is a schematic diagram showing superposition of waveforms to be transmitted when a Natto envelope waveform is used in an embodiment of the present invention.
  • 51 is a time domain waveform and a frequency domain waveform diagram of a triangular envelope waveform in an embodiment of the present invention
  • 53 is a schematic diagram of superposition of waveforms to be transmitted when a triangular envelope waveform is used in an embodiment of the present invention
  • Figure 54 is a time-domain waveform and a frequency domain waveform diagram of a Barson envelope waveform in an embodiment of the present invention.
  • 55 is an envelope waveform diagram of each time after the Barson window is shifted in an embodiment of the present invention.
  • FIG. 56 is a schematic diagram showing superposition of waveforms to be transmitted when a Barson envelope waveform is used in an embodiment of the present invention.
  • 57 is a time-domain waveform and a frequency domain waveform diagram of a graph envelope waveform in an embodiment of the present invention.
  • FIG. 58 is an envelope waveform diagram of each time after the base window is shifted according to an embodiment of the present invention.
  • 60 is a time-domain waveform and a frequency domain waveform diagram of a Taylor envelope waveform in an embodiment of the present invention.
  • 61 is an envelope waveform diagram of each time after a Taylor window is shifted according to an embodiment of the present invention.
  • the inventors found that the increase in transmission power is mainly related to the spectrum of the multiplexed signal (ie, the modulation window function), not the shape and bandwidth of the multiplexed signal spectrum as theoretically assumed. Any request.
  • window functions there are many window functions in the prior art, it is theoretically free to use various window functions to modulate the transmitted symbols, but since the rectangular window is easier and less expensive to generate, design, and apply than other window functions, At present, the rectangular window is preferentially used in signal modulation, and the spectral bandwidth of the rectangular wave is wide, and the performance of the multiplexed waveform system is poor, resulting in high required transmission power and bit error rate.
  • the input digital signal sequence is modulated by using a window function superior to the rectangular wave when applying the overlapping time division multiplexing technique.
  • the overlapping time division multiplexing system includes a signal transmitter A01 and a receiver A02.
  • the transmitter A01 includes an overlapping time division multiplexing modulation device 301 and a transmitting device 302.
  • the overlapping time division multiplexing modulation device 301 is configured to generate a complex modulation envelope waveform carrying an output signal sequence; the transmitting device 102 is configured to transmit the complex modulation envelope waveform to the receiver A02.
  • the receiver A02 includes a receiving device 303 and a sequence detecting device 305.
  • the receiving device 303 is configured to receive the complex modulation envelope waveform transmitted by the transmitting device 302.
  • the sequence detecting device 305 is configured to perform data sequence detection in the time domain on the received complex modulation envelope waveform for decision output.
  • the receiver A02 further includes a pre-processing device 304 disposed between the receiving device 303 and the sequence detecting device 305 for assisting in forming a sequence of synchronously received digital signals within each frame.
  • a pre-processing device 304 disposed between the receiving device 303 and the sequence detecting device 305 for assisting in forming a sequence of synchronously received digital signals within each frame.
  • the input digital signal sequence forms a plurality of transmission signals in which the symbols overlap each other in the time domain by the overlapping time division multiplexing modulation means 301, and the transmission signal is transmitted from the transmitting means 302 to the receiver A02.
  • the receiving device 303 of the receiver A02 receives the signal transmitted by the transmitting device 302, and passes the pre-
  • the processing device 304 forms a digital signal suitable for the sequence detection device 305 to detect and receive, and the sequence detecting device 305 detects the data sequence in the time domain of the received signal, thereby outputting a decision.
  • the structure of the corresponding transmitter and receiver in the overlapping frequency division multiplexing system is as shown in FIG. 3B, and the transmitter B1 includes an overlapping frequency division multiplexing modulation device 310 and a transmitting device 320, wherein the overlapping frequency division multiplexing modulation
  • the device 310 is configured to modulate and generate a complex modulation envelope waveform carrying an output signal sequence
  • the transmitting device 320 is configured to transmit the complex modulation envelope waveform to the receiver B2.
  • the receiver B2 includes a receiving device 330 and an overlapping frequency division multiplexing demodulating device 340, wherein the receiving device 330 is configured to receive the complex modulation envelope waveform transmitted by the transmitting device 320, and the overlapping frequency division multiplexing demodulating device 340 is configured to The received complex modulation envelope waveform is demodulated and decoded.
  • the overlapping time division multiplexing modulation device 301 (OvTDM modulation device) in FIG. 3A includes a waveform generation module 301, a shift module 302, a multiplication module 303, and a superposition module 304.
  • the waveform generation module 301 is configured to generate an initial envelope waveform of the waveform smoothing in the time domain according to the design parameters.
  • the shifting module 302 is configured to shift the initial envelope waveform by a predetermined shift interval in the time domain according to the number of overlapping multiplexing to obtain an offset envelope waveform of the transmitted signal at each moment.
  • Modulation module 305 is operative to convert the input digital signal sequence into a sequence of positive and negative symbols.
  • the multiplication module 303 is configured to multiply the converted sequence of positive and negative symbols with the offset envelope waveform of the signal transmitted at each time after the offset to obtain a modulation envelope waveform at each moment.
  • the superposition module 304 is configured to superimpose the modulation envelope waveforms at respective moments in the time domain to obtain a complex modulation envelope waveform carrying the output signal sequence.
  • the overlapping frequency division multiplexing modulation apparatus 310 includes a waveform generation module 411, a shifting module 412, a conversion module 413, a multiplication module 414, a superposition module 415, and a transformation module 416. .
  • the waveform generation module 411 is configured to generate an initial envelope waveform of the waveform smoothing in the frequency domain according to the design parameters.
  • the design parameters include at least a bandwidth width of the initial envelope waveform.
  • the shifting module 412 is configured to shift the initial envelope waveform in a frequency domain according to a predetermined spectral interval according to the number of overlapping multiplexing to obtain an envelope waveform of each subcarrier.
  • the conversion module 413 is configured to convert the input digital signal sequence into a sequence of positive and negative symbols.
  • the multiplication module 414 is configured to multiply the symbols in the sequence of positive and negative symbols and the corresponding subcarrier envelope waveforms to obtain a modulation envelope waveform of each subcarrier.
  • the superimposing module 415 is configured to superimpose the modulation envelope waveforms of the subcarriers in the frequency domain to obtain Complex modulation envelope waveform in the frequency domain.
  • the transform module 416 is configured to transform the complex modulation envelope waveform in the frequency domain to a complex modulation envelope waveform in the time domain.
  • the transform module 416 can transform the complex modulation envelope waveform in the frequency domain into a complex modulation envelope waveform in the time domain by using an inverse Fourier transform.
  • the complex modulation envelope waveform generated by the modulation carries an output signal sequence corresponding to the converted positive and negative symbol sequence, and the output signal sequence is composed of output signals of each spectral interval, and the output signals of each spectral interval are within each spectral interval.
  • the overlapping time division multiplexing modulation apparatus 101 is further described below in conjunction with the overlapping time division multiplexing modulation method.
  • the overlapping time division multiplexing modulation method includes the following steps:
  • the waveform generation module 401 generates an initial envelope waveform h(t) of the waveform smoothing in the time domain based on the design parameters.
  • the user can input the design parameters to achieve flexible configuration according to system performance indicators in the actual system.
  • the design parameters include the window length L of the initial envelope waveform, such as when the initial envelope waveform is a Bartlett envelope waveform.
  • the design parameters include the window length L of the initial envelope waveform and the sidelobe attenuation r, such as when the initial envelope waveform is a Chebyshev envelope waveform.
  • the design parameters can be determined according to the characteristics of the corresponding initial envelope waveform.
  • the shifting module 402 shifts the initial envelope waveform by a predetermined shift interval in the time domain according to the number of overlapping multiplexing times K to obtain an offset envelope waveform h (ti* ⁇ T) of the transmitted signal at each moment. .
  • the shift interval is a time interval ⁇ T
  • the modulation module 405 converts the input digital signal sequence into a sequence of positive and negative symbols.
  • the modulation module 405 converts 0 in the input digital signal sequence to +A, 1 to -A, and A to a non-zero arbitrary number to obtain a sequence of positive and negative symbols. For example, when A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK (Binary Phase Shift Keying) modulation.
  • BPSK Binary Phase Shift Keying
  • the multiplication module 403 multiplies the converted sequence of positive and negative symbols x i by the offset envelope waveform h (ti* ⁇ T) of the signal transmitted at each time after the offset to obtain a modulation envelope waveform x at each time. i h(ti* ⁇ T).
  • the superposition module 404 superimposes the modulation envelope waveform x i h(ti* ⁇ T) at each time in the time domain to obtain a complex modulation envelope waveform carrying the output signal sequence, that is, the transmitted signal.
  • the signal sent can be expressed as follows:
  • the frequency domain bandwidth is narrower, and the superposed waveform is smoother and limited to a narrower bandwidth, thereby improving the spectrum utilization rate and transmission rate of the system, and reducing the bit error rate of the system. .
  • the overlapping time division multiplexing modulation device 301 can be implemented by the following hardware unit.
  • the overlapping time division multiplexing modulation device 301 includes a digital waveform generator 501, a shift register 502, a modulator 503, a multiplier 504, and an adder 505.
  • an in-phase waveform of the first initial envelope waveform is digitally formed by the digital waveform generator 501, the initial envelope waveform is smoothed in the time domain; and the first initial generated by the digital waveform generator 401 by the shift register 502
  • the in-phase waveform of the envelope waveform is shifted to generate an offset envelope waveform of the transmitted signal at each moment; then, modulator 503 converts the input digital signal sequence into a sequence of positive and negative symbols, and multiplier 504 converts the converted positive signal.
  • the negative symbol sequence is multiplied by the offset envelope waveform of the transmitted signal at each time after the offset to obtain the modulated envelope waveform at each moment; finally, the adder 505 superimposes the modulated envelope waveform at each moment in the time domain.
  • a transmit signal is formed.
  • the overlapping frequency division multiplexing demodulation device includes a spectrum module 51, a frequency segmentation module 52, a convolutional coding module 53 and a data detection module 54. .
  • the spectrum module 51 is configured to transform the received symbol sequence in the above time domain to form a received signal spectrum.
  • the spectral module 51 transforms the received symbol sequence in the time domain into a received signal spectrum using a Fourier transform.
  • the frequency segmentation module 52 is configured to segment the received signal spectrum in the frequency domain by the subcarrier spectral interval ⁇ B to obtain a received signal segmentation spectrum.
  • the convolutional coding module 53 is configured to perform convolutional coding on the received signal segmentation spectrum in each subcarrier spectral interval ⁇ B to obtain a sequence between the received signal spectrum and the positive and negative symbol sequence converted by the input digital signal sequence in the transmitter.
  • convolutional coding module 53 is configured to perform convolutional coding on the received signal segmentation spectrum in each subcarrier spectral interval ⁇ B to obtain a sequence between the received signal spectrum and the positive and negative symbol sequence converted by the input digital signal sequence in the transmitter.
  • the data detecting module 54 is configured to detect the sequence of positive and negative symbols according to the one-to-one correspondence.
  • FIG. 6 is a block diagram of a preprocessing apparatus of a receiver A02 in an overlapping time division system according to an embodiment of the present invention.
  • the pre-processing device includes a synchronizer 501, a channel estimator 502, and a digitizer 503.
  • the synchronizer 501 forms symbol time synchronization in the receiver for the received signal; the channel estimator 502 then estimates the channel parameters; the digitizer 503 digitizes the received signal in each frame to form a suitable sequence detecting device. The sequence detects the received digital signal sequence.
  • FIG. 7 is a block diagram of a sequence detecting apparatus 202 of a receiver A02 in an embodiment of the overlapping time division system according to the present invention.
  • the sequence detecting means includes an analyzing unit memory 701, a comparator 702, and a plurality of reserved path memories 703 and an Euclidean distance memory 704 or a weighted Euclidean distance memory (not shown).
  • the analysis unit memory 701 makes a complex convolutional coding model and a trellis diagram of the overlapping time division multiplexing system, and lists all states of the overlapping time division multiplexing system, and stores them; and the comparator 702 according to the analysis unit memory 701 a trellis diagram that searches for a path that is at least a minimum Euclidean distance or a weighted minimum Euclidean distance from the received digital signal; and the reserved path memory 703 and the Euclidean distance memory 704 or the weighted Euclidean distance memory are used to store the comparator 702, respectively.
  • the reserved path and Euclidean distance or weighted Euclidean distance of the output need to be prepared for each of the stable states.
  • the length of the reserved path memory 703 may preferably be 4K to 5K.
  • the Euclidean distance memory 604 or the weighted Euclidean distance memory preferably stores only relative distances.
  • the initial envelope waveforms used in overlapping time division/frequency division multiplexing modulation methods, apparatus, and systems may include Chebyshev, Gaussian, Hamming, Hann, Blackman. ), Blackman-Harris, Bartlett, Bartlett-Hanning, Bohman, Flat Top, Nuttall, Multiplexed waveforms such as Parzen, Taylor, Tukey, Kaiser, and Triangular, and evolutionary waveforms based on them.
  • the initial envelope waveform is a Chebyshev envelope waveform
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • the Chebyshev envelope waveform h(t) of the transmitted signal is generated according to the design parameters.
  • the Chebyshev window in the time domain waveform starts from approximately 0, and the frequency domain sidelobe attenuation is 80 dB.
  • 0 in the input digital signal sequence can be converted to +A, 1 is converted to -A, and A is taken.
  • the value is non-zero any number to get a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the signal sent can be expressed as:
  • the output signal sequence is determined by:
  • the operation value of the modulation envelope waveform is +A
  • the modulation envelope waveform is multiplied by the negative symbol and the envelope waveform at the time
  • FIG. 11 is a schematic diagram of the principle of K-way waveform multiplexing, which has a parallelogram shape.
  • Each row represents a waveform to be transmitted x i h(ti* ⁇ T) obtained by multiplying a symbol x i to be transmitted with an envelope waveform h (ti* ⁇ T) at a corresponding time.
  • a 0 to a k-1 represent coefficient values of each part obtained by K-segmentation of each window function waveform (envelope waveform), specifically, coefficients regarding amplitude values.
  • FIG. 12 is a schematic diagram showing the principle of the symbol superposition process of the K-way waveform. In the superimposition process of Fig.
  • the third digit on the left side of the first row indicates the first input symbol +1
  • the third digit on the left side of the second row indicates the second input symbol +1
  • the third digit on the left of the third row indicates the third input.
  • the middle 3 digits of the 1st line represent the 4th input symbol -1
  • the middle 3 digits of the 2nd row represent the 5th input symbol -1
  • the 3rd row of the 3rd row represents the 6th input symbol + 1.
  • the third number on the right side of the first line indicates the seventh input symbol -1
  • the third number on the right side of the second line indicates the eighth input symbol +1. Therefore, after the three waveforms are superimposed, the resulting output symbol is ⁇ +1 +2 +1 -1 -3 -1 +1 ⁇ .
  • the Chebyshev envelope waveform starts from 0 (0.0028, close to 0) in the time domain and has a smooth waveform, the superimposed waveform is smoother and the frequency domain bandwidth is narrower, so that the superimposed waveform has higher spectral efficiency and is transmitted. The transmission power required for the signal is low. Because the Chebyshev envelope waveform can design the sidelobe attenuation by itself, it can be flexibly configured according to the system performance index in the actual system.
  • the signal receiving process includes the following steps:
  • the received signal is synchronized, including carrier synchronization, frame synchronization, symbol time synchronization, and the like.
  • the received signal in each frame is digitized.
  • the symbol sequence is based on the tree diagram of the input-output relationship of FIG. 7 and the node state transition relationship diagram of FIG. 8, and the front-to-back comparison between the symbols is performed to obtain the node transition path.
  • the node state transition in this case is shown by the bold black line in Figure 13. Since the first symbol of s(t) is +1, the node transition path is: +1->a->a->b ->d->d->c->b->c, according to this transfer relationship, the input symbol sequence can be found as ⁇ +1 +1 -1 -1 -1 +1 -1 +1 ⁇ .
  • FIG. 15 is a time domain and frequency domain waveform diagram of a rectangular wave.
  • the initial envelope waveform selects a rectangular wave envelope waveform
  • the respective signals generated according to the above signal generation process and the superimposed waveform diagram are as shown in FIG. 16, wherein three different broken lines represent three waveform diagrams, and solid lines indicate superposition. After the waveform.
  • the rectangular wave starts from 1 in the time domain and has a wide bandwidth.
  • the sidelobe attenuation is slow in the frequency domain, so the waveform after time domain superposition is not smooth, the frequency domain bandwidth is wide, and the effective signal And invalid letter
  • the numbers are indistinguishable, so that the transmission power required in the process of transmitting and receiving signals is increased, and the accuracy and codec capability of waveform cutting during reception of signals are reduced.
  • the transmission rate is the same and the spectrum efficiency is the same in an actual system, the transmission power and the bit error rate required when a rectangular wave is used are high.
  • the Chebyshev window used in this embodiment starts from 0 (0.0028, close to 0) in the time domain, the side lobe attenuates faster, the waveform after signal superposition is smooth, and the frequency domain bandwidth is narrower, which improves the waveform cutting process.
  • the accuracy rate and the error correction capability of the codec process reduce the transmission power of the signal, so that when the spectrum efficiency is constant, a higher transmission rate can be achieved by using a lower transmission power.
  • the Chebyshev window can design the sidelobe attenuation by itself, it can be flexibly configured according to system performance indicators in the actual system.
  • the initial envelope waveform may also select various envelope waveforms of functions that evolved by Chebyshev window function, including the sum of Chebyshev pulse shaping, the derivatives of each order, and the sum of the derivatives of each order.
  • the envelope waveforms of the functions, these envelope waveforms also have the characteristics of waveform smoothing in the time domain, so the effects of the Chebyshev envelope waveform can be achieved by using these envelope waveforms.
  • This embodiment can also be applied to an overlapping frequency division multiplexing system, except that the Chebyshev envelope waveform is a function waveform in the frequency domain, that is, the left image in FIG. 8 is a sampling in the frequency domain, and the right image is The normalization function on the time domain.
  • the rest of the modulation and demodulation method steps are similar and will not be described again.
  • the signal generation process includes the following steps:
  • the Blackman first derivative of the transmitted signal and the envelope waveform h(t) corresponding to the Blackman-Harris first derivative are generated.
  • the envelope waveform of the first derivative of Blackman starts from approximately 0 in the time domain, and the amplitude becomes negative in the latter half.
  • the waveform approaches the sine wave and the frequency domain sidelobe attenuation It is about 40dB.
  • the envelope waveform of the first derivative of Blackman-Harris starts from approximately 0 in the time domain, and becomes negative in the latter half.
  • the waveform approaches the sine wave, and the frequency domain
  • the sidelobe attenuation is about 100 dB.
  • ⁇ (n) 0.42-0.5cos(2 ⁇ n/(N-1))+0.08cos(4 ⁇ n/(N-1))
  • the obtained waveform is the first half of the Blackman window, and for the waveform of the latter half of the Blackman window (that is, when M ⁇ n ⁇ N-1), It and the first half
  • ⁇ (n) a0-a1cos(2 ⁇ n/(N-1))+a2cos(4 ⁇ n/(N-1))+a3cos(6 ⁇ n/(N-1))
  • ⁇ (n) a0-a1cos2 ⁇ n/N+a2cos4 ⁇ n/N+a3cos6 ⁇ n/N
  • N is the window length
  • a0 0.35875
  • a1 0.48829
  • a2 0.14128
  • a3 0.01168. It should be noted that n in the above formula only represents the function variable in the formula.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A takes a value of 1
  • the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the signal sent can be expressed as:
  • the output signal sequence is determined by:
  • the operation value of the modulation envelope waveform is +A
  • the modulation envelope waveform is multiplied by the negative symbol and the envelope waveform at the time
  • the signal receiving process is the same as the signal receiving process in the case where the Chebyshev envelope waveform is used in the first embodiment, and therefore, the description is not repeated herein.
  • Blackman's first derivative and Blackman-Harris first derivative multiplexed waveform are smoother in the time domain and the side lobes are attenuated faster, the required transmission power is lower, and the waveform is cut with higher precision.
  • the received symbol sequence is more accurate.
  • the characteristic of the rectangular wave is that the main lobe is concentrated.
  • the disadvantage is that the side lobes are high and have negative side lobes, which leads to high frequency interference and leakage in the transformation, and even negative spectrum phenomenon, and the amplitude identification accuracy is the lowest.
  • the Blackman first-order derivative and the Blackman-Harris first-order derivative multiplexed waveform are characterized by a main lobe width, a low side lobes, the highest amplitude recognition accuracy, and better selectivity.
  • the OvTDM process with the first derivative of Blackman and the first derivative of Blackman-Harris as the multiplexed waveform the signal in the signal transmission process, the time domain waveform is smooth, the frequency domain bandwidth is narrow, and the transmission power required for transmitting the signal is low, and Both spectrum utilization and transmission rates are high.
  • the waveform is smooth in the time domain, the accuracy of cutting the waveform is higher, and the error rate of the system is lowered. System performance is greatly improved over rectangular waves.
  • the initial envelope waveform may also select various Blackman window prototypes, or envelope waveforms of other functions evolved by the Blackman window function, including the addition of Blackman pulse shaping, the derivatives of each order,
  • the envelope waveforms of the functions such as the sum of the derivatives of each order can be approximated by the first derivative of the Blackman waveform by using these envelope waveforms.
  • the initial envelope waveform selects various Blackman-Harris window prototypes, or envelope waveforms of other functions evolved by the Blackman-Harris window function, including the multiples of Blackman-Harris pulse shaping, the derivatives of each order Envelope waveforms of functions such as sums of derivatives of the order, these envelope waveforms also have waveform smoothing characteristics in the time domain. Therefore, the first derivative of the Blackman-Harris waveform can be achieved by using these envelope waveforms. Approximate effect.
  • This embodiment can also be applied to an overlapping frequency division multiplexing system, except that the Blackman envelope waveform is a function waveform in the frequency domain, that is, the left image in FIGS. 17 and 18 is a sampling in the frequency domain, and the right image is shown. Is the normalization function on the time domain. The rest of the modulation and demodulation method steps are similar and will not be described again.
  • This embodiment performs modulation and demodulation with the Bartlett envelope waveform in the OvTDM system.
  • the initial envelope waveform in this embodiment is the envelope waveform of the Bartlett envelope waveform or its evolution window function.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • a Bartlett envelope waveform h(t) of a transmission signal is generated based on design parameters.
  • the Bartlett window in the time domain starts at 0 and has an attenuation of nearly 30 dB out of band in the frequency domain.
  • Bartlett window function can be expressed by the following formula:
  • the Bartlett envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 21 is the sampling in the frequency domain, and the right graph is the time domain.
  • the normalization function on The rest of the modulation and demodulation method steps are similar and will not be described again.
  • 0 in the input digital signal sequence can be converted to +1, and 1 is converted to -1 to obtain a sequence of positive and negative symbols.
  • the input ⁇ 0, 1 ⁇ bit sequence is BPSK-modulated into a ⁇ +1, -1 ⁇ symbol sequence.
  • the method for receiving the signal is similar to the method of the previous embodiment, and details are not described herein again.
  • the initial envelope waveform as a Gaussian envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • a Gaussian envelope waveform h(t) of a transmission signal is generated according to design parameters.
  • the Gaussian window in the time domain waveform starts from approximately 0, and the frequency domain sidelobe attenuation is nearly 50 dB.
  • N is the window length, -N/2 ⁇ n ⁇ N/2, ⁇ is a preset parameter, and FIG. 24 shows the time domain waveform and frequency of the Gaussian window with ⁇ , 2.5, 1.5, and 0.5, respectively. Domain waveform.
  • the Gaussian envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 24 is the sampling in the frequency domain, and the right graph is in the time domain. Normalized function. The rest of the modulation and demodulation method steps are similar and will not be described again.
  • the transfer envelope waveform is shown in Figure 25.
  • 0 in the input digital signal sequence can be converted to +A, and 1 can be converted to -A to obtain a sequence of positive and negative symbols.
  • a takes a value of 1 the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the method for receiving the signal is similar to the method of the previous embodiment, and details are not described herein again.
  • the initial envelope waveform is the envelope waveform of the Hann envelope waveform or its evolution window function.
  • the following is a further description of the Hann envelope waveform with the initial envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • the Hann envelope waveform h(t) of the transmitted signal is generated according to the design parameters.
  • the Hann window in the time domain waveform starts from 0 o'clock, and the frequency domain sidelobe attenuation is approximately 80 dB.
  • the time domain waveform and the frequency domain waveform of the Hann window obtained by symmetric sampling and periodic sampling are respectively shown in FIG.
  • n in the above formula only represents a function variable in a general expression.
  • the Hanning envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 27 is the sampling in the frequency domain, and the right graph is the time domain.
  • the normalization function on The rest of the modulation and demodulation method steps are similar and will not be described again.
  • the transfer envelope waveform is shown in Figure 28.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the method for receiving the signal is similar to the method of the previous embodiment, and details are not described herein again.
  • the application will be further described below with the initial envelope waveform as the Kaiser envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • a Kaiser envelope waveform h(t) of the transmitted signal is generated according to the design parameters.
  • the beta increases, the start of the time domain waveform gradually approaches 0, and the waveform becomes smoother. The faster the sidelobe attenuation of the frequency domain waveform, the better the performance after superposition in the later steps.
  • the time domain waveform and the frequency domain waveform of the Kaiser window when beta is 0.5, 2, and 5 are shown in FIG. 30, respectively.
  • I 0 ( ⁇ ) is the first type of deformed zero-order Bessel function
  • ⁇ is the shape parameter of the window function, which is determined by:
  • ⁇ is the difference (dB) between the main lobe value and the side lobe value of the Kaiser window function.
  • the Kaiser envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 30 is the sampling in the frequency domain, and the right graph is the time domain.
  • the normalization function on The rest of the modulation and demodulation method steps are similar and will not be described again.
  • the transfer envelope waveform is shown in Figure 31.
  • 0 in the input digital signal sequence can be converted to +A, and 1 can be converted to -A to obtain a sequence of positive and negative symbols.
  • the input ⁇ 0, 1 ⁇ bit sequence is BPSK-modulated into a ⁇ +1, -1 ⁇ symbol sequence.
  • the application will be further described below with the initial envelope waveform as the Hamming envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • a Hamming envelope waveform h(t) of a transmission signal is generated based on design parameters.
  • the Hamming window in the time domain waveform starts at approximately 0 (0.08) and the frequency domain sidelobe attenuation is approximately 50 dB.
  • the time domain waveform and the frequency domain waveform of the Hamming window obtained by symmetric sampling and periodic sampling are respectively shown in FIG.
  • n in the above formula only represents the function variable in the formula.
  • the Hamming envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 33 is the sampling in the frequency domain, and the right graph is the time domain.
  • the normalization function on The rest of the modulation and demodulation method steps are similar and will not be described again.
  • the transfer envelope waveform is shown in Figure 34.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is passed through BPSK. The modulation is converted into a sequence of ⁇ +1, -1 ⁇ symbols.
  • the application will be further described below with the initial envelope waveform as the Bartlett-Hanning envelope waveform.
  • the input symbol x i ⁇ +1 +1 -1 -1 -1 +1 -1 +1 ⁇ is taken as an example to illustrate the signal transmission and reception of OvTDM. process.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • the Bartlett-Hanning envelope waveform h(t) of the transmitted signal is generated according to the design parameters.
  • the Bartlett-Hanning window in the time domain waveform starts from 0 o'clock and the frequency domain sidelobe attenuation is nearly 40 dB.
  • n in the above formula only represents the function variable in the formula.
  • the Bartlett-Hanning envelope waveform is a function waveform in the frequency domain, that is, the left image in FIG. 36 is the sampling in the frequency domain, and the right image. Is the normalization function on the time domain. The rest of the modulation and demodulation method steps are similar and will not be described again.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the application is further described below with the initial envelope waveform as the Blackman envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • a Blackman envelope waveform h(t) that transmits a signal is generated according to design parameters.
  • the Blackman window in the time domain waveform starts from 0 o'clock and the frequency domain sidelobe attenuation is nearly 80 dB.
  • ⁇ (n) 0.42-0.5cos(2 ⁇ n/(N-1))+0.08cos(4 ⁇ n/(N-1))
  • the time domain waveform and the frequency domain waveform of the Blackman window obtained by symmetric sampling and periodic sampling are respectively shown in FIG.
  • the Blackman envelope waveform is a function waveform in the frequency domain, that is, the left image in FIG. 39 is the sampling in the frequency domain, and the right image is the time domain.
  • the normalization function on The rest of the modulation and demodulation method steps are similar and will not be described again.
  • the transfer envelope waveform is shown in Figure 40.
  • 0 in the input digital signal sequence can be converted to +A, 1 is converted to -A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the application will be further described below with the initial envelope waveform as the Berman envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • a Bohman envelope waveform h(t) of a transmission signal is generated based on design parameters.
  • the Bohman window in the time domain waveform starts from 0 o'clock and the out-of-band attenuation in the frequency domain is nearly 60 dB.
  • Bohman window function (symmetric function)
  • ⁇ (x) (1-
  • the Berman envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 42 is the sampling in the frequency domain, and the right graph is the time domain.
  • the normalization function on The rest of the modulation and demodulation method steps are similar and will not be described again.
  • the transfer envelope waveform is shown in Figure 43.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the initial envelope waveform as a Flat Top envelope waveform.
  • the input symbol x i ⁇ +1 +1 -1 -1 -1 +1 -1 +1 ⁇ is taken as an example to illustrate the signal transmission and reception of OvTDM. process.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • a flat top envelope waveform h(t) of the transmission signal is generated according to the design parameters.
  • the flat top window in the time domain waveform starts from a point of approximately 0 (-0.0004), and the frequency domain sidelobe attenuation is nearly 100 dB.
  • the time domain waveform and the frequency domain waveform of the Flat Top window obtained by symmetric sampling and periodic sampling are respectively shown in FIG.
  • ⁇ (n) a 0 -a 1 cos(2 ⁇ n/N)+a 2 cos(4 ⁇ n/N)-a 3 cos(6 ⁇ n/N)+a 4 cos(8 ⁇ n/N)
  • n in the above formula only represents the function variable in the formula.
  • the flat top envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 45 is the sampling in the frequency domain, and the right graph is the time domain.
  • the normalization function on The rest of the modulation and demodulation method steps are similar and will not be described again.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the present application will be further described below with the initial envelope waveform as the Nuttall envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • the Nuttall window in the time domain waveform starts from an approximate 0 (0.0004) point, and the frequency domain sidelobe attenuation is nearly 100 dB.
  • the time domain waveform and the frequency domain waveform of the Nuttall window obtained by symmetric sampling and periodic sampling are respectively shown in FIG.
  • ⁇ (n) a 0 -a 1 cos(2 ⁇ n/(N-1))+a 2 cos(4 ⁇ n/(N-1))-a 3 cos(6 ⁇ n/(N-1))
  • ⁇ (n) a 0 -a 1 cos(2 ⁇ n/N)+a 2 cos(4 ⁇ n/N)-a 3 cos(6 ⁇ n/N)
  • n 0, 1, 2, 3, ..., N-1. It should be noted that n in the above formula only represents the function variable in the formula.
  • the Nator envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 48 is the sampling in the frequency domain, and the right graph is the time graph.
  • the normalization function on the domain The rest of the modulation and demodulation method steps are similar and will not be described again.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the initial envelope waveform as a triangular envelope waveform.
  • the input symbol x i ⁇ +1 +1 -1 -1 -1 +1 -1 +1 ⁇ is taken as an example to illustrate the signal transmission and reception of OvTDM. process.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • a triangular (Triangular) envelope waveform h(t) of the transmitted signal is generated according to the design parameters.
  • the Triangular window in the time domain waveform starts at 0 and the frequency domain sidelobe attenuation is nearly 30 dB.
  • Triangular window function For a Triangular window function, it can be expressed by the following formula:
  • n in the above formula only represents the function variable in the formula.
  • the triangular envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 51 is the sampling in the frequency domain, and the right graph is in the time domain. Normalized function. The rest of the modulation and demodulation method steps are similar and will not be described again.
  • the transfer envelope waveform is shown in Figure 52.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the application will be further described below with the initial envelope waveform as the Parzen envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • a Parzen envelope waveform h(t) of a transmission signal is generated based on design parameters.
  • the Parzen window in the time domain waveform starts from 0 o'clock and has an attenuation of nearly 60 dB in the frequency domain.
  • the Barson envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 54 is the sampling in the frequency domain, and the right graph is the time domain.
  • the normalization function on The rest of the modulation and demodulation method steps are similar and will not be described again.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the initial envelope waveform as the Tukey envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • the Tukey envelope waveform h(t) of the transmitted signal is generated according to the design parameters.
  • R is the ratio of the cone to the constant value, and the value is 0 to 1.
  • the Tukey window will evolve into other ordinary windows.
  • the starting point of the time domain waveform starts from 0.
  • the tapered area becomes more and more, and the waveform becomes more and more smooth.
  • the side-valve attenuation of the frequency domain waveform becomes faster and faster, so after superposition Better performance.
  • Tukey window function For the Tukey window function, it can be expressed by the following formula:
  • ⁇ in the formula is the above R value. It should be noted that x in the above formula only represents the function variable in the formula.
  • the graph envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 57 is the sampling in the frequency domain, and the right graph is the time domain.
  • the normalization function on The rest of the modulation and demodulation method steps are similar and will not be described again.
  • the transfer envelope waveform is shown in Figure 58.
  • 0 in the input digital signal sequence can be converted to +A, and 1 can be converted to -A to obtain a sequence of positive and negative symbols.
  • the input ⁇ 0, 1 ⁇ bit sequence is BPSK-modulated into a ⁇ +1, -1 ⁇ symbol sequence.
  • the application will be further described below with the initial envelope waveform as the Taylor envelope waveform.
  • the length of the input symbol refers to the length of the signal transmitted by one frame.
  • the signal generation process includes the following steps:
  • the Taylor envelope waveform h(t) of the transmitted signal is generated according to the design parameters.
  • nbar affects the starting position of the time domain waveform
  • sll affects the frequency domain sidelobe attenuation value
  • the Taylor envelope waveform is a function waveform in the frequency domain, that is, the left graph in FIG. 60 is the sampling in the frequency domain, and the right graph is in the time domain. Normalized function. The rest of the modulation and demodulation method steps are similar and will not be described again.
  • 0, 1 in the input digital signal sequence can be converted to ⁇ A, and A is a non-zero arbitrary number to obtain a sequence of positive and negative symbols.
  • A is 1, the input ⁇ 0, 1 ⁇ bit sequence is converted into a ⁇ +1, -1 ⁇ symbol sequence by BPSK modulation.
  • the overlapping time division/frequency division multiplexing modulation method, device and system provided by the invention smoothes the initial envelope waveform in the time domain (or frequency domain), so that the superposed waveform is smooth, and the transmission power of the system grows linearly and slowly. Indirectly improves spectrum utilization and transmission rate.
  • the overlapping time division multiplexing modulation method, device and system can be applied to wireless communication systems such as mobile communication, satellite communication, microwave line-of-sight communication, scattering communication, atmospheric light communication, infrared communication, underwater acoustic communication, etc., and can be applied to large capacity. Wireless transmission can also be applied to small-capacity lightweight radio systems.

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Abstract

本发明涉及一种重叠复用调制方法、装置和系统，根据设计参数生成一个时域或频域内波形平滑的初始包络波形；根据重叠复用次数将所述初始包络波形在时域或频域上按预定的频谱间隔进行移位，得到各子载波包络波形；将输入的数字信号序列转换成正负符号序列；将所述正负符号序列中的符号与各自对应的子载波包络波形相乘，得到各子载波的调制包络波形；将所述各子载波的调制包络波形在时域或频域上进行叠加，得到时域或频域上的复调制包络波形；将所述时域或频域上的复调制包络波形进行变换，得到时域或频域上的复调制包络波形。所得到的复调制包络波形，在时域或频域上波形平滑，时域或频域上能量集中且持续时间短，因此频谱利用和信号传输速率高，传输功率和误码率低。

Description

一种重叠复用调制方法、装置和系统 技术领域
本发明涉及通信领域，具体涉及一种重叠复用调制方法、装置及重叠复用系统。
背景技术
时间分割(以下简称时分)复用(TDM：Time Division Multiplexing)是一种在数字通信中让多个占据较窄时间持续期的信号符号共享一个较宽时间持续期的技术。而频分复用FDM(Frequency Division Multiplexing)是一种让多个占据较窄带宽的信号共享一个较宽带宽的技术。各被利用的信号带宽分别为B1，B2，B3，B4，….，当然它们也可以占据相同带宽，△B为最小保护带宽，实际保护带宽可以宽裕一些。△B应大于所使用的解复用滤波器的过渡带宽加上系统的最大频率漂移及信道的最大频率扩散量。这是最常见的频分复用技术，现有的绝大多数的广播系统、通信系统和雷达系统等都采用的是这种技术。这种技术的最大特点是被利用的信号频谱之间是相互隔离的，不会存在相互干扰。
如图1A所示为常规的时分复用技术的示意图；图1A中各被复用信号符号的时间持续期(工程上称之为时隙宽度)分别为T1，T2，T3，T4，...，在工程上通常让它们占据相同的时隙宽度，ΔT为最小保护时隙，实际保护时隙宽度应该宽裕一些。ΔT应大于所使用解复用门电路的过渡时间宽度加上系统的最大时间抖动量。这是最常见的时分复用技术。现有绝大多数的多路数字广播系统、多路数字通信等系统采用的都是这种技术。
而图1B则对应于频分复用技术的示意图。各被利用的信号带宽分别为B1，B2，B3，B4，….，当然它们也可以占据相同带宽，△B为最小保护带宽，实际保护带宽可以宽裕一些。△B应大于所使用的解复用滤波器的过渡带宽加上系统的最大频率漂移及信道的最大频率扩散量。这是最常见的频分复用技术，现有的绝大多数的广播系统、通信系统和雷达系统等都采用的是这种技术。这种技术的最大特点是被利用的信号频谱之间是相互隔离的，不会存在相互干扰。
这种技术应用于数字通信时的最大特点是被复用信号符号之间在时间上是完全相互隔离的，决不会存在相互干扰，对被复用的信号符号没有任何限制，各个信号的符号持续期(时隙宽度)可以有不同的宽度，也能适用于不同的通信体制，只要它们的时隙相互不重叠交叉就可以了，因此使用最为广泛。但是这种复用，复用本身对改善系统的频谱效率毫无作用。
所以，传统的观点是相邻信道之间在时域上不重叠，以避免相邻信道之间产生干扰，但这种技术制约了频谱效率的提高。现有技术的时分复用技术的观点是各信道之间不但不需要相互隔离，而且可以有很强的相互重叠，如图2A所示， 现有技术将信道之间的重叠视为一种新的编码约束关系，并根据该约束关系提出了相应的调制和解调技术，因此称之为重叠时分复用(OvTDM：Overlapped Time Division Multiplexing)，这种技术使得频谱效率随重叠次数K成比例的增加；而在频域上则对应为重叠频分复用(Overlapped Frequency Division Multiplexing)，其对应于图2B所示。
理论上，当采用重叠时分复用技术或重叠频分复用进行数据传输时，重叠次数K可无限地增加，因此频谱效率也可无限地增加，但在实验室研究阶段却发现随着重叠次数K的增加，虽然频谱效率得到增加，但是传输功率随之也增长，而传输功率的增长反过来在一定程度上也限制了重叠次数K的增加，从而也限制了频谱效率的增加。
发明内容
根据本申请的第一方面，本申请提供了一种重叠时分复用调制方法,包括：
根据设计参数生成在时域内波形平滑的初始包络波形；
根据重叠复用次数将初始包络波形在时域内按预定的移位间隔进行移位，以得到各个时刻发送信号的偏移包络波形；
将输入的数字信号序列转换成正负符号序列；
将转换后的正负符号序列与偏移后各个时刻发送信号的偏移包络波形相乘，以得到各个时刻的调制包络波形；
将各个时刻的调制包络波形在时域上进行叠加，以得到携带输出信号序列的复调制包络波形。
根据本申请的第二方面，本申请还提供了一种重叠时分复用调制装置,包括：
波形生成模块，用于根据设计参数生成在时域内波形平滑的初始包络波形；
移位模块，用于根据重叠复用次数将初始包络波形在时域内按预定的移位间隔进行移位，以得到各个时刻发送信号的偏移包络波形；
调制模块，用于将输入的数字信号序列转换成正负符号序列；
乘法模块，用于将输入的的正负符号序列与偏移后各个时刻发送信号的偏移包络波形相乘，以得到各个时刻的调制包络波形；
叠加模块，用于将各个时刻的调制包络波形在时域上进行叠加，以得到携带输出信号序列的复调制包络波形。
本发明的第三个方面，提供提供一种重叠频分复用调制方法，包括以下步骤：
根据设计参数生成一个频域内波形平滑的初始包络波形；
根据重叠复用次数将所述初始包络波形在频域上按预定的频谱间隔进行移位，得到各子载波包络波形；
将输入的数字信号序列转换成正负符号序列；
将所述正负符号序列中的符号与各自对应的子载波包络波形相乘，得到各子 载波的调制包络波形；
将所述各子载波的调制包络波形在频域上进行叠加，得到频域上的复调制包络波形；
将所述频域上的复调制包络波形进行变换，得到时域上的复调制包络波形。
本发明的第四个方面，提供一种重叠频分复用调制装置，包括：
波形生成模块，用于生成一个频域内波形平滑的初始包络波形；
移位模块，用于根据重叠复用次数将所述初始包络波形在频域上按预定的频谱间隔进行移位，得到各子载波包络波形；
转换模块，用于将输入的数字信号序列转换成正负符号序列；
乘法模块，用于将所述正负符号序列中的符号与各自对应的子载波包络波形相乘，得到各子载波的调制包络波形；
叠加模块，用于将所述各子载波的调制包络波形在频域上进行叠加，得到频域上的复调制包络波形；
变换模块，用于将所述频域上的复调制包络波形进行变换，得到时域上的复调制包络波形。
本发明提供的重叠时分复用调制方法、装置及系统中，由于初始包络波形的时域波形较平滑，频域带宽较窄，叠加后的波形较平滑且限定在较窄带宽内，因此提高了系统的频谱利用率和传输速率，降低系统的误码率；而重叠频分复用调制方法、装置及系统，由于生成的初始包络波形在频域内波形平滑，相应地，其在时域内能量集中且持续时间较短，因此经过其调制形成的复调调包络波形在时域能量集中且持续时间较短，因此其频谱利用率高，信号传输速率也高，并且只需要较低的传输功率，被解调时具有较低的误码率。
附图说明
图1A为常规的时分复用技术的示意图；
图1B为常规的频分复用技术的示意图；
图2A为重叠时分复用原理示意图；
图2B为重叠频分复用原理示意图；
图3A为本发明一种实施例中重叠时分复用系统的结构示意图；
图3B为本发明一种实施例中重叠频分复用系统的结构示意图；图4A为本发明一种实施例中重叠时分复用调制装置的结构示意图；
图4B为本发明一种实施例中重叠频分复用调制装置的结构示意图；
图5A为本发明一种实施例中重叠时分复用调制装置的硬件结构示意图；
图5B为本发明一种实施例中重叠频分复用调制装置的硬件结构示意图；
图6为本发明一种实施例中接收机预处理装置的结构示意图；
图7为本发明一种实施例中接收机序列检测装置的结构示意图；
图8为本发明一种实施例中切比雪夫包络波形的时域波形和频域波形图；
图9为本发明一种实施例中切比雪夫窗经移位后各个时刻的包络波形图；
图10为本发明一种实施例中采用切比雪夫包络波形时待发送波形的叠加示意图；
图11为K路波形复用的原理示意图；
图12为K路波形的符号叠加过程原理示意图；
图13为K＝3时重叠时分复用系统的输入-输出关系树图；
图14为节点状态转移关系图；
图15为矩形波的时域和频域波形图；
图16为包络波形选择矩形波包络波形时生成的各个信号和叠加后的波形图；
图17为本发明一种实施例中布莱克曼一阶导数包络波形的时域波形和频域波形图；
图18为本发明一种实施例中布莱克曼-哈里斯一阶导数包络波形的时域波形和频域波形图；
图19为本发明一种实施例中采用布莱克曼一阶导数包络波形时待发送波形的叠加示意图；
图20为本发明一种实施例中采用布莱克曼-哈里斯一阶导数包络波形时待发送波形的叠加示意图。
图21为本发明一种实施例中巴特莱特包络波形的时域波形和频域波形图；
图22为本发明一种实施例中巴特莱特窗经移位后各个时刻的包络波形图；
图23为本发明一种实施例中采用巴特莱特包络波形时待发送波形的叠加示意图；
图24为本发明一种实施例中高斯包络波形的时域波形和频域波形图；
图25为本发明一种实施例中高斯窗经移位后各个时刻的包络波形图；
图26为本发明一种实施例中采用高斯包络波形时待发送波形的叠加示意图；
图27为本发明一种实施例中汉宁包络波形的时域波形和频域波形图；
图28为本发明一种实施例中汉宁窗经移位后各个时刻的包络波形图；
图29为本发明一种实施例中采用汉宁包络波形时待发送波形的叠加示意图；
图30为本发明一种实施例中凯塞包络波形的时域波形和频域波形图；
图31为本发明一种实施例中凯塞窗经移位后各个时刻的包络波形图；
图32A为本发明一种实施例中beta＝0.5时采用凯塞包络波形时待发送波形的叠加示意图；
图32B为本发明一种实施例中beta＝2时采用凯塞包络波形时待发送波形的叠加示意图；
图32C为本发明一种实施例中beta＝5时采用凯塞包络波形时待发送波形的叠 加示意图；
图33为本发明一种实施例中汉明包络波形的时域波形和频域波形图；
图34为本发明一种实施例中汉明窗经移位后各个时刻的包络波形图；
图35为本发明一种实施例中采用汉明包络波形时待发送波形的叠加示意图；
图36为本发明一种实施例中巴特莱特-汉宁包络波形的时域波形和频域波形图；
图37为本发明一种实施例中巴特莱特-汉宁窗经移位后各个时刻的包络波形图；
图38为本发明一种实施例中采用巴特莱特-汉宁包络波形时待发送波形的叠加示意图；
图39为本发明一种实施例中布莱克曼包络波形的时域波形和频域波形图；
图40为本发明一种实施例中布莱克曼窗经移位后各个时刻的包络波形图；
图41为本发明一种实施例中采用布莱克曼包络波形时待发送波形的叠加示意图；
图42为本发明一种实施例中伯曼包络波形的时域波形和频域波形图；
图43为本发明一种实施例中伯曼窗经移位后各个时刻的包络波形图；
图44为本发明一种实施例中采用伯曼包络波形时待发送波形的叠加示意图；
图45为本发明一种实施例中平顶包络波形的时域波形和频域波形图；
图46为本发明一种实施例中平顶窗经移位后各个时刻的包络波形图；
图47为本发明一种实施例中采用平顶包络波形时待发送波形的叠加示意图；
图48为本发明一种实施例中纳托尔包络波形的时域波形和频域波形图；
图49为本发明一种实施例中纳托尔窗经移位后各个时刻的包络波形图；
图50为本发明一种实施例中采用纳托尔包络波形时待发送波形的叠加示意图；
图51为本发明一种实施例中三角形包络波形的时域波形和频域波形图；
图52为本发明一种实施例中三角形窗经移位后各个时刻的包络波形图；
图53为本发明一种实施例中采用三角形包络波形时待发送波形的叠加示意图；
图54为本发明一种实施例中巴尔森包络波形的时域波形和频域波形图；
图55为本发明一种实施例中巴尔森窗经移位后各个时刻的包络波形图；
图56为本发明一种实施例中采用巴尔森包络波形时待发送波形的叠加示意图；
图57为本发明一种实施例中图基包络波形的时域波形和频域波形图；
图58为本发明一种实施例中图基窗经移位后各个时刻的包络波形图；
图59A为本发明一种实施例中R＝0.1时采用图基包络波形时待发送波形的叠加示意图；
图59B为本发明一种实施例中R＝0.5时采用图基包络波形时待发送波形的叠加示意图；
图59C为本发明一种实施例中R＝0.9时采用图基包络波形时待发送波形的叠加示意图；
图60为本发明一种实施例中泰勒包络波形的时域波形和频域波形图；
图61为本发明一种实施例中泰勒窗经移位后各个时刻的包络波形图；
图62A为本发明一种实施例中nbar＝4，sll＝-30时采用泰勒包络波形时待发送波形的叠加示意图；
图62B为本发明一种实施例中nbar＝6，sll＝-50时采用泰勒包络波形时待发送波形的叠加示意图；
图62C为本发明一种实施例中nbar＝8，sll＝-80时采用泰勒包络波形时待发送波形的叠加示意图；
具体实施方式
下面通过具体实施方式结合附图对本发明作进一步详细说明。
在对重叠时分复用技术研究中，发明人发现传输功率的增长主要跟被复用信号(即调制窗函数)的频谱有关，并非如理论上所设想的对复用信号频谱的形状、带宽没有任何要求。虽然现有技术中存在很多窗函数，理论上可自由采用各种窗函数对传输符号进行调制，但由于矩形窗相较于其它窗函数在产生、设计和应用上更容易、成本更低，因此目前在进行信号调制时优先采用矩形窗，而矩形波的频谱带宽较宽，复用波形系统性能很差，导致所需的传输功率和误码率都很高。
基于上述发现，在本发明实施例中，在应用重叠时分复用技术时采用一种优于矩形波的窗函数对输入的数字信号序列进行调制。
请参考图3A，重叠时分复用系统包括信号发射机A01和接收机A02。
发射机A01包括重叠时分复用调制装置301和发射装置302。重叠时分复用调制装置301用于生成携带输出信号序列的复调制包络波形；发射装置102用于将该复调制包络波形发射到接收机A02。
接收机A02包括接收装置303和序列检测装置305。接收装置303用于接收发射装置302发射的复调制包络波形；序列检测装置305用于对接收的复调制包络波形进行时域内的数据序列检测，以进行判决输出。
优选的，接收机A02还包括设置在接收装置303和序列检测装置305之间的预处理装置304，用于辅助形成每一帧内的同步接收数字信号序列。
在发射机A01中，输入的数字信号序列通过重叠时分复用调制装置301形成多个符号在时域上相互重叠的发射信号，再由发射装置302将该发射信号发射到接收机A02。接收机A02的接收装置303接收发射装置302发射的信号，经过预 处理装置304形成适合序列检测装置305进行检测接收的数字信号，序列检测装置305对接收信号进行时域内的数据序列检测，从而输出判决。
而对应的发射机和接收机在重叠频分复用系统中的结构则如图3B所示，发射机B1包括重叠频分复用调制装置310和发射装置320，其中，重叠频分复用调制装置310用于调制生成携带输出信号序列的复调制包络波形，发射装置320用于将上述复调制包络波形发射到接收机B2。接收机B2包括接收装置330和重叠频分复用解调装置340，其中，接收装置330用于接收发射装置320发送的上述复调制包络波形，重叠频分复用解调装置340用于对接收的复调制包络波形进行解调译码。
请参考图4A，图3A中的重叠时分复用调制装置301(OvTDM调制装置)包括波形生成模块301、移位模块302、乘法模块303和叠加模块304。
波形生成模块301用于根据设计参数生成在时域内波形平滑的初始包络波形。
移位模块302用于根据重叠复用次数将初始包络波形在时域内按预定的移位间隔进行移位，以得到各个时刻发送信号的偏移包络波形。
调制模块305用于将输入的数字信号序列转换成正负符号序列。
乘法模块303用于将转换后的正负符号序列与偏移后各个时刻发送信号的偏移包络波形相乘，以得到各个时刻的调制包络波形。
叠加模块304用于将各个时刻的调制包络波形在时域上进行叠加，以得到携带输出信号序列的复调制包络波形。
而在图4B所述的重叠频分复用调制装置中，重叠频分复用调制装置310包括波形生成模块411、移位模块412、转换模块413、乘法模块414、叠加模块415和变换模块416。
波形生成模块411用于根据设计参数生成一个频域内波形平滑的初始包络波形。在一实施例中，设计参数至少包括初始包络波形的带宽宽度。
移位模块412用于根据重叠复用次数将初始包络波形在频域上按预定的频谱间隔进行移位，得到各子载波包络波形。在一实施例中，频谱间隔为子载波频谱间隔△B，其中子载波频谱间隔△B＝B/K，B为初始包络波形的带宽，K为重叠复用次数。
转换模块413用于将输入的数字信号序列转换成正负符号序列。在一实施例中，转换模块413将输入的数字信号序列转换成正负符号序列具体为：将输入的数字信号序列中的0转换为+A，数字信号序列中的1转换为-A，以形成正负符号序列并输出。例如，取A＝1，在一具体实施例中，转换模块413采用BPSK调制方式，将输入的{0，1}比特序列经过调制转换成{+1，-1}的符号序列。
乘法模块414用于将上述正负符号序列中的符号与各自对应的子载波包络波形相乘，得到各子载波的调制包络波形。
叠加模块415用于将上述各子载波的调制包络波形在频域上进行叠加，得到 频域上的复调制包络波形。
变换模块416用于将上述频域上的复调制包络波形变换到到时域上的复调制包络波形。在一具体实施例中，变换模块416可以采用傅氏反变换，将上述频域上的复调制包络波形变换成时域上的复调制包络波形。
上述调制生成的复调制包络波形携带有与转换得到的正负符号序列对应的输出信号序列，此输出信号序列由各频谱间隔的输出信号组成，各频谱间隔的输出信号为各频谱间隔内的调制包络波形的运算值叠加后的结果，当调制包络波形由正符号与子载波包络波形相乘得到时，其运算值为+1，由负符号与子载波包络波形相乘得到时，其运算值为-1。
回到图4A，下面结合重叠时分复用调制方法，对重叠时分复用调制装置101做进一步说明，重叠时分复用调制方法包括下面步骤：
(1)波形生成模块401根据设计参数生成在时域内波形平滑的初始包络波形h(t)。
在生成初始包络波形时，可以通过用户输入设计参数，以实现在实际系统中根据系统性能指标灵活配置。
在某些实施例中，当初始包络波形的旁瓣衰减已经确定时，设计参数包括初始包络波形的窗长度L，例如当初始包络波形为巴特莱特包络波形时。
在某些实施例中，设计参数包括初始包络波形的窗长度L和旁瓣衰减r，例如当初始包络波形为切比雪夫包络波形时。
当然，当初始包络波形为其他形式时，可以根据相应初始包络波形的特点确定设计参数。
(2)移位模块402根据重叠复用次数K将初始包络波形在时域内按预定的移位间隔进行移位，以得到各个时刻发送信号的偏移包络波形h(t-i*△T)。
其中，移位间隔为时间间隔△T，时间间隔△T为：△T＝L/K。
另外，还需要保证△T不小于系统采样率的倒数。
i的取值与输入符号长度N有关，且i取0到N-1的整数。例如，当N＝8时，i取0至7的整数。
(3)调制模块405将输入的数字信号序列转换成正负符号序列。
具体的，调制模块405将输入的数字信号序列中的0转换为+A，1转换为-A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK(Binary Phase Shift Keying，移相键控)调制转换成{+1、-1}符号序列。
(4)乘法模块403将转换后的正负符号序列xi与偏移后各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，以得到各个时刻的调制包络波形xih(t-i*△T)。
(5)叠加模块404将各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
发送的信号可以如下表示：
Figure PCTCN2016111405-appb-000001
由于初始包络波形的时域波形较平滑，频域带宽较窄，叠加后的波形较平滑且限定在较窄带宽内，因此提高了系统的频谱利用率和传输速率，降低系统的误码率。
请参考图5A，具体的，重叠时分复用调制装置301可通过下面硬件单元实现。重叠时分复用调制装置301包括数字波形发生器501、移位寄存器502、调制器503、乘法器504及加法器505。
首先由数字波形发生器501以数字方式形成第一个初始包络波形的同相波形，该初始包络波形在时域内平滑；再由移位寄存器502将数字波形发生器401产生的第一个初始包络波形的同相波形进行移位，以产生各个时刻发送信号的偏移包络波形；接着，调制器503将输入的数字信号序列转换成正负符号序列，乘法器504则将转换后的正负符号序列与偏移后各个时刻发送信号的偏移包络波形相乘，以得到各个时刻的调制包络波形；最后由加法器505将各个时刻的调制包络波形在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，形成发射信号。
而对于重叠频分复用系统的解调装置，其结构如图5B所示，重叠频分复用解调装置包括频谱模块51、频率分段模块52、卷积编码模块53和数据检测模块54。
频谱模块51用于将上述时间域上的接收符号序列进行变换以形成接收信号频谱。在一具体实施例中，频谱模块51采用傅氏变换，将上述时间域上的接收符号序列变换成接收信号频谱。
频率分段模块52用于将接收信号频谱在频域以子载波频谱间隔△B进行分段得到接收信号分段频谱。
卷积编码模块53用于对各子载波频谱间隔△B内的接收信号分段频谱进行卷积编码，得到接收信号频谱与发射机中经输入的数字信号序列转换成的正负符号序列之间的一一对应关系。
数据检测模块54用于根据上述一一对应关系，检测出上述正负符号序列。
请参考图6，为本发明实施例，重叠时分系统中接收机A02的预处理装置的框图。
预处理装置包括同步器501、信道估计器502和数字化处理器503。其中同步器501对接收信号在接收机内形成符号时间同步；接着信道估计器502对信道参数进行估计；数字化处理器503对每一帧内的接收信号进行数字化处理，从而形成适合序列检测装置进行序列检测接收的数字信号序列。
请参考图7，为本发明，重叠时分系统实施例中接收机A02的序列检测装置202的框图。
序列检测装置包括分析单元存储器701、比较器702及多个保留路径存储器703和欧氏距离存储器704或加权欧氏距离存储器(图中未示出)。在检测过程中，分析单元存储器701做出重叠时分复用系统的复数卷积编码模型及格状图，并列出重叠时分复用系统的全部状态，并存储；而比较器702根据分析单元存储器701中的格状图，搜索出与接收数字信号最小欧氏距离或加权最小欧氏距离的路径；而保留路径存储器703和欧氏距离存储器704或加权欧氏距离存储器则分别用于存储比较器702输出的保留路径和欧氏距离或加权欧氏距离。其中，保留路径存储器703和欧氏距离存储器704或加权欧氏距离存储器需要为每一个稳定状态各准备一个。保留路径存储器703长度可以优选为4K～5K。欧氏距离存储器604或加权欧氏距离存储器优选为只存储相对距离。
重叠时分/频分复用调制方法、装置及系统中采用的初始包络波形可以包括切比雪夫(Chebyshev)、高斯(Gaussian)、汉明(Hamming)、汉宁(Hann)、布莱克曼(Blackman)、布莱克曼-哈里斯(Blackman-Harris)、巴特莱特(Bartlett)、巴特莱特-汉宁(Bartlett-Hanning)、伯曼(Bohman)、平顶(Flat Top)、纳托尔(Nuttall)、巴尔森(Parzen)、泰勒(Taylor)、图基(Tukey)、凯塞(Kaiser)、三角形(Triangular)等复用波形及以其为基础的演变波形。
实施例一
本实施例以初始包络波形为切比雪夫包络波形，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
请参考图5，信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的切比雪夫包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，旁瓣衰减r＝80dB，其时域波形和频域波形如附图8所示。从图8中可以看出，时域波形中切比雪夫窗是由近似0点开始，频域旁瓣衰减为80dB。
(2)将(1)所设计的切比雪夫包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图9所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0转换为+A，1转换为-A，，A取 值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图10所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。发射信号波形图如图10中的实线波形所示。
发送的信号可以表示为：
Figure PCTCN2016111405-appb-000002
具体的，输出信号序列通过下面方式确定：
当调制包络波形由正符号与该时刻包络波形相乘得到时，令该调制包络波形的运算值为+A，当调制包络波形由负符号与该时刻包络波形相乘得到时，令该调制包络波形的运算值为-A。对于每个移位间隔，将位于该移位间隔内的调制包络波形的运算值叠加，得出该移位间隔的输出信号，从而形成输出信号序列。
故，本实施例中，A取值为1时，叠加后的输出符号(输出信号序列)即为：s(t)＝{+1 +2 +1 -1 -3 -1 -1 +1}。
请参考图11，为K路波形复用的原理示意图，其呈平行四边形形状。其中，每一行表示一个所要发送的符号xi与相应时刻的包络波形h(t-i*△T)相乘后得到的待发送信号波形xih(t-i*△T)。a0～ak-1表示对每个窗函数波形(包络波形)进行K次分段得到的每部分的系数值，具体为关于幅度值的系数。
由于将输入的数字信号序列转换成正负符号序列时，将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数以得到正负符号序列。例如，A取值为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列，以得到正负符号序列。所以图12所示即为K路波形的符号叠加过程原理示意图。图12叠加过程中，第1行左边3个数表示第1个输入符号+1，第2行左边3个数表示第2个输入符号+1，第3行左边3个数表示第3个输入符号-1，第1行中间3个数表示第4个输入符号-1，第2行中间3个数表示第5个输入符号-1，第3行中间3个数表示第6个输入符号+1，第1行右边3个数表示第7个输入符号-1，第2行右边3个数表示第8个输入符号+1。因此，三个波形叠加后，得到的输出符号为{+1 +2 +1 -1 -3 -1 -1 +1}。
当然，如果输入符号的长度为其他数值时，可以按照图11和图12所示的方 式进行叠加，以得到输出符号。
由于切比雪夫包络波形在时域由0(0.0028，接近0)开始，具有平滑的波形，因此叠加后的波形较平滑，频域带宽较窄，使得叠加后的波形频谱效率较高，发送信号所需的传输功率较低。又因为切比雪夫包络波形可以自行设计旁瓣衰减，因此在实际系统中可根据系统性能指标灵活配置。
请参考图6和图7，信号接收过程包括下面步骤：
(1)首先对接收信号进行同步，包括载波同步、帧同步、符号时间同步等。
(2)根据取样定理，对每一帧内的接收信号进行数字化处理。
(3)对接收到的波形按照波形发送时间间隔切割。
(4)对接收的信号进行时间域内的数据序列检测，以进行判决输出，即按照一定的译码算法对切割后的波形进行译码。
经过上述(1)～(2)的预处理步骤后，波形切割后得到的接收符号序列为：s(t)＝{+1 +2 +1 -1 -3 -1 -1 +1}，对符号序列根据图7输入-输出关系的树图和图8节点状态转移关系图，进行符号之间的前后比较，得到节点转移路径。
图13中，向上的树枝为+1输入，向下的树枝为-1输入。在第三枝以后该树图就变成重复的了，因为凡是从标记为a的节点辐射出的树枝都有同样的输出，该结论对节点b、c、d也同样适用。它们不外乎是如图14所示的几种可能，从图14中可以看出从节点a只能转移到(经输入+1)节点a及(经输入-1)节点b，同时b只能到(输入+1)c及(输入-1)d，c只能到(输入+1)a及(输入-1)b，d只能到(输入+1)c及(输入-1)d。产生这种现象的原因很简单，因为只有相邻K(具体到本例是3)个符号才会形成相互干扰。所以当第K位数据输入到信道时，最早来的第1位数据已经移出最右边的一个移位单元了。因此信道的输出除了取决于现时刻数据的输入，还决定于前K-1个数据的输入。
本案例中的节点状态转移如图13中的加黑粗线所示，由于s(t)的第一个符号为+1，所以节点转移路径为：+1->a->a->b->d->d->c->b->c，根据此转移关系即可求出输入的符号序列为{+1 +1 -1 -1 -1 +1 -1 +1}。
本实施例中，由于切比雪夫包络波形在时域上较平滑，且旁瓣衰减较快，因此所需的传输功率较低，对波形进行切割时精度更高，接收到的符号序列准确度更好。
请参考图15，为矩形波的时域和频域波形图。当初始包络波形选择矩形波包络波形时，那么根据上述信号生成过程生成的各个信号和叠加后的波形图如图16所示，其中三条不同的虚线表示三个波形图，实线表示叠加后的波形图。
从图16中可以看出，矩形波在时域上由1开始，并且带宽较宽，在频域上旁瓣衰减缓慢，因此时域叠加后的波形不平滑，频域带宽较宽，有效信号和无效信 号难以区分，使得发送和接收信号过程中所需要的传输功率增加，接收信号过程中波形切割的准确率和编解码能力降低。在实际系统中传输速率相同和频谱效率相同的情况下，使用矩形波时所需的传输功率和误码率都很高。
然而本实施例中采用的切比雪夫窗在时域的起点由0(0.0028，接近0)开始，旁瓣衰减较快，信号叠加后的波形平滑，频域带宽较窄，提高了波形切割过程的准确率和编解码过程的纠错能力，降低了信号的传输功率，使得在频谱效率一定时，使用较低的传输功率就能达到较高的传输速率。又因为切比雪夫窗可以自行设计旁瓣衰减，因此在实际系统中可根据系统性能指标灵活配置。
另外，在其他实施例中，初始包络波形还可以选择各种以切比雪夫窗函数演变的函数的包络波形，包括切比雪夫脉冲成型的连乘、各阶导数、各阶导数之和等函数的包络波形，这些包络波形在时域上同样具有波形平滑的特点，因此采用这些包络波形后均可以达到与采用切比雪夫包络波形相近似的效果。
该实施例也可以应用在重叠频分复用系统中，区别仅在于，切比雪夫包络波形为频域上的函数波形，即图8中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
实施例二
本实施例以初始包络波形分别为布莱克曼一阶导数、布莱克曼-哈里斯一阶导数复用波形，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。
同样请参考图5，信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的布莱克曼一阶导数、布莱克曼-哈里斯一阶导数对应的包络波形h(t)。
本实施例中设计参数中的窗长度L＝63，其对应的时域波形和频域波形分别如图17和图18所示。
从图17中可以看出，布莱克曼一阶导数的包络波形在时域中是由近似0点开始，在后半部分幅值变为负数，波形趋近于正弦波，频域旁瓣衰减为40dB左右。
从图18中可以看出，布莱克曼-哈里斯一阶导数的包络波形在时域中是由近似0点开始，在后半部分幅值变为负数，波形趋近于正弦波，频域旁瓣衰减为100dB左右。
具体的，对于布莱克曼窗函数，其可通过下面公式表示：
ω(n)＝0.42-0.5cos(2πn/(N-1))+0.08cos(4πn/(N-1))
其中，N为窗长度，0≤n≤M-1，当N为偶数时，M＝N/2，当N为奇数时，M＝(N+1)/2。
需要说明的是，由于上述公式中，0≤n≤M-1，即得到的波形为前半部分布莱克曼窗，对于后半部分布莱克曼窗的波形(即M≤n≤N-1时)，其与前半部分 的波形以直线n＝M呈轴对称，即将前半部分波形沿直线n＝M水平翻转后即可得到。
具体的，对于布莱克曼-哈里斯窗函数(对称函数)，其可通过下面公式表示：
ω(n)＝a0-a1cos(2πn/(N-1))+a2cos(4πn/(N-1))+a3cos(6πn/(N-1))
对于布莱克曼-哈里斯窗函数(周期函数)，其可通过下面公式表示：
ω(n)＝a0-a1cos2πn/N+a2cos4πn/N+a3cos6πn/N
其中，N为窗长度，0≤n≤N-1，a0＝0.35875，a1＝0.48829，a2＝0.14128，a3＝0.01168。需要说明的是，上述公式中的n仅表示公式中的函数变量。
(2)将(1)所设计的布莱克曼一阶导数、布莱克曼-哈里斯一阶导数包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，A取值为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})中的符号与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形分别如图19、20所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。发射信号波形图分别如图19和图20中的实线波形所示。
发送的信号可以表示为：
Figure PCTCN2016111405-appb-000003
具体的，输出信号序列通过下面方式确定：
当调制包络波形由正符号与该时刻包络波形相乘得到时，令该调制包络波形的运算值为+A，当调制包络波形由负符号与该时刻包络波形相乘得到时，令该调制包络波形的运算值为-A。对于每个移位间隔，将位于该移位间隔内的调制包络波形的运算值叠加，得出该移位间隔的输出信号，从而形成输出信号序列。
故，本实施例中，A取值为1时，叠加后的输出符号(输出信号序列)即为：s(t)＝{+1 +2 +1 -1 -3 -1 -1 +1}。
初始包络波形分别为布莱克曼一阶导数、布莱克曼-哈里斯一阶导数复用波形时，波形复用原理示及符号叠加过程原理与实施例一相同，请参考附图11和图 12。
本实施例中，信号接收过程与实施例一采用切比雪夫包络波形时的信号接收过程相同，因此，本实施例不再赘述。
由于布莱克曼一阶导数、布莱克曼-哈里斯一阶导数复用波形在时域上较平滑，且旁瓣衰减较快，因此所需的传输功率较低，对波形进行切割时精度更高，接收到的符号序列准确度更好。
矩形波的特点是主瓣比较集中，缺点是旁瓣较高，并有负旁瓣，导致变换中带进了高频干扰和泄漏，甚至出现负频谱现象，幅值识别精度最低。布莱克曼一阶导数和布莱克曼-哈里斯一阶导数复用波形的特点是主瓣宽，旁瓣比较低，幅值识别精度最高，有更好的选择性。
以布莱克曼一阶导数、布莱克曼-哈里斯一阶导数为复用波形的OvTDM过程，信号发送过程中，时域波形平滑，频域带宽较窄，发送信号所需的传输功率较低，且频谱利用率和传输速率都较高。接收信号过程中，由于波形在时域较平滑，因此在对波形切割时准确度更高，降低了系统的误码率。系统性能较矩形波得到了很大的改善。
另外，在其他实施例中，初始包络波形还可以选择各种布莱克曼窗原型，或以布莱克曼窗函数演变的其他函数的包络波形，包括布莱克曼脉冲成型的连乘、各阶导数、各阶导数之和等函数的包络波形，采用这些包络波形后均可以达到与采用布莱克曼波形一阶导数相近似的效果。
或者，初始包络波形选择各种布莱克曼-哈里斯窗原型，或以布莱克曼-哈里斯窗函数演变的其他函数的包络波形，包括布莱克曼-哈里斯脉冲成型的连乘、各阶导数、各阶导数之和等函数的包络波形，这些包络波形在时域上同样具有波形平滑的特点，因此采用这些包络波形后均可以达到与采用布莱克曼-哈里斯波形一阶导数相近似的效果。
该实施例也可以应用在重叠频分复用系统中，区别仅在于，布莱克曼包络波形为频域上的函数波形，即图17、18中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
实施例三
本实施例以OvTDM系统中的巴特莱特包络波形进行调制解调。本实施例中初始包络波形为巴特莱特(Bartlett)包络波形或其演变窗函数的包络波形。
下面则以初始包络波形为巴特莱特(Bartlett)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的巴特莱特(Bartlett)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图21所示。从图21中可以看出，时域中巴特莱特(Bartlett)窗是由0开始，频域带外衰减近30dB。
具体的，巴特莱特(Bartlett)窗函数可以通过下面公式表示：
Figure PCTCN2016111405-appb-000004
其中，巴特莱特(Bartlett)窗的窗长度L＝N+1。需要说明的是，上述公式中的n仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，巴特莱特包络波形为频域上的函数波形，即图21中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的巴特莱特(Bartlett)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图22所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0转换为+1，1转换为-1，以得到正负符号序列。例如，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图23所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。发射信号波形图如图23中的实线波形所示。
而信号的接收方法与前面实施例的方法相似，在此不再赘述。
实施例四
下面则以初始包络波形为高斯(Gaussian)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1  -1+1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的高斯(Gaussian)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，Alpha＝2.5，其时域波形和频域波形如附图24所示。从图24中可以看出，时域波形中高斯(Gaussian)窗是由近似0点开始，频域旁瓣衰减近50dB。
具体的，对于高斯(Gaussian)窗函数，其可通过下面公式表示：
Figure PCTCN2016111405-appb-000005
其中，N为窗长度，-N/2≤n≤N/2，α为预设参数，图24中示出了α分别为2.5、1.5、0.5时高斯(Gaussian)窗的时域波形和频域波形。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，高斯包络波形为频域上的函数波形，即图24中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的高斯(Gaussian)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图25所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0转换为+A，1转换为-A，以得到正负符号序列。例如，A取值为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图26所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。发射信号波形图如图26中的实线波形所示。
而信号的接收方法与前面实施例的方法相似，在此不再赘述。
实施例五
本实施例中，初始包络波形为汉宁(Hann)包络波形或其演变窗函数的包络波形。
下面则以初始包络波形为汉宁(Hann)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的汉宁(Hann)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图27所示。从图27中可以看出，时域波形中汉宁(Hann)窗是由0点开始的，频域旁瓣衰减近似为80dB。另外，图27中分别示出了采用symmetric抽样和periodic抽样得到的汉宁(Hann)窗的时域波形和频域波形。
具体的，对于汉宁(Hann)窗函数，其可通过下面公式表示：
ω(n)＝0.5(1-cos(2πn/N))
其中，0≤n≤N，窗长度L＝N+1。需要说明的是，上述公式中的n仅代表一般表达式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，汉宁包络波形为频域上的函数波形，即图27中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的汉宁(Hann)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图28所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图29所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
而信号的接收方法与前面实施例的方法相似，在此不再赘述。
实施例六
下面则以初始包络波形为凯塞(Kaiser)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的凯塞(Kaiser)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，beta分别为0.5、2、5，其时域波形和频域波形如附图30所示。从图30中可以看出，随着beta的增加，时域波形起点逐渐趋近于0，波形越来越平滑；频域波形旁瓣衰减越快，因此后面步骤中叠加后的性能更优。图30中分别示出了beta为0.5、2、5时，凯塞(Kaiser)窗的时域波形和频域波形。
具体的，对于凯塞(Kaiser)窗函数，其可通过下面公式表示：
Figure PCTCN2016111405-appb-000006
其中，I0(β)是第一类变形零阶贝塞尔函数，β是窗函数的形状参数，其由下式确定：
Figure PCTCN2016111405-appb-000007
α为凯塞(Kaiser)窗函数的主瓣值和旁瓣值之间的差值(dB)，改变β的值，可以对产瓣宽度和旁瓣衰减进行自由选择。β值越大，窗函数频谱的旁瓣值就越小，而其主瓣宽度就越宽。需要说明的是，上述公式中的n仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，凯塞包络波形为频域上的函数波形，即图30中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的凯塞(Kaiser)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图31所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0转换为+A，1转换为-A，以得到正负符号序列。例如，取A＝1，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图32(图32A～图32C)所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例七
下面则以初始包络波形为汉明(Hamming)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的汉明(Hamming)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图33所示。从图33中可以看出，时域波形中汉明(Hamming)窗是由近似0(0.08)点开始，频域旁瓣衰减为近50dB。另外，图33中分别示出了采用symmetric抽样和periodic抽样得到的汉明(Hamming)窗的时域波形和频域波形。
具体的，对于汉明(Hamming)窗函数，其可通过下面公式表示：
ω(n)＝0.54-0.46cos(2πn/N)
其中，0≤n≤N，窗长度L＝N+1。需要说明的是，上述公式中的n仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，汉明包络波形为频域上的函数波形，即图33中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的汉明(Hamming)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图34所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK 调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图35所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例八
下面则以初始包络波形为巴特莱特-汉宁(Bartlett-Hanning)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的巴特莱特-汉宁(Bartlett-Hanning)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图36所示。从图36中可以看出，时域波形中巴特莱特-汉宁(Bartlett-Hanning)窗是由0点开始，频域旁瓣衰减为近40dB。
具体的，对于汉明窗函数，其可通过下面公式表示：
ω(n)＝0.62-0.48|n/N-0.5|+0.38cos(2π(n/N-0.5))
其中，0≤n≤N，窗长度L＝N+1。需要说明的是，上述公式中的n仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，巴特莱特-汉宁包络波形为频域上的函数波形，即图36中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的巴特莱特-汉宁(Bartlett-Hanning)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图37所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包 络波形xih(t-i*△T)；形成后的波形如图38所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例九
下面则以初始包络波形为布莱克曼(Blackman)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的布莱克曼(Blackman)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图39所示。从图39中可以看出，时域波形中布莱克曼(Blackman)窗是由0点开始，频域旁瓣衰减为近80dB。
具体的，对于布莱克曼(Blackman)窗函数，其可通过下面公式表示：
ω(n)＝0.42-0.5cos(2πn/(N-1))+0.08cos(4πn/(N-1))
其中，N为窗长度，0≤n≤M-1，当N为偶数时，M＝N/2，当N为奇数时，M＝(N+1)/2。需要说明的是，上述公式中的n仅表示公式中的函数变量。
需要说明的是，由于上述公式中，0≤n≤M-1，即得到的波形为前半部分布莱克曼(Blackman)窗，对于后半部分布莱克曼(Blackman)窗的波形(即M≤n≤N-1时)，其与前半部分的波形以直线n＝M呈轴对称，即将前半部分波形沿直线n＝M水平翻转后即可得到。另外，图39中分别示出了采用symmetric抽样和periodic抽样得到的布莱克曼(Blackman)窗的时域波形和频域波形。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，布莱克曼包络波形为频域上的函数波形，即图39中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的布莱克曼(Blackman)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图40所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0转换为+A，1转换为-A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2) 生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图41所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例十
下面则以初始包络波形为伯曼包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的伯曼(Bohman)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图42所示。从图42中可以看出，时域波形中伯曼(Bohman)窗是由0点开始，频域带外衰减为近60dB。
具体的，对于伯曼(Bohman)窗函数(对称函数)，其可通过下面公式表示：
ω(x)＝(1-|x|)cos(π|x|)+(1/π)sin(π|x|)
其中，-1≤x≤1。需要说明的是，上述公式中的x仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，伯曼包络波形为频域上的函数波形，即图42中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的伯曼(Bohman)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图43所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图44所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例十一
下面则以初始包络波形为平顶(Flat Top)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的平顶(Flat Top)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图45所示。从图45中可以看出，时域波形中平顶(Flat Top)窗是由近似0(-0.0004)点开始，频域旁瓣衰减为近100dB。另外，图45中分别示出了采用symmetric抽样和periodic抽样得到的平顶(Flat Top)窗的时域波形和频域波形。
具体的，对于平顶(Flat Top)窗函数，其可通过下面公式表示：
ω(n)＝a0-a1cos(2πn/N)+a2cos(4πn/N)-a3cos(6πn/N)+a4cos(8πn/N)
其中，0≤n≤N，窗长度L＝N+1，a0＝0.21557895，a1＝0.41663185，a2＝0.277263185，a3＝0.083578947，a4＝0.006947368。需要说明的是，上述公式中的n仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，平顶包络波形为频域上的函数波形，即图45中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的平顶(Flat Top)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图46所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图47所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例十二
下面则以初始包络波形为纳托尔(Nuttall)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的纳托尔(Nuttall)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图48所示。从图48中可以看出，时域波形中纳托尔(Nuttall)窗是由近似0(0.0004)点开始，频域旁瓣衰减为近100dB。另外，图48中分别示出了采用symmetric抽样和periodic抽样得到的纳托尔(Nuttall)窗的时域波形和频域波形。
具体的，对于纳托尔(Nuttall)窗函数(对称函数)，其可通过下面公式表示：
ω(n)＝a0-a1cos(2πn/(N-1))+a2cos(4πn/(N-1))-a3cos(6πn/(N-1))
具体的，对于纳托尔(Nuttall)窗函数(周期函数)，其可通过下面公式表示：
ω(n)＝a0-a1cos(2πn/N)+a2cos(4πn/N)-a3cos(6πn/N)
其中，n＝0,1,2,3,…,N-1。需要说明的是，上述公式中的n仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，纳托尔包络波形为频域上的函数波形，即图48中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的纳托尔(Nuttall)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图49所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图50所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例十三
下面则以初始包络波形为三角形(Triangular)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的三角形(Triangular)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图51所示。从图51中可以看出，时域波形中三角形(Triangular)窗是由0点开始，频域旁瓣衰减为近30dB。
具体的，对于三角形(Triangular)窗函数，其可通过下面公式表示：
当窗长度L为奇数时，
Figure PCTCN2016111405-appb-000008
当窗长度L为偶数时，
Figure PCTCN2016111405-appb-000009
需要说明的是，上述公式中的n仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，三角包络波形为频域上的函数波形，即图51中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的三角形(Triangular)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图52所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图53所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例十四
下面则以初始包络波形为巴尔森(Parzen)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的巴尔森(Parzen)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，其时域波形和频域波形如附图54所示。从图54中可以看出，时域波形中巴尔森(Parzen)窗是由0点开始，频域带外衰减近60dB。
具体的，对于巴尔森(Parzen)窗函数，其可通过下面公式表示：
Figure PCTCN2016111405-appb-000010
其中，-(N-1)/2≤n≤(N-1)/2。需要说明的是，上述公式中的n仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，巴尔森包络波形为频域上的函数波形，即图54中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的巴尔森(Parzen)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取 值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图55所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图56所示，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例十五
下面则以初始包络波形为图基(Tukey)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的图基(Tukey)包络波形h(t)。
本实施例中设计参数中，窗长度L＝63，以R分别为0.1、0.5、0.9为例，其时域波形和频域波形如附图57所示。其中，R是锥形区域到恒定值的比例，取值为0～1，当R取极值时，图基(Tukey)窗就会演变为其他的普通窗。R＝1，图基(Tukey)窗等效为汉宁窗；R＝0，图基(Tukey)窗等效为矩形窗。
从图57中可以看出，时域波形起点由0开始，随着R的增加，锥形区域越来越多，波形越来越平滑；频域波形旁瓣衰减越来越快，因此叠加后的性能更优。
具体的，对于图基(Tukey)窗函数，其可通过下面公式表示：
Figure PCTCN2016111405-appb-000011
其中，式中的α即为上述R值。需要说明的是，上述公式中的x仅表示公式中的函数变量。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，图基包络波形为频域上的函数波形，即图57中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的图基(Tukey)包络波形h(t)在时域内按预定的移位间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图58所示。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0转换为+A，1转换为-A，以得到正负符号序列。例如，取A＝1，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图59所示(图59A～图59C)，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
实施例十六
下面则以初始包络波形为泰勒(Taylor)包络波形来对本申请做进一步说明。其中，重叠复用次数K＝3，输入符号长度N＝8，输入符号xi＝{+1 +1 -1 -1 -1 +1 -1 +1}为例来说明OvTDM的信号发送和接收过程。其中，输入符号长度是指发送一帧信号的长度。
信号生成过程包括下面步骤：
(1)首先根据设计参数生成发送信号的泰勒(Taylor)包络波形h(t)。
本实施例使用matlab的泰勒(Taylor)函数，w＝taylorwin(n,nbar,sll)来生成需要的泰勒(Taylor)窗，本实施例中设计参数中，窗长度L＝63，以nbar＝4，sll＝-30，；nbar＝6，sll＝-50；nbar＝8，sll＝-80三组参数为例，其时域波形和频域波形如附图60所示，从频域图中可以看出旁瓣衰减对应为sll的值，分别为30dB、50dB、80dB，随着nbar的增加，时域波形起点越来越趋近于0，最高点的值越来越大，波形越来越平滑，因此叠加后的性能更优。其中，nbar影响着时域波形的起点位置，sll影响频域旁瓣衰减值。
基于前面的实施例可知，若该实施例应用在OvFDM系统中时，泰勒包络波形为频域上的函数波形，即图60中的左图为频域上的采样，右图为时域上的归一化函数。其余的调制解调方法步骤相似，再次不再赘述。
(2)将(1)所设计的泰勒(Taylor)包络波形h(t)在时域内按预定的移位 间隔进行移位，其中，移位间隔为时间间隔△T(△T＝L/K＝21)。移位后，形成各个时刻发送信号的偏移包络波形h(t-i*△T)(由于N＝8，因此i为整数且取值为0～7)，移位后各个时刻发送信号的偏移包络波形图如图61所示(nbar＝4，sll＝-30)。
(3)将输入的数字信号序列转换成正负符号序列。
具体的，可以将输入的数字信号序列中的0,1转换为±A，A取值为非0任意数，以得到正负符号序列。例如，取A为1时，将输入的{0,1}比特序列经过BPSK调制转换成{+1、-1}符号序列。
(4)将正负符号序列xi(本实施例中xi＝{+1 +1 -1 -1 -1 +1 -1 +1})与(2)生成的各个时刻发送信号的偏移包络波形h(t-i*△T)相乘，得到各个时刻的调制包络波形xih(t-i*△T)；形成后的波形如图62所示(图62A～图62C)，其中三条不同的虚线表示相乘后的三个波形图。
(5)将(4)所形成的各个时刻的调制包络波形xih(t-i*△T)在时域上进行叠加，以得到携带输出信号序列的复调制包络波形，即发送的信号。
本发明提供的重叠时分/频分复用调制方法、装置及系统由于初始包络波形在时域(或频域)内平滑，使得叠加后的波形平滑，从而系统的传输功率呈线性缓慢增长，间接提高了频谱利用率和传输速率。该重叠时分复用调制方法、装置及系统可以应用到移动通信、卫星通信、微波视距通信、散射通信、大气层光通信、红外通信、水声通信等无线通信系统中，既可以应用于大容量无线传输，也可以应用于小容量的轻型无线电系统。
本领域技术人员可以理解，上述实施方式中各种方法的全部或部分步骤可以通过程序来指令相关硬件完成，该程序可以存储于一计算机可读存储介质中，存储介质可以包括：只读存储器、随机存储器、磁盘或光盘等。
以上应用了具体个例对本发明进行阐述，只是用于帮助理解本发明，并不用以限制本发明。对于本发明所属技术领域的技术人员，依据本发明的思想，还可以做出若干简单推演、变形或替换。

Claims (22)

  1. 一种重叠时分复用调制方法,其特征在于，包括：
    生成在时域内波形平滑的初始包络波形；
    根据重叠复用次数将初始包络波形在时域内按预定的移位间隔进行移位，以得到各个时刻发送信号的偏移包络波形；
    将输入的数字信号序列转换成正负符号序列；
    将转换后的正负符号序列与偏移后各个时刻发送信号的偏移包络波形相乘，以得到各个时刻的调制包络波形；
    将各个时刻的调制包络波形在时域上进行叠加，以得到携带输出信号序列的复调制包络波形。
  2. 如权利要求1所述的方法，其特征在于，所述初始包络波形为切比雪夫包络波形、高斯包络波形、汉明包络波形、汉宁包络波形、布莱克曼包络波形、布莱克曼-哈里斯包络波形、巴特莱特包络波形、巴特莱特-汉宁包络波形、伯曼包络波形、平顶包络波形、纳托尔包络波形、巴尔森包络波形、泰勒包络波形、图基包络波形、凯塞包络波形、三角形包络波形中的一种。
  3. 如权利要求1所述的方法，其特征在于，所述移位间隔为时间间隔△T，时间间隔△T为：
    △T＝L/K
    其中，K为重叠复用次数，取值为非0正数；L为初始包络波形的窗长度。
  4. 如权利要求1所述的方法，其特征在于，将输入的数字信号序列转换成正负符号序列具体为：将输入的数字信号序列中的0转换为+A，1转换为-A，以得到正负符号序列，其中A的取值为非0任意数。
  5. 如权利要求1所述的方法，其特征在于，所述输出信号序列通过下面方式确定：
    当调制包络波形由正符号与该时刻的偏移包络波形相乘得到时，令该调制包络波形的运算值为+A，当调制包络波形由负符号与该时刻的包络波形相乘得到时，令该调制包络波形的运算值为-A；其中A的取值为非0任意数；
    对于每个移位间隔，将位于该移位间隔内的调制包络波形的运算值叠加，得出该移位间隔的输出信号，从而形成输出信号序列。
  6. 一种重叠时分复用调制装置,其特征在于，包括：
    波形生成模块，用于生成在时域内波形平滑的初始包络波形；
    移位模块，用于根据重叠复用次数将初始包络波形在时域内按预定的移位间隔进行移位，以得到各个时刻发送信号的偏移包络波形；
    调制模块，用于将输入的数字信号序列转换成正负符号序列；
    乘法模块，用于将转换后的正负符号序列与偏移后各个时刻发送信号的偏移包络波形相乘，以得到各个时刻的调制包络波形；
    叠加模块，用于将各个时刻的调制包络波形在时域上进行叠加，以得到携带输出信号序列的复调制包络波形。
  7. 权利要求6所述的装置，其特征在于，所述初始包络波形为切比雪夫包络波形、高斯包络波形、汉明包络波形、汉宁包络波形、布莱克曼包络波形、布莱克曼-哈里斯包络波形、巴特莱特包络波形、巴特莱特-汉宁包络波形、伯曼包络波形、平顶包络波形、纳托尔包络波形、巴尔森包络波形、泰勒包络波形、图基包络波形、凯塞包络波形、三角形包络波形中的一种。
  8. 权利要求6所述的装置，其特征在于，所述移位间隔为时间间隔△T，时间间隔△T为：
    △T＝L/K
    其中，K为重叠复用次数，取值为非0正数；L为初始包络波形的窗长度。
  9. 权利要求6所述的装置，其特征在于，调制模块用于将输入的数字信号序列转换成正负符号序列时：调制模块用于将输入的数字信号序列中的0转换为+A，1转换为-A，A取值为非0任意数，以得到正负符号序列。
  10. 权利要求6所述的装置，其特征在于，复调制包络波形携带的输出信号序列由各移位间隔的输出信号组成，各移位间隔的输出信号为各移位间隔内的调制包络波形的运算值叠加后的结果，当调制包络波形由正符号与该时刻的包络波形相乘得到时，其运算值为+A，由负符号与该时刻的包络波形相乘得到时，其运算值为-A，A取值为非0任意数，。
  11. 一种重叠时分复用调制解调系统，其特征在于，包括发射机和 接收机；
    所述发射机包括：
    如权利要求6-10任一项所述的重叠时分复用调制装置，用于生成携带输出信号序列的复调制包络波形；
    发射装置，用于将所述复调制包络波形发射到接收机；
    所述接收机包括：
    接收装置，用于接收所述发射装置发射的复调制包络波形；
    序列检测装置，用于对接收的复调制包络波形进行时域内的数据序列检测，以进行判决输出。
  12. 一种重叠频分复用调制方法,其特征在于，包括以下步骤：
    生成一个频域内波形平滑的初始包络波形；
    根据重叠复用次数将所述初始包络波形在频域上按预定的频谱间隔进行移位，得到各子载波包络波形；
    将输入的数字信号序列转换成正负符号序列；
    将所述正负符号序列中的符号与各自对应的子载波包络波形相乘，得到各子载波的调制包络波形；
    将所述各子载波的调制包络波形在频域上进行叠加，得到频域上的复调制包络波形；
    将所述频域上的复调制包络波形变换为时域上的复调制包络波形。
  13. 如权利要求12所述的重叠频分复用调制方法，其特征在于，所述频谱间隔为子载波频谱间隔△B，其中子载波频谱间隔△B＝B/K，B为所述初始包络波形的带宽，K为重叠复用次数，取值为非0正数。
  14. 如权利要求12所述的重叠频分复用调制方法，其特征在于，将输入的数字信号序列转换成正负符号序列具体为：将输入的数字信号序列中的0转换为+A，1转换为-A，以形成正负符号序列，其中A的取值为非0任意数。
  15. 如权利要求12所述的重叠频分复用调制方法，其特征在于，所述初始包络波形为切比雪夫包络波形、高斯包络波形、汉明包络波形、汉宁包络波形、布莱克曼包络波形、布莱克曼-哈里斯包络波形、巴特莱特包络波形、巴特莱特-汉宁包络波形、伯曼包络波形、平顶包络波形、纳托尔包络波形、巴尔森包络波形、泰勒包络波形、图基包络波形、凯塞包络波形、三角形包络波形中的一种。
  16. 如权利要求12所述的重叠频分复用调制方法，其特征在于，所述复调制包络波形携带的输出信号序列通过以下步骤确定：
    当调制包络波形由正符号与子载波包络波形相乘得到时，令该调制包络波形的运算值为+A，当调制包络波形由负符号与子载波包络波形相乘得到时，令该调制包络波形的运算值为-A；其中A的取值为非0任意数；
    对于每个频谱间隔，将位于该频谱间隔内的调制包络波形的运算值叠加，得出该频谱间隔的输出信号，从而形成输出信号序列。
  17. 一种重叠频分复用调制装置，其特征在于，包括：
    波形生成模块，用于生成一个频域内波形平滑的初始包络波形；
    移位模块，用于根据重叠复用次数将所述初始包络波形在频域上按预定的频谱间隔进行移位，得到各子载波包络波形；
    转换模块，用于将输入的数字信号序列转换成正负符号序列；
    乘法模块，用于将所述正负符号序列中的符号与各自对应的子载波包络波形相乘，得到各子载波的调制包络波形；
    叠加模块，用于将所述各子载波的调制包络波形在频域上进行叠加，得到频域上的复调制包络波形；
    变换模块，用于将所述频域上的复调制包络波形变换为时域上的复调制包络波形。
  18. 如权利要求17所述的重叠频分复用调制装置，其特征在于，所述频谱间隔为子载波频谱间隔△B，其中子载波频谱间隔△B＝B/K，B为所述初始包络波形的带宽，K为重叠复用次数，取值为非0正数。
  19. 如权利要求17所述的重叠频分复用调制装置，其特征在于，所述转换模块将输入的数字信号序列转换成正负符号序列具体为：将输入的数字信号序列中的0,1转换为±A,A取值为非0任意数。
  20. 如权利要求17所述的重叠频分复用调制装置，其特征在于，所述波形生成模块生成的初始包络波形为切比雪夫包络波形、高斯包络波形、汉明包络波形、汉宁包络波形、布莱克曼包络波形、布莱克曼-哈里斯包络波形、巴特莱特包络波形、巴特莱特-汉宁包络波形、伯曼包络波形、平顶包络波形、纳托尔包络波形、巴尔森包络波形、泰勒包络波形、图基包络波形、凯塞包络波形、三角形包络波形中的一种。
  21. 如权利要求17所述的重叠频分复用调制装置，其特征在于，所述复调制包络波形携带的输出信号序列由各频谱间隔的输出信号组成，各频谱间隔的输出信号为各频谱间隔内的调制包络波形的运算值叠加后的结果，当调制包络波形由正符号与子载波包络波形相乘得到时，其运算值为+A，由负符号与子载波包络波形相乘得到时，其运算值为-A,A取值为非0任意数。
  22. 一种重叠频分复用系统，其特征在于，包括发射机和接收机；
    所述发射机包括：
    如权利要求17至21中任一项所述的重叠频分复用调制装置，用于调制生成携带输出信号序列的复调制包络波形；
    发射装置，用于将所述复调制包络波形发射到接收机；
    所述接收机包括：
    接收装置，用于接收所述复调制包络波形；
    重叠频分复用解调装置，用于对接收的复调制包络波形进行解调，最终经过译码得到最终的输入比特序列。
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KR20180094043A (ko) 2018-08-22
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