Classifications

• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
  • H01Q3/26—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture
  • H01Q3/2682—Time delay steered arrays
  • H01Q3/2694—Time delay steered arrays using also variable phase-shifters
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q15/00—Devices for reflection, refraction, diffraction or polarisation of waves radiated from an antenna, e.g. quasi-optical devices
  • H01Q15/14—Reflecting surfaces; Equivalent structures
  • H01Q15/16—Reflecting surfaces; Equivalent structures curved in two dimensions, e.g. paraboloidal
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q21/00—Antenna arrays or systems
  • H01Q21/0006—Particular feeding systems
  • H01Q21/0037—Particular feeding systems linear waveguide fed arrays
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q21/00—Antenna arrays or systems
  • H01Q21/0006—Particular feeding systems
  • H01Q21/0037—Particular feeding systems linear waveguide fed arrays
  • H01Q21/0068—Dielectric waveguide fed arrays
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q21/00—Antenna arrays or systems
  • H01Q21/06—Arrays of individually energised antenna units similarly polarised and spaced apart
  • H01Q21/061—Two dimensional planar arrays
  • H01Q21/065—Patch antenna array
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
  • H01Q3/26—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture
  • H01Q3/30—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array
  • H01Q3/32—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array by mechanical means
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
  • H01Q3/26—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture
  • H01Q3/30—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array
  • H01Q3/34—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array by electrical means
  • H01Q3/36—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the relative phase or relative amplitude of energisation between two or more active radiating elements; varying the distribution of energy across a radiating aperture varying the relative phase between the radiating elements of an array by electrical means with variable phase-shifters
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
  • H01Q3/44—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the electric or magnetic characteristics of reflecting, refracting, or diffracting devices associated with the radiating element
  • H01Q3/443—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the electric or magnetic characteristics of reflecting, refracting, or diffracting devices associated with the radiating element varying the phase velocity along a leaky transmission line
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q5/00—Arrangements for simultaneous operation of antennas on two or more different wavebands, e.g. dual-band or multi-band arrangements
  • H01Q5/30—Arrangements for providing operation on different wavebands
  • H01Q5/307—Individual or coupled radiating elements, each element being fed in an unspecified way
  • H01Q5/342—Individual or coupled radiating elements, each element being fed in an unspecified way for different propagation modes
  • H01Q5/35—Individual or coupled radiating elements, each element being fed in an unspecified way for different propagation modes using two or more simultaneously fed points
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q9/00—Electrically-short antennas having dimensions not more than twice the operating wavelength and consisting of conductive active radiating elements
  • H01Q9/04—Resonant antennas
  • H01Q9/0407—Substantially flat resonant element parallel to ground plane, e.g. patch antenna
  • H01Q9/0428—Substantially flat resonant element parallel to ground plane, e.g. patch antenna radiating a circular polarised wave
  • H01Q9/0435—Substantially flat resonant element parallel to ground plane, e.g. patch antenna radiating a circular polarised wave using two feed points

Definitions

• This patent relates to series-fed phased array antennas and in particular to
• 20 terminal antennas are either large/bulky (dish/positioners) or high cost (phased
• Phased array antennas have many applications in radio broadcast, military, space, radar, sonar, weather satellite, optical and other communication systems.
• a phased array is an array of radiating elements where the relative phases of respective 25 signals feeding the elements may be varied. As a result, the radiation pattern of the array can be reinforced in a desired direction and suppressed in undesired directions. The relative amplitudes of the signals radiated by the individual elements, through constructive and destructive interference effects, determines the effective radiation pattern.
• a phased array may be designed to point continuously in a fixed direction, or to scan rapidly in azimuth or elevation.
• series fed arrays are typically frequency sensitive therefore leading to bandwidth constraints. This is because when the operational frequency is changed, the phase between the radiating elements changes proportionally to the length of the feedline section. As a result the beam in a standard series-fed array tilts in a nonlinear manner.
• a series fed antenna array may utilize a number of coupling elements.
• the coupling elements extract a portion of the transmission power for each radiator from one, or preferably two, waveguides.
• Controlled phase shifters may optionally be placed at each coupler, to delay the amount of transmission power to each one of the respective phased array elements.
• the transmission line may also be terminated with a dummy load at the end opposite the feed to avoid reflections.
• this shortcoming is avoided by using a waveguide having a variable wave propagation constant.
• a waveguide having a variable wave propagation constant In one example of a circularly polarized array implemented with such a waveguide, a single line of dual polarization couplers, or a pair of waveguides are used. Coupling between the variable dielectric waveguide and the antenna elements can be individually controlled providing accurate phasing of each element while keeping the Standing Wave Ratio (SWR) relatively low.
• multiple radiation modes may be used to extend a field of regard. Each of the radiation modes may be optimized for operation within a certain range of frequencies.
• progressive delay elements can be embedded in the waveguide couplers.
• coupler walls are placed along the variable dielectric waveguide.
• the coupler walls may be curved. These curved walls form focusing dielectric mirrors. These cause the energy entering the coupler to travel back and forth between the mirrors, accumulating delay, and thus effecting a further phase shift.
• the variable propogation constant of the waveguide is provided by adjusting an air gap between layers in the waveguide.
• the waveguide is generally configured as an elongated slab with a top surface, a bottom surface, a feed end, and a load end.
• the waveguide may be formed from dielectric material layers such as silicon nitride, silicon dioxide, magnesium fluoride, titanium dioxide or other materials suitable for propagation at the desired frequency of operation. Adjacent layers may be formed of materials with different dielectric constants.
• a control element is also provided to adjust a size of the gaps.
• the control element may be, for example, a piezoelectric, electroactive material or a mechanical position control. Such gaps may further be used to control the beamwidth and direction of the array.
• delay elements for a number of feed points are positioned along the waveguide and fed with progressive delay elements.
• the delay elements may be embedded into or on the waveguide.
• plated-through holes are formed along the waveguide orthogonal to the reconfigurable gap structure. Pins positioned in the plated-through holes allow the gap structure to mechanically slide up and down as the actuator gap changes size.
• a 2-D circular or a rectangular travelling wave array is fed by waveguide(s) with multiple layers and actuator controlled gaps to provide high gain, hemispherical coverage.
• the antenna solution described herein addresses the need for a steerable, wide bandwidth, low-profile antenna by using a variable effective dielectric traveling wave antenna array.
• a variable effective dielectric traveling wave antenna array By applying propagation velocity control to the traveling wave array technology, an efficient passive antenna array results which is linear, naturally bidirectional, and has no active, complex and expensive electronic devices.
• the dielectric traveling wave structure provides excitation of antenna elements to yield a cost-effective, high gain microwave antenna array that can handle the necessary high power levels. There are a multitude of possible applications for this phased array technology.
• Fig. 1 is a isometric view of of a unit cell waveguide coupler.
• Fig. 2 is a side view of the unit cell.
• Fig. 3 is a cross-section end view of the unit cell in an embodiment using a pair of variable dielectric waveguides feeding a patch radiator.
• Fig. 4 is a embodiment using a single waveguide, with couplers for each array element; the couplers include matched reflection phase shifters as may be implemented with a quadrature hybrid.
• Fig. 5 is a top view of an embodiment using a pair of waveguides with a constant phase shift provided by using dual quadrature couplers for each element.
• Fig. 6 is a more detailed top view of one cell of the two-waveguide embodiment of Fig. 5.
• Fig. 7 is a cross-sectional view of the unit cell for that same embodiment of Figs. 5 and 6.
• Fig. 8 is a isometric, partial cutaway view showing detail of the same embodiment showing the coupled waveguide walls formed as plates.
• Fig. 9 is another isometric view of the same embodiment with the walls implemented using pins.
• Fig. 10 is a high level schematic diagram of a series fed phased array.
• Fig. 11 illustrates e-field magnitude for different air gap sizes.
• Fig. 12 illustrates scanning the array by charging the effective dielectric of the waveguide at a selected frequency.
• Fig. 13 shows a microactuator in more detail.
• Fig. 14 shows (A) epsilon versus scan angle; (B) effective epsilon versus air gap size and (C) dieletric constant versus frequency for a two layer dielectric waveguide.
• Fig. 15 shows a gain versus scan angle plot.
• Fig. 16 shows the zero directivity loss scanning through broadside.
• Fig. 17 shows a feed arrangement for both right hand (RH) and left hand (LH) circularly polarizd (CP) elements.
• Fig. 18 is a plot of gain and axial ratio for three frequencies.
• Fig. 19 is a resulting elevation pattern for a 30 x 85 element array.
• Fig. 20 is a parameter table for a Ka-band implementation.
• Fig. 21 is an expected gain pattern.
• Fig. 22 shows effective dielectric constant versus scan angle for three radiation modes.
• Fig. 23 illustrates gain versus angle when multiple radiation modes are employed to extend a field of regard.
• Figs. 24 and 25 are an isometric and cutaway side view of an implementation using curved walls disposed perpendicular to the propogation axis of the waveguide.
• Fig. 26A illustrates a waveguide with variable effective propagation constant and crossed dupole radiator.
• Fig. 26B illustrates an electrical connection diagram
• Fig. 27 is an exploded top view of a multilayer waveguide where the waveguide sidewalls are defined using sliding pins with plated through holes.
• Fig. 28 is a side cross-sectional view of the Fig. 27 embodiment.
• Fig. 29 is a bottom view of the same embodiment.
• Fig. 30A is a top view of the same implementation.
• Fig. 30B is a side view, again of the same implementation.
• Figs. 31A, 31B, and 31C are cross-sectional, top and side views of the another implementation using circular array elements.
• a transmission line (which may be a waveguide or any other Transverse Electromagnetic Mode (TEM) line) contains all of the antenna element tap points which control power division and sidelobe levels, as well as the phase shifters which control the scan angle of the array.
• TEM Transverse Electromagnetic Mode
• this simplification can be provided by performing the phase shift function by varying the wave propagation velocity of the transmission line, thereby inducing a change in electrical length between the elements.
• the resulting electrical length is given by (Equations 1 , 2)
• the characteristic impedance of the transmission line is thus a fundamental parameter of the implementation, affecting power distribution, efficiency, input Voltage Standing Wave Ration (VSWR) and the like.
• VSWR Voltage Standing Wave Ration
• line impedance and velocity are coupled in this way is typically considered a fundamental limitation of the series fed array.
• scan angle and power bandwidth are coupled together; two parameters that are normally independent in other antenna systems.
• the variable waveguide / transmission line appears as a reflection type function, the desired phase shift may still be achieved using the same
• a quadrature coupler uses coaxial holes and an L-shaped probe to feed each radiating antenna element in a linear array. This arrangement solves the problem of how to control the coupling between the variable dielectric waveguide and the antenna elements to achieve accurate weighting of the antenna elements, while still keeping the Voltage Standing Wave Ratio (VSWR) low enough to eliminate the photonic band gap null for broad side angles.
• VSWR Voltage Standing Wave Ratio
• One embodiment of such a waveguide coupler 101 is coupled to a variable dielectric waveguide 102 below it via several slots 103 formed in the broad walls of the main variable dielectric waveguide 102 and the coupler 101.
• the slots 103 may be provided in various orientations, numbers and sizes which control the coupling level into and/or out of the coupled waveguide.
• Fig. 1 illustrates a unit waveguide coupler 101; each element of a multielement array requires one such unit coupler.
• the unit waveguide couplers 101 are periodically spaced along a main axis of the waveguide 102 according to the desired radiating element spacing on the top layer.
• the unit waveguide coupler 101 is formed in a Printed Circuit Board (PCB) with walls defined by vias or metal plates, but the unit coupler 101 can also be formed in a traditional waveguide structure.
• the waveguide coupler 101 need only be relatively short in length, as it is used to transfer a guided mode from the main waveguide structure 102, up to the radiating element.
• the variable waveguide(s) 102 are formed from a dielectric material or mechanical configuration for which the propogration constant can be varied, either by using materials where dielectric constant is changed via a bias voltage, or through mechanical layer separation in multilayer waveguides. See the discussion below, as well as our related U.S. Patent Publication 2012/0206310 for more details of adjustable waveguide structures.
• Fig. 2 shows a side view of the unit cell 101 geometry.
• the coupler On one end of the coupler (the end which feeds a patch antenna radiating element 104) there is a shorted pin 106 (via) that passes through a coaxial hole in the top of the waveguide, up through substrate layers and lands on an L-shaped probe 105 under the patch element 104.
• a shorted pin 106 On the other side of the coupler 101 is another pin, serving as a matched load 107. Because the coupler 101 is directional, very little energy is dissipated in the matched load 107.
• the L-probe 105 sits another substrate 108 and on top of that the patch radiator element 104.
• the L-probe 105 is capacitively coupled to the patch radiator 104.
• the shunt capacitance between the L-probe and ground plane is cancelled with the series inductance provided by the load pin 107.
• Fig. 3 shows further details of the geometry of the feed for an embodiment with two waveguides 102-1, 102-2 arranged in parallel.
• two respective L- probes 105-1, 105-2, waveguide couplers 101-1, 101-2, and main variable dielectric waveguides 102-1, 102-2 are situated with a single radiating patch 104 (as per Figs. 3 and 5), each radiating patch radiates a very wide, highly efficient antenna pattern as shown in Fig. 10. Any polarization can be achieved by controlling the phase shift and amplitude for the inputs to the two variable dielectric waveguides, as described below for certain example configurations.
• phase shift between two feeds changes along with change in a variable dielectric used to implant the main waveguide(s) 102.
• this phase shift between the scatterers or couplers 101 varies with the imaginary component of gamma (and velocity of propagation).
• the impact of this variable phase shift causes the axial ratio of a Circularly Polarized (CP) antenna to degrade because the axial ratio has a term for phase difference in it.
• CP Circularly Polarized
• FIG. 5 shows the two waveguides 102-1, 102-2 having a relative constant phase shift 110 placed before the feed. In the CP antenna example, this would be a constant phase shift of 90 degrees leading into one of the waveguides. In this way, the phase shift between pairs of scatterers or couplers 101 is fixed, and the change in propagation constant in the waveguide does not affect this phase shift (only the L- probes 105 are shown in Fig. 5 for the sake of clarity; it is understood that unit couplers 101 are associated with each radiating element 104 in this embodiment as were shown in Fig. 3).
• the two waveguides 102-2, 102-2 can feed a single line of dual polarization, dual input radiators as per Fig. 4, or each waveguide can feed an individual line of single polarization radiators, as per Fig. 5.
• This implementation solves an impedance mismatch when changing transmission line velocity.
• this implementation a) inserts an impedance transformer between each radiating element of the array and the following device; and 2) places two equivalent variable transmission lines on quadrature hybrid ports and using combined reflected waves at a fourth port as output.
• (vp) is changed to steer the beam.
• Fig. 4 The advantage of the Fig. 4 approach is that the addition of impedance transformer eliminates VSWR buildup; in addition, the reflectionless phase shifter decouples Zo and Vp.
• the impedance at the junction of each antenna element and the rest of the array can be made to equal 50 ohms by making the parallel combination of the element and feedline impedance 50 ohms. This is done by increasing the feedline impedance by using a quarter wave transformer, or other methods.
• Fig. 5 is a top cutaway view of one implementation of the two waveguide implementation.
• Fig. 6 shows the detail for one unit cell from a top view.
• a circular radiating element is implemented as a patch antenna 104.
• Two waveguide couplers 101-1, 101-2 feed the patch element 104 in quadrature as mentioned previously.
• the walls defining each of the unit waveguide couplers 101 may be implemented with a "picket fence" of via pins 130 disposed, as shown, in a rectangular region about the unit cell. Also visible are the L-probes 105-1, 105-2, load pins 107-, 107-2, and coupling slots 103-1, 103-2.
• Fig. 7 is a more detailed cross-sectional side view of the unit cell 101 for this embodiment showing the radiating patch, L- shaped probe 105, coaxial holes 112 that accommodate L-shaped probe 105, shorting pin 107, and section of the coupled waveguide 102.
• Example dimenstions and materials are also listed in Fig. 7 (in this view the vertical axes of the L-shaped probe 105 and shorting pin 107 are seen aligned with one another).
• Figs. 8 and 9 are futher isometric views of a two waveguide embodiment showing the several radiating patches and unit couplers.
• Fig. 8 uses metal plates to define the unit cell walls; the Fig. 9 arrangement instead uses pins to accomplish the same end.
• a single transmission line (waveguide or TEM line) contains all the antenna element tap points which control the power division and sidelobe levels, and the phase shifters which control the scan angle of the array. This is a great savings in electronic circuitry over a corporate feed structure which would require many 2-way power dividers to perform the same function. In some instances there is a further simplification by performing the phase shifting function by varying the transmission line wave propagation velocity as mentioned above.
• the DTWA array of Figs. 5, 6, 10 and elsewhere herein is a completely passive electronically scanned array, which provides numerous advantages over an active electronically scanned array.
• a complete Ka-Band DTWA Tx/Rx array assembly including electronic components located behind the array can be fit into in the same housing.
• AESA Active Electronically Steered Array
• PESA Passive Electronically Steered Array
• the PESA implementation preferred herein is a much better approach due to its lower cost, ruggedness, and simplicity.
• the approach provides the same performance characteristics as a full AESA due to the unique architecture, but without the complexity and cost of active electronic modules.
• G/T receive
• LNA's microwave Low Noise Amplifiers
• Electromagnetic Interference (EMI) environment of several systems operating close together The filtering must be placed at the very input and therefore has a negative effect on Signal to Noise Ration (SNR), so size and weight are increased if the filters are to be low loss.
• SNR Signal to Noise Ration
• the problem of phase and gain matching of each channel is greatly exacerbated by the filter/amplifier combination, especially at Ka-Band.
• AESA systems therefore must employ correctional phase shifters with enough resolution to accommodate these phase errors and are calibrated in software requiring large amounts of data storage. This calibration varies with temperature and time, which greatly complicates the system operation and makes control of the phase shifters a major networking challenge. Calibration times can even limit system concept of operation.
• AM/PM conversion is much more of an issue on transmit, especially if amplitude tapering is attempted in order to reduce transmit pattern sidelobes.
• the amplifiers must therefore be highly linear which means they are inefficient and output filtering to reduce broadband cosite noise further reduces the system efficiency. All of these components add size, weight and power. The fact that they are operating at a high power level and temperature reduces Mean Time Between Failures (MTBF). Even though AESA's are theoretically capable of "graceful degradation,” it has been observed that PESA's are always more reliable. Passive electronically steered arrays (PESAs), especially of the type described herein, have absolutely none of the AESA disadvantages.
• the preferred PESA implementation uses a micro-actuated control of the delay of a waveguide transmission medium feeding the elements to steer the beam of the array.
• Array gain is achieved by coherently combining element outputs in very low loss, broadband waveguide directional couplers.
• the circuitry is passive and bi-directional, the same antenna array is used on receive and transmit with no difference in performance. In fact, full duplex operation is possible, which has been range tested and proven at Ku-Band.
• Fig. 1 1 illustrates a modeled E-field magnitude for the DTWA for two different air gap thicknesses.
• Fig. 12 illustrates gain patterns versus elevation angle for different effective dielectrics (selected by chosing the size of the gap between dielectric layers).
• variable effective dielectric waveguides 102-1 , 102-2 provide series phase shifting for the radiating elements.
• ⁇ represents the scan angle along the array axis.
• wg is controlled by the variation of an air gap within the waveguide, changed by micro- actuators (d is element separation), m is described as an integer radiation mode number and can be any integer and ⁇ is element spacing. As higher radiation modes are used, higher dielectric material are used to support such a slow wave.
• variable effective dielectric waveguides 102 each may contain two (2) layers of dielectric.
• the space between the two (2) boards forms a single air gap in each waveguide, which is controlled by the micro-actuation.
• the upper board remains fixed to a multi-layer PCB above it, while only the lower board is moved down to control the air gap height. As the air gap thickness is increased, wg increases, causing a change in ⁇ . This method has been described previously for phase shifter
• Fig. 14 illustrates further results of an HFSS model of the single gap embodiment for Ka-band, showing (A) epsilon versus scan angle; (B) effective epsilon versus air gap size and (C) dielectric constant versus cutoff frequency.
• a particular waveguide dispersion may cause very small beam squint; as well, element spacing may also cause beam squint. Those can be corrected by
• Circular Polarization For most Ka-Band SOTM applications, Circular Polarization (CP) is desirable.
• CP Circular Polarization
• two (2) variable dielectric waveguides fed in quadrature feeding a single line of radiating patches as per Fig. 5.
• This approach provides better axial ratio over Field of Regard (FoR) than a single waveguide, as the Theta, Phi gain angle deltas are held constant at 90°, over the entire FoR.
• the waveguide may be arranged to ensure the fundamental guided mode propagates and all other guided modes remain in cutoff for the entire frequency band of operation - and through the entire air gap range.
• the open stop band also known as photonic band gap, is eliminated in this traveling wave antenna because the return loss of each directional waveguide coupler is so low, the coherent sum of reflected power does not significantly increase the VSWR.
• Fig. 16 illustrates the resulting zero directivity loss in a plot of RHCP gain versus theta for different gap sizes.
• the coupler maintains directivity, coupling value, and return loss throughout system bandwidth and waveguide characteristic impedence.
• Fig. 17 illustrates a RH/LH CP feed circuit.
• the waveguide-fed unit cell patch may also have a broad 3 dB elevation pattern to reduce scan loss over the required Field of Regard (FoR), and low axial ratio over a broad elevation pattern for maximum signal efficiency. Additionally, the patch fed from the waveguide directional coupler has extremely low return loss and high efficiency, allowing it to achieve a peak gain of 8.5 dBiC throughout the Ka- Band.
• the unit cell RHCP gain, LHCP gain and axial ratio patterns for three (3) frequencies are shown in Fig. 18, with performance represenative of the entire Rx Ka-Band. Depending on G/T and gain margin required over FoR, it is possible to customize the unit-cell pattern to fit the gain over FoR requirement more closely.
• the array gain over FoR presented below has excess G/T at broadside for the Ka-Band SOTM application, and meets the G/T requirement at the FoR edges. If the array size/gain over FoR trade shows the G/T may not roll off below the requirement at the FoR edge, this unit-cell would be modified to have a lower peak gain, but broader elevation pattern, which ultimately smooths out the G/T over scan angle, meeting the requirement over all scan angles, while reducing the required array size.
• a directional coupler adapted from work done by H. J. Riblet in "A New Type of Waveguide Directional Coupler," Proceedings of the IRE, 1948, pp. 61-63, was designed to provide: 1) a controllable coupling value for amplitude taper implementation, and 2) extremely low return loss in the waveguide, eliminating the photonic band stop passing through broadside. Additionally, the directional coupler directly feeds the patch feed, eliminating any lossy intermediary feedline. The photonic band gap phenomenon which results in large gain variation with frequency is alleviated by this method.
• waveguide directional couplers solves two (2) issues: 1) elimination of the photonic band gap effect, which causes a gain dropout at broadside on typical half-wave traveling wave arrays, and 2) allowing precise amplitude illumination for sidelobe level/beamwidth control.
• the directional couplers have extremely low return loss, essentially eliminating any reflection in the main waveguide, which is the source of the photonic band gap effect.
• the size and shape of the coupler elements control the level of energy coupled into a guided mode in a PCB integrated waveguide which in turn feeds the patch above it.
• Fig. 19 the sidelobe level control that is a result of a modified- Taylor amplitude series of waveguide directional couplers to form the desired current distribution along the array is shown.
• SLL and beamwidth control are critical because communications satellites are tightly placed in the geostationary orbital positions, as close as 1 degree. Strict sidelobe level and beamwidth compliance is required by 47 CFR FCC 25.209 and similar military requirements exist as well to prevent interference with adjacent orbital positions.
• the guided mode within the dielectric waveguide may be excited or received directly from a PA, LNA or down/upconverter.
• a short coaxial cable connects the antenna to these devices.
• a coaxial feed was designed to interface with the waveguide.
• a mode conversion from a coaxial cable to a waveguide guided mode, whose impedance and wavenumber is changing rapidly (over a factor 2: 1) as the beam is scanned through the FoR is achieved over the Rx Ka-Band bandwidth (19.2 - 21.2 GHz) using the waveguide feed developed for the Ka-Band DTWA.
• the feed achieves ⁇ 2:1 VSWR and ⁇ 0.6 dB insertion loss over all air gaps and throughout the band.
• a Wideband Global Satellite (WGS) transponder a typical Ka-Band transponder
• potential communications channel bandwidth 125 MHz
• a beacon signal typically accompanies the various transponder signals, either above, below, or within the band.
• a separate beacon receiver is used to monitor the beacon signal, and provide input to the core terminal on tracking movements.
• this beacon signal could be separated in frequency from the main channel bandwidth, requiring an effective instantaneous bandwidth greater than the communications channel itself. For this reason, it may be necessary to extend the instantaneous bandwidth of the antenna above what is actually required for the communications channel to accommodate this tracking method.
• a single directional coupler per line can be placed either between elements or on the underside of the waveguide to couple energy from a subset of the line array into a beacon receiver.
• a lower risk implementation trades size, weight, power and cost to reduce the instantaneous bandwidth to only what is needed for the communications channel. With the addition of a lower gain beacon DTWA and beacon receiver, adjacent to the communications channel DTWA, tracking is performed side by side with the main array, at the cost of additional surface area.
• An end fed array provides angle scanning by varying the propagation velocity of the transmission line feeding the antenna elements, thereby controlling the differential phase between elements. This also has the result that angle scanning occurs as the operating frequency is changed, which is an undesirable effect for our application.
• antennas constructed from arrays of waveguide-fed slots exist more because of their ease of fabrication than by a desire to synthesize such a structure. In any case, the problem comes about due to the vectorial summation of element outputs; the direction in which an end fed array has maximum gain is a direct function of the phase through the medium of propagation between elements.
• the relative phase is a function of frequency.
• corporate feed structures means are provided to equalize these delays so that array pointing is independent of frequency.
• provisions have to be made to delay the arrival of signals from elements closer to the output to equal the delay from elements further away. For example, the delay from the sixth element to the output is six (6) times the delay from the first element to the output, where there is an excess phase shift of:
• the far-field peak beam direction may only vary over a very small angle across the RF bandwidth. This undesired beam scanning with frequency causes a slight distortion in the gain over frequency curve, and the severity of that distortion depends on the beamwidth. This method is acceptable up to a 2.5% bandwidth, given the beamwidth is not extremely narrow.
• a progressive delay approach allows equalization of delays and far-field pattern alignment over a 10% bandwidth.
• a delay element is inserted between the coupled waveguide and the radiating element.
• the delay element is designed N times for different delay values, and each is implemented separately along the line array.
• the limiting factor in the progressive delay element approach is loss per unit delay. As with the waveguide, loss in the delay element must be kept to a minimum.
• Various circuit and material combination can realize the delay as a several section ( ⁇ 5) hairpin resonator coupled-line filter, implemented in microstrip on fused quartz is optimal. With this approach, loss per delay has been simulated as less than 1.0 dB/nS. 3.
• Dielectric wedge approach - A dielectric wedge may be placed on top of the array, and is integrated as part of the radome.
• the dielectric constant and shape of the wedge creates the progressive delay required along the line array.
• the advantage of the wedge is that it can be implemented in a low loss, high epsilon dielectric, providing a high delay to loss per unit length ratio. For this reason, it can achieve the highest bandwidth, >10 .
• DTWA antenna One specific implementation of the DTWA antenna was designed to meet the requirements shown in Fig. 20. These requirements were derived from a system level performance analysis for Ka-Band SOTM on move applications. 6. Multiple radiation modes extend field of regard in a traveling wave antenna.
• ⁇ is the scan angle
• ⁇ is the free space wavelength
• s is the line array element spacing
• ⁇ is the free space propagation constant
• ⁇ is the adjustable waveguide propagation constant
• the x axis represents theta (scan angle), and the y-axis represents an "effective dielectric constant" which is related to beta.
• a solution to the equation is shown for three frequencies ( at the operating frequency band edges and at a middle frequency) for an element spacing of .525 ⁇ .
• beta the waveguide propagation constant
• the solution to the equation scans along theta.
• HFSS High Frequency Structured Simulator
• progressive delay elements may be embedded in or withthe waveguide couplers 101.
• One possible geometry is shown in Figs. 24 and 25.
• the input and output coupler faces 140 lying transverse to the axis of the variable dielectric waveguide 101 may be curved to form a pair of focusing dielectric mirrors 145.
• the energy entering the coupler 101 then travels back and forth (as shown by dashed lines 147) between the mirrors 145 much like the mirrors in a laser.
• the far- field beam direction may only scan over a very small angle across the bandwidth. This beam scanning with frequency causes a slight distortion in the gain over frequency curve, and the severity of that distortion depends on the beamwidth. This method is acceptable up to a 2.5% bandwidth, given the beamwidth is not extremely narrow.
• the progressive delay approach allows equalization of delays and far-field pattern alignment over a 10% bandwidth.
• a delay element can be inserted between the coupled waveguide and the radiating element.
• the delay element is designed N times for different delay values, and each one is implemented separately along the line array.
• the limiting factor in the progressive delay element approach is loss per unit delay. As with the waveguide, loss in the delay element must be kept to a minimum.
• Dielectric wedge approach A dielectric wedge may be placed atop the array, and integrated as part of the radome.
• the dielectric constant and shape of the wedge performs time delay beamforming for each progressive element.
• the advantage of the wedge is that it can be implemented in a low loss, high epsilon dielectric, providing a high delay to loss per unit length ratio. For this reason, it can achieve the highest relative bandwidth, >10%.
• An array of antenna elements here consisting of crossed bow ties 1504, are placed along the length of the top surface of the waveguide 1502.
• the antenna elements 1504 may each be fed with a quadrature hybrid combiner as for the other embodiments (not shown).
• the key to the wide band operation is a delay line 1525 embedded in or with each antenna element along the array.
• the delay line 1525 is a compact helical HE11 mode line using a high dielectric constant material such as titanium dioxide or barium tetratitinate.
• the additional delay is provided by changing the propagation constant in the waveguide with a gap structure.
• a waveguide has plated-through holes provided with a reconfigurable gap structure, with pins positioned in the plated-through holes. The pins allow the structure to slide up and down as the actuator gap changes size.
• a 2-D gap structure may utilize layers of dielectric slabs 1602 with rows of periodically spaced plated through holes 1610 and actuator strips 1620 of piezoelectric or electro active material.
• the rows of plated through holes define side walls of individual waveguide sections 1502.
• the slab waveguide 1600 arrangement is shown in Fig. 27.
• Pins 1630 are placed along the actuator strips to:
• Strips of conducting material can be deposited on both sides of the piezoelectric layers 1620 to enable control voltages to be impressed upon the piezoelectric actuators through the pins 1630.
• the control voltages can be applied separately to each row or applied to the entire array by connecting the conducing strips together at one end of the structure.
• Fig. 28 shows a side view of the same structure 1600 with an exciting horn antenna (feed) 1650 at one end.
• an exciting horn antenna (feed) 1650 at one end.
• each horn is fed with a progressive phase shift.
• the radiationg patch(es) are placed in a layer 1650 above the slabs 1602.
• Fig. 29 shows a bottom view of the same slab waveguide structure 1603 with the array of horn antennas 1650 now visible at one end.
• the reconfigurable gaps 1603 and the waveguide pins 1630 are also seen.
• the lower surface may have a printed circuit board 1680 that provides control and power circuits to the actuators which allows for control of the gap size(s).
• the control of the gaps changes the effective dielectric of the slab which allows for scanning of the beam without a change of frequency in the traveling wave array.
• 2-D circular and rectangular travelling wave arrays are fed by slab waveguides with multiple layers and actuator controlled gaps to provide high gain hemispherical coverage.
• traveling wave arrays would typically require a separate waveguide to provide exitation to each row of a 2-D traveling wave array.
• a single waveguide provides an elevation steerable line array of elements with the line arrays configured side-by-side.
• a separate conventional feed system is used to excite each- line array with the proper phase or time delay to provide steerabiility in the azimuthal plane.
• the elevation steering of the traveling wave line arrays is accomplished by actuator controls gaps in the dielectric to control the propagation constant.
• the two geometries to be considered are (A) a Cartesian geometry using rectangular slabs and (B) a circularly symmetric geometry using circular slabs.
• a square slab waveguide 1600 (again, formed of multiple dielectric layers as per Fig. 16 (27?) is used in which the exciting elements 1910 are mounted along the sides of the waveguide.
• the exciting elements (vertically polarized) 1940 of two adjacent sides are used to generate a plane wave excitation in the slab as shown by the dotted line 1960 in Fig. 30A.
• a plane wave 1620 in any direction can be generated by the use of the exciting elements 1910 on the appropriate two adjacent sides.
• the exciting elements 1910 should have beam widths of 90° to guarantee uniform coverage over the azimuthal plane.
• Mounted on the top surface of the slab waveguide 1600 are so-called scattering elements 1940 which intercept a small amount of the plane wave excitation and reradiate the power. The system thus operates as a leaky wave structure.
• the scattering elements 1940 which should exhibit hemispherical patterns, can be circularly polarized crossed dipoles are arranged in a Cartesian grid pattern, as shown.
• Figs. 31 A, 3 IB and 31C provide circular symmetry as : 1) a "flat” circular slab version and 2) a “conical wedge” version.
• the flat circular case in Figs. 31A and 3 IB uses a circular slab waveguide with a hole in the center for the exciting elements, a commutator, and a beam former.
• the beam former feeds a sector of exciting vertically polarized elements 2010 to obtain a narrow beam in the direction of that sector, while the commutator 2020 selects the sector direction.
• the scattering elements are configured in concentric circles 2030 (only partially shown for clarity), keeping the number of elements in each concentric circle constant.
• the elevation angle of the beam is determined by the propagation constant of the slab waveguide 2002 with configurable gaps 2003 as determined by the gap width, which is controlled by the gap actuators.
• the azimuthal angle of the beam is determined by the position of the commutator 2020.
• the scattering elements 2050 should have a pattern providing hemispherical coverage.
• the wedge version shown in Fig. 31C provides wideband coverage using a conical wedge 2080 as a progressive delay element.
• the wedge 2080 is situated on top of the circular slab waveguide 2090 with configurable gaps 2092.
• An exponential coupling layer 2095 is introduced between the wedge and the slab waveguide.
• the exponential layer 2095 is needed to generate a uniform plane wave across the wedge 2080.
• No scattering elements are needed since the layer and the high dielectric constant of the wedge provide a leaky structure.
• the elevation angle of the beam is, as in the flat slab version of Figs. 31A and 31B, determined by the propagation constant of the slab waveguide as determined by the gap width. Since no scattering elements are used, arbitrary polarization can be provided in the main beam by introducing circularly polarized exciting elements 2099, or combine vertical and horizontal elements such as crossed bowties.

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• 2015
  • 2015-03-20 EP EP15719878.9A patent/EP3271966A1/en not_active Withdrawn
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