WO2016103981A1 - Δς modulator, and transmitter - Google Patents

Δς modulator, and transmitter Download PDF

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Publication number
WO2016103981A1
WO2016103981A1 PCT/JP2015/082225 JP2015082225W WO2016103981A1 WO 2016103981 A1 WO2016103981 A1 WO 2016103981A1 JP 2015082225 W JP2015082225 W JP 2015082225W WO 2016103981 A1 WO2016103981 A1 WO 2016103981A1
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signal
path
modulator
quantized data
output
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PCT/JP2015/082225
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French (fr)
Japanese (ja)
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前畠　貴
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住友電気工業株式会社
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Priority claimed from JP2015122752A external-priority patent/JP6447386B2/en
Application filed by 住友電気工業株式会社 filed Critical 住友電気工業株式会社
Priority to US15/527,034 priority Critical patent/US10090855B2/en
Publication of WO2016103981A1 publication Critical patent/WO2016103981A1/en

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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03M—CODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M3/00—Conversion of analogue values to or from differential modulation
    • H03M3/02—Delta modulation, i.e. one-bit differential modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/02—Transmitters
    • H04B1/04—Circuits

Definitions

  • the present invention relates to a ⁇ modulator and a transmitter.
  • Patent Document 1 discloses a ⁇ modulator.
  • the ⁇ modulator has, as a basic configuration, a loop filter, a quantizer, and an internal path that connects the loop filter and the quantizer.
  • the ⁇ modulator performs ⁇ modulation on the input signal to generate quantized data.
  • Quantized data generated by the ⁇ modulator includes an analog signal component such as an RF signal at a predetermined frequency (target frequency).
  • the analog signal component may have distortion due to ⁇ modulation.
  • Patent Document 1 discloses that the pulse waveform of a pulse train output from a ⁇ modulator affects the signal characteristics of an analog signal expressed by the pulse train output from the ⁇ modulator.
  • the pulse waveform of the pulse train output from the ⁇ modulator causes distortion in the analog signal represented by the pulse train.
  • Patent Document 1 discloses that processing is performed on a pulse train output from a ⁇ modulator.
  • the present invention from one aspect includes a loop filter, a quantizer that generates quantized data based on an output of the loop filter, an internal path connected to the loop filter or the quantizer,
  • a delta-sigma modulator comprising: a compensator that applies a compensation signal for distortion generated in a frequency component of a target frequency among frequency components of a pulse train corresponding to quantized data to the internal path.
  • the ⁇ modulator itself can compensate for distortion occurring in the frequency component of the target frequency.
  • a ⁇ modulator includes a loop filter, a quantizer that generates quantized data based on an output of the loop filter, and an internal path connected to the loop filter or the quantizer And a compensator that provides a compensation signal for distortion generated in the frequency component of the target frequency among the frequency components of the pulse train corresponding to the quantized data to the internal path. Since the ⁇ modulator includes a compensator, the ⁇ modulator itself can compensate for distortion occurring in the frequency component of the target frequency.
  • the distortion includes distortion generated in a frequency component of the target frequency due to asymmetry between the rising edge and the falling edge of the pulse in the pulse train corresponding to the quantized data.
  • the quantized data output from the ⁇ modulator is compensated for the distortion generated in the frequency component of the target frequency due to the asymmetry. It will be a thing.
  • the compensator includes a detector that detects a change in the quantized data, and a generator that generates a compensation signal based on the change in the quantized data detected by the detector. preferable.
  • a change in quantized data it is possible to detect the occurrence of asymmetry that occurs at the rise and fall of the pulse in the pulse train. Therefore, by generating the compensation signal based on the change in the quantized data, the compensation signal can be generated based on the occurrence of asymmetry.
  • the detector outputs a detection signal at a timing when the quantized data changes.
  • the generator can generate a compensation signal based on the detection signal output at the timing when the quantized data changes.
  • the generator includes a fractional delay, and the fractional delay generates the compensation signal based on the detection signal.
  • the fractional delay makes it easy to generate a compensation signal that compensates for distortion that occurs in the frequency component of the target frequency due to asymmetry.
  • the quantizer is preferably a 1-bit quantizer.
  • the asymmetry of the rise and fall of the pulse train is likely to be a problem, and it is advantageous to perform compensation with the ⁇ modulator itself.
  • the internal path includes a first path for feeding back the quantized data to the loop filter, and the compensator provides the compensation signal to the first path.
  • the compensation signal By providing the compensation signal to the first path for feeding back to the loop filter, the operation is easily stabilized.
  • the internal path may include a second path for supplying the output of the loop filter to the quantizer, and the compensator may supply the compensation signal to the second path. In this case, compensation using a compensation signal is possible.
  • the internal path includes a second path for supplying the output of the loop filter to the quantizer, and the compensator supplies the compensation signal to the first path and the second path. preferable. In this case, wideband distortion compensation is possible.
  • the transmitter according to the embodiment outputs a pulse corresponding to the quantized data as a transmission signal.
  • FIG. 1 shows a transmitter 10.
  • the transmitter 10 includes a digital signal processing unit 12 and an analog filter 16.
  • the digital signal processing unit 12 outputs a digital signal (1-bit pulse train) that represents an RF (Radio Frequency) signal that is an analog signal.
  • the RF signal is a signal radiated into the space as a radio wave, for example, an RF signal for mobile communication or an RF signal for broadcasting service.
  • the pulse train output from the digital signal processing unit 12 is given to an analog filter (bandpass filter or lowpass filter) 16.
  • the analog signal represented by the pulse train output from the digital signal processing unit 12 includes a frequency component other than the frequency (target frequency) of the RF signal as a noise component.
  • the noise component is removed by the analog filter.
  • the signal transmission path 14 between the digital signal processing unit 12 and the analog filter 16 may be a signal wiring formed on a circuit board, or a communication cable such as an optical fiber or an electric cable.
  • the digital signal processing unit 12 includes a baseband unit 18 that outputs a baseband signal (IQ signal) that is a transmission signal, a modulator (orthogonal modulator) 20 that modulates the baseband signal, a processing unit 22, and ⁇ modulation.
  • a device 24 and a controller 26 are provided.
  • the baseband unit 18 outputs an IQ baseband signal (I signal, Q signal) as digital data.
  • the quadrature modulator 20 is configured as a digital quadrature modulator that performs quadrature modulation on an IQ baseband signal by digital signal processing.
  • the processing unit 22 performs digital signal processing on the quadrature modulation signal output from the quadrature modulation 20 and outputs a digital IF signal.
  • the digital signal processing performed by the processing unit 22 includes, for example, digital pre-distortion (DPD), crest factor reduction (Crest Factor Reduction; CFR), and digital up-conversion (Digital Up Conversion; DUC).
  • the digital RF signal output from the processing unit 22 is given to the ⁇ modulator 24.
  • the ⁇ modulator 24 performs ⁇ modulation on the digital RF signal, quantizes the digital RF signal, and outputs quantized data (pulse train).
  • the pulse train output from the ⁇ modulator 24 represents an analog RF signal.
  • the transmitter 10 transmits this pulse train as a transmission signal.
  • the ⁇ modulator 24 passes a signal component of a desired frequency (target frequency f 0 ), and causes noise in a band near the target frequency f 0 to shift out of the band. Perform shaping.
  • the ⁇ modulator 24 includes a loop filter 30, a quantizer 36 that outputs quantized data, an internal path 42 connected to the loop filter 30 or the quantizer 36, and a compensator 38. And an adder 40.
  • the internal path 42 includes a first path (feedback path) 42 a for feeding back the quantized data output from the quantizer 36 to the loop filter 30, and a first path for supplying the output of the loop filter 30 to the quantizer 36.
  • the path through which the signal flows only to the outside of the ⁇ modulator 24 is substantially the same as the external path of the ⁇ modulator 24 and is not included in the internal path 42.
  • the internal path 42 is a path that can reach the quantizer 36 that generates the output of the ⁇ modulator 24.
  • the loop filter 30 has two inputs and one output, and receives an input signal (digital RF signal) to the ⁇ modulator 24 and a feedback signal from the quantizer 36 side.
  • the loop filter 30 includes a first adder 32, an L (z) transfer function block 33, a second adder 34, and a feedforward path 35.
  • the first adder 32 adds the input signal to the ⁇ modulator 24 and the feedback signal from the quantizer 36 side.
  • the feedback signal is given to the first adder 32 through the first path 42a.
  • the transfer function L (z) of the transfer function block 33 determines the characteristics of the ⁇ modulator 24 and is determined based on a desired signal transfer function and noise transfer function.
  • the second adder 34 adds the output of the transfer function block 33 and the input signal to the ⁇ modulator 24.
  • the input signal is given to the second adder 34 via the feedforward path 35.
  • the feedforward path 35 and the second adder 34 may be omitted.
  • the output of the second adder 34 that is, the output of the loop filter 30 is given to the quantizer 36 through the second path 42b.
  • the quantizer 36 is a 1-bit quantizer, and outputs 1-bit quantized data obtained by quantizing the output of the loop filter 30 into 1 bit.
  • the quantized data (pulse train) output from the quantizer 36 is output from the ⁇ modulator 24 as an output of the ⁇ modulator 24 and is fed back to the loop filter 30 via the first path 42a.
  • the compensator 38 outputs a compensation signal C for distortion generated in the analog RF signal (frequency component of the target frequency f 0 ) having the frequency f 0 expressed by the pulse train output from the ⁇ modulator 24.
  • the compensation signal C is for canceling or suppressing distortion generated in the RF signal.
  • the compensation signal C output from the compensator 38 is given to the first path 42a by the adder 40 provided in the first path 42a.
  • the compensation signal C for canceling or suppressing the distortion is supplied to the internal path 42 of the ⁇ modulator 24, so that the distortion is compensated inside the ⁇ modulator 24.
  • the quantized data output from the ⁇ modulator 24 represents an RF signal (frequency component of the target frequency f 0 ) whose distortion has been compensated.
  • the ⁇ modulator 24 Since the ⁇ modulator 24 has the first path 42a that is a feedback path, even if a compensation signal is given to any position of the internal path 42, the compensation signal passes through the transfer function block 33. Therefore, ⁇ modulation having a desired characteristic can be performed in a state where distortion compensation by the compensation signal is performed. However, when the compensation signal is given to the first path 42a as shown in FIG. 2A, the compensation signal passes through the transfer function block 33 before the quantizer 36, and thus the operation is easily stabilized.
  • the compensation signal is given to the first path 42a, but may be given to another internal path 42 (for example, the second path 42b). That is, as shown in FIG. 2B, the compensation signal C output from the compensator 38 is not only supplied to the first path 42a, but also by a second adder (adder / subtracter) 34 provided in the second path 42b. It may be given to the second path 42b.
  • the compensation signal C given to the second path 42b is given to the second path 42b so as to subtract the compensation signal C from the signal flowing through the second path 42b.
  • the output V of the ⁇ modulator 24 is expressed by the following equation (1).
  • the first term on the right side of the equation (1) is the input signal U, and the second term represents the quantization noise E multiplied by the filter characteristic by the noise transfer function NTF of the ⁇ modulator 24. If only the first term and the second term on the right side of Equation (1) are used, the output V of the normal ⁇ modulator is obtained.
  • the third term on the right side of Expression (1) is generated.
  • the compensation signal C multiplied by the filter characteristic of L (z) / (1 + L (z)) is reflected in the output V.
  • the frequency characteristic of L (z) / (1 + L (z)) is a band pass as shown in FIG. 2C. That is, as shown in FIG. 2C, a band-pass filter having a pass band from L (z) / (1 + L (z)) 950 MHz to 1050 MHz. For this reason, only the component in the passband of the bandpass filter is reflected in the output V of the compensation signal C, and distortion compensation is performed in a relatively narrow band.
  • the ⁇ modulator 24 has a driver (not shown) in order to output a pulse train corresponding to the quantized data.
  • the driver has a switching element and the like, and the rise and fall of the pulse are formed by the ON / OFF operation of the switching element.
  • the rise time and fall time of the pulse formed by the driver do not coincide with each other, and asymmetry occurs between the rise and fall of the pulse. This asymmetric component degrades the RF signal.
  • the asymmetrical components of the rise and fall of the pulse are defined.
  • the pulse train S out (t) output from the ⁇ modulator 24 is defined as the following equation (A).
  • the quantized data d k takes +1 as a value corresponding to the high level of the pulse and ⁇ 1 as a value corresponding to the low level of the pulse.
  • U (t) is a unit step function.
  • the second term of the equation (A) indicates the difference between S out (t) corresponding to the actual waveform and the ideal waveform S Ideal .
  • F (t ⁇ kt) in the second term is defined as in the following formula (C). Sign is a sign function.
  • (C-1) is positive in sign of a value indicating a difference between a certain quantized data value d k and a temporally previous quantized data value d k ⁇ 1.
  • the case where the pulse corresponding to the quantized data d k rises is shown.
  • (C-2) if the sign of the value that indicates the difference between the value d k-1 value d k and temporally preceding quantized data of a quantized data is negative, i.e., quantization It shows a case where pulse corresponding to the signal d k falls.
  • (C-3) is a case where a value indicating a difference between a certain quantized data value d k and a temporally previous quantized data value d k ⁇ 1 is zero, that is, a pulse value. This is the case when there is no change.
  • f rise (t) and f fall (t) are the rising waveform and falling waveform of the pulse, respectively.
  • f rise (t) and f fall (t) can be decomposed into a symmetric component f sym (t) and an asymmetric component f Asym (t) as shown in equation (D).
  • the asymmetric component f Asym (t) can be obtained from the following formula (E) from the formula (D).
  • Equation (E) shows that the asymmetric component f Asym (t) disappears when the rising waveform f rise (t) and the falling waveform f fall (t) have the relationship of the following equation (F). Is shown.
  • FIG. 3 shows a pulse waveform (asymmetric waveform having an asymmetric component) that does not satisfy the equation (F).
  • FIG. 3A shows an eye pattern of the asymmetric waveform S out (t). This eye pattern is asymmetric with respect to the time axis.
  • the asymmetric waveform shown in FIG. 3 is a waveform in which the pulse fall time is longer than the pulse rise time.
  • FIG. 3B shows a time axis waveform of the asymmetric waveform S out (t)
  • FIG. 3C shows an ideal waveform S Ideal (t) for the asymmetric waveform
  • FIG. 3E shows the symmetric component f sym (t) in the rising waveform f rise (t) and the falling waveform f fall (t) in the asymmetric waveform
  • FIG. 3E shows the rising waveform f rise (t) and the rising in the asymmetric waveform.
  • the asymmetric component f Asym (t) in the falling waveform f fall (t) is shown.
  • the asymmetric waveform is distorted with respect to the ideal waveform S Ideal (t) and has a distortion component.
  • the pulse rising waveform f rise (t) has a distortion component (first distortion component)
  • the pulse falling waveform f fall (t) has a distortion component (second distortion component).
  • the distortion component has an asymmetric component f Asym (t) together with a symmetric component f sym (t) (see FIGS. 3D and 3E ).
  • the presence of the symmetric component f sym (t) has little influence on the characteristics of the RF signal (for example, adjacent channel leakage power (ACLR)), but the asymmetric component f Asym (t) is a characteristic of the RF signal. (See Patent Document 1).
  • the shape of the pulse output from the ⁇ modulator 24 affects the RF signal (frequency component of the target frequency f 0 ) that is to be processed by the ⁇ modulator 24.
  • the distortion of the RF signal caused by the waveform distortion (asymmetrical component) of the pulse train is compensated in advance by the compensation signal inside the ⁇ modulator 24 before the pulse train is output. Therefore, even if the pulse waveform output from the ⁇ modulator 24 has an asymmetric component, degradation of the ACLR of the RF signal is suppressed.
  • FIG. 4 shows an example of a compensator 38 suitable for compensating for distortion due to an asymmetric component of the rise and fall of the pulse train.
  • the compensator 38 includes a detector 44 and a compensation signal generator 46.
  • the detector 44 detects a change in quantized data (rising or falling of the pulse train). Since the asymmetric component occurs at the rise or fall of the pulse train, the occurrence of the asymmetric component can be detected by detecting the rise or fall of the pulse train.
  • the detector 44 is supplied with the quantized data (pulse train) output from the quantizer 36 as an input.
  • the detector 44 outputs a detection signal (pulse detection signal) at the timing when the quantized data changes.
  • the pulse train output from the ⁇ modulator 24 becomes as shown in FIG. 5B.
  • the asymmetric component occurs at the rise and fall of the pulse train in FIG.
  • the detector 44 outputs a detection signal (quantized data change detection signal) in synchronization with the generation timing of the asymmetric component.
  • the detector 44 generates a detection signal shown in FIG. 5D, so that the delay element 48, the adder 50, the sign function unit 52, the Abs (absolute value) function unit 54, have.
  • the adder (difference unit) 50 of the detector 44 obtains a difference between the quantized data at a certain sampling clock and the quantized data of the clock immediately before the sampling clock.
  • the delay element 48 provides the adder 50 with the quantized data of the previous clock before the sampling clock.
  • the adder 50 outputs 0 when the quantized data in a certain sampling clock and the quantized data of the clock previous to the sampling clock match, and when they do not match (the quantized data has changed). ), A value other than 0 is output.
  • the sign function unit 52 outputs +1, ⁇ 1, or 0 according to the sign of the output from the adder 50.
  • the Abs function unit 54 outputs the absolute value of the output of the sign function unit 52. That is, the Abs function unit 54 outputs 1 when the quantized data has changed from the quantized data of the previous sampling clock in each sampling clock, and outputs 0 when the quantized data does not change. Is output. Therefore, the detector 44 can output a detection signal as shown in FIG.
  • the compensation signal generator 46 generates a compensation signal that suppresses the asymmetric component (see FIG. 5C) based on the detection signal indicating the change in the quantized data.
  • the compensation signal generator 46 is configured by a fractional delay. This fractional delay has the same configuration as a finite impulse response (FIR) filter. That is, the compensation signal generator 46 includes a plurality of delay elements 56a, 56b, 56c, and 56d, a plurality of gain control elements 58a, 58b, 58c, 58d, and 58e, and an adder 60. Has an FIR filter structure.
  • the compensation signal generator 46 of FIG. 4 has a 4-tap digital filter configuration.
  • the compensation signal generator 46 acts as a filter on the pulse-like detection signal, and generates a compensation signal for suppressing an asymmetric component that causes distortion in the RF signal that is the frequency component of the target frequency f 0 . Since the pulse-like detection signal has a wide frequency component, it is easy to generate a compensation signal by a filter action.
  • the detection signal only needs to have a frequency component necessary for the compensation signal, and is not limited to a pulse shape.
  • FIG. 6 shows a method of determining the coefficients (gains) Ci of the gain control elements 58a to 58e.
  • a digital RF signal (a coefficient determination test signal) is input to the ⁇ modulator 24 so that quantized data (pulse train) is output from the ⁇ modulator 24.
  • the process of FIG. 6 is performed.
  • step S1 the coefficients of all gain control elements 58a to 58e are set to zero.
  • the compensation signal is also zero (no compensation signal).
  • the ACLR of the output (RF signal) of the ⁇ modulator 24 is changed while changing the value of the coefficient C1 within a predetermined search range (for example, ⁇ 0.2 to 0.2). taking measurement.
  • the value with the best ACLR is determined as the value of the coefficient C1 (step S3).
  • FIG. 7A shows the ACLR (vertical axis) when the coefficient C1 (horizontal axis) is changed between ⁇ 0.2 and 0.2.
  • the coefficient C2 is changed between ⁇ 0.2 and 0.2, and the ALCR is measured.
  • C3, C4, and C5 all the coefficients C1 to C5 can be determined.
  • the coefficients C1 to C5 may be dynamically changed at a necessary time.
  • FIG. 8 shows a configuration for dynamically changing the coefficients C1 to C5.
  • the controller 26 can change the coefficients C1 to C5 of the gain control elements 58a to 58e. Further, the controller 26 is configured to acquire a pulse train (an RF signal expressed by) corresponding to the quantized data output from the ⁇ modulator 24.
  • the controller 26 configured as described above can change the coefficients C1 to C5 by executing the processing of FIG. In this case, when the asymmetric component of the pulse train changes with time, the distortion can be appropriately compensated by updating the coefficients C1 to C5.
  • FIGS. 10A and 11A show the ACLR when the compensation by the compensator 38 is not performed.
  • FIGS. 10B and 11B show ACLRs when compensation by the compensator 38 is performed. In FIG. 10B and FIG. 11B, the adjacent channel leakage power is lower than in FIG. 10A and FIG. Therefore, the effect of compensation by the compensator 38 is recognized.
  • the compensator 38 is not limited to that shown in FIG. 4, and may be any one that outputs a compensation signal for compensating for distortion of the RF signal expressed by the pulse train.
  • the compensator 38 is not limited to one that compensates for distortion occurring in the target frequency f 0 due to an asymmetric component generated in the pulse train (pulse train corresponding to quantized data) directly output by the ⁇ modulator 24, and other devices the asymmetric component occurring in the output pulse train (pulse train corresponding to the quantized data), may be configured to compensate for the distortion of the target frequency f 0.
  • the pulse train output by the receiver Compensation for distortion by the asymmetrical component may be performed by the ⁇ modulator 24 on the transmitter side.
  • FIG. 12 shows a dual band ⁇ modulator (multiband ⁇ modulator) 24.
  • the dual band ⁇ modulator (multiband ⁇ modulator) 24 is disclosed in Japanese Patent Application Laid-Open No. 2014-165846.
  • the dual-band ⁇ modulator 24 can input two (a plurality of) input signals U 1 and U 2 having different frequencies, and a plurality of loop filters (a first loop filter 30a and a second loop filter 30b), An adder 15 that adds the outputs of the filters 30a and 30b and a quantizer 36 that quantizes the output of the adder 15 are provided.
  • the quantizer 36 of the ⁇ modulator 24 outputs a single output signal (quantized data) including two input signals U 1 and U 2 .
  • the ⁇ modulator 24 has, as internal paths, a first path 42a-1 for feeding back the quantized data output from the quantizer 36 to the first loop filter 30a, and a quantization output from the quantizer 36. A first path 42a-2 for feeding back data to the second loop filter 30b.
  • the ⁇ modulator 24 further includes, as an internal path, a second path 42 b-1 connected from the first loop filter 30 a to the adder 15 for supplying the output of the first loop filter 30 a to the quantizer 36, and a second loop In order to provide the output of the filter 30 b to the quantizer 36, a second path 42 b-2 connected from the second loop filter 30 b to the adder 15 is provided.
  • the first loop filter 30a includes a first adder 32a, a transfer function block 33a of L 1 (z), a second adder 34a, and a feedforward path 35a. ing.
  • the second loop filter 30b also includes a first adder 32b, a transfer function block 33b of L 2 (z), a second adder 34b, and a feedforward path 35b. ing.
  • the first input signal U 1, which is input to the first loop filter 30a is, for example, a first 1RF signal center frequency f 1 (first target frequency f 1).
  • the second input signal U 2 which is input to the second loop filter 30b is a second 2RF signal center frequency f 2 (first target frequency f 2).
  • a plurality of (two) compensators 38 a corresponding to the number of input signals U 1 and U 2 . 38b is provided.
  • the plurality of compensators 38a and 38b include a first compensator 38a and a second compensator 38b.
  • the first compensator 38a outputs a first compensation signal.
  • the first compensation signal is for compensating for distortion generated in the first RF signal (frequency component of the first target frequency f 1 ) having the frequency f 1 expressed by the pulse train output from the ⁇ modulator 24.
  • the first compensation signal is given to the first path 42a-1 for feeding back the quantized data output from the quantizer 36 to the first loop filter 30a.
  • the first compensation signal is given to the first path 42a-1 by the adder 40a provided in the first path 42a-1.
  • the second compensator 38b outputs a second compensation signal.
  • the second compensation signal is for compensating for distortion generated in the second RF signal (frequency component of the second target frequency f 2 ) having the frequency f 2 expressed by the pulse train output from the ⁇ modulator 24.
  • the second compensation signal is given to the first path 42a-2 for feeding back the quantized data output from the quantizer 36 to the second loop filter 30b.
  • the second compensation signal is given to the first path 42a-2 by the adder 40b provided in the first path 42a-2.
  • the first compensation signal may be given to the second path 42b-1.
  • the second compensation signal may be given to the second path 42b-2.

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  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
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Abstract

This ΔΣ modulator is provided with: a loop filter 30; a quantizer 36 that generates quantized data on the basis of the output of the loop filter 30; an internal pathway 42 that is connected to the loop filter 30 or the quantizer 36; and a compensator 38 that applies, to the internal pathway 42, a compensation signal for compensation of distortion occurring in the frequency component at target frequencies among frequency components of a pulse sequence corresponding to the quantized data.

Description

ΔΣ変調器及び送信機ΔΣ modulator and transmitter
　本発明は、ΔΣ変調器及び送信機に関するものである。 The present invention relates to a ΔΣ modulator and a transmitter.
　特許文献１は、ΔΣ変調器を開示している。ΔΣ変調器は、基本構成として、ループフィルタ、量子化器、及び、ループフィルタと量子化器とを接続する内部経路、を有している。ΔΣ変調器は、入力信号をΔΣ変調し、量子化データを生成する。 Patent Document 1 discloses a ΔΣ modulator. The ΔΣ modulator has, as a basic configuration, a loop filter, a quantizer, and an internal path that connects the loop filter and the quantizer. The ΔΣ modulator performs ΔΣ modulation on the input signal to generate quantized data.
特開２０１４－１４０５９号公報JP 2014-14059 A
　ΔΣ変調器が生成する量子化データは、所定の周波数（ターゲット周波数）において、ＲＦ信号などのアナログ信号成分を含む。しかし、アナログ信号成分は、ΔΣ変調に起因して、歪を持つ場合がある。 Quantized data generated by the ΔΣ modulator includes an analog signal component such as an RF signal at a predetermined frequency (target frequency). However, the analog signal component may have distortion due to ΔΣ modulation.
　例えば、特許文献１は、ΔΣ変調器から出力されるパルス列のパルス波形が、ΔΣ変調器から出力されるパルス列によって表現されるアナログ信号の信号特性に影響を与えることを開示している。Δ変調器から出力されるパルス列のパルス波形が、パルス列によって表現されるアナログ信号に歪を生じさせる。
　アナログ信号の信号特性劣化を補償するために、特許文献１は、ΔΣ変調器から出力されたパルス列に対して処理を行うことを開示している。 For example, Patent Document 1 discloses that the pulse waveform of a pulse train output from a ΔΣ modulator affects the signal characteristics of an analog signal expressed by the pulse train output from the ΔΣ modulator. The pulse waveform of the pulse train output from the Δ modulator causes distortion in the analog signal represented by the pulse train.
In order to compensate for signal characteristic degradation of an analog signal, Patent Document 1 discloses that processing is performed on a pulse train output from a ΔΣ modulator.
　特許文献１では、ΔΣ変調に起因する歪を、ΔΣ変調の後に行われる処理で補償している。しかし、ΔΣ変調の後処理だけに頼らず、ΔΣ変調器自体で、ターゲット周波数の周波数成分に生じる歪の補償が行えると有利である。 In Patent Document 1, distortion caused by ΔΣ modulation is compensated by processing performed after ΔΣ modulation. However, it is advantageous if the distortion generated in the frequency component of the target frequency can be compensated by the ΔΣ modulator itself without relying only on the post-processing of ΔΣ modulation.
　一の観点からみた本発明は、ループフィルタと、前記ループフィルタの出力に基づいて、量子化データを生成する量子化器と、前記ループフィルタ又は前記量子化器に接続された内部経路と、前記量子化データに対応したパルス列の周波数成分のうちターゲット周波数の周波数成分に生じる歪の補償信号を、前記内部経路に与える補償器と、を備えるΔΣ変調器である。 The present invention from one aspect includes a loop filter, a quantizer that generates quantized data based on an output of the loop filter, an internal path connected to the loop filter or the quantizer, A delta-sigma modulator comprising: a compensator that applies a compensation signal for distortion generated in a frequency component of a target frequency among frequency components of a pulse train corresponding to quantized data to the internal path.
　本発明によれば、ΔΣ変調器自体で、ターゲット周波数の周波数成分に生じる歪の補償が行える。 According to the present invention, the ΔΣ modulator itself can compensate for distortion occurring in the frequency component of the target frequency.
送信機（通信システム）のブロック図である。It is a block diagram of a transmitter (communication system). ΔΣ変調器のブロック図である。It is a block diagram of a ΔΣ modulator. ΔΣ変調器の他の例を示すブロック図である。It is a block diagram which shows the other example of a delta-sigma modulator. Ｌ（ｚ）／（１＋Ｌ（ｚ））の周波数特性図である。It is a frequency characteristic figure of L (z) / (1 + L (z)). 非対称波形の波形図である。It is a wave form diagram of an asymmetrical waveform. 補償器のブロック図である。It is a block diagram of a compensator. 非対称波形と検出信号とを示す波形図である。It is a wave form diagram which shows an asymmetrical waveform and a detection signal. 係数決定処理のフローチャートである。It is a flowchart of a coefficient determination process. 係数決定処理中のＡＣＬＲを示す図である。It is a figure which shows ACLR in a coefficient determination process. 補償信号生成部の他の例を示すブロック図である。It is a block diagram which shows the other example of a compensation signal production | generation part. ＲＦ信号のアイパターンである。It is an eye pattern of an RF signal. ＲＦ信号の周波数スペクトラムである。This is the frequency spectrum of the RF signal. 図１０よりも広帯域の周波数スペクトラムである。The frequency spectrum is wider than that in FIG. デュアルバンドΔΣ変調器のブロック図である。It is a block diagram of a dual band ΔΣ modulator.
　以下、実施形態について図面を参照しながら説明する。 Hereinafter, embodiments will be described with reference to the drawings.
［１．実施形態の概要］
（１）実施形態に係るΔΣ変調器は、ループフィルタと、前記ループフィルタの出力に基づいて、量子化データを生成する量子化器と、前記ループフィルタ又は前記量子化器に接続された内部経路と、前記量子化データに対応したパルス列の周波数成分のうちターゲット周波数の周波数成分に生じる歪の補償信号を、前記内部経路に与える補償器と、を備える。ΔΣ変調器が、補償器を備えていることで、ΔΣ変調器自体が、ターゲット周波数の周波数成分に生じる歪を補償することができる。 [1. Outline of Embodiment]
(1) A ΔΣ modulator according to an embodiment includes a loop filter, a quantizer that generates quantized data based on an output of the loop filter, and an internal path connected to the loop filter or the quantizer And a compensator that provides a compensation signal for distortion generated in the frequency component of the target frequency among the frequency components of the pulse train corresponding to the quantized data to the internal path. Since the ΔΣ modulator includes a compensator, the ΔΣ modulator itself can compensate for distortion occurring in the frequency component of the target frequency.
（２）前記歪は、前記量子化データに対応したパルス列におけるパルスの立ち上がりと立ち下がりとの非対称性によって、前記ターゲット周波数の周波数成分に生じる歪を含むのが好ましい。この場合、パルス列におけるパルスの立ち上がりと立ち下がりとの非対称性があっても、ΔΣ変調器から出力される量子化データは、非対称性に起因してターゲット周波数の周波数成分に生じる歪が補償されたものとなる。 (2) It is preferable that the distortion includes distortion generated in a frequency component of the target frequency due to asymmetry between the rising edge and the falling edge of the pulse in the pulse train corresponding to the quantized data. In this case, even if there is an asymmetry between the rising edge and the falling edge of the pulse in the pulse train, the quantized data output from the ΔΣ modulator is compensated for the distortion generated in the frequency component of the target frequency due to the asymmetry. It will be a thing.
（３）前記補償器は、前記量子化データの変化を検出する検出器と、前記検出器によって検出された前記量子化データの変化に基づいて補償信号を生成する生成器と、を備えるのが好ましい。量子化データの変化を検出することで、パルス列におけるパルスの立ち上がりと立ち下がりで生じる非対称性の発生を検出することができる。したがって、量子化データの変化に基づいて補償信号を生成することで、非対称性の発生に基づいて補償信号を生成することができる。 (3) The compensator includes a detector that detects a change in the quantized data, and a generator that generates a compensation signal based on the change in the quantized data detected by the detector. preferable. By detecting a change in quantized data, it is possible to detect the occurrence of asymmetry that occurs at the rise and fall of the pulse in the pulse train. Therefore, by generating the compensation signal based on the change in the quantized data, the compensation signal can be generated based on the occurrence of asymmetry.
（４）前記検出器は、前記量子化データが変化したタイミングで検出信号を出力するのが好ましい。この場合、生成器は、量子化データが変化したタイミングで出力された検出信号に基づいて補償信号を生成することができる。 (4) It is preferable that the detector outputs a detection signal at a timing when the quantized data changes. In this case, the generator can generate a compensation signal based on the detection signal output at the timing when the quantized data changes.
（５）前記生成器は、フラクショナルディレイを含み、前記フラクショナルディレイは、前記検出信号に基づいて前記補償信号を生成するのが好ましい。フラクショナルディレイによって、非対称性に起因してターゲット周波数の周波数成分に生じる歪を補償する補償信号を生成するのが容易となる。 (5) Preferably, the generator includes a fractional delay, and the fractional delay generates the compensation signal based on the detection signal. The fractional delay makes it easy to generate a compensation signal that compensates for distortion that occurs in the frequency component of the target frequency due to asymmetry.
（６）前記量子化器は、１ｂｉｔ量子化器であるのが好ましい。１ｂｉｔ量子化データの場合、パルス列の立ち上がりと立ち下がりの非対称性が問題となり易いため、ΔΣ変調器自体で補償を行うことが有利である。 (6) The quantizer is preferably a 1-bit quantizer. In the case of 1-bit quantized data, the asymmetry of the rise and fall of the pulse train is likely to be a problem, and it is advantageous to perform compensation with the ΔΣ modulator itself.
（７）前記内部経路は、前記量子化データを前記ループフィルタへフィードバックするための第１経路を含み、前記補償器は、前記補償信号を前記第１経路に与えるのが好ましい。補償信号をループフィルタへフィードバックするための第１経路に与えることで、動作が安定し易くなる。 (7) Preferably, the internal path includes a first path for feeding back the quantized data to the loop filter, and the compensator provides the compensation signal to the first path. By providing the compensation signal to the first path for feeding back to the loop filter, the operation is easily stabilized.
（８）前記内部経路は、前記ループフィルタの出力を前記量子化器へ与えるための第２経路を含み、前記補償器は、前記補償信号を前記第２経路に与えてもよい。この場合、補償信号による補償が可能である。
（９）前記内部経路は、前記ループフィルタの出力を前記量子化器へ与えるための第２経路を含み、前記補償器は、前記補償信号を前記第１経路及び前記第２経路に与えるのが好ましい。この場合、広帯域の歪補償が可能である。 (8) The internal path may include a second path for supplying the output of the loop filter to the quantizer, and the compensator may supply the compensation signal to the second path. In this case, compensation using a compensation signal is possible.
(9) The internal path includes a second path for supplying the output of the loop filter to the quantizer, and the compensator supplies the compensation signal to the first path and the second path. preferable. In this case, wideband distortion compensation is possible.
（１０）実施形態に係る送信機は、前記量子化データに対応したパルスを、送信信号として出力する。 (10) The transmitter according to the embodiment outputs a pulse corresponding to the quantized data as a transmission signal.
［２．実施形態の詳細］
［２．１　送信機の構成］
　図１は、送信機１０を示している。この送信機１０は、デジタル信号処理部１２と、アナログフィルタ１６と、を有している。デジタル信号処理部１２は、アナログ信号であるＲＦ（Ｒａｄｉｏ　Ｆｒｅｑｕｅｎｃｙ）信号を表現するデジタル信号（１ｂｉｔパルス列）を出力する。ＲＦ信号は、無線波として空間に放射される信号であり、例えば、移動体通信のためのＲＦ信号、又は放送サービスのためのＲＦ信号である。 [2. Details of Embodiment]
[2.1 Transmitter configuration]
FIG. 1 shows a transmitter 10. The transmitter 10 includes a digital signal processing unit 12 and an analog filter 16. The digital signal processing unit 12 outputs a digital signal (1-bit pulse train) that represents an RF (Radio Frequency) signal that is an analog signal. The RF signal is a signal radiated into the space as a radio wave, for example, an RF signal for mobile communication or an RF signal for broadcasting service.
　デジタル信号処理部１２から出力されたパルス列は、アナログフィルタ（バンドパスフィルタ又はローパスフィルタ）１６に与えられる。デジタル信号処理部１２から出力されたパルス列が表現するアナログ信号は、ＲＦ信号の周波数（ターゲット周波数）以外の周波数の成分を雑音成分として含んでいる。その雑音成分は、アナログフィルタによって除去される。 The pulse train output from the digital signal processing unit 12 is given to an analog filter (bandpass filter or lowpass filter) 16. The analog signal represented by the pulse train output from the digital signal processing unit 12 includes a frequency component other than the frequency (target frequency) of the RF signal as a noise component. The noise component is removed by the analog filter.
　デジタル信号処理部１２とアナログフィルタ１６との間の信号伝送路１４は、回路基板に形成された信号配線であってもよいし、光ファイバー又は電気ケーブルなどの通信ケーブルであってもよい。 The signal transmission path 14 between the digital signal processing unit 12 and the analog filter 16 may be a signal wiring formed on a circuit board, or a communication cable such as an optical fiber or an electric cable.
　デジタル信号処理部１２は、送信信号であるベースバンド信号（ＩＱ信号）を出力するベースバンド部１８と、ベースバンド信号を変調する変調器（直交変調器）２０と、処理部２２と、ΔΣ変調器２４と、コントローラ２６と、を備えている。 The digital signal processing unit 12 includes a baseband unit 18 that outputs a baseband signal (IQ signal) that is a transmission signal, a modulator (orthogonal modulator) 20 that modulates the baseband signal, a processing unit 22, and ΔΣ modulation. A device 24 and a controller 26 are provided.
　ベースバンド部１８は、ＩＱベースバンド信号（Ｉ信号、Ｑ信号）をデジタルデータとして出力する。直交変調器２０は、ＩＱベースバンド信号を、デジタル信号処理で直交変調を行うデジタル直交変調器として構成されている。処理部２２は、直交変調２０から出力された直交変調信号に対して、デジタル信号処理を施し、デジタルＩＦ信号を出力する。処理部２２が行うデジタル信号処理は、例えば、デジタル前置歪補償（Ｄｉｇｉｔａｌ　Ｐｒｅ－Ｄｉｓｔｏｒｔｉｏｎ；ＤＰＤ）、クレストファクタリダクション（Ｃｒｅｓｔ　Ｆａｃｔｏｒ　Ｒｅｄｕｃｔｉｏｎ；ＣＦＲ）、デジタルアップコンバージョン（Ｄｉｇｉｔａｌ　Ｕｐ　Ｃｏｎｖｅｒｓｉｏｎ；ＤＵＣ）を含む。 The baseband unit 18 outputs an IQ baseband signal (I signal, Q signal) as digital data. The quadrature modulator 20 is configured as a digital quadrature modulator that performs quadrature modulation on an IQ baseband signal by digital signal processing. The processing unit 22 performs digital signal processing on the quadrature modulation signal output from the quadrature modulation 20 and outputs a digital IF signal. The digital signal processing performed by the processing unit 22 includes, for example, digital pre-distortion (DPD), crest factor reduction (Crest Factor Reduction; CFR), and digital up-conversion (Digital Up Conversion; DUC).
　処理部２２から出力されるデジタルＲＦ信号は、中心周波数がｆ０であり、所定の帯域を持つ。処理部２２から出力されるデジタルＲＦ信号は、ΔΣ変調器２４に与えられる。ΔΣ変調器２４は、デジタルＲＦ信号に対してΔΣ変調を行って、デジタルＲＦ信号を量子化し、量子化データ（パルス列）を出力する。ΔΣ変調器２４から出力されたパルス列は、アナログＲＦ信号を表現したものとなっている。送信機１０は、このパルス列を送信信号として送信する。 Digital RF signal output from the processor 22, the center frequency is f 0, having a predetermined bandwidth. The digital RF signal output from the processing unit 22 is given to the ΔΣ modulator 24. The ΔΣ modulator 24 performs ΔΣ modulation on the digital RF signal, quantizes the digital RF signal, and outputs quantized data (pulse train). The pulse train output from the ΔΣ modulator 24 represents an analog RF signal. The transmitter 10 transmits this pulse train as a transmission signal.
［２．２　ΔΣ変調］
　ΔΣ変調器２４がバンドパス型である場合、ΔΣ変調器２４は、所望の周波数（ターゲット周波数ｆ０）の信号成分を通過させ、ターゲット周波数ｆ０近傍の帯域の雑音を帯域外に移行させるノイズシェイピングを行う。図２Ａに示すように、ΔΣ変調器２４は、ループフィルタ３０と、量子化データを出力する量子化器３６と、ループフィルタ３０又は量子化器３６に接続された内部経路４２と、補償器３８と、加算器４０と、を備えている。内部経路４２は、量子化器３６から出力された量子化データをループフィルタ３０へフィードバックするための第１経路（フィードバック経路）４２ａと、ループフィルタ３０の出力を量子化器３６に与えるための第２経路４２ｂと、を含む。ΔΣ変調器２４の外部のみへ信号が流れ出る経路は、ΔΣ変調器２４の外部経路と実質的に同じであるため、内部経路４２には含まれない。内部経路４２は、ΔΣ変調器２４の出力を生成する量子化器３６に至ることができる経路である。 [2.2 ΔΣ modulation]
When the ΔΣ modulator 24 is a bandpass type, the ΔΣ modulator 24 passes a signal component of a desired frequency (target frequency f 0 ), and causes noise in a band near the target frequency f 0 to shift out of the band. Perform shaping. As shown in FIG. 2A, the ΔΣ modulator 24 includes a loop filter 30, a quantizer 36 that outputs quantized data, an internal path 42 connected to the loop filter 30 or the quantizer 36, and a compensator 38. And an adder 40. The internal path 42 includes a first path (feedback path) 42 a for feeding back the quantized data output from the quantizer 36 to the loop filter 30, and a first path for supplying the output of the loop filter 30 to the quantizer 36. 2 paths 42b. The path through which the signal flows only to the outside of the ΔΣ modulator 24 is substantially the same as the external path of the ΔΣ modulator 24 and is not included in the internal path 42. The internal path 42 is a path that can reach the quantizer 36 that generates the output of the ΔΣ modulator 24.
　ループフィルタ３０は、２入力１出力であり、ΔΣ変調器２４への入力信号（デジタルＲＦ信号）と、量子化器３６側からのフィードバック信号と、が入力される。ループフィルタ３０は、第１加算器３２と、Ｌ（ｚ）の伝達関数ブロック３３と、第２加算器３４と、フィードフォワード経路３５と、を備えている。 The loop filter 30 has two inputs and one output, and receives an input signal (digital RF signal) to the ΔΣ modulator 24 and a feedback signal from the quantizer 36 side. The loop filter 30 includes a first adder 32, an L (z) transfer function block 33, a second adder 34, and a feedforward path 35.
　第１加算器３２は、ΔΣ変調器２４への入力信号と、量子化器３６側からのフィードバック信号と、を加算する。フィードバック信号は、第１経路４２ａ介して、第１加算器３２へ与えられる。伝達関数ブロック３３の伝達関数Ｌ（ｚ）は、Δ変調器２４としての特性を決定するものであり、所望の信号伝達関数及び雑音伝達関数に基づいて、決定される。第２加算器３４は、伝達関数ブロック３３の出力とΔΣ変調器２４への入力信号と、を加算する。入力信号は、フィードフォワード経路３５を介して、第２加算器３４に与えられる。なお、フィードフォワード経路３５及び第２加算器３４は省略してもよい。 The first adder 32 adds the input signal to the ΔΣ modulator 24 and the feedback signal from the quantizer 36 side. The feedback signal is given to the first adder 32 through the first path 42a. The transfer function L (z) of the transfer function block 33 determines the characteristics of the Δ modulator 24 and is determined based on a desired signal transfer function and noise transfer function. The second adder 34 adds the output of the transfer function block 33 and the input signal to the ΔΣ modulator 24. The input signal is given to the second adder 34 via the feedforward path 35. The feedforward path 35 and the second adder 34 may be omitted.
　第２加算器３４の出力、つまりループフィルタ３０の出力、は、第２経路４２ｂを介して、量子化器３６に与えられる。量子化器３６は、１ｂｉｔ量子化器であり、ループフィルタ３０の出力を１ｂｉｔに量子化した１ｂｉｔ量子化データを出力する。量子化器３６から出力された量子化データ（パルス列）は、ΔΣ変調器２４の出力として、ΔΣ変調器２４から出力されるとともに、第１経路４２ａを介して、ループフィルタ３０へフィードバックされる。 The output of the second adder 34, that is, the output of the loop filter 30 is given to the quantizer 36 through the second path 42b. The quantizer 36 is a 1-bit quantizer, and outputs 1-bit quantized data obtained by quantizing the output of the loop filter 30 into 1 bit. The quantized data (pulse train) output from the quantizer 36 is output from the ΔΣ modulator 24 as an output of the ΔΣ modulator 24 and is fed back to the loop filter 30 via the first path 42a.
　補償器３８は、ΔΣ変調器２４から出力されるパルス列が表現する周波数ｆ０のアナログＲＦ信号（ターゲット周波数ｆ０の周波数成分）に生じる歪の補償信号Ｃを出力する。補償信号Ｃは、ＲＦ信号に生じる歪を打消し又は抑制するためのものである。補償器３８から出力された補償信号Ｃは、第１経路４２ａに設けられた加算器４０によって、第１経路４２ａに与えられる。歪を打消し又は抑制するための補償信号ＣがΔΣ変調器２４の内部経路４２に与えられることで、Δ変調器２４の内部で、歪の補償が行われる。この結果、ΔΣ変調器２４から出力された量子化データは、歪が補償されたＲＦ信号（ターゲット周波数ｆ０の周波数成分）を表現するものとなる。 The compensator 38 outputs a compensation signal C for distortion generated in the analog RF signal (frequency component of the target frequency f 0 ) having the frequency f 0 expressed by the pulse train output from the ΔΣ modulator 24. The compensation signal C is for canceling or suppressing distortion generated in the RF signal. The compensation signal C output from the compensator 38 is given to the first path 42a by the adder 40 provided in the first path 42a. The compensation signal C for canceling or suppressing the distortion is supplied to the internal path 42 of the ΔΣ modulator 24, so that the distortion is compensated inside the Δ modulator 24. As a result, the quantized data output from the ΔΣ modulator 24 represents an RF signal (frequency component of the target frequency f 0 ) whose distortion has been compensated.
　ΔΣ変調器２４は、フィードバック経路である第１経路４２ａを有しているため、内部経路４２のいずれの位置に補償信号を与えても、その補償信号は、伝達関数ブロック３３をいずれ通過することになるため、補償信号による歪補償がなされた状態で、所望の特性のΔΣ変調を行うことができる。ただし、図２Ａに示すように補償信号を第１経路４２ａに与えると、その補償信号は、量子化器３６よりも前に伝達関数ブロック３３を通過することになるため、動作が安定し易い。 Since the ΔΣ modulator 24 has the first path 42a that is a feedback path, even if a compensation signal is given to any position of the internal path 42, the compensation signal passes through the transfer function block 33. Therefore, ΔΣ modulation having a desired characteristic can be performed in a state where distortion compensation by the compensation signal is performed. However, when the compensation signal is given to the first path 42a as shown in FIG. 2A, the compensation signal passes through the transfer function block 33 before the quantizer 36, and thus the operation is easily stabilized.
　図２Ａでは、補償信号は、第１経路４２ａに与えられるが、他の内部経路４２（例えば、第２経路４２ｂ）に与えられても良い。すなわち、図２Ｂに示すように、補償器３８が出力する補償信号Ｃは、第１経路４２ａに与えられるだけでなく、第２経路４２ｂに設けられた第２加算器（加減算器）３４によって、第２経路４２ｂに与えられても良い。第２経路４２ｂに与えられる補償信号Ｃは、第２経路４２ｂを流れる信号から補償信号Ｃを減算するように、第２経路４２ｂに与えられる。 In FIG. 2A, the compensation signal is given to the first path 42a, but may be given to another internal path 42 (for example, the second path 42b). That is, as shown in FIG. 2B, the compensation signal C output from the compensator 38 is not only supplied to the first path 42a, but also by a second adder (adder / subtracter) 34 provided in the second path 42b. It may be given to the second path 42b. The compensation signal C given to the second path 42b is given to the second path 42b so as to subtract the compensation signal C from the signal flowing through the second path 42b.
　補償信号Ｃを第１経路４２ａ及び第２経路４２ｂに与えた場合、歪補償を広帯域に行うことができる。すなわち、図２Ａに示すΔΣ変調器２４の場合、ΔΣ変調器２４の出力Ｖは、次の式（１）によって表される。
Figure JPOXMLDOC01-appb-M000001
 　式（１）の右辺の第１項は入力信号Ｕであり、第２項はΔΣ変調器２４の雑音伝達関数ＮＴＦによるフィルタ特性が掛けられた量子化雑音Ｅを示す。式（１）の右辺の第１項及び第２項だけであれば、通常のΔΣ変調器の出力Ｖとなる。図２Ａに示すΔΣ変調器２４では、補償信号Ｃを第１経路４２ａに与えているため、式（１）の右辺の第３項が発生する。式（１）では、補償信号ＣにＬ（ｚ）／（１＋Ｌ（ｚ））のフィルタ特性が掛けられたものが、出力Ｖに反映されることになる。Ｌ（ｚ）／（１＋Ｌ（ｚ））の周波数特性は、図２Ｃに示すように、バンドパスとなる。つまり、図２Ｃに示すように、Ｌ（ｚ）／（１＋Ｌ（ｚ））９５０ＭＨｚから１０５０ＭＨｚを通過帯域として持つバンドパスフィルタとなる。このため、補償信号Ｃは、バンドパスフィルタの通過帯域における成分しか、出力Ｖに反映されず、比較的狭帯域での歪補償となる。 When the compensation signal C is given to the first path 42a and the second path 42b, distortion compensation can be performed over a wide band. That is, in the case of the ΔΣ modulator 24 shown in FIG. 2A, the output V of the ΔΣ modulator 24 is expressed by the following equation (1).
Figure JPOXMLDOC01-appb-M000001
 The first term on the right side of the equation (1) is the input signal U, and the second term represents the quantization noise E multiplied by the filter characteristic by the noise transfer function NTF of the ΔΣ modulator 24. If only the first term and the second term on the right side of Equation (1) are used, the output V of the normal ΔΣ modulator is obtained. In the ΔΣ modulator 24 shown in FIG. 2A, since the compensation signal C is given to the first path 42a, the third term on the right side of Expression (1) is generated. In equation (1), the compensation signal C multiplied by the filter characteristic of L (z) / (1 + L (z)) is reflected in the output V. The frequency characteristic of L (z) / (1 + L (z)) is a band pass as shown in FIG. 2C. That is, as shown in FIG. 2C, a band-pass filter having a pass band from L (z) / (1 + L (z)) 950 MHz to 1050 MHz. For this reason, only the component in the passband of the bandpass filter is reflected in the output V of the compensation signal C, and distortion compensation is performed in a relatively narrow band.
　これに対し、図２Ｂに示すように、補償信号ｃを第２経路４２ｂにも与えると、ΔΣ変調器２４の出力は、式（１）におけるＣの係数は１となり、次の式（２）によって表される。
Figure JPOXMLDOC01-appb-M000002
 　式（２）の右辺の第３項は、補償信号Ｃの全周波数成分が出力Ｖから減算されるため、広帯域での歪補償が可能である。 On the other hand, as shown in FIG. 2B, when the compensation signal c is also applied to the second path 42b, the output of the ΔΣ modulator 24 has a C coefficient of 1 in equation (1), and the following equation (2) Represented by
Figure JPOXMLDOC01-appb-M000002
 In the third term on the right side of Equation (2), since all the frequency components of the compensation signal C are subtracted from the output V, distortion compensation in a wide band is possible.
［２．３　パルス列の波形歪によって生じるＲＦ信号の歪とその補償］
　本実施形態では、補償器３８から出力される補償信号によって補償される歪の一例として、ΔΣ変調器２４から出力されるパルス列の波形歪によってＲＦ信号に生じる歪を想定する。ΔΣ変調器２４は、量子化データをパルス列として出力するため、そのパルス列の波形が歪んでいると、そのパルス列が表現するＲＦ信号に歪が生じる。具体的には、特許文献１に示すように、量子化データに対応したパルス列におけるパルスの立ち上がりと立ち下がりとの非対称性によって、ＲＦ信号（ターゲット周波数ｆ０の周波数成分）に歪が生じる。 [2.3 Distortion of RF signal caused by waveform distortion of pulse train and its compensation]
In the present embodiment, as an example of the distortion compensated by the compensation signal output from the compensator 38, the distortion generated in the RF signal due to the waveform distortion of the pulse train output from the ΔΣ modulator 24 is assumed. Since the ΔΣ modulator 24 outputs the quantized data as a pulse train, if the waveform of the pulse train is distorted, the RF signal represented by the pulse train is distorted. Specifically, as shown in Patent Document 1, distortion occurs in the RF signal (frequency component of the target frequency f 0 ) due to asymmetry between the rising edge and the falling edge of the pulse in the pulse train corresponding to the quantized data.
　ΔΣ変調器２４は、量子化データに対応するパルス列を出力するために、図示しないドライバを有している。ドライバは、スイッチング素子などを有しており、スイッチング素子のＯＮ／ＯＦＦ動作によってパルスの立ち上がりと立ち下がりが形成される。ドライバによって形成されるパルスの立ち上がり時間と立ち下がり時間とは一致しないことが一般的であり、パルスの立ち上がりと立ち下がりとの非対称性が生じる。この非対称成分がＲＦ信号を劣化させる。以下、パルスの立ち上がりと立ち下がりの非対称成分について定義する。 The ΔΣ modulator 24 has a driver (not shown) in order to output a pulse train corresponding to the quantized data. The driver has a switching element and the like, and the rise and fall of the pulse are formed by the ON / OFF operation of the switching element. Generally, the rise time and fall time of the pulse formed by the driver do not coincide with each other, and asymmetry occurs between the rise and fall of the pulse. This asymmetric component degrades the RF signal. In the following, the asymmetrical components of the rise and fall of the pulse are defined.
　まず、ΔΣ変調器２４から出力されるパルス列Ｓｏｕｔ（ｔ）は、下記式（Ａ）のように定義される。
Figure JPOXMLDOC01-appb-M000003
 First, the pulse train S out (t) output from the ΔΣ modulator 24 is defined as the following equation (A).
Figure JPOXMLDOC01-appb-M000003
　式（Ａ）の第１項であるＳＩｄｅａｌは、量子化データｄｋ（＝±１）を理想的な矩形波で表現したものであり、式（Ｂ）のように定義される。ここでは、量子化データｄｋは、パルスのＨｉｇｈレベルに対応した値として＋１をとり、パルスのＬｏｗレベルに対応した値として－１をとる。Ｕ（ｔ）は、単位ステップ関数である。
Figure JPOXMLDOC01-appb-M000004
 S Ideal , which is the first term of the equation (A), represents the quantized data d k (= ± 1) with an ideal rectangular wave, and is defined as the equation (B). Here, the quantized data d k takes +1 as a value corresponding to the high level of the pulse and −1 as a value corresponding to the low level of the pulse. U (t) is a unit step function.
Figure JPOXMLDOC01-appb-M000004
　式（Ａ）の第２項は、実際の波形に相当するＳｏｕｔ（ｔ）と、理想的な波形ＳＩｄｅａｌとの差を示している。第２項におけるｆ（ｔ－ｋｔ）は、下記式（Ｃ）のように定義される。Ｓｉｇｎは、符号関数である。
Figure JPOXMLDOC01-appb-M000005
Figure JPOXMLDOC01-appb-M000006
 The second term of the equation (A) indicates the difference between S out (t) corresponding to the actual waveform and the ideal waveform S Ideal . F (t−kt) in the second term is defined as in the following formula (C). Sign is a sign function.
Figure JPOXMLDOC01-appb-M000005
Figure JPOXMLDOC01-appb-M000006
　式（Ｃ）において、（Ｃ－１）は、ある量子化データの値ｄｋと時間的に一つ前の量子化データの値ｄｋ－１との差分を示す値の符号がプラスである場合、すなわち、量子化データｄｋに対応したパルスが、立ち上がる場合を示す。
　（Ｃ－２）は、ある量子化データの値ｄｋと時間的に一つ前の量子化データの値ｄｋ－１との差分を示す値の符号がマイナスである場合、すなわち、量子化信号ｄｋに対応したパルスが立ち下がる場合を示す。
　（Ｃ－３）は、ある量子化データの値ｄｋと時間的に一つ前の量子化データの値ｄｋ－１との差分を示す値がゼロである場合、すなわち、パルスの値に変化がない場合である。 In the expression (C), (C-1) is positive in sign of a value indicating a difference between a certain quantized data value d k and a temporally previous quantized data value d k−1. In other words, the case where the pulse corresponding to the quantized data d k rises is shown.
(C-2), if the sign of the value that indicates the difference between the value d k-1 value d k and temporally preceding quantized data of a quantized data is negative, i.e., quantization It shows a case where pulse corresponding to the signal d k falls.
(C-3) is a case where a value indicating a difference between a certain quantized data value d k and a temporally previous quantized data value d k−1 is zero, that is, a pulse value. This is the case when there is no change.
　ｆｒｉｓｅ（ｔ）とｆｆａｌｌ（ｔ）は、それぞれ、パルスの立ち上がり波形と立ち下がり波形である。ｆｒｉｓｅ（ｔ）とｆｆａｌｌ（ｔ）は、式（Ｄ）に示すように、対称成分ｆｓｙｍ（ｔ）と非対称成分ｆＡｓｙｍ（ｔ）に分解することができる。
　非対称成分ｆＡｓｙｍ（ｔ）は、式（Ｄ）より、下記式（Ｅ）によって求めることができる。
Figure JPOXMLDOC01-appb-M000007
 f rise (t) and f fall (t) are the rising waveform and falling waveform of the pulse, respectively. f rise (t) and f fall (t) can be decomposed into a symmetric component f sym (t) and an asymmetric component f Asym (t) as shown in equation (D).
The asymmetric component f Asym (t) can be obtained from the following formula (E) from the formula (D).
Figure JPOXMLDOC01-appb-M000007
　式（Ｅ）は、立ち上がり波形ｆｒｉｓｅ（ｔ）と立ち下がり波形ｆｆａｌｌ（ｔ）とが、下記式（Ｆ）の関係を有している場合に、非対称成分ｆＡｓｙｍ（ｔ）が無くなることを示している。
Figure JPOXMLDOC01-appb-M000008
 Equation (E) shows that the asymmetric component f Asym (t) disappears when the rising waveform f rise (t) and the falling waveform f fall (t) have the relationship of the following equation (F). Is shown.
Figure JPOXMLDOC01-appb-M000008
　図３は、式（Ｆ）を満たさないパルス波形（非対称成分を有する非対称波形）を示している。図３（ａ）は、非対称波形Ｓｏｕｔ（ｔ）のアイパターンを示している。このアイパターンは、時間軸に対して非対称となっている。具体的には、図３に示す非対称波形は、パルスの立ち上がり時間よりも、パルスの立ち下がり時間の方が長い波形となっている。 FIG. 3 shows a pulse waveform (asymmetric waveform having an asymmetric component) that does not satisfy the equation (F). FIG. 3A shows an eye pattern of the asymmetric waveform S out (t). This eye pattern is asymmetric with respect to the time axis. Specifically, the asymmetric waveform shown in FIG. 3 is a waveform in which the pulse fall time is longer than the pulse rise time.
　図３（ｂ）は、非対称波形Ｓｏｕｔ（ｔ）の時間軸波形を示し、図３（ｃ）は、非対称波形についての理想的な波形ＳＩｄｅａｌ（ｔ）を示し、図３（ｄ）は、非対称波形における立ち上がり波形ｆｒｉｓｅ（ｔ）と立ち下がり波形ｆｆａｌｌ（ｔ）における対称成分ｆｓｙｍ（ｔ）を示し、図３（ｅ）は、非対称波形における立ち上がり波形ｆｒｉｓｅ（ｔ）と立ち下がり波形ｆｆａｌｌ（ｔ）における非対称成分ｆＡｓｙｍ（ｔ）を示している。 3B shows a time axis waveform of the asymmetric waveform S out (t), FIG. 3C shows an ideal waveform S Ideal (t) for the asymmetric waveform, and FIG. FIG. 3E shows the symmetric component f sym (t) in the rising waveform f rise (t) and the falling waveform f fall (t) in the asymmetric waveform, and FIG. 3E shows the rising waveform f rise (t) and the rising in the asymmetric waveform. The asymmetric component f Asym (t) in the falling waveform f fall (t) is shown.
　図３に示すように、非対称波形は、理想的な波形ＳＩｄｅａｌ（ｔ）に対して歪んでおり、歪成分を有する。具体的には、パルスの立ち上がり波形ｆｒｉｓｅ（ｔ）に歪成分（第１の歪成分）を有するとともに、パルスの立ち下がり波形ｆｆａｌｌ（ｔ）に歪成分（第２の歪成分）を有する。 As shown in FIG. 3, the asymmetric waveform is distorted with respect to the ideal waveform S Ideal (t) and has a distortion component. Specifically, the pulse rising waveform f rise (t) has a distortion component (first distortion component), and the pulse falling waveform f fall (t) has a distortion component (second distortion component). .
　式（Ｆ）を満たさない場合、歪成分は、対称成分ｆｓｙｍ（ｔ）とともに、非対称成分ｆＡｓｙｍ（ｔ）を有する（図３（ｄ）、図３（ｅ）参照）。歪成分のうち、対称成分ｆｓｙｍ（ｔ）の存在は、ＲＦ信号の特性（例えば、隣接チャネル漏洩電力（ＡＣＬＲ））に及ぼす影響は少ないが、非対称成分ｆＡｓｙｍ（ｔ）はＲＦ信号の特性に影響を及ぼす（特許文献１参照）。つまり、ΔΣ変調器２４が出力するパルスの形状が、ΔΣ変調器２４によって処理される対象であるＲＦ信号（ターゲット周波数ｆ０の周波数成分）に影響を及ぼす。 When Expression (F) is not satisfied, the distortion component has an asymmetric component f Asym (t) together with a symmetric component f sym (t) (see FIGS. 3D and 3E ). Among the distortion components, the presence of the symmetric component f sym (t) has little influence on the characteristics of the RF signal (for example, adjacent channel leakage power (ACLR)), but the asymmetric component f Asym (t) is a characteristic of the RF signal. (See Patent Document 1). In other words, the shape of the pulse output from the ΔΣ modulator 24 affects the RF signal (frequency component of the target frequency f 0 ) that is to be processed by the ΔΣ modulator 24.
　本実施形態では、パルス列の波形歪（非対称成分）によって生じることになるＲＦ信号の歪が、パルス列が出力される前に、ΔΣ変調器２４内部で、補償信号によって予め補償される。したがって、ΔΣ変調器２４から出力されるパルス波形が非対称成分を有していても、ＲＦ信号のＡＣＬＲの劣化が抑制される。 In this embodiment, the distortion of the RF signal caused by the waveform distortion (asymmetrical component) of the pulse train is compensated in advance by the compensation signal inside the ΔΣ modulator 24 before the pulse train is output. Therefore, even if the pulse waveform output from the ΔΣ modulator 24 has an asymmetric component, degradation of the ACLR of the RF signal is suppressed.
　図４は、パルス列の立ち上がりと立ち下がりの非対称成分による歪を補償するのに適した補償器３８の例を示している。補償器３８は、検出器４４と、補償信号生成器４６と、を有している。 FIG. 4 shows an example of a compensator 38 suitable for compensating for distortion due to an asymmetric component of the rise and fall of the pulse train. The compensator 38 includes a detector 44 and a compensation signal generator 46.
　検出器４４は、量子化データの変化（パルス列の立ち上がり又は立ち下がり）を検出する。非対称成分は、パルス列の立ち上がり又は立ち下がりにおいて生じるため、パルス列の立ち上がり又は立ち下がりを検出することで、非対称成分の発生を検出することができる。検出器４４には、量子化器３６から出力された量子化データ（パルス列）が入力として与えられる。検出器４４は、量子化データが変化したタイミングで検出信号（パルス状検出信号）を出力する。 The detector 44 detects a change in quantized data (rising or falling of the pulse train). Since the asymmetric component occurs at the rise or fall of the pulse train, the occurrence of the asymmetric component can be detected by detecting the rise or fall of the pulse train. The detector 44 is supplied with the quantized data (pulse train) output from the quantizer 36 as an input. The detector 44 outputs a detection signal (pulse detection signal) at the timing when the quantized data changes.
　例えば、ΔΣ変調器２４における１サンプリングクロック毎の量子化データが図５（ａ）に示すように変化する場合、ΔΣ変調器２４から出力されるパルス列は、図５（ｂ）に示すようになる。非対称成分は、図５（ｃ）に示すように、図５（ｂ）のパルス列の立ち上がりと立ち下がりで生じる。図５（ｄ）に示すように、検出器４４は、非対称成分の発生タイミングに合わせて検出信号（量子化データ変化検出信号）を出力する。 For example, when the quantized data for each sampling clock in the ΔΣ modulator 24 changes as shown in FIG. 5A, the pulse train output from the ΔΣ modulator 24 becomes as shown in FIG. 5B. . As shown in FIG. 5C, the asymmetric component occurs at the rise and fall of the pulse train in FIG. As shown in FIG. 5D, the detector 44 outputs a detection signal (quantized data change detection signal) in synchronization with the generation timing of the asymmetric component.
　図４に戻り、検出器４４は、図５（ｄ）に示す検出信号を生成するため、遅延素子４８と、加算器５０と、符号関数部５２と、Ａｂｓ（絶対値）関数部５４と、を有している。検出器４４の加算器（差分器）５０は、あるサンプリングクロックにおける量子化データと、そのサンプリングクロックよりも一つ前のクロックの量子化データと、の差分を求める。遅延素子４８は、サンプリングクロックよりも一つ前のクロックの量子化データを、加算器５０に与える。加算器５０は、あるサンプリングクロックにおける量子化データと、そのサンプリングクロックよりも一つ前のクロックの量子化データと、が一致する場合、０を出力し、一致しない場合（量子化データが変化した場合）、０以外の値を出力する。符号関数部５２は、加算器５０の出力の符号に応じて、＋１，－１，又は０を出力する。Ａｂｓ関数部５４は、符号関数部５２の出力の絶対値を出力する。つまり、Ａｂｓ関数部５４は、各サンプリングクロックにおいて、量子化データが一つ前のサンプリングクロックの量子化データから変化した場合には、１を出力し、量子化データが変化しない場合には、０を出力する。したがって、検出器４４は、図５（ｄ）に示すような検出信号を出力することができる。 Returning to FIG. 4, the detector 44 generates a detection signal shown in FIG. 5D, so that the delay element 48, the adder 50, the sign function unit 52, the Abs (absolute value) function unit 54, have. The adder (difference unit) 50 of the detector 44 obtains a difference between the quantized data at a certain sampling clock and the quantized data of the clock immediately before the sampling clock. The delay element 48 provides the adder 50 with the quantized data of the previous clock before the sampling clock. The adder 50 outputs 0 when the quantized data in a certain sampling clock and the quantized data of the clock previous to the sampling clock match, and when they do not match (the quantized data has changed). ), A value other than 0 is output. The sign function unit 52 outputs +1, −1, or 0 according to the sign of the output from the adder 50. The Abs function unit 54 outputs the absolute value of the output of the sign function unit 52. That is, the Abs function unit 54 outputs 1 when the quantized data has changed from the quantized data of the previous sampling clock in each sampling clock, and outputs 0 when the quantized data does not change. Is output. Therefore, the detector 44 can output a detection signal as shown in FIG.
　補償信号生成器４６は、量子化データの変化を示す検出信号に基づいて、非対称成分（図５（ｃ）参照）を抑制する補償信号を生成する。補償信号生成器４６は、フラクショナルディレイによって構成されている。このフラクショナルディレイは、有限インパルス応答（ＦＩＲ）フィルタと同様の構成を有している。つまり、補償信号生成器４６は、複数の遅延素子５６ａ，５６ｂ，５６ｃ，５６ｄと、複数のゲイン制御素子５８ａ，５８ｂ，５８ｃ，５８ｄ，５８ｅと、加算器６０と、を有して複数タップのＦＩＲフィルタ構造を持つ。図４の補償信号生成器４６は、４タップのデジタルフィルタ構成となっている。 The compensation signal generator 46 generates a compensation signal that suppresses the asymmetric component (see FIG. 5C) based on the detection signal indicating the change in the quantized data. The compensation signal generator 46 is configured by a fractional delay. This fractional delay has the same configuration as a finite impulse response (FIR) filter. That is, the compensation signal generator 46 includes a plurality of delay elements 56a, 56b, 56c, and 56d, a plurality of gain control elements 58a, 58b, 58c, 58d, and 58e, and an adder 60. Has an FIR filter structure. The compensation signal generator 46 of FIG. 4 has a 4-tap digital filter configuration.
　補償信号生成器４６は、パルス状の検出信号に対して、フィルタとして作用し、ターゲット周波数ｆ０の周波数成分であるＲＦ信号に歪を生じさせる非対称成分を抑制するための補償信号を生成する。パルス状の検出信号は広い周波数成分を有しているため、フィルタ作用によって補償信号を生成するのが容易である。なお、検出信号は、補償信号に必要な周波数成分を有していればよく、パルス状に限定されるものではない。 The compensation signal generator 46 acts as a filter on the pulse-like detection signal, and generates a compensation signal for suppressing an asymmetric component that causes distortion in the RF signal that is the frequency component of the target frequency f 0 . Since the pulse-like detection signal has a wide frequency component, it is easy to generate a compensation signal by a filter action. The detection signal only needs to have a frequency component necessary for the compensation signal, and is not limited to a pulse shape.
　検出信号から適切な補償信号を生成するには、各ゲイン制御素子５８ａ～５８ｅの係数（ゲイン）Ｃｉ（ｉ＝１～Ｎ；Ｎはゲイン制御素子の数；図４ではＮ＝５）を適切に設定すればよい。非対称成分は、ΔΣ変調器２４（パルス列を出力するドライバ）によってばらつきがあるため、予め、適切な補償信号を生成できる係数Ｃｉを決定して、各ゲイン制御素子５８ａ～５８ｅに設定しておく。 In order to generate an appropriate compensation signal from the detection signal, the coefficient (gain) Ci (i = 1 to N; N is the number of gain control elements; N = 5 in FIG. 4) of each gain control element 58a to 58e is appropriate. Should be set. Since the asymmetric component varies depending on the ΔΣ modulator 24 (driver that outputs a pulse train), a coefficient Ci that can generate an appropriate compensation signal is determined in advance and set in each of the gain control elements 58a to 58e.
　図６は、各ゲイン制御素子５８ａ～５８ｅの係数（ゲイン）Ｃｉを決定する方法を示している。まず、ΔΣ変調器２４にデジタルＲＦ信号（係数決定用のテスト信号）を入力し、ΔΣ変調器２４から量子化データ（パルス列）が出力されている状態にする。この状態で、図６の処理が行われる。ステップＳ１において、全てのゲイン制御素子５８ａ～５８ｅの係数がゼロに設定される。全ての係数がゼロに設定されると補償信号もゼロ（補償信号なし）となる。そして、係数Ｃ１からＣ５まで順番に、係数を決定していく（ステップＳ２～Ｓ５）。具体的には、まず、係数Ｃ１（ｉ＝１）を決定する。係数Ｃ１の決定のために、所定の探索範囲（例えば、－０．２～０．２）の間で、係数Ｃ１の値を変化させつつ、Δ変調器２４の出力(ＲＦ信号)のＡＣＬＲを測定する。ＡＣＬＲが最良となる値を、係数Ｃ１の値として決定する（ステップＳ３）。 FIG. 6 shows a method of determining the coefficients (gains) Ci of the gain control elements 58a to 58e. First, a digital RF signal (a coefficient determination test signal) is input to the ΔΣ modulator 24 so that quantized data (pulse train) is output from the ΔΣ modulator 24. In this state, the process of FIG. 6 is performed. In step S1, the coefficients of all gain control elements 58a to 58e are set to zero. When all the coefficients are set to zero, the compensation signal is also zero (no compensation signal). Then, the coefficients are determined in order from the coefficients C1 to C5 (steps S2 to S5). Specifically, first, the coefficient C1 (i = 1) is determined. In order to determine the coefficient C1, the ACLR of the output (RF signal) of the Δ modulator 24 is changed while changing the value of the coefficient C1 within a predetermined search range (for example, −0.2 to 0.2). taking measurement. The value with the best ACLR is determined as the value of the coefficient C1 (step S3).
　図７（ａ）は、係数Ｃ１（横軸）を、－０．２から０．２の間で変化させた場合のＡＣＬＲ（縦軸）を示している。図７（ａ）は、Ｃ１＝－０．０７において、ＡＣＬＲ＝４０．４９［ｄＢ］となり、最良となっていることを示す。したがって、Ｃ１＝－０．０７に決定される。 FIG. 7A shows the ACLR (vertical axis) when the coefficient C1 (horizontal axis) is changed between −0.2 and 0.2. FIG. 7A shows that ACLR = 40.49 [dB] when C1 = −0.07, which is the best. Therefore, C1 = −0.07 is determined.
　次に、Ｃ１＝－０．０７の状態で、係数Ｃ２を決定する。係数Ｃ２（ｉ＝２）の決定のために、Ｃ１＝－０．０７の状態で、係数Ｃ２を－０．２～０．２の間で変化させ、ＡＬＣＲを測定する。図７（ｂ）は、Ｃ１＝－０．０７の状態で、係数Ｃ２（横軸）を、－０．２～０．２の間で変化させた場合のＡＣＬＲ（縦軸）を示している。図７（ｂ）は、Ｃ２＝０．０７において、ＡＣＬＲ＝５１．８６［ｄＢ］となり、最良となっていることを示す。したがって、Ｃ２＝０．０７に決定される。
　同様に、Ｃ３，Ｃ４，Ｃ５も決定することで、全ての係数Ｃ１～Ｃ５を決定できる。 Next, the coefficient C2 is determined with C1 = −0.07. In order to determine the coefficient C2 (i = 2), with the condition of C1 = −0.07, the coefficient C2 is changed between −0.2 and 0.2, and the ALCR is measured. FIG. 7B shows the ACLR (vertical axis) when the coefficient C2 (horizontal axis) is changed between −0.2 and 0.2 in a state where C1 = −0.07. . FIG. 7B shows that ACLR = 51.86 [dB] at C2 = 0.07, which is the best. Therefore, C2 = 0.07 is determined.
Similarly, by determining C3, C4, and C5, all the coefficients C1 to C5 can be determined.
　図６の処理は、ΔΣ変調器２４又はΔΣ変調器２４の量子化データに対応するパルス列を出力する機器（送信機１０など）の出荷前に行っても良いし、ΔΣ変調器２４の稼働時における必要な時点で行って、係数Ｃ１～Ｃ５を動的に変更してもよい。図８は、係数Ｃ１～Ｃ５を動的に変更するための構成を示している。コントローラ２６は、ゲイン制御素子５８ａ～５８ｅの係数Ｃ１～Ｃ５を変更することができる。さらに、コントローラ２６は、ΔΣ変調器２４から出力された量子化データに対応するパルス列（が表現するＲＦ信号）を取得するよう構成されている。以上のように構成されたコントローラ２６が、図６の処理を実行することで、係数Ｃ１～Ｃ５を変更できる。この場合、パルス列の非対称成分が経時的に変化する場合には、係数Ｃ１～Ｃ５を更新することで、適切に歪を補償できる。 6 may be performed before shipment of the ΔΣ modulator 24 or a device (such as the transmitter 10) that outputs a pulse train corresponding to the quantized data of the ΔΣ modulator 24, or when the ΔΣ modulator 24 is in operation. The coefficients C1 to C5 may be dynamically changed at a necessary time. FIG. 8 shows a configuration for dynamically changing the coefficients C1 to C5. The controller 26 can change the coefficients C1 to C5 of the gain control elements 58a to 58e. Further, the controller 26 is configured to acquire a pulse train (an RF signal expressed by) corresponding to the quantized data output from the ΔΣ modulator 24. The controller 26 configured as described above can change the coefficients C1 to C5 by executing the processing of FIG. In this case, when the asymmetric component of the pulse train changes with time, the distortion can be appropriately compensated by updating the coefficients C1 to C5.
　図９～図１１は、補償信号によって非対称成分に起因するＲＦ信号の歪が補償され、ＲＦ信号のＡＣＬＲが向上したことを示している。ΔΣ変調器２４の出力は、図９のアイパターンに示すように、パルスの立ち上がり時間と立ち下がり時間とが異なり、非対称成分を含んでいる。図１０及び図１１は、ΔΣ変調器２４から出力されるパルス列（図９）によるＲＦ信号のＡＣＬＲを示している。なお、ＲＦ信号は、中心周波数（ターゲット周波数）が１０００ＭＨｚである。図１０（ａ）及び図１１（ａ）は、補償器３８による補償を行わなかった場合のＡＣＬＲを示している。一方、図１０（ｂ）及び図１１（ｂ）は、補償器３８による補償を行った場合のＡＣＬＲを示している。図１０（ｂ）及び図１１（ｂ）では、図１０（ａ）及び図１１（ａ）に比べて、隣接チャネル漏洩電力が低下している。したがって、補償器３８による補償の効果が認められる。 9 to 11 show that the distortion of the RF signal caused by the asymmetric component is compensated by the compensation signal, and the ACLR of the RF signal is improved. As shown in the eye pattern of FIG. 9, the output of the ΔΣ modulator 24 has a pulse rise time and a fall time that are different from each other and includes an asymmetric component. 10 and 11 show the ACLR of the RF signal by the pulse train (FIG. 9) output from the ΔΣ modulator 24. The RF signal has a center frequency (target frequency) of 1000 MHz. FIGS. 10A and 11A show the ACLR when the compensation by the compensator 38 is not performed. On the other hand, FIGS. 10B and 11B show ACLRs when compensation by the compensator 38 is performed. In FIG. 10B and FIG. 11B, the adjacent channel leakage power is lower than in FIG. 10A and FIG. Therefore, the effect of compensation by the compensator 38 is recognized.
　補償器３８は、図４に示すものに限られず、パルス列によって表現されるＲＦ信号の歪を補償するための補償信号を出力するものであれば足りる。
　また、補償器３８は、ΔΣ変調器２４が直接出力するパルス列（量子化データに対応するパルス列）に生じる非対称成分によって、ターゲット周波数ｆ０に生じる歪を補償するものに限られず、他の機器が出力するパルス列（量子化データに対応するパルス列）に生じる非対称成分によって、ターゲット周波数ｆ０に生じる歪を補償するものであってもよい。例えば、ΔΣ変調器２４を有する送信機から出力されたパルス列（例えば、光信号のパルス列）を受信した受信機が、光信号に対応した電気信号のパルス列を出力する場合、受信機の出力するパルス列の非対称成分による歪の補償を、送信機側のΔΣ変調器２４によって行っても良い。 The compensator 38 is not limited to that shown in FIG. 4, and may be any one that outputs a compensation signal for compensating for distortion of the RF signal expressed by the pulse train.
The compensator 38 is not limited to one that compensates for distortion occurring in the target frequency f 0 due to an asymmetric component generated in the pulse train (pulse train corresponding to quantized data) directly output by the ΔΣ modulator 24, and other devices the asymmetric component occurring in the output pulse train (pulse train corresponding to the quantized data), may be configured to compensate for the distortion of the target frequency f 0. For example, when a receiver that receives a pulse train (for example, a pulse train of an optical signal) output from a transmitter having a ΔΣ modulator 24 outputs a pulse train of an electrical signal corresponding to the optical signal, the pulse train output by the receiver Compensation for distortion by the asymmetrical component may be performed by the ΔΣ modulator 24 on the transmitter side.
［２．４　デュアルバンドΔΣ変調器における歪補償］
　図１２は、デュアルバンドΔΣ変調器（マルチバンドΔΣ変調器）２４を示している。デュアルバンドΔΣ変調器（マルチバンドΔΣ変調器）２４は、特開２０１４－１６５８４６号公報に開示されている。デュアルバンドΔΣ変調器２４は、周波数の異なる２つ(複数の)の入力信号Ｕ１，Ｕ２を入力可能であり、複数のループフィルタ（第１ループフィルタ３０ａ及び第２ループフィルタ３０ｂ）と、各フィルタ３０ａ，３０ｂの出力を加算する加算器１５と、加算器１５の出力を量子化する量子化器３６と、を備えている。ΔΣ変調器２４の量子化器３６は、２つの入力信号Ｕ１，Ｕ２が含まれる単一の出力信号（量子化データ）を出力する。 [2.4 Distortion compensation in dual-band ΔΣ modulator]
FIG. 12 shows a dual band ΔΣ modulator (multiband ΔΣ modulator) 24. The dual band ΔΣ modulator (multiband ΔΣ modulator) 24 is disclosed in Japanese Patent Application Laid-Open No. 2014-165846. The dual-band ΔΣ modulator 24 can input two (a plurality of) input signals U 1 and U 2 having different frequencies, and a plurality of loop filters (a first loop filter 30a and a second loop filter 30b), An adder 15 that adds the outputs of the filters 30a and 30b and a quantizer 36 that quantizes the output of the adder 15 are provided. The quantizer 36 of the ΔΣ modulator 24 outputs a single output signal (quantized data) including two input signals U 1 and U 2 .
　ΔΣ変調器２４は、内部経路として、量子化器３６から出力された量子化データを第１ループフィルタ３０ａへフィードバックするための第１経路４２ａ－１と、量子化器３６から出力された量子化データを第２ループフィルタ３０ｂへフィードバックするための第１経路４２ａ－２と、を備えている。
　ΔΣ変調器２４は、内部経路として、更に、第１ループフィルタ３０ａの出力を量子化器３６へ与えるために第１ループフィルタ３０ａから加算器１５に繋がる第２経路４２ｂ－１と、第２ループフィルタ３０ｂの出力を量子化器３６へ与えるために第２ループフィルタ３０ｂから加算器１５に繋がる第２経路４２ｂ－２と、を備えている。 The ΔΣ modulator 24 has, as internal paths, a first path 42a-1 for feeding back the quantized data output from the quantizer 36 to the first loop filter 30a, and a quantization output from the quantizer 36. A first path 42a-2 for feeding back data to the second loop filter 30b.
The ΔΣ modulator 24 further includes, as an internal path, a second path 42 b-1 connected from the first loop filter 30 a to the adder 15 for supplying the output of the first loop filter 30 a to the quantizer 36, and a second loop In order to provide the output of the filter 30 b to the quantizer 36, a second path 42 b-2 connected from the second loop filter 30 b to the adder 15 is provided.
　第１ループフィルタ３０ａは、図２Ａのループフィルタ３０と同様に、第１加算器３２ａ、Ｌ１（ｚ）の伝達関数ブロック３３ａと、第２加算器３４ａと、フィードフォワード経路３５ａと、を備えている。第２ループフィルタ３０ｂも、図２Ａのループフィルタ３０と同様に、第１加算器３２ｂ、Ｌ２（ｚ）の伝達関数ブロック３３ｂと、第２加算器３４ｂと、フィードフォワード経路３５ｂと、を備えている。 Similar to the loop filter 30 in FIG. 2A, the first loop filter 30a includes a first adder 32a, a transfer function block 33a of L 1 (z), a second adder 34a, and a feedforward path 35a. ing. Similarly to the loop filter 30 of FIG. 2A, the second loop filter 30b also includes a first adder 32b, a transfer function block 33b of L 2 (z), a second adder 34b, and a feedforward path 35b. ing.
　第１ループフィルタ３０ａに入力される第１入力信号Ｕ１は、例えば、中心周波数ｆ１（第１ターゲット周波数ｆ１）の第１ＲＦ信号である。第２ループフィルタ３０ｂに入力される第２入力信号Ｕ２は、中心周波数ｆ２（第１ターゲット周波数ｆ２）の第２ＲＦ信号である。 The first input signal U 1, which is input to the first loop filter 30a is, for example, a first 1RF signal center frequency f 1 (first target frequency f 1). The second input signal U 2 which is input to the second loop filter 30b is a second 2RF signal center frequency f 2 (first target frequency f 2).
　周波数の異なる２つ(複数の)の入力信号Ｕ１，Ｕ２が入力されるΔΣ変調器２４の場合、入力信号Ｕ１，Ｕ２の数に応じた複数（２つ）の補償器３８ａ，３８ｂが設けられている。複数の補償器３８ａ，３８ｂは、第１補償器３８ａと第２補償器３８ｂとを含む。第１補償器３８ａは、第１補償信号を出力する。第１補償信号は、ΔΣ変調器２４から出力されるパルス列が表現する周波数ｆ１の第１ＲＦ信号（第１ターゲット周波数ｆ１の周波数成分）に生じる歪を補償するためのものである。第１補償信号は、量子化器３６から出力された量子化データを第１ループフィルタ３０ａへフィードバックするための第１経路４２ａ－１に与えられる。第１補償信号は、第１経路４２ａ－１に設けられた加算器４０ａによって、第１経路４２ａ－１に与えられる。 In the case of the ΔΣ modulator 24 to which two (a plurality of) input signals U 1 and U 2 having different frequencies are input, a plurality of (two) compensators 38 a, corresponding to the number of input signals U 1 and U 2 . 38b is provided. The plurality of compensators 38a and 38b include a first compensator 38a and a second compensator 38b. The first compensator 38a outputs a first compensation signal. The first compensation signal is for compensating for distortion generated in the first RF signal (frequency component of the first target frequency f 1 ) having the frequency f 1 expressed by the pulse train output from the ΔΣ modulator 24. The first compensation signal is given to the first path 42a-1 for feeding back the quantized data output from the quantizer 36 to the first loop filter 30a. The first compensation signal is given to the first path 42a-1 by the adder 40a provided in the first path 42a-1.
　第２補償器３８ｂは、第２補償信号を出力する。第２補償信号は、ΔΣ変調器２４から出力されるパルス列が表現する周波数ｆ２の第２ＲＦ信号（第２ターゲット周波数ｆ２の周波数成分）に生じる歪を補償するためのものである。第２補償信号は、量子化器３６から出力された量子化データを第２ループフィルタ３０ｂへフィードバックするための第１経路４２ａ－２に与えられる。第２補償信号は、第１経路４２ａ－２に設けられた加算器４０ｂによって、第１経路４２ａ－２に与えられる。第１補償信号は、例えば、第２経路４２ｂ－１に与えられても良い。第２補償信号は、例えば、第２経路４２ｂ－２に与えられても良い。 The second compensator 38b outputs a second compensation signal. The second compensation signal is for compensating for distortion generated in the second RF signal (frequency component of the second target frequency f 2 ) having the frequency f 2 expressed by the pulse train output from the ΔΣ modulator 24. The second compensation signal is given to the first path 42a-2 for feeding back the quantized data output from the quantizer 36 to the second loop filter 30b. The second compensation signal is given to the first path 42a-2 by the adder 40b provided in the first path 42a-2. For example, the first compensation signal may be given to the second path 42b-1. For example, the second compensation signal may be given to the second path 42b-2.
　図１２のΔΣ変調器２４によれば、ΔΣ変調器２４から出力されるパルス波形が非対称成分を有していても、第１ＲＦ信号及び第２ＲＦ信号のＡＣＬＲの劣化が抑制される。 12, even if the pulse waveform output from the ΔΣ modulator 24 has an asymmetric component, deterioration of the ACLR of the first RF signal and the second RF signal is suppressed.
［３．付記］
　なお、今回開示された実施の形態はすべての点で例示であって制限的なものではないと考えられるべきである。本発明の範囲は、上記した意味ではなく、特許請求の範囲によって示され、特許請求の範囲と均等の意味、及び範囲内でのすべての変更が含まれることが意図される。 [3. Addendum]
The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the meanings described above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.
　１０　送信機
　１２　デジタル信号処理部
　１４　信号伝送路
　１５　加算器
　１６　アナログフィルタ（アナログＢＰＦ）
　１８　ベースバンド部
　２０　デジタル直交変調器
　２２　処理部
　２４　ΔΣ変調器
　２６　コントローラ
　３０　ループフィルタ
　３０ａ　第１ループフィルタ
　３０ｂ　第２ループフィルタ
　３２　第１加算器
　３２ａ　第１加算器
　３２ｂ　第１加算器
　３３　伝達関数ブロック
　３３ａ　伝達関数ブロック
　３３ｂ　伝達関数ブロック
　３４　第２加算器
　３４ａ　第２加算器
　３４ｂ　第２加算器
　３５　フィードフォワード経路
　３５ａ　フィードフォワード経路
　３５ｂ　フィードフォワード経路
　３６　量子化器
　３８　補償器
　３８ａ　第１補償器
　３８ｂ　第２補償器
　４０　加算器
　４０ａ　加算器
　４０ｂ　加算器
　４２　内部経路
　４２ａ　第１経路
　４２ｂ　第２経路
　４２ａ－１　第１経路
　４２ｂ－１　第２経路
　４２ａ－２　第１経路
　４２ｂ－２　第２経路
　４４　検出器
　４６　補償信号生成器
　４８　遅延素子
　５０　加算器
　５２　符号関数部
　５４　Ａｂｓ関数部
　５６ａ，５６ｂ，５６ｃ，５６ｄ　遅延素子
　５８ａ，５８ｂ，５８ｃ，５８ｄ，５８ｅ　ゲイン制御素子
　６０　加算器 DESCRIPTION OF SYMBOLS 10 Transmitter 12 Digital signal processing part 14 Signal transmission line 15 Adder 16 Analog filter (analog BPF)
18 baseband unit 20 digital quadrature modulator 22 processing unit 24 ΔΣ modulator 26 controller 30 loop filter 30a first loop filter 30b second loop filter 32 first adder 32a first adder 32b first adder 33 transfer function block 33a transfer function block 33b transfer function block 34 second adder 34a second adder 34b second adder 35 feed forward path 35a feed forward path 35b feed forward path 36 quantizer 38 compensator 38a first compensator 38b second Compensator 40 Adder 40a Adder 40b Adder 42 Internal path 42a First path 42b Second path 42a-1 First path 42b-1 Second path 42a-2 First path 42b-2 Second path 44 Detector 46 Compensation signal generator 48 Slow Element 50 adder 52 sign function unit 54 Abs function unit 56a, 56b, 56c, 56d delay elements 58a, 58b, 58c, 58d, 58e gain control element 60 adders

Claims (10)

  1. 　ループフィルタと、
    　前記ループフィルタの出力に基づいて、量子化データを生成する量子化器と、
    　前記ループフィルタ又は前記量子化器に接続された内部経路と、
    　前記量子化データに対応したパルス列の周波数成分のうちターゲット周波数の周波数成分に生じる歪の補償信号を、前記内部経路に与える補償器と、
    　を備えるΔΣ変調器。     A loop filter,
    A quantizer for generating quantized data based on the output of the loop filter;
    An internal path connected to the loop filter or the quantizer;
    A compensator that provides a compensation signal for distortion generated in a frequency component of a target frequency among frequency components of a pulse train corresponding to the quantized data to the internal path;
    A ΔΣ modulator comprising:
  2. 　前記歪は、前記量子化データに対応したパルス列におけるパルスの立ち上がりと立ち下がりとの非対称性によって、前記ターゲット周波数の周波数成分に生じる歪を含む
    　請求項１に記載のΔΣ変調器。     The ΔΣ modulator according to claim 1, wherein the distortion includes distortion generated in a frequency component of the target frequency due to asymmetry between a rising edge and a falling edge of a pulse in a pulse train corresponding to the quantized data.
  3. 　前記補償器は、
    　　前記量子化データの変化を検出する検出器と、
    　　前記検出器によって検出された前記量子化データの変化に基づいて補償信号を生成する生成器と、
    　を備える請求項２記載のΔΣ変調器。     The compensator is
    A detector for detecting a change in the quantized data;
    A generator for generating a compensation signal based on a change in the quantized data detected by the detector;
    A ΔΣ modulator according to claim 2.
  4. 　前記検出器は、前記量子化データが変化したタイミングで検出信号を出力する
    　請求項３記載のΔΣ変調器。     The ΔΣ modulator according to claim 3, wherein the detector outputs a detection signal at a timing when the quantized data changes.
  5. 　前記生成器は、フラクショナルディレイを含み、
    　前記フラクショナルディレイは、前記検出信号に基づいて前記補償信号を生成する
    　請求項４記載のΔΣ変調器。     The generator includes a fractional delay;
    The ΔΣ modulator according to claim 4, wherein the fractional delay generates the compensation signal based on the detection signal.
  6. 　前記量子化器は、１ｂｉｔ量子化器である
    　請求項１～５のいずれか１項に記載のΔΣ変調器。     The ΔΣ modulator according to any one of claims 1 to 5, wherein the quantizer is a 1-bit quantizer.
  7. 　前記内部経路は、前記量子化データを前記ループフィルタへフィードバックするための第１経路を含み、
    　前記補償器は、前記補償信号を前記第１経路に与える
    　請求項１～６のいずれか１項に記載のΔΣ変調器。     The internal path includes a first path for feeding back the quantized data to the loop filter;
    7. The ΔΣ modulator according to claim 1, wherein the compensator gives the compensation signal to the first path.
  8. 　前記内部経路は、前記ループフィルタの出力を前記量子化器へ与えるための第２経路を含み、
    　前記補償器は、前記補償信号を前記第２経路に与える
    　請求項１～６のいずれか１項に記載のΔΣ変調器。     The internal path includes a second path for providing the output of the loop filter to the quantizer;
    7. The ΔΣ modulator according to claim 1, wherein the compensator gives the compensation signal to the second path.
  9. 　前記内部経路は、前記ループフィルタの出力を前記量子化器へ与えるための第２経路を含み、
    　前記補償器は、前記補償信号を前記第２経路に更に与える
    　請求項７に記載のΔΣ変調器。     The internal path includes a second path for providing the output of the loop filter to the quantizer;
    The ΔΣ modulator according to claim 7, wherein the compensator further provides the compensation signal to the second path.
  10. 　請求項１～９のいずれか１項に記載のΔΣ変調器を備え、
    　前記量子化データに対応したパルスを、送信信号として出力する送信機。     A ΔΣ modulator according to any one of claims 1 to 9,
    A transmitter that outputs a pulse corresponding to the quantized data as a transmission signal.
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