WO2015078367A1 - 一种基于模块化多电平电压源型换流器的损耗确定方法 - Google Patents

一种基于模块化多电平电压源型换流器的损耗确定方法 Download PDF

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WO2015078367A1
WO2015078367A1 PCT/CN2014/092231 CN2014092231W WO2015078367A1 WO 2015078367 A1 WO2015078367 A1 WO 2015078367A1 CN 2014092231 W CN2014092231 W CN 2014092231W WO 2015078367 A1 WO2015078367 A1 WO 2015078367A1
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sub
loss
bipolar transistor
insulated gate
gate bipolar
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English (en)
French (fr)
Inventor
于鹏
王海田
汤广福
孔明
杨杰
马巍巍
夏方
李小
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State Grid Smart Grid Research Institute of SGCC
Dalian Power Supply Co State Grid Liaoning Electric Power Co Ltd
State Grid Corp of China SGCC
C Epri Electric Power Engineering Co Ltd
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State Grid Smart Grid Research Institute of SGCC
Dalian Power Supply Co State Grid Liaoning Electric Power Co Ltd
State Grid Corp of China SGCC
C Epri Electric Power Engineering Co Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/42—Conversion of DC power input into AC power output without possibility of reversal
    • H02M7/44—Conversion of DC power input into AC power output without possibility of reversal by static converters
    • H02M7/48—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/42—Conversion of DC power input into AC power output without possibility of reversal
    • H02M7/44—Conversion of DC power input into AC power output without possibility of reversal by static converters
    • H02M7/48—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/483—Converters with outputs that each can have more than two voltages levels
    • H02M7/4835—Converters with outputs that each can have more than two voltages levels comprising two or more cells, each including a switchable capacitor, the capacitors having a nominal charge voltage which corresponds to a given fraction of the input voltage, and the capacitors being selectively connected in series to determine the instantaneous output voltage
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to a method for determining a field of flexible direct current transmission (VSC-HVDC), in particular to a method for determining a loss based on a modular multi-level voltage source type converter.
  • VSC-HVDC flexible direct current transmission
  • Flexible DC transmission technology has been widely used in various fields due to its flexibility in power transmission control.
  • Flexible DC transmission technology based on modular multi-level converter is a new type of flexible DC transmission technology. There is no commutation failure, and it can supply power to the passive network without reactive compensation and can flexibly control the output.
  • the power and flexible DC transmission technology is especially suitable for small-capacity load power supply in remote areas and islands, new energy collection and grid connection, and urban power supply applications. Therefore, flexible DC transmission technology has broad application prospects.
  • the inverter output voltage harmonic component is high.
  • PWM Pulse Width Modulation
  • the switching frequency is high, generally 20 to 40 times the fundamental frequency, and the switching frequency is too high.
  • the increase of the energy transmission efficiency of the flexible direct current transmission system is reduced.
  • the harmonic component of the output voltage of the converter is low, and the amount of harmonic components decreases as the number of output levels of the converter increases.
  • the controllable voltage source type converter can adopt a lower switching frequency. Generally no more than 3 times the fundamental frequency, thereby reducing the loss and improving the energy transmission efficiency of the flexible direct current transmission system.
  • the switching device used in flexible DC transmission technology is mainly a fully controlled device - insulated gate bipolar transistor (IGBT), which is used in a flexible DC converter in a high-voltage large-capacity flexible DC transmission system.
  • the gate bipolar transistor IGBT exceeds two thousand, so the stable operation capability of the insulated gate bipolar transistor IGBT is one of the key factors affecting the stable operation of the flexible DC converter valve.
  • the overheating of the insulated gate bipolar transistor IGBT due to overload or excessive energy consumption is the most important cause of failure. Therefore, the accuracy of the junction temperature calculation of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD (free-wheeling diode) seriously affects the reliability of the flexible DC converter.
  • the loss calculation of the flexible DC converter insulated gate bipolar transistor IGBT and freewheeling diode FWD under various operating conditions is the only effective means to solve the corresponding junction temperature calculation.
  • the loss diagram of the flexible DC converter valve is shown in Figure 1. It mainly includes:
  • Loss of insulated gate bipolar transistor IGBT which mainly includes conduction loss, turn-on loss, turn-off loss, and drive circuit loss. The sum of the turn-on loss and the turn-off loss is called the switching loss.
  • Loss of freewheeling diode FWD which mainly includes conduction loss, reverse recovery loss, turn-on loss and cutoff loss. Among them, because the turn-on time is very short, the turn-on loss can be neglected; the diode reverse cut-off current is small, and the turn-off loss is negligible.
  • the switching loss of the insulated gate bipolar transistor IGBT and the reverse recovery loss of the freewheeling diode FWD can be referred to the corresponding parameters provided by the manufacturer. However, the operating conditions of these devices are different from those corresponding to the manufacturer parameters, and further conversion is required.
  • the conduction loss of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD can also be expressed by parameters such as the corresponding conduction threshold voltage, on-resistance, and on-current.
  • the bridge arm current has two paths in the sub-module, that is, it may flow through the sub-module upper half-arm insulated gate bipolar transistor IGBT/lower half-bridge freewheeling diode FWD or upper half-bridge freewheeling diode FWD/ Half-bridge insulated gate bipolar transistor IGBT. Therefore, the calculation of the conduction loss also has the problem of determining the expression of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD current.
  • the calculation of the loss of the voltage source converter is mainly for the switching type voltage source converter, but the current and controllable voltage of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD of the switching voltage source converter
  • the insulated gate bipolar transistor IGBT of the source voltage source converter is essentially different from the current in the freewheeling diode FWD.
  • the existing loss calculation method for the switching voltage source converter cannot be applied to the controllable Loss calculation of voltage source type voltage source converter; some research results for loss calculation of controllable voltage source converter, but almost all of the insulated gate bipolar transistor IGBT and freewheeling using inverter The average current in the diode FWD does not relate to the effective current in the insulated gate bipolar transistor IGBT and the freewheeling diode FWD of the converter. With the rapid application of flexible DC transmission engineering, the research on the calculation procedure of flexible DC converter loss has important theoretical and engineering significance.
  • the object of the invention is to provide a loss determination method based on a modular multi-level voltage source converter, which can reflect the loss and flexible DC transmission of an insulated gate bipolar transistor IGBT and a freewheeling diode FWD in a submodule
  • the relationship between the modulation factor, power factor, and active transmission power of the system visualizes complex problems.
  • the invention provides a loss determining method based on a modular multi-level voltage source type converter, wherein the modular multi-level voltage source type converter is composed of three phases, and each phase is composed of the same structure of the upper and lower two.
  • the bridge arm is formed; the intermediate end of the upper and lower arms is connected to the AC end of the modular multilevel converter;
  • Each of the upper and lower arms includes a reactor and N sub-modules of the same structure; the sub-modules of each bridge are cascaded and the end is communicated with the modular multi-level converter through a reactor. End connection; each sub-module of the sub-module is cascaded and the other end is connected to one end of the cascaded sub-module of the other two-phase bridge arm to form a positive end of the DC end of the modular multi-level voltage source converter Negative busbar
  • the improvement is that the method comprises the following steps:
  • Step 1 Determine the current of the single-phase voltage source converter
  • Step 2 Determine the conduction loss of the voltage source type converter
  • Step 3 Determine the switching loss of the voltage source converter
  • Step 4 Determine the total three-phase total loss of the voltage source converter.
  • step 1 includes the following sub-steps:
  • Step 1.1 Determine the bridge arm current of the voltage source converter, the expression is as follows:
  • I dc is the DC line current
  • I m is the i-phase current peak
  • ⁇ is the fundamental wave angle frequency
  • Is the angle of the i-phase current lag i voltage
  • i is phase A, phase B or phase C;
  • Step 1.2 Determine the current of the upper half arm and the lower half arm of the submodule; the currents flowing through the upper half arm and the lower half arm of the submodule are:
  • m is the modulation factor, ie the phase voltage peak The ratio of the DC line to ground voltage absolute value U dcN ;
  • Step 1.3 Determine the average currents I T1_avg , I T2_avg and the effective currents I T1_rms and I T2_rms of the insulated gate bipolar transistor IGBTs in the upper half and the lower half of the submodule , respectively; the expressions are as follows:
  • Step 1.4 Determine the average currents I D1_avg , I D2_avg and the effective currents I D1_rms and I D2_rms of the freewheeling diode FWD in the upper half and the lower half of the submodule , respectively; the expressions are as follows:
  • the device parameters provided by the switching device manufacturer and the average current and the effective current of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD in step 1 are used to determine the conduction of the voltage source converter. Loss, including:
  • the average conduction losses of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD of the upper half arm and the lower half arm are:
  • U T1_0 , U T2_0 are the conduction threshold voltages of the insulated gate bipolar transistor IGBT in the upper half of the sub-module and the lower half of the sub-bridge;
  • r T1_0 , r T2_0 are the forward conduction resistances of the insulated gate bipolar transistor IGBTs in the upper half of the submodule and the lower half of the bridge;
  • U D1_0 and U D2_0 are respectively the conduction threshold voltages of the freewheeling diode FWD in the upper half arm and the lower half of the submodule;
  • r D1_0 and r D2_0 are the forward conduction resistances of the freewheeling diode FWD in the upper half of the submodule and the lower half of the sub-bridge.
  • step 3 includes the following sub-steps:
  • Step 3.1 respectively determine the switching loss of the insulated gate bipolar transistor IGBT and the reverse recovery loss of the freewheeling diode FWD in the upper half arm and the lower half arm of the submodule;
  • P swT1 and P swT2 are the switching losses of the insulated gate bipolar transistor IGBT in the upper half of the sub-module and the lower half of the sub-bridge, respectively;
  • P recD1 and P recD2 are the reverse recovery losses of the freewheeling diode FWD in the upper half of the submodule and the lower half of the sub-bridge;
  • E on_T1 (E on_T2 ) and E off_T1 (E off_T2 ) are respectively the insulated gate bipolar transistor IGBT of the upper half or lower half of the submodule at the reference voltage u ref_T1 (u ref_T2 ), reference current i ref_T1 (i ref_T2 ) and the turn-on and turn-off losses of the maximum operating junction temperature;
  • E rec_D1 (E rec_D2 ) is the reversal of the freewheeling diode FWD in the upper half or lower half of the submodule at the reference voltage u ref_D1 (u ref_D2 ), the reference current i ref_D1 (i ref_D2 ) and the maximum operating junction temperature, respectively. Recovery loss; the maximum working junction temperature is taken as 125 ° C;
  • U c is the sub-module capacitor voltage
  • Step 3.2 respectively determine the switching loss of the insulated gate bipolar transistor IGBT in the upper half arm or the lower half of the submodule, and the reverse recovery loss of the freewheeling diode FWD;
  • K T1 and K T2 are the junction temperature coefficients of the insulated gate bipolar transistor IGBT of the upper half-arm and the lower half of the sub-module;
  • K D1 and K D2 are the junction temperature coefficients of the freewheeling diode FWD in the upper half-arm and the lower half of the sub-module, respectively;
  • Step 3.3 Determine the junction temperature coefficient of the insulated gate bipolar transistor IGBT and the junction temperature coefficient of the freewheeling diode FWD by an iterative method.
  • the expressions are as follows:
  • t is the working junction temperature
  • E 1swT1 and E 2swT1 are the switching losses of the upper half-bridge insulated gate bipolar transistor IGBT at 125 ° C and 25 ° C, respectively;
  • E 1swT2 and E 2swT2 are the switching losses of the sub-module lower half-arm insulated gate bipolar transistor IGBT at 125 ° C and 25 ° C, respectively;
  • E 1recD1 and E 2recD1 are the reverse recovery losses of the upper half-bridge freewheeling diode FWD of the submodule at 125 ° C and 25 ° C, respectively;
  • E 1recD2 and E 2recD2 are the reverse recovery losses of the sub-module lower half-arm freewheeling diode FWD at 125 ° C and 25 ° C, respectively.
  • N is the number of submodules of each bridge arm of the inverter
  • P T1drive and P T2drive are the losses of the drive circuit of the insulated gate bipolar transistor IGBT in the upper half-arm and the lower half of the sub-module.
  • the invention systematically analyzes the upper half bridge arm and the lower half bridge insulated gate bipolar transistor IGBT and freewheeling two of the submodule
  • the effective current in the pole tube FWD provides a calculation method for the loss of the controllable voltage source type converter.
  • the calculation results are basically consistent with the actual flexible DC transmission project. Therefore, the loss calculation method has a good guiding significance for the loss evaluation of the flexible DC converter and the configuration of the converter cooling system;
  • Insulated gate bipolar transistor IGBT and freewheeling diode FWD manufacturers use the physical conductivity of the device to control the threshold voltage, average current, forward conduction resistance and effective current to calculate the upper half of the sub-bridge and the lower half of the sub-module
  • the conduction loss of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD has clear physical meaning
  • the loss calculation method is simple and reliable
  • Figure 1 is a schematic diagram showing the structure of a sub-module loss
  • FIG. 2 is a schematic structural view of a modular multilevel converter
  • FIG. 3 is a diagram showing the relationship between switching loss and collector current of a 5SNA 1500E330305ABBHiPak TM type insulated gate bipolar transistor IGBT according to the present invention
  • FIG. 4 is a diagram showing the relationship between the reverse recovery loss, current, power and forward current of a freewheeling diode in a 5SNA 1500E330305ABBHiPak TM type insulated gate bipolar transistor IGBT according to the present invention
  • FIG. 5 is a flow chart of a method for determining loss of a modular multi-level voltage source converter according to the present invention.
  • this patent effectively solves the inverter half-bridge upper insulated gate bipolar transistor IGBT/lower half-bridge freewheeling diode FWD.
  • the average current and effective current of the upper half bridge freewheeling diode FWD/lower half bridge insulated gate bipolar transistor IGBT a loss determination method of the modular multilevel voltage source converter is proposed.
  • the characteristic of the loss determination method is that it can reflect the relationship between the loss of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD in the submodule and the modulation factor, power factor and active transmission power of the flexible direct current transmission system, and visualize the complicated problem. .
  • the invention provides a method for determining the loss of a modular multi-level voltage source converter, wherein a schematic diagram of a modular multi-level voltage source converter is shown in FIG. 2, and a modular multi-level voltage source is used.
  • the flow device consists of three phases, each phase is connected in series
  • the upper and lower arms of the same structure are constructed; the intermediate ends of the upper and lower arms are connected to the AC end of the modular multilevel converter; each of the upper and lower arms includes 1 reactor and N Sub-modules with the same structure; the sub-modules of each bridge arm are cascaded and connected to the AC end of the modular multi-level converter through a reactor; the sub-modules of each bridge arm are cascaded and the other end is connected to the other two phases.
  • the cascaded sub-modules of the bridge arms are connected at one end to form positive and negative bus bars of the DC end of the modular multi-level voltage source type converter; the sub-module is composed of a half bridge and a capacitor branch connected in parallel thereto, the half bridge being composed of The upper half arm and the lower half arm are composed of an insulated gate bipolar transistor IGBT and a freewheeling diode FWD connected in parallel therewith.
  • FIG. 5 A flow chart of the method for determining the loss of a modular multi-level voltage source converter is shown in FIG. 5, and includes the following steps:
  • Embodiment The method of the present invention will be described by taking the upper arm of the inverter A phase in an inverted state as an example.
  • Step 1 Taking the upper arm of the inverter A phase in the inverter state as an example, the composition of the inverter and the reference direction of each current are as shown in FIG. 2, and the current of the controllable voltage source type converter is calculated;
  • Step 1.1 Calculate the bridge arm current of the controllable voltage source type converter
  • I dc is the DC line current
  • I m is the peak value of phase A current
  • ⁇ is the fundamental angle frequency
  • the angle of the A voltage is delayed by the A phase current.
  • Step 1.2 Calculate the current of the upper and lower tubes of the submodule; the currents flowing through the upper and lower tubes of the submodule are:
  • m is the modulation factor, ie the phase voltage peak The ratio of the DC line to ground voltage absolute value U dcN .
  • Step 1.3 Calculate the average currents I T1_avg , I T2_avg and the effective currents I T1_rms and I T2_rms of the insulated gate bipolar transistor IGBTs in the upper half and lower half of the submodule , respectively, as follows:
  • Step 1.4 Determine the average currents I D1_avg , I D2_avg and the effective currents I D1_rms and I D2_rms of the freewheeling diode FWD in the upper half and the lower half of the submodule , respectively; the expressions are as follows:
  • Step 2 Calculate the conduction loss of the controllable voltage source inverter by calculating the device parameters provided by the switching device manufacturer and calculating the average current and effective current of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD in step 1;
  • the average conduction loss of the insulated gate bipolar transistor IGBT and the freewheeling diode FWD of the upper half arm and the lower half arm are:
  • U T1_0 , U T2_0 are the conduction threshold voltages of the insulated gate bipolar transistor IGBT in the upper half of the sub-module and the lower half of the sub-bridge;
  • r T1_0 , r T2_0 are the forward conduction resistances of the insulated gate bipolar transistor IGBTs in the upper half of the submodule and the lower half of the bridge;
  • U D1_0 and U D2_0 are respectively the conduction threshold voltages of the freewheeling diode FWD in the upper half arm and the lower half of the submodule;
  • r D1_0 and r D2_0 are the forward conduction resistances of the freewheeling diode FWD in the upper half of the submodule and the lower half of the sub-bridge.
  • Step 3 Calculate the switching loss of the controllable voltage source type converter
  • Step 3.1 Calculate the switching loss of the upper and lower bridge insulated gate bipolar transistor IGBT and the reverse recovery loss of the freewheeling diode FWD.
  • P swT1 and P swT2 are respectively switching losses of the sub-module upper and lower bridge arm insulated gate bipolar transistor IGBT;
  • P recD1 and P recD2 are the reverse recovery losses of the freewheeling diode FWD of the upper and lower arms of the submodule;
  • E on_T1 (E on_T2 ) and E off_T1 (E off_T2 ) are the sub-module upper (lower) bridge arm insulated gate bipolar transistor IGBT at reference voltage u ref_T1 (u ref_T2 ), reference current i ref_T1 (i ref_T2 ) and maximum Turn-on and turn-off losses of the operating junction temperature (often taking 125 ° C);
  • E rec_D1 (E rec_D2 ) is the inverse of the sub-module (lower) bridge arm freewheeling diode FWD at the reference voltage u ref_D1 (u ref_D2 ), the reference current i ref_D1 (i ref_D2 ) and the maximum operating junction temperature (usually 125 ° C) Recovery loss
  • U c is the submodule capacitor voltage.
  • Step 3.2 Calculate the switching loss of the insulated gate bipolar transistor IGBT and the reverse recovery loss of the freewheeling diode FWD, respectively;
  • K T1 and K T2 are the junction temperature coefficients of the insulated gate bipolar transistor IGBT in the upper half of the sub-module and the lower half of the sub-bridge;
  • K D1 and K D2 are the junction temperature coefficients of the freewheeling diode FWD in the upper half-arm and the lower half of the sub-module, respectively.
  • the invention adopts the relationship between the switching loss of the 5SNA 1500E330305ABBHiPak TM type insulated gate bipolar transistor IGBT and the collector current, and the relationship between the reverse recovery loss of the freewheeling diode, the current, the electric quantity and the forward current of the insulated gate bipolar transistor IGBT, respectively. As shown in Figure 3 and Figure 4.
  • Step 3.2 Calculate the junction temperature coefficient of the insulated gate bipolar transistor IGBT and the junction temperature coefficient of the freewheeling diode FWD by an iterative method
  • t is the working junction temperature
  • E 1swT1 and E 2swT1 are the switching losses of the upper half-bridge insulated gate bipolar transistor IGBT at 125 ° C and 25 ° C, respectively;
  • E 1swT2 and E 2swT2 are the switching losses of the sub-module lower half-arm insulated gate bipolar transistor IGBT at 125 ° C and 25 ° C, respectively;
  • E 1recD1 and E 2recD1 are the reverse recovery losses of the upper half-bridge freewheeling diode FWD of the submodule at 125 ° C and 25 ° C, respectively;
  • E 1recD2 and E 2recD2 are the reverse recovery losses of the lower half-bridge freewheeling diode FWD of the submodule at 125 ° C and 25 ° C, respectively.
  • Step 4 Calculate the total three-phase total loss of the controllable voltage source converter.
  • N is the number of submodules of each bridge arm of the inverter
  • P T1drive and P T2drive are the losses of the drive circuit of the upper half-arm and the lower half-bridge insulated gate bipolar transistor IGBT of the sub-module.
  • the upper half of the rectifier sub-module and the lower half The current of the insulated gate bipolar transistor IGBT in the bridge arm can be approximated (ignoring the loss of the converter station) equivalent to the current of the freewheeling diode FWD in the upper half arm and the lower half arm of the inverter submodule; on the submodule
  • the current of the freewheeling diode FWD in the half bridge arm and the lower half arm can be approximately equivalent to the current of the insulated gate bipolar transistor IGBT in the upper half arm and the lower half arm of the inverter submodule. Therefore, the calculation of the switching device loss in the rectified state of the converter can be referred to the calculation of the loss of the switching device in its inverter state.
  • the invention systematically analyzes the effective current in the upper half-bridge, the lower half-bridge insulated gate bipolar transistor IGBT and the freewheeling diode FWD of the sub-module, and provides calculation for the loss program calculation of the controllable voltage source type converter. method.
  • the calculation results of this program are basically consistent with the actual flexible DC transmission project. Therefore, the loss calculation program has a good guiding significance for the loss evaluation of the flexible DC converter and the configuration of the converter cooling system.

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Abstract

一种基于模块化多电平技术的柔性直流换流器的损耗确定方法，该方法包括下述步骤：步骤1：确定单相电压源型换流器的电流；步骤2：确定电压源型换流器的导通损耗；步骤3：确定电压源型换流器的开关损耗；步骤4：确定电压源型换流器三相总损耗。该方法有效求解出换流器子模块上半桥臂绝缘栅双极型晶体管IGBT/下半桥臂续流二极管FWD或上半桥臂续流二极管FWD/下半桥臂绝缘栅双极型晶体管IGBT的平均电流和有效电流，能反映出子模块中绝缘栅双极型晶体管IGBT与续流二极管FWD的损耗与柔性直流输电系统的调制因数、功率因素、有功传输功率等关系，将复杂问题直观化。

Description

一种基于模块化多电平电压源型换流器的损耗确定方法 技术领域
本发明涉及柔性直流输电(VSC-HVDC)领域的确定方法，具体涉及一种基于模块化多电平电压源型换流器的损耗确定方法。
背景技术
柔性直流输电技术因其功率传输控制的灵活性而在各种领域中得到了广泛的应用。基于模块化多电平换流器的柔性直流输电技术是一种新型的柔性直流输电技术，不存在换相失败，具有可向无源网络供电、无需无功补偿、能够灵活控制输出的有功无功功率，柔性直流输电技术特别适合偏远地区、岛屿等小容量负荷供电、新能源集电并网以及城市供电等应用场合。因此，柔性直流输电技术有着广阔的应用前景。
在开关型柔性直流输电系统中，由于换流器输出电平数低，换流器输出电压谐波成分高。为了改善柔性直流换流器输出波形质量，通常使用脉宽调制技术PWM(Pulse Width Modulation)，且开关频率较高，一般都在20倍～40倍基频，过高的开关频率带来开关损耗的增加，降低了柔性直流输电系统能量的传输效率。在可控电压源型柔性直流输电系统中，由于采用了模块化多电平技术，换流器输出电压谐波成分低，且谐波成分量随换流器输出电平数的增加而降低。与开关型柔性直流输电技术相比，由于换流阀每相上桥臂和下桥臂每次投入或切出1个子模块，所以可控电压源型换流器可以采用较低的开关频率，一般不超过3倍基频，从而降低了损耗，提高了柔性直流输电系统的能量传输效率。
柔性直流输电技术采用的开关器件主要是全控型器件——绝缘栅双极型晶体管IGBT(insulated gate bipolar transistor)，在高压大容量柔性直流输电系统中，一个柔性直流换流器中采用的绝缘栅双极型晶体管IGBT超过两千只，所以绝缘栅双极型晶体管IGBT的稳定运行能力是影响柔性直流换流阀稳定运行的关键因数之一。而过载或能耗过大等原因引起绝缘栅双极型晶体管IGBT结温过高是导致其失效的最主要原因。因此，绝缘栅双极型晶体管IGBT和续流二极管FWD(free-wheeling diode)结温计算的准确度严重影响柔性直流换流器的可靠性。柔性直流换流器绝缘栅双极型晶体管IGBT和续流二极管FWD在各类运行状态下的损耗计算是解决对应结温计算的唯一有效手段。柔性直流换流阀的损耗示意图如图1所示，主要包括：
(1)绝缘栅双极型晶体管IGBT的损耗，其主要包括导通损耗、开通损耗、关断损耗和驱动电路损耗。其中开通损耗和关断损耗之和称之为开关损耗。
(2)续流二极管FWD的损耗，其主要包括导通损耗、反向恢复损耗、开通损耗和截止 损耗。其中，因开通时间很短，开通损耗可以忽略；二极管反向截止电流很小，截止损耗也可忽略。
绝缘栅双极型晶体管IGBT的开关损耗和续流二极管FWD的反向恢复损耗可参照厂家提供的对应参数，但这些器件的工况与厂家参数相关损耗对应的工况不同，仍需进一步折算。绝缘栅双极型晶体管IGBT和续流二极管FWD的导通损耗也可采用对应的导通阈值电压、导通电阻、导通电流等参数表示。但是桥臂电流在子模块中存在两条通路，即可能流经子模块上半桥臂绝缘栅双极型晶体管IGBT/下半桥臂续流二极管FWD或上半桥臂续流二极管FWD/下半桥臂绝缘栅双极型晶体管IGBT。因此，计算导通损耗还存在确定绝缘栅双极型晶体管IGBT与续流二极管FWD电流表达式的难题。
目前，有关电压源换流器的损耗计算主要还是针对开关型电压源换流器，但开关型电压源换流器的绝缘栅双极型晶体管IGBT与续流二极管FWD中的电流与可控电压源型电压源换流器的绝缘栅双极型晶体管IGBT与续流二极管FWD中的电流有着本质的区别，因此，现有的有关开关型电压源换流器的损耗计算方法不能适用于可控电压源型电压源换流器的损耗计算；针对可控电压源型换流器的损耗计算已有一些研究成果，但几乎都只采用了换流器的绝缘栅双极型晶体管IGBT与续流二极管FWD中的平均电流，而未涉及到换流器的绝缘栅双极型晶体管IGBT与续流二极管FWD中的有效电流。随着柔性直流输电工程快速推广应用，对柔性直流换流器损耗的计算程序的研究有着重要的理论与工程意义。
发明内容
针对上述现有技术中提到的换流器的损耗计算中没有采用子模块上半桥臂、下半桥臂的绝缘栅双极型晶体管IGBT、续流二极管FWD中的有效电流的不足，本发明的目的是提供一种基于模块化多电平电压源型换流器的损耗确定方法，本发明能反映出子模块中绝缘栅双极型晶体管IGBT与续流二极管FWD的损耗与柔性直流输电系统的调制因数、功率因素、有功传输功率等关系，将复杂问题直观化。
本发明的目的是采用下述技术方案实现的：
本发明提供一种基于模块化多电平电压源型换流器的损耗确定方法，所述模块化多电平电压源型换流器由三相构成，每相由串联的结构相同的上下两桥臂构成；上下两桥臂的中点处连接模块化多电平换流器的交流端；
所述上下两桥臂中每个桥臂包括1个电抗器和N个结构相同的子模块；每个桥臂的子模块级联后一端通过电抗器与模块化多电平换流器的交流端连接；每个桥臂的子模块级联后另一端与另两相桥臂的级联的子模块一端连接，形成模块化多电平电压源型换流器直流端的正 负极母线；
其改进之处在于，所述方法包括下述步骤：
步骤1：确定单相电压源型换流器的电流；
步骤2：确定电压源型换流器的导通损耗；
步骤3：确定电压源型换流器的开关损耗；
步骤4：确定电压源型换流器三相总损耗。
进一步地，所述步骤1包括下述子步骤：
步骤1.1：确定电压源型换流器的桥臂电流，表达式如下：
Figure PCTCN2014092231-appb-000001
其中：Idc为直流线路电流；Im为i相电流峰值；ω为基波角频率；
Figure PCTCN2014092231-appb-000002
为i相电流滞后i电压的角度；i为A相、B相或C相；
步骤1.2：确定子模块中上半桥臂和下半桥臂的电流；流经子模块上半桥臂和下半桥臂的电流分别为：
Figure PCTCN2014092231-appb-000003
其中：m为调制因数，即相电压峰值
Figure PCTCN2014092231-appb-000004
与直流线路对地电压绝对值UdcN的比值；
步骤1.3：分别确定子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的平均电流IT1_avg、IT2_avg及有效电流IT1_rms、IT2_rms；表达式分别如下：
Figure PCTCN2014092231-appb-000005
Figure PCTCN2014092231-appb-000006
Figure PCTCN2014092231-appb-000007
Figure PCTCN2014092231-appb-000008
步骤1.4：分别确定子模块上半桥臂、下半桥臂中续流二极管FWD的平均电流ID1_avg、ID2_avg及有效电流ID1_rms、ID2_rms；表达式分别如下：
Figure PCTCN2014092231-appb-000009
Figure PCTCN2014092231-appb-000010
Figure PCTCN2014092231-appb-000011
Figure PCTCN2014092231-appb-000012
进一步地，所述步骤2中，利用开关器件厂家提供的器件参数与步骤1中绝缘栅双极型晶体管IGBT与续流二极管FWD的平均电流与有效电流，确定电压源型换流器的导通损耗，包括：
在一个基波周期内，上半桥臂、下半桥臂的绝缘栅双极型晶体管IGBT和续流二极管FWD平均导通损耗分别为：
Figure PCTCN2014092231-appb-000013
Figure PCTCN2014092231-appb-000014
Figure PCTCN2014092231-appb-000015
Figure PCTCN2014092231-appb-000016
其中：
Figure PCTCN2014092231-appb-000017
分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的平均导通损耗；
UT1_0、UT2_0分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的导通阈值电压；
rT1_0、rT2_0分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的正向导通电阻；
Figure PCTCN2014092231-appb-000018
分别为子模块上半桥臂、下半桥臂中续流二极管FWD的平均导通损耗；
UD1_0、UD2_0分别为子模块上半桥臂、下半桥臂中续流二极管FWD的导通阈值电压；
rD1_0、rD2_0分别为子模块上半桥臂、下半桥臂中续流二极管FWD的正向导通电阻。
进一步地，所述步骤3包括下述子步骤：
步骤3.1：分别确定子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的开关损耗和续流二极管FWD的反向恢复损耗；
Figure PCTCN2014092231-appb-000019
Figure PCTCN2014092231-appb-000020
Figure PCTCN2014092231-appb-000021
Figure PCTCN2014092231-appb-000022
Prec＝PrecD1KD1+PrecD2KD2          (11)；
其中：
PswT1、PswT2分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的开关损耗；
PrecD1、PrecD2分别为子模块上半桥臂、下半桥臂中续流二极管FWD的反向恢复损耗；
fp为载波频率；
Eon_T1(Eon_T2)、Eoff_T1(Eoff_T2)分别为子模块上半桥臂或下半桥臂中绝缘栅双极型晶体管IGBT在参考电压uref_T1(uref_T2)、参考电流iref_T1(iref_T2)且最大工作结温的开通与关断损耗；
Erec_D1(Erec_D2)分别为子模块上半桥臂或下半桥臂中续流二极管FWD在参考电压uref_D1(uref_D2)、参考电流iref_D1(iref_D2)且最大工作结温的反向恢复损耗；所述最大工作结温取125℃；
Uc为子模块电容器电压；
步骤3.2：分别确定子模块上半桥臂或下半桥臂中绝缘栅双极型晶体管IGBT的开关损耗、续流二极管FWD的反向恢复损耗；
Psw＝PswT1KT1+PswT2KT2       (12)；
Prec＝PrecD1KD1+PrecD2KD2    (13)；
其中：KT1、KT2分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的结温系数；
KD1、KD2分别为子模块上半桥臂、下半桥臂中续流二极管FWD的结温系数；
步骤3.3：通过迭代法分别确定绝缘栅双极型晶体管IGBT的结温系数和续流二极管FWD的结温系数，表达式分别如下：
Figure PCTCN2014092231-appb-000023
Figure PCTCN2014092231-appb-000024
Figure PCTCN2014092231-appb-000025
Figure PCTCN2014092231-appb-000026
其中：
t为工作结温；
E1swT1、E2swT1分别为子模块上半桥臂绝缘栅双极型晶体管IGBT在125℃和25℃时的开关损耗；
E1swT2、E2swT2分别为子模块下半桥臂绝缘栅双极型晶体管IGBT在125℃和25℃时的开关损耗；
E1recD1、E2recD1分别为子模块上半桥臂续流二极管FWD在125℃和25℃时的反向恢复损耗；
E1recD2、E2recD2分别为子模块下半桥臂续流二极管FWD在125℃和25℃时的反向恢复损耗。
进一步地，所述步骤4中电压源型换流器三相总损耗用下述表达式表示：
Figure PCTCN2014092231-appb-000027
其中：N为换流器每个桥臂的子模块个数；
PT1drive、PT2drive分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的驱动电路的损耗。
与现有技术比，本发明达到的有益效果是：
1、本发明系统地分析了子模块上半桥臂、下半桥臂绝缘栅双极型晶体管IGBT与续流二 极管FWD中的有效电流，为可控电压源型换流器的损耗提供了计算方法。该计算结果与实际柔性直流输电工程基本吻合。因此，该损耗计算方法对柔性直流换流器的损耗评估与换流器冷却系统的配置具有较好的指导意义；
2、采用绝缘栅双极型晶体管IGBT与续流二极管FWD厂家惯用的利用器件导通阈值电压、平均电流、正向导通电阻与有效电流等物理量计算子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT、续流二极管FWD的导通损耗，物理意义明确；
3、依据子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT、续流二极管FWD的平均电流与有效电流的解析形式，损耗计算方法简单可靠；
4、子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT、续流二极管FWD的导通损耗与柔性直流输电系统调制因数、功率因数、传输有功功率的关系直观化。
附图说明
图1为子模块损耗的构成示意图；
图2为模块化多电平换流器结构示意图；
图3为本发明采用5SNA 1500E330305ABBHiPakTM型绝缘栅双极型晶体管IGBT的开关损耗与集电极电流的关系；
图4为本发明采用5SNA 1500E330305ABBHiPakTM型绝缘栅双极型晶体管IGBT中续流二极管反向恢复损耗、电流、电量与正向电流的关系；
图5为本发明提供的模块化多电平电压源型换流器的损耗确定方法的流程图。
具体实施方式
下面结合附图对本发明的具体实施方式作进一步的详细说明。
为解决绝缘栅双极型晶体管IGBT与续流二极管FWD损耗计算这一问题，本专利有效求解出换流器子模块上半桥臂绝缘栅双极型晶体管IGBT/下半桥臂续流二极管FWD或上半桥臂续流二极管FWD/下半桥臂绝缘栅双极型晶体管IGBT的平均电流和有效电流，提出了一种模块化多电平电压源型换流器的损耗确定方法。
该损耗确定方法的特点是能反映出子模块中绝缘栅双极型晶体管IGBT与续流二极管FWD的损耗与柔性直流输电系统的调制因数、功率因素、有功传输功率等关系，将复杂问题直观化。
本发明提供的模块化多电平电压源型换流器的损耗确定方法，其中模块化多电平电压源型换流器的结构示意图如图2所示，模块化多电平电压源型换流器由三相构成，每相由串联 的结构相同的上下两桥臂构成；上下两桥臂的中点处连接模块化多电平换流器的交流端；所述上下两桥臂中每个桥臂包括1个电抗器和N个结构相同的子模块；每个桥臂的子模块级联后一端通过电抗器与模块化多电平换流器的交流端连接；每个桥臂的子模块级联后另一端与另两相桥臂的级联的子模块一端连接，形成模块化多电平电压源型换流器直流端的正负极母线；所述子模块由半桥与其并联的电容器支路构成，所述半桥由上半桥臂和下半桥臂构成，所述上半桥臂和下半桥臂均由绝缘栅双极型晶体管IGBT以及与其并联的续流二极管FWD组成。
模块化多电平电压源型换流器的损耗确定方法的流程图如图5所示，包括下述步骤：
实施例：以逆变状态下的换流器A相上桥臂为例对本发明的方法进行描述。
步骤1：以逆变状态下的换流器A相上桥臂为例，换流器的构成以及各电流的参考方向如图2所示，计算可控电压源型换流器的电流；
步骤1.1：计算可控电压源型换流器桥臂电流；
Figure PCTCN2014092231-appb-000028
其中：
Idc为直流线路电流；
Im为A相电流峰值；
ω为基波角频率；
Figure PCTCN2014092231-appb-000029
为A相电流滞后A电压的角度。
步骤1.2：计算子模块上管、下管的电流；流经子模块上管、下管的电流分别为：
Figure PCTCN2014092231-appb-000030
其中
m为调制因数，即相电压峰值
Figure PCTCN2014092231-appb-000031
与直流线路对地电压绝对值UdcN的比值。
步骤1.3：分别计算子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的平均电流IT1_avg、IT2_avg及有效电流IT1_rms、IT2_rms表达式分别如下：
Figure PCTCN2014092231-appb-000032
Figure PCTCN2014092231-appb-000033
Figure PCTCN2014092231-appb-000034
Figure PCTCN2014092231-appb-000035
步骤1.4：分别确定子模块上半桥臂、下半桥臂中续流二极管FWD的平均电流ID1_avg、ID2_avg及有效电流ID1_rms、ID2_rms；表达式分别如下：
Figure PCTCN2014092231-appb-000036
Figure PCTCN2014092231-appb-000037
Figure PCTCN2014092231-appb-000038
Figure PCTCN2014092231-appb-000039
步骤2：利用开关器件厂家提供的器件参数与步骤1中计算绝缘栅双极型晶体管IGBT与续流二极管FWD的平均电流与有效电流，计算可控电压源型换流器的导通损耗；
在一个基波周期内，上半桥臂、下半桥臂的绝缘栅双极型晶体管IGBT、续流二极管FWD平均导通损耗分别为：
Figure PCTCN2014092231-appb-000040
Figure PCTCN2014092231-appb-000041
Figure PCTCN2014092231-appb-000042
Figure PCTCN2014092231-appb-000043
其中：
Figure PCTCN2014092231-appb-000044
为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的平均导通损耗；
UT1_0、UT2_0分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的导通阈值电压；
rT1_0、rT2_0分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的正向导通电阻；
Figure PCTCN2014092231-appb-000045
为子模块上半桥臂、下半桥臂中续流二极管FWD的平均导通损耗；
UD1_0、UD2_0分别为子模块上半桥臂、下半桥臂中续流二极管FWD的导通阈值电压；
rD1_0、rD2_0分别为子模块上半桥臂、下半桥臂中续流二极管FWD的正向导通电阻。
步骤3：计算可控电压源型换流器的开关损耗；
步骤3.1：分别计算上下桥臂绝缘栅双极型晶体管IGBT的开关损耗、续流二极管FWD的反向恢复损耗；
Figure PCTCN2014092231-appb-000046
Figure PCTCN2014092231-appb-000047
Figure PCTCN2014092231-appb-000048
Figure PCTCN2014092231-appb-000049
Prec＝PrecD1KD1+PrecD2KD2      (11)；
其中：
PswT1、PswT2分别为子模块上下桥臂绝缘栅双极型晶体管IGBT的开关损耗；
PrecD1、PrecD2为子模块上下桥臂续流二极管FWD的反向恢复损耗；
fp为载波频率；
Eon_T1(Eon_T2)、Eoff_T1(Eoff_T2)分别为子模块上(下)桥臂绝缘栅双极型晶体管IGBT在参考电压uref_T1(uref_T2)、参考电流iref_T1(iref_T2)和最大工作结温(常取125℃)的开通与关断损耗；
Erec_D1(Erec_D2)为子模块上(下)桥臂续流二极管FWD在参考电压uref_D1(uref_D2)、参考电流iref_D1(iref_D2)和最大工作结温(常取125℃)的反向恢复损耗；
Uc为子模块电容器电压。
步骤3.2：分别计算绝缘栅双极型晶体管IGBT的开关损耗、续流二极管FWD的反向恢复损耗；
Psw＝PswT1KT1+PswT2KT2      (12)；
Prec＝PrecD1KD1+PrecD2KD2     (13)；
KT1、KT2分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的结温系数；
KD1、KD2分别为子模块上半桥臂、下半桥臂中续流二极管FWD的结温系数。
本发明采用5SNA 1500E330305ABBHiPakTM型绝缘栅双极型晶体管IGBT的开关损耗与集电极电流的关系以及绝缘栅双极型晶体管IGBT中续流二极管反向恢复损耗、电流、电量与正向电流的关系分别如图3和图4所示。
步骤3.2：通过迭代法分别计算绝缘栅双极型晶体管IGBT的结温系数和续流二极管FWD的结温系数；
Figure PCTCN2014092231-appb-000050
Figure PCTCN2014092231-appb-000051
Figure PCTCN2014092231-appb-000052
Figure PCTCN2014092231-appb-000053
其中：
t为工作结温；
E1swT1、E2swT1分别为子模块上半桥臂绝缘栅双极型晶体管IGBT在125℃、25℃时的开关损耗；
E1swT2、E2swT2分别为子模块下半桥臂绝缘栅双极型晶体管IGBT在125℃、25℃时的开关损耗；
E1recD1、E2recD1分别为子模块上半桥臂续流二极管FWD在125℃、25℃时的反向恢复损耗；
E1recD2、E2recD2分别为子模块下半桥臂续流二极管FWD在125℃、25℃时的反向恢复损耗。
步骤4：计算可控电压源型换流器三相总损耗。
所述可控电压源型换流器三相总损耗的计算公式为：
Figure PCTCN2014092231-appb-000054
其中：
N为换流器每个桥臂的子模块个数；
PT1drive、PT2drive分别为子模块上半桥臂、下半桥臂绝缘栅双极型晶体管IGBT的驱动电路的损耗。
若换流器以整流状态运行时，由于流经桥臂的实际直流电流分量与整流状态下的桥臂直流电流分量方向相反，而交流电流分量相同，则整流器子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的电流可近似(忽略换流站的损耗)等效为逆变器子模块上半桥臂、下半桥臂中续流二极管FWD的电流；子模块上半桥臂、下半桥臂中续流二极管FWD的电流可近似等效为逆变器子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的电流。因此，换流器在整流状态下的开关器件损耗的计算可参照其逆变状态下开关器件损耗的计算。
本发明系统地分析了子模块上半桥臂、下半桥臂绝缘栅双极型晶体管IGBT与续流二极管FWD中的有效电流，为可控电压源型换流器的损耗程序计算提供了计算方法。该程序的计算结果与实际柔性直流输电工程基本吻合。因此，该损耗计算程序对柔性直流换流器的损耗评估与换流器冷却系统的配置具有较好的指导意义。
最后应当说明的是：以上实施例仅用以说明本发明的技术方案而非对其限制，尽管参照上述实施例对本发明进行了详细的说明，所属领域的普通技术人员应当理解：依然可以对本发明的具体实施方式进行修改或者等同替换，而未脱离本发明精神和范围的任何修改或者等同替换，其均应涵盖在本发明的权利要求范围当中。

Claims (5)

  1. 一种基于模块化多电平电压源型换流器的损耗确定方法，所述模块化多电平电压源型换流器由三相构成，每相由串联的结构相同的上下两桥臂构成；上下两桥臂的中点处连接模块化多电平换流器的交流端；
    所述上下两桥臂中每个桥臂包括1个电抗器和N个结构相同的子模块；每个桥臂的子模块级联后一端通过电抗器与模块化多电平换流器的交流端连接；每个桥臂的子模块级联后另一端与另两相桥臂的级联的子模块一端连接，形成模块化多电平电压源型换流器直流端的正负极母线；
    其特征在于，所述方法包括下述步骤：
    步骤1：确定单相电压源型换流器的电流；
    步骤2：确定电压源型换流器的导通损耗；
    步骤3：确定电压源型换流器的开关损耗；
    步骤4：确定电压源型换流器三相总损耗。
  2. 如权利要求1所述的损耗确定方法，其特征在于，所述步骤1包括下述子步骤：
    步骤1.1：确定电压源型换流器的桥臂电流，表达式如下：
    Figure PCTCN2014092231-appb-100001
    其中：Idc为直流线路电流；Im为i相电流峰值；ω为基波角频率；
    Figure PCTCN2014092231-appb-100002
    为i相电流滞后i电压的角度；i为A相、B相或C相；
    步骤1.2：确定子模块中上半桥臂和下半桥臂的电流；流经子模块上半桥臂和下半桥臂的电流分别为：
    Figure PCTCN2014092231-appb-100003
    其中：m为调制因数，即相电压峰值
    Figure PCTCN2014092231-appb-100004
    与直流线路对地电压绝对值UdcN的比值；
    步骤1.3：分别确定子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的平均电流IT1_avg、IT2_avg及有效电流IT1_rms、IT2_rms；表达式分别如下：
    Figure PCTCN2014092231-appb-100005
    Figure PCTCN2014092231-appb-100006
    步骤1.4：分别确定子模块上半桥臂、下半桥臂中续流二极管FWD的平均电流ID1_avg、ID2_avg及有效电流ID1_rms、ID2_rms；表达式分别如下：
    Figure PCTCN2014092231-appb-100007
  3. 如权利要求1所述的损耗确定方法，其特征在于，所述步骤2中，利用开关器件厂家提供的器件参数与步骤1中绝缘栅双极型晶体管IGBT与续流二极管FWD的平均电流与有效电流，确定电压源型换流器的导通损耗，包括：
    在一个基波周期内，上半桥臂、下半桥臂的绝缘栅双极型晶体管IGBT和续流二极管FWD平均导通损耗分别为：
    Figure PCTCN2014092231-appb-100008
    Figure PCTCN2014092231-appb-100009
    Figure PCTCN2014092231-appb-100010
    Figure PCTCN2014092231-appb-100011
    其中：
    Figure PCTCN2014092231-appb-100012
    分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的平均导通损耗；
    UT1_0、UT2_0分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的导通阈值电压；
    rT1_0、rT2_0分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的正向导通电阻；
    Figure PCTCN2014092231-appb-100013
    分别为子模块上半桥臂、下半桥臂中续流二极管FWD的平均导通损耗；
    UD1_0、UD2_0分别为子模块上半桥臂、下半桥臂中续流二极管FWD的导通阈值电压；
    rD1_0、rD2_0分别为子模块上半桥臂、下半桥臂中续流二极管FWD的正向导通电阻。
  4. 如权利要求1所述的损耗确定方法，其特征在于，所述步骤3包括下述子步骤：
    步骤3.1：分别确定子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的开关损耗和续流二极管FWD的反向恢复损耗；
    Figure PCTCN2014092231-appb-100014
    Figure PCTCN2014092231-appb-100015
    Figure PCTCN2014092231-appb-100016
    Figure PCTCN2014092231-appb-100017
    Prec＝PrecD1KD1+PrecD2KD2                (11)；
    其中：
    PswT1、PswT2分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的开关损耗；
    PrecD1、PrecD2分别为子模块上半桥臂、下半桥臂中续流二极管FWD的反向恢复损耗；
    fp为载波频率；
    Eon_T1(Eon_T2)、Eoff_T1(Eoff_T2)分别为子模块上半桥臂或下半桥臂中绝缘栅双极型晶体管IGBT在参考电压uref_T1(uref_T2)、参考电流iref_T1(iref_T2)且最大工作结温的开通与关断损耗；
    Erec_D1(Erec_D2)分别为子模块上半桥臂或下半桥臂中续流二极管FWD在参考电压uref_D1(uref_D2)、参考电流iref_D1(iref_D2)且最大工作结温的反向恢复损耗；所述最大工作结温取125℃；
    Uc为子模块电容器电压；
    步骤3.2：分别确定子模块上半桥臂或下半桥臂中绝缘栅双极型晶体管IGBT的开关损耗、续流二极管FWD的反向恢复损耗；
    Psw＝PswT1KT1+PswT2KT2        (12)；
    Prec＝PrecD1KD1+PrecD2KD2        (13)；
    其中：KT1、KT2分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的结温系数；
    KD1、KD2分别为子模块上半桥臂、下半桥臂中续流二极管FWD的结温系数；
    步骤3.3：通过迭代法分别确定绝缘栅双极型晶体管IGBT的结温系数和续流二极管FWD的结温系数，表达式分别如下：
    Figure PCTCN2014092231-appb-100018
    Figure PCTCN2014092231-appb-100019
    Figure PCTCN2014092231-appb-100020
    Figure PCTCN2014092231-appb-100021
    其中：
    t为工作结温；
    E1swT1、E2swT1分别为子模块上半桥臂绝缘栅双极型晶体管IGBT在125℃和25℃时的开关损耗；
    E1swT2、E2swT2分别为子模块下半桥臂绝缘栅双极型晶体管IGBT在125℃和25℃时的开关损耗；
    E1recD1、E2recD1分别为子模块上半桥臂续流二极管FWD在125℃和25℃时的反向恢复损耗；
    E1recD2、E2recD2分别为子模块下半桥臂续流二极管FWD在125℃和25℃时的反向恢复损耗。
  5. 如权利要求1所述的损耗确定方法，其特征在于，所述步骤4中电压源型换流器三相总损耗用下述表达式表示：
    Figure PCTCN2014092231-appb-100022
    其中：N为换流器每个桥臂的子模块个数；
    PT1drive、PT2drive分别为子模块上半桥臂、下半桥臂中绝缘栅双极型晶体管IGBT的驱动电路的损耗。
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