WO2015002658A1 - Antenne de surface à impédance artificielle orientable électroniquement - Google Patents

Antenne de surface à impédance artificielle orientable électroniquement Download PDF

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Publication number
WO2015002658A1
WO2015002658A1 PCT/US2013/050412 US2013050412W WO2015002658A1 WO 2015002658 A1 WO2015002658 A1 WO 2015002658A1 US 2013050412 W US2013050412 W US 2013050412W WO 2015002658 A1 WO2015002658 A1 WO 2015002658A1
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WIPO (PCT)
Prior art keywords
dielectric substrate
impedance
wave
antenna
artificial impedance
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Application number
PCT/US2013/050412
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English (en)
Inventor
Daniel J. Gregoire
Joseph S. Colburn
Original Assignee
Hrl Laboratories, Llc
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Filing date
Publication date
Priority claimed from US13/934,553 external-priority patent/US9466887B2/en
Application filed by Hrl Laboratories, Llc filed Critical Hrl Laboratories, Llc
Priority to EP13888596.7A priority Critical patent/EP3017504B1/fr
Priority to CN201380077921.1A priority patent/CN105379011B/zh
Publication of WO2015002658A1 publication Critical patent/WO2015002658A1/fr

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Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/20—Non-resonant leaky-waveguide or transmission-line antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/28—Non-resonant leaky-waveguide or transmission-line antennas; Equivalent structures causing radiation along the transmission path of a guided wave comprising elements constituting electric discontinuities and spaced in direction of wave propagation, e.g. dielectric elements or conductive elements forming artificial dielectric
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q15/00—Devices for reflection, refraction, diffraction or polarisation of waves radiated from an antenna, e.g. quasi-optical devices
    • H01Q15/0006—Devices acting selectively as reflecting surface, as diffracting or as refracting device, e.g. frequency filtering or angular spatial filtering devices
    • H01Q15/006—Selective devices having photonic band gap materials or materials of which the material properties are frequency dependent, e.g. perforated substrates, high-impedance surfaces
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
    • H01Q3/44—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the electric or magnetic characteristics of reflecting, refracting, or diffracting devices associated with the radiating element
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q3/00—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system
    • H01Q3/44—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the electric or magnetic characteristics of reflecting, refracting, or diffracting devices associated with the radiating element
    • H01Q3/443—Arrangements for changing or varying the orientation or the shape of the directional pattern of the waves radiated from an antenna or antenna system varying the electric or magnetic characteristics of reflecting, refracting, or diffracting devices associated with the radiating element varying the phase velocity along a leaky transmission line

Definitions

  • This disclosure relates to artificial impedance surface antennas (AISAs) . More specifically, it presents a low cost, 2D, electronically steerable, artificial impedance, surface antenna.
  • phased array antennas have complex electronics and are quite costly.
  • AISAs electronically steered artificial impedance surface antennas
  • U.S. Patent Nos . 7,245,269, 7,071,888, and 7,253,780 to Sievenpiper These antennas are useful for some applications, but are not suitable for all applications that need two dimensional steering.
  • mechanical steering can be used to provide steering of a ID electronically steered antenna in a second dimension.
  • the antennas described by Sievenpiper also require vias for providing voltage control to varactors.
  • AISAS Artificial impedance surface antennas
  • AIS artificial impedance surface
  • references [1] - [6] describe artificial impedance surface antennas (AISA) formed from modulated artificial impedance surfaces (AIS) .
  • AISA artificial impedance surface antennas
  • AIS modulated artificial impedance surfaces
  • Patel [1] demonstrated a scalar AISA using an end-fire, flare- fed one-dimensional, spatially-modulated AIS consisting of a linear array of metallic strips on a grounded dielectric.
  • Sievenpiper, Colburn and Fong [2] - [4] have demonstrated scalar and tensor AISAs on both flat and curved surfaces using waveguide- or dipole-fed, two-dimensional, spatially-modulated AIS consisting of a grounded dielectric topped with a grid of metallic patches.
  • Gregoire [5] -[6] has examined the dependence of AISA operation on its design properties.
  • the basic principle of AISA operation is to use the grid momentum of the modulated AIS to match the wave vectors of an excited surface-wave front to a desired plane wave.
  • this can be expressed as
  • k a is the radiation's free-space wavenumber at the design freguency
  • ⁇ 0 is the angle of the desired radiation with respect to the AIS normal
  • n Q is the surface wave's refractive index averaged over the AIS modulation.
  • the SW impedance is typically chosen to have a pattern that modulates the SW impedance sinusoidally along the SWG according to
  • p is the period of the modulation
  • X is the mean impedance
  • M is the modulation amplitude. X, M and are chosen such that the angle of the radiation ⁇ in the x-z plane w.r.t the z axis is determined by
  • n 0 is related to Z (x) by
  • the AISA impedance modulation of Eqn. (2) can be generalized for an AISA of any shape as
  • the AIS can be realized as a grid of metallic patches on a grounded dielectric.
  • modulation is produced by varying the size of the patches according to a function that correlates the patch size to the surface wave index.
  • the correlation between index and patch size can be determined using simulations, calculation and/or measurement techniques. For example, Colburn [3] and Fong [4] use a combination of HFSS unit-cell eigenvalue simulations and near field measurements of test boards to determine their correlation function. Fast approximate methods presented by Luukkonen [7] can also be used to calculate the correlation. However, empirical correction factors are often applied to these methods. In many regimes, these methods agree very well with HFSS eigenvalue simulations and near-field measurements. They break down when the patch size is large compared to the substrate thickness, or when the surface-wave phase shift per unit cell approaches 180°.
  • the AIS is a grid of metallic patches on a dielectric substrate.
  • the surface-wave impedance is locally controlled at each position on the AIS by applying a variable voltage to voltage-variable varactors connected between each of the patches.
  • an AIS's SW impedance can be tuned with capacitive loads inserted between impedance elements [8] , [9] .
  • Each patch is electrically connected to neighboring patches on all four sides with voltage-variable varactor capacitor. The voltage is applied to the varactors though electrical vias connected to each impedance element patch.
  • Half of the patches are electrically connected to the groundplane with vias that run from the center of each patch down through the dielectric substrate.
  • the rest of the patches are electrically connected to voltage sources that run through the substrates, and through holes in the ground plane to the voltage sources .
  • a steerable artificial impedance surface antenna steerable in phi and theta angles comprises dielectric substrate, a plurality of metallic strips on a first surface of the dielectric
  • the metallic strips spaced apart across a length of the dielectric substrate and each metallic strip extending along a width of the dielectric substrate, and Surface wave feeds spaced apart along the width of the dielectric substrate near an edge of the dielectric substrate, wherein the dielectric substrate is substantially in an X-Y plane defined by an X axis and a Y axis, wherein the phi angle is an angle in the X-Y plane relative to the X axis, and wherein theta angle is an angle relative to a Z axis orthogonal to the X-Y plane .
  • a steerable artificial impedance surface antenna steerable in phi and theta angles comprises a dielectric substrate, a plurality of metallic strips on a first surface of the dielectric
  • the dielectric substrate is substantially in an X-Y plane defined by an X axis and a Y axis, wherein the phi angle is an angle in the X-Y plane relative to the X axis, and wherein the theta angle is an angle relative to a Z axis orthogonal to the X-Y plane.
  • FIG. 1 shows surface waves propagating outward from a source interact with the modulated impedance to produce radiation in a narrow beam in accordance with the prior art
  • FIG. 2A shows an electronically steered artificial impedance surface antenna (AISA)
  • FIG. 2B shows a side elevation view of an AISA in accordance with the present disclosure ,-
  • FIG. 3 is a diagram of a spherical coordinate system showing the angles and the transformations to Cartesian coordinates in accordance with the prior art,-
  • FIG. 4 shows another electronically steered artificial impedance surface antenna (AISA) in accordance with the present disclosure ,-
  • FIG. 5 shows yet another electronically steered artificial impedance surface antenna (AISA) in accordance with the present disclosure
  • FIG. 6 shows another side elevation view of an AISA in accordance with the present disclosure
  • FIG. 7 shows yet another side elevation view of an AISA in accordance with the present disclosure.
  • FIG. 2A shows an electronically steered artificial impedance surface antenna (AISA) in accordance with the present disclosure that is relatively low cost and capable of steering in both theta ⁇ ) and phi ( ⁇ p) directions.
  • FIG. 3 is a diagram of a spherical coordinate system showing the theta
  • the phi ( ) angle is the angle in the x-y plane
  • the theta ⁇ ) angle is the angle from the z axis.
  • AISA in accordance with the present disclosure is capable of steering in both theta ⁇ ) and phi ( ⁇ ) directions, those skilled in the art refer to it as a 2D electronically steered artificial impedance surface antenna (AISA) .
  • the electronically steered artificial impedance surface antenna (AISA) of FIG. 2A includes a tunable
  • AISA artificial impedance surface antenna
  • voltage control network 102 voltage control network
  • the tunable artificial impedance surface antenna (AISA) 101 is in the X-Y plane of FIG. 3, the steering of the primary gain lobe of the electronically steered artificial impedance surface antenna (AISA) is controlled in the phi ( ⁇ ) direction by changing the relative phase difference between the RF surface wave feeds 108 of the ID RF feed network 103.
  • the theta steering is controlled by varying or modulating the surface wave impedance of the tunable artificial impedance surface antenna (AISA) 101.
  • the artificial impedance surface antenna (AISA) 101 in the embodiment of FIG. 2A includes a dielectric substrate
  • the impedance of the AISA 101 may be varied or modulated by controlling voltages to the metallic strips 107 on the tunable artificial impedance surface antenna (AISA) 101.
  • the voltages on the metallic strips 107 change the capacitance of varactors 109 between the metallic strips
  • the voltage control network 102 applies direct current (DC) voltages to the metallic strips 107 on the AISA structure.
  • Control bus 105 provides control for the voltage control network 102.
  • the control bus 105 may be from a microprocessor, central processing unit, or any computer or processor .
  • Control bus 104 provides control for the ID RF feed network 103.
  • the control bus 104 may be from a
  • microprocessor central processing unit, or any computer or processor.
  • FIG. 2B shows a side elevation view of FIG. 2A.
  • varactors 109 are between the metallic strips 107, which are on the surface of the dielectric substrate 106.
  • the dielectric substrate 106 may or may not have a ground plane 119 on a surface opposite to the surface upon which the metallic strips 107 are located.
  • varactors are not between the metallic strips 107.
  • the dielectric substrate 106 may further include a material 404 with tunable electrical properties, such as a liquid crystal.
  • the impedance elements such as the metallic strips 107, which may be formed, deposited, printed, or pasted onto the dielectric substrate 106
  • the properties of the dielectric substrate 106, or the material 404 with tunable electrical properties may change.
  • the dielectric constant may change, thereby changing the impedance between the metallic strips 107, and thereby steering a beam in the theta direction.
  • a varactor is a type of diode whose capacitance varies as a function of the voltage applied across its terminals, which makes it useful for tuning applications.
  • varactors 109 are used between the metallic strips 107, as shown in FIG. 2A, by controlling the voltage applied to the varactors 109 via the metallic strips 107, the capacitances of the varactors 109 vary, which in turn varies or modulates the capacitive coupling and the impedance between the metallic strips 107 to steer a beam in the theta direction.
  • the polarities of the varactors 109 are aligned such that all the varactor connections to any one of the metallic strips 107 are connected with the same polarity.
  • One terminal on a varactor may be referred to as an anode, and the other terminal as a cathode.
  • some of the metallic strips 107 are only connected to anodes of varactors 109, and other metallic strips 107 are only connected to cathodes of varactors 109.
  • adjacent metallic strips 107 on the AISA 101 alternate in being connected to anodes or cathodes of varactors 109.
  • the spacing of the metallic strips 107 in one dimension of the AISA may be a fraction of the RF surface wave (SW) wavelength of the RF waves that propagate across the AISA from the RF surface wave feeds 108.
  • the spacing of the metallic strips 107 may be at most 1/5 of the RF surface wave (SW) wavelength of the RF waves.
  • the fraction may be only about 1/10 of the RF surface wave
  • the spacing between varactors 109 connected to the metallic strips 107 in a second dimension of the AISA is typically about the same as the spacing between metallic strips.
  • the RF SW feeds 108 may be a phased array corporate feed structure, or may be conformal surface wave feeds, which are integrated into the AISA, such as by using micro-strips.
  • Conformal surface wave feeds that may be used include those described in U.S. Patent Application Serial No. 13/242,102 filed September 23, 2011, or those described in "Directional Coupler for Transverse-Electric Surface Waves", published in IP.com Prior Art Database Disclosure No. IPCO 000183639D, May 29, 2009, which are incorporated herein by reference as though set forth in full.
  • the spacing between the RF SW feeds 108 in the second dimension of the AISA or the y dimension of FIG. 3, may be based on rules of thumb for phased array antennas that dictate they be no farther apart than 1/2 of the free-space wavelength for the highest frequency signal to be transmitted or received.
  • the thickness of the dielectric substrate 106 is determined by its permittivity and the frequency of radiation to be transmitted or received. The higher the permittivity, the thinner the substrate can be.
  • the capacitance values of the varactors 109 are determined by the range necessary for the desired AISA impedance modulations to obtain the various angles of radiation.
  • An AISA operating at about 10 GHz may use for the dielectric substrate 106, a 50-mil thick Rogers Corp 3010 circuit board material with a relative permittivity equal to 11.2.
  • the metallic strips 107 may be spaced 2 millimeters (mm) to 3 mm apart on the dielectric substrate 106.
  • the RF surface wave feeds 108 may be spaced 1.5 centimeters (cm) apart and the varactors 109 may be spaced 2 mm to 3 mm apart.
  • the varactors 109 vary in capacitance from 0.2 to 2.0 pico farads (pF) . Designs for different radiation frequencies or designs using different substrates will vary accordingly.
  • the transmit/receive module 110 is connected to the feed network 103.
  • the feed network 103 can be of any type that is known to those skilled in the state of the art of phased array antennas.
  • the feed network 103 shown in FIG. 2A includes a series of RF transmission lines 111 connected to the transmit /receive module 110, power dividers 112, and phase shifters 113.
  • the phase shifters 113 are controlled by voltage control lines 118 from a digital to analog converter (DAC) 114 that receives digital control signals 104 to control the steering in the phi ( ) direction.
  • DAC digital to analog converter
  • the antenna main lobe is steered in the phi direction by using the feed network 103 to impose a phase shift between each of the RF SW feeds 108. If the RF SW feeds 108 are spaced uniformly, then the phase shift between adjacent RF SW feeds 108 is constant.
  • the relation between the phi ( ⁇ ) steering angle, and the phase shift may be calculated using standard phased array methods, according to equation,
  • ⁇ is the radiation wavelength
  • d is the spacing between SW feeds 108
  • ⁇ is the phase shift between SW feeds 108.
  • the RF SW feeds 108 may also be spaced non-uniformly, and the phase shifts adjusted accordingly.
  • the antenna lobe is steered in the theta ( ⁇ ) direction by applying voltages to the varactors 109 between the metallic strips 107 such that AISA 101 has surface-wave impedance Z sw , that is modulated or varied periodically with the distance (x) away from the SW feeds 108, according to equation,
  • X and M are the mean impedance and the amplitude of its modulation respectively, and p is the modulation period.
  • the variation of the surface-wave impedance Z sw may be modulated sinusoidally .
  • the theta steering angle ⁇ is related to the impedance modulation by the equation,
  • ⁇ is the wavelength of the radiation
  • the beam is steered in the theta direction by tuning the varactor voltages such that X, M, and p result in the desired theta ⁇ .
  • the dependence of the surface wave (SW) impedance on the varactor capacitance is calculated using transcendental equations resulting from the transverse resonance method or by using full-wave numerical simulations.
  • voltages are applied to the varactors 109 by grounding alternate metallic strips 107 to ground 120 and applying tunable voltages via voltage control lines 116 to the rest of the strips 107.
  • the voltage applied to each voltage control line 116 is a function of the desired theta ( ⁇ ) , and may be different for each voltage control line 116.
  • the voltages may be applied from a digital- to-analog converter (DAC) 117 that receives digital controls 105 from a controller for steering in the theta direction.
  • the controller may be a microprocessor, central processing unit (CPU) or any computer, processor or controller.
  • An advantage of grounding half of the metallic strips 107 is that only half as many voltage control lines 116 are required as there are metallic strips 107.
  • a disadvantage is that the spatial resolution of the voltage control and hence the impedance modulation is limited to twice the. spacing between metallic strips.
  • FIG. 4 shows another electronically steered artificial impedance surface antenna (AISA) in accordance with the present disclosure that is essentially the same as the embodiment described with reference to FIG. 2A, except in the embodiment of FIG. 4, a voltage is applied to each of the metallic strips 207 by voltage control lines 216. Twice as many control voltages are required compared to the embodiment of FIG. 2A, however, the spatial resolution of the impedance modulation is doubled.
  • the voltage applied to each voltage control line 216 is a function of the desired theta ( ⁇ ) angle, and may be different for each voltage control line 216.
  • the voltages are applied from a digital-to-analog converter (DAC) 217 that receives digital controls 205 from an outside source, which may be a microprocessor, central processing unit (CPU) or any computer or processor, for steering in the theta direction.
  • DAC digital-to-analog converter
  • the antenna main lobe is steered in the phi direction by using the feed network 203 to impose a phase shift between each of the RF SW feeds 208 in the same manner as described with reference to FIG. 1.
  • FIG. 5 illustrates a preferred embodiment where the theta ⁇ angle control DACs 117 and 217 of FIGs, 2A and 4 are replaced by a single control voltage from a variable voltage source 350.
  • source 350 As the voltage of variable voltage: source 350 is varied, the AISA radiation angle varies between a minimum and maximum theta angle that is determined by the details of the AISA design.
  • the voltage is applied though voltage control lines 352 and 354 to the metallic strips 340 on the surface of the AISA.
  • Voltage control line 354 may be a ground with the voltage control line 352 being a variable voltage. Across the x dimension, the metallic strips 340 are alternately tied to voltage control line 352 or to voltage control line 354.
  • One or more varactors diodes 309 may be in each gap between adjacent metallic strips 340 and electrically connected to the metallic strips in the same manner as shown in FIG. 2A.
  • the metallic strips may have centers that are equally spaced in the x dimension, with the widths of the metallic strips 340 periodically varying with a period p 346.
  • the number of metallic strips in a period 346 can be any number, although 10 to 20 is reasonable for most designs.
  • the width variation is designed to produce surface-wave impedance with a periodic modulation in the X-direction with period p 346, for example, the sinusoidal variation of equation (8) above .
  • the surface-wave impedance at each point on the AISA is determined by the width of the metallic strips and the voltage applied to the varactors 309. The relation between the surface-wave impedance and these parameters is well understood and documented in the references [1] - [9] .
  • the capacitance of the diode varactors 309 varies with the applied voltage. When the voltage is 0 volts, the diode capacitance is at its maximum value of C max . The capacitance decreases as the voltage is increased until it reaches a minimum value of C m i n . As the diode capacitance is varied, the impedance modulation parameters, X and M in Eqn.
  • the substrate 401 which may be used for dielectric substrates 106, 206 or 306, is a material whose electrical permittivity is varied with application of an electric field. As described above, no varactors 109, 209 or 309 are used in this embodiment.
  • a voltage is applied to metallic strips 402 on the AISA, an electric field is produced between adjacent strips and also between the strips and the substrate ground plane 403. The electric field changes the permittivity of the substrate material, which results in a change in the capacitance between adjacent metallic strips 402. As in the other embodiments, the capacitance between adjacent metallic strips 402 determines the surface-wave impedance.
  • variable material 404 may be any electrically variable material, such as liquid crystal material or barium strontium titanate (3 ⁇ 4ST) . It may be necessary, especially in the case of using liquid crystals, to embed the variable material 404 in pockets within an inert substrate 405, as shown in FIG. 7.
  • the antenna main lobe is steered in the phi direction by using the feed network 303 to impose a phase shift between each of the RF S feeds 308 in the same manner as described with reference to FIG. 2A.
  • a steerable artificial impedance surface antenna steerable in phi and theta angles comprising:
  • the dielectric substrate is substantially in an X-Y plane defined by an X axis and a Y axis;
  • phi angle is an angle in the: X-Y plane relative to the X axis
  • theta angle is an angle relative to a Z axis orthogonal to the X-Y plane.
  • the steerable artificial impedance surface antenna of concept 1 or 17 further comprising:
  • the tunable element comprises a plurality of varactors coupled between each adjacent pair of metallic strips.
  • Concept 4 The steerable artificial impedance surface antenna of concept 3 wherein:
  • each respective varactor coupled to a respective metallic strip has a same polarity of the respective varactor coupled to the respective metallic strip.
  • the tunable element comprises an electrically variable material between adjacent metallic strips.
  • the electrically variable material comprises a liquid crystal material or barium strontium titanate (BST) .
  • the dielectric substrate is an inert substrate; and the electrically variable material is embedded within an inert substrate.
  • the steerable artificial impedance surface antenna of concept 8 further comprising:
  • RF radio frequency
  • the steerable artificial impedance surface antenna of concept 9 wherein the radio frequency (RF) feed network comprises:
  • phase shifters coupled to the transmit/receive module and to a respective surface wave feed
  • phase shift controller coupled to the phase shifters.
  • alternating metallic strips of the plurality of metallic strips are coupled to a ground
  • each metallic strip not coupled to ground is coupled to a respective voltage from a voltage source
  • each metallic strip is coupled to a voltage source
  • the steerable artificial impedance surface antenna of concept 1 further comprising:
  • a ground plane on a second surface of the dielectric substrate opposite the first surface of the dielectric substrate .
  • the metallic strips have centers spaced apart by a fraction of a wavelength of a surface wave propagated across the dielectric substrate;
  • fraction is less than or equal to 0.2.
  • the tunable elements are varactors
  • a spacing between adjacent varactors coupled between two adjacent metallic strips is approximately the same as the spacing between centers of adjacent metallic strips.
  • the artificial impedance surface antenna has a surface- wave impedance Z sw , that is modulated or varied periodically by applying voltages to the metallic strips such that at distance ⁇ x) away from the surface wave feeds the surface wave impedance varies according to : where X and M are a mean impedance and an amplitude of modulation respectively, and p is a modulation period; and the theta angle is related to the surface wave impedance modulation by
  • ⁇ is a wavelength of a surface wave propagated across the dielectric substrate
  • a steerable artificial impedance surface antenna steerable in phi and theta angles comprising:
  • phi angle is an angle in the X-Y plane relative to the X axis
  • the steerable artificial impedance surface antenna of concept 17 further comprising:
  • a ground plane on a second surface of the dielectric substrate opposite the first surface of the dielectric substrate .
  • alternating metallic strips of the pplurality of metallic strips are coupled to a first terminal of a variable voltage source
  • each metallic strip not coupled to the first terminal is coupled to a second terminal of the variable voltage source; wherein the surface wave impedance of the artificial impedance surface antenna is varied by changing a voltage between the first and second terminals of the variable voltage source .
  • the tunable element comprises an electrically variable material between adjacent metallic strips.
  • the electrically variable material comprises a liquid crystal material or barium strontium titanate (T3ST) .
  • the dielectric substrate is an inert substrate; and the electrically variable material is embedded within an inert substrate .
  • the surface wave feeds are configured so that a relative phase difference between each surface wave feed determines the phi angle for a primary gain lobe of the electronically steered artificial impedance surface antenna (AISA) .
  • AISA electronically steered artificial impedance surface antenna
  • a ground plane on a second surface of the dielectric substrate opposite the first surface of the dielectric substrate.
  • alternating metallic strips of the plurality of metallic strips are coupled to a first terminal of a variable voltage source
  • each metallic strip not coupled to the first terminal is coupled to a second terminal of the variable voltage source
  • the surface wave impedance of the artificial impedance surface antenna is varied by changing a voltage between the first and second terminals of the variable voltage source.
  • the steerable artificial impedance surface antenna of claim 17 further comprising:
  • RF radio frequency
  • This invention relates to an electrically tunable surface impedance structure with a suppressed backward wave.
  • Surface impedance structures are a tunable electrically tunable surface impedance structure is taught by US Patent Nos. 6,538,621 and 7,068,234. This disclosure relates to a technique for reducing the propensity of the structures taught by US Patent Nos. 6,538,621 and 7,068,234 to generate a backward wave.
  • Fig. la depicts a conceptual view of a frequency selective surface 20 without varactor diodes (which varactor diodes or other variable capacitance devices can be used to realize an electrically steerable surface wave antenna - see Fig 2a).
  • the surface 20 of Fig. la comprises a plane of periodic metal patches 22 separated from a ground plane 26 by a dielectric layer 21 (not shown in Fig. lb, but see, for example, Figs. 2a and 2b).
  • An antenna (not shown) is typically mounted directly on the frequency selective surface 20. See, e.g., US Patent No. 7,068,234 issued June 27, 2006.
  • the thickness of the dielectric layer 26 can be less than 0.1 of a wavelength of operational frequency of the non-shown antenna.
  • This surface 20 supports a fundamental TM surface wave as shown in its dispersion diagram (frequency vs. propagation constant) of Fig. lb.
  • the surface impedance of any TM surface wave structure can be calculated by using:
  • Fig. la depicts the basic structure that supports a fundamental TM surface wave mode.
  • a dielectric substrate 21 (see Figs. 2a and 2b, not shown in Fig. la for ease of illustration) between the plane of metallic patches 22 and the ground plane 26 provides structural support and is also a parameter that determines the dispersion of the structure.
  • This structure can be made using printed circuit board technology, with a 2-D array of metallic patches 26 formed on one major surface of the printed circuit board and a metallic ground plane 26 formed on an opposing major surface of the printed circuits board, with the dielectric of the printed circuit board providing structural support.
  • the equivalent circuit model of the structure is superimposed over the physical elements of Fig.
  • a series inductance (LR) is due to current flow on the patch 22
  • a shunt capacitance (CR) is due to voltage potential from patch 22 to ground plane 26
  • a series capacitance (CL) is due to fringing fields between the gaps between the patches 22.
  • the dispersion diagram of Fig. lb shows that a fundamental TM forward wave mode (since the slope is positive) is supported.
  • the gap capacitance (between neighboring metal patches 22) can be electrically controlled by the use of varactor diodes 30.
  • the varactor diodes 30 are disposed in the gap between each patch 22 and are connected to neighboring patches 22 as shown in Fig. 2a.
  • the structure of Fig. la has been modified to include not only varactor diodes 30 but also a biasing network supplying biasing voltages Vi, V 2 , . .. V u .
  • Fig. 2b shows a cross-sectional view of the structure of Fig.
  • Fig. 3a depicts a model similar to that of Fig. 1 a, but showing the effect of introducing the bias network of Figs. 2a and 2b by a shunt inductance L L .
  • TM backward wave is supported when a series capacitance and a shunt inductance are present, the latter of which is contributed by the bias via 28.
  • the backward wave decreases the frequency/ impedance range of the surface wave structure since one can couple to only a forward wave or to a backward wave at a given frequency.
  • variable capacitors such as, for example, varactor diodes
  • Sievenpiper see, for example, US Patent No 7,068,234
  • the present invention provides a method of delaying the onset of a backward wave mode in a frequency selective surface having a two dimensional array of conductive patches and an RF ground plane, the two dimensional array of patches being interconnected by variable capacitors, the method including separating grounds associated with the variable capacitors from the RF ground plane and providing a separate conductive mesh structure as a control voltage ground for the variable capacitors.
  • the present invention provides a tunable impedance surface having : (a) a RF ground plane; (b) a plurality of elements disposed in an array a distance from the ground plane; (c) a capacitor arrangement for controllably varying capacitance between at least selected ones of adjacent elements in said array; and (d) a grounding mesh associated with said capacitor arrangement for providing a control voltage ground to capacitors in said capacitor arrangement, the grounding mesh being spaced from the RF ground plane by a dielectric.
  • the present invention provides a method of tuning a high impedance surface for reflecting a radio frequency signal comprising: arranging a plurality of generally spaced-apart conductive surfaces in an array disposed essentially parallel to and spaced from a conductive RF ground plane and varying the capacitance between at least selected ones of adjacent conductive surfaces in to thereby tune the impedance of said high impedance surface using control voltages , the control voltages being referenced to a control voltage ground supplied via a grounding mesh which is isolated from said RF ground plane by a layer of dielectric material.
  • the present invention provides a tunable impedance surface for reflecting a radio frequency beam, the tunable surface comprising: (a) a ground plane; (b) a plurality of elements disposed in an array a distance from the ground plane, the distance being less than a wavelength of the radio frequency beam; (c) a capacitor arrangement for controllably varying the impedance along said array; and (d) means for suppressing a formation of a backward wave by said tunable impedance surface.
  • the present invention provides a tunable impedance surface comprising: (a) a ground plane; (b) a plurality of discreet elements disposed in a two- dimensional array a distance from the ground plane; and (c) a plurality of capacitors coupling neighboring ones of the elements in said two dimensional array for controllably varying capacitive coupling between the neighboring ones of said elements in said two-dimensional array while at the same time suppressing a formation of a backward wave by the tunable impedance surface.
  • Fig . 1 a depicts a perspective view of a prior art frequency selective surface consisting of a plane of periodic metal patches or elements separated from a ground plane by a dielectric layer;
  • Fig. lb is a graph of frequency vs. propagation constant for the surface of Fig. la;
  • Fig . 2a is a top view of a prior art selective frequency surface with variable capacitors in the form of varactors, added to tunably control the impedance of the surface;
  • Fig . 2b is a side elevational view of the surface if Fig . 2a;
  • Fig. 3a depicts in a model similar to that of Fig. la, but showing the effect of introducing the bias network for controlling the varactors of Figs. 2a and 2b;
  • Fig. 3b is a graph of frequency vs. propagation constant for the surface of Fig. 3a:
  • Figs . 4a and 4b are plan and side elevational views of an embodiment of a frequency selective surface with variable capacitors to control surface impedance of the surface and a RF ground plane which is separated from a ground mesh used with the variable capacitors;
  • Fig. 5 is a graph of the numerical dispersion diagram of tunable surface wave impedance structure based on conventional biasing network as shown in Figs. 2a and 2b.
  • Fig. 6 is a graph of the numerical dispersion diagram of tunable surface wave impedance structure based on biasing network as shown in Figs. 4a and 4b.
  • Surface wave impedance goes beyond j250 Ohm and is extended out to j310 Ohm and higher.
  • Patch size and the dielectric layer between patch/RF ground are the same as used to generate Fig. 5.
  • This invention prevents a backward wave mode from occurring in a frequency selective surface while allowing for biasing of the varactor diodes used to control the dispersion and thus the surface impedance of the frequency selective surface at a fixed frequency.
  • This improved frequency selective surface is realized by separating a RF ground plane from the bias network ground.
  • Figs. 4a and 4b show that the RF ground plane 26 has been separated from an open mesh-like arrangement 25 of conductors connecting the bias grounding vias 24 to a common potential.
  • the ground plane 26 is located above the mesh-like arrangement 25 of conductors in Fig. 4b so that from a radio frequency perspective, the ground plane 26 serves as a RF ground for the conductive patches or elements 22 without undue interference from their associated conductive control vias 24, 28 which penetrate the ground plane 26 at penetrations 32.
  • the conductive control vias 24 are connected to the common potential (bias voltage ground 27) associated with the biasing voltages Vi, V 2 , ... V n , via the conductive mesh 25 while conductive vias 28 are connected to the biasing voltages Vi , V 2 , ... V n themselves. So the bias voltage ground 27 is separated from the RF ground 26.
  • the substrate 21 is preferably formed as a multi-layer substrate with, for example, three layers 21-1 , 21-2, and 21-3 of dielectric material (as such, for example, a multi-layer printed circuit board) .
  • the conductive patches or elements 22 are preferably formed by metal patches or elements disposed on layer 21-1 of a multi-layer printed circuit board.
  • the bias ground network or mesh 25 preferably takes the form of a meshed structure , in which the connection lines 25 are disposed diagonally, in plan view, with respect to the conductive patches or elements 22 as shown in Fig. 4a.
  • Relatively thin wires 25 are preferably used in the meshed bias network to provide a high impedance at RF frequencies of interest and are preferably printed between layers 21-2 and 21-3 of the multi-layer printed circuit board.
  • Penetration 32 is designed to be small enough to provide a suitable RF ground at the RF frequencies of interest but large enough to avoid contacting conductive vias 24 and 28 - in other words, the penetrations 32 should appear as essentially a short circuit at the RF frequencies of interest and as essentially an open circuit at the switching frequencies of the bias voltages Vi, V 2 , .. . V n .
  • the RF return current follows the path of least impedance which, in the present invention, is provided by the RF ground plane 26 which is preferably formed as a layer of a conductor, such a copper, with openings 32 formed therein.
  • the bias ground network 25 need not necessarily assume the meshed structure shown in Fig. 4a as other arrangements of the wires making up the meshed structure will likely prove to be satisfactory in presenting a suitably high impedance at the RF frequencies of interest so that the RF frequencies of interest will not treat the bias ground network 25 as an RF ground.
  • the bias ground network 25 begins to appear more like an RF ground, the less effective the present invention is in suppressing the backward wave. So ideally the bias ground network 25 should have as high an impedance as possible at the RF frequencies of interest consistent with the need to provide a bias ground 27 for the bias voltages Vi , V 2 , ... V n (which are at or near DC compared to the RF references of interest).
  • the bias ground network 25 is depicted as being located below the RF ground plane 26 so that it is further from the array of conductive patches or elements 22 than is the RF ground plane 26. This location is believed to be preferable compared to switching the positions RF ground plane 26 and the bias ground network 25; but if the bias ground network 25 has a suitably high impedance at the RF frequencies of interest, it may function suitably even if it is located closer to the array of conductive patches or elements 22 than is the RF ground plane 26. Testing and/or simulation should be able to verify whether or not this is correct.
  • wires which make up the meshed structure of the bias ground network 25 is used without implication as to shape or material. While the wires are preferably provided by electrically conductive strips disposed on a printed circuit board, they might alternatively individual wires, they might be round or flat, coiled or straight and they might be formed by conductive regions on or in a semiconductor substrate.
  • the patch plane comprises a 2-D array of conductive patches or elements 22 of a type A cell (Cell A) and a type B cell (Cell B) forms; a type A cell is connected to the bias ground network 25 while a type B cell is connected to a separate bias voltage network of voltages Vi, V 2 , ... V n . Only two cells are marked with dashed lines designating the cell types for ease of illustration in Fig. 4b, but they preferably repeat in a checkerboard fashion.
  • a cell includes its patch/element 22, its associated portion of the RF ground plane 26, and its associated control electrode or via (via 24 for a type A cell or via 28 for a type B cell). As can be seen from Figs. 4a and 4b, generally speaking the immediate neighbors of a type A cell are four type B cells and the immediate neighbors of a type B cell are four type A cells.
  • Dielectric layer 21-1 separates the conductive patches or elements 22 from the RF ground plane 26 and preferably provides structural support for surface 20.
  • size and dielectric nature of the dielectric layer 21-1 is a parameter that dictates the RF properties of the structure 20.
  • RF ground plane 26 provides a return path for the RF current; holes 32 are introduced in the RF ground plane 26 to allow the via 24 of Cell A type cells to connect to the meshed DC ground plane 25 and to allow the via 28 Cell B type cells to connect to the bias voltage network.
  • Dielectric layer 21-2 preferably acts a support structure for the bias ground network or mesh 25 and the bias voltage network.
  • An optional dielectric layer 21-3 can be added beneath dielectric layer 21-1 and mesh 25 to provide additional power and/or signal connections for vias 28.
  • Dielectric layers 21-1 , 21-2 and 21-3 can each consist of multiple dielectric substrates sandwiched together, if desired.
  • the mesh DC ground plane 25 preferably comprises diagonal cross connections which are made up of thin metal traces for presenting high impedance from a RF standpoint.
  • the via 24 of Cell A connects directly to the mesh DC ground plane 25.
  • the ground plane 25 can likely take other forms than a mesh like structure, but the mesh structure shown in Fig. 4a is believed to yield a structure which is easy to manufacture and which will present a high impedance to the surface at RF frequencies of interest.
  • the bias voltage network 25 connects to the conductive vias 28 of Cells B.
  • Fig. 5 is a graph of the numerical dispersion diagram of tunable surface wave impedance structure based on conventional biasing network as shown in Figs. 2a and 2b.
  • Fig. 5 shows that by changing the varactor diode's capacitance (a range of 0.1 pF to 0.2 pF is shown), the surface impedance can be varied at fixed frequencies.
  • the surface impedance range is limited to j250 Ohms after which a backward wave mode appears, which the source propagating wave cannot couple to. So after j250 Ohms, the mode appears to be cut-off due to the onset of backward wave propagation.
  • Fig. 6 is a graph of the numerical dispersion diagram of tunable surface wave impedance structure based on biasing network as shown in Figs. 4a and 4b.
  • Surface wave impedance goes beyond j250 ⁇ and is extended out to j31 ⁇ and higher.
  • Patch size and the dielectric layer between patches 22 and the RF ground 26 are the same as used to generate Fig. 5.
  • surface impedance tuning is also possible by changing the varactor diode's capacitance (a range of 0.1 pF to 0.3 pF is shown in Fig. 6) and the surface impedance range is increased; the surface impedance range is extended to j310 ⁇ and above.
  • MEMS capacitors and optically controlled varactors may be used in lieu of the voltage controlled capacitors (varactors) discussed above. If such optically controlled varactors need to be supplied with a bias voltage, then the conductive vias 24 and 28 discussed above are still needed, but a common bias voltage may be substituted for the bias voltages VI , V2, ... Vn discussed above as the optically controlled varactors would be controlled, in terms of varying their capacitance, by optical fibers preferably routed through penetrations in substrate 21 located, for example, directly under the varactors 30 shown in Fig. 4a.
  • variable capacitors are varactors.
  • a tunable impedance surface comprising:
  • the tunable impedance surface of claim 5 further including a substrate having at least first and second layers, said first layer being a first dielectric layer facing said ground plane on a first major surface thereof and facing said plurality of elements on a second major surface thereof and said second layer being a second dielectric layer and providing said dielectric material.
  • a method of tuning a high impedance surface for reflecting a radio frequency signal comprising:
  • bias voltages being referenced to a bias voltage ground supplied via a grounding mesh which is isolated from said RF ground plane by a layer of dielectric material.
  • step varying the capacitance between adjacent conductive surfaces in said array includes connecting variable capacitors between said at least selected ones of adjacent conductive surfaces.
  • the capacitance is varied between all adjacent elements.
  • the step of varying the capacitance between at least selected ones of adjacent conductive surfaces includes applying said bias voltages to selected ones of said conductive surfaces and applying said bias voltage ground to other ones of said conductive surfaces.
  • spacing of each conductive surface from the RF ground plane is less than a wavelength of a radio frequency signal impinging said surface, and preferably less than one tenth of a wavelength of a radio frequency signal impinging said surface.
  • a tunable impedance surface for reflecting a radio frequency beam comprising:
  • (d) means for suppressing a formation of a backward wave by said tunable impedance surface.
  • a tunable impedance surface comprising: (a) a ground plane; (b) a plurality of discreet elements disposed in a two-dimensional array a distance from the ground plane; and (c) a plurality of capacitors coupling neighboring ones of said elements in said two dimensional array for controllably varying capacitive coupling between said neighboring ones of said elements in said two-dimensional array while at the same time suppressing a formation of a backward wave by said tunable impedance surface.
  • a tunable impedance surface comprises a RF ground plane; a plurality of patches or elements disposed in an array a distance from the ground plane; a capacitor arrangement for controllably varying capacitance between at least selected ones of adjacent patches or elements in the array; and a grounding mesh associated with the capacitor arrangement for providing a control voltage ground to capacitors in the capacitor arrangement, the grounding mesh being spaced from the RF ground plane by dielectric material.
  • a conformal surface wave feed provides a transition from a coaxial line or other transmission line to surface wave transmission that can be used to launch a surface wave onto surface-wave media.
  • a Conformal Surface Wave Feed (CSWF) is believed to be unknown in the art.
  • the closest prior art may be a low-profile waveguide (LPWG) surface-wave coupler (see Fig. lb) that has been used to feed previous conformal Artificial Impedance Surface (AIS) antennas.
  • LPWG low-profile waveguide
  • AIS Artificial Impedance Surface
  • the present invention relates to CSWF that can be used to feed an AIS antenna or in other applications.
  • the CSWF provides a transition from a coaxial line or other transmission line to surface wave transmission that can be used to launch a surface wave onto surface-wave media of an AIS antenna, for example.
  • a wave is launched from a transmission line (typically a 50 ⁇ eoax-to- microstrip adaptor) into a tapered microstrip (MS) line that spreads the wave energy out into a broad phase front, and then into a surface- wave medium (SWM) .
  • the MS is tapered such that the insertion loss is preferably minimized from one end of the taper to the other.
  • the permittivity of the MS substrate is lower than the permittivity of the SWM substrate in order to match the wave speeds between the MS and the surface wave, thus minimizing insertion loss from the MS to the SWM.
  • the present invention provides a transmission line feed for a surface wave medium having a dielectric substrate with an array of electrically conductive patches formed thereon.
  • the transmission line feed includes: (a) a microstrip substrate, the microstrip substrate having a first permittivity which is lower than a second permittivity of the dielectric substrate of the surface wave medium, the microstrip substrate abutting against the dielectric substrate of the surface wave medium; (b) a tapered microstrip disposed on the microstrip substrate, the tapered microstrip tapering from a relatively narrow end to a relatively wide end, the relative wide end terminating where the microstrip substrate abuts against the surface wave substrate; and (c) an adapter for coupling a transmission line to the relatively narrow end of the tapered microstrip.
  • the present invention provides a method of feeding RF energy to a surface wave medium having a dielectric substrate with an array of electrically conductive patches formed thereon, the RF energy being fed to said surface via a coaxial transmission line feed.
  • the method includes: providing a microstrip substrate having a first permittivity which is lower than a second permittivity of the dielectric substrate of the surface wave medium; butting the microstrip substrate against the dielectric substrate of the surface wave medium; forming a tapered microstrip on the microstrip substrate, the tapered microstrip tapering from a relatively narrow end to a relatively wide end, the relative wide end terminating where the microstrip substrate abuts against the surface wave substrate; and providing an adapter for coupling the coaxial transmission line to the relatively narrow end of the tapered microstrip.
  • Fig. la depicts an embodiment of a CSWF; the CSWF 10 includes a microstrip whose width tapers from a relatively narrow end at a coax-to-MS adaptor (not shown in Fig. la) to relatively wider end at a surface- wave medium (SWM - not shown in Fig. la).
  • the CSWF launches a surface wave with a broad phase front into the surface-wave medium and at least a portion of which may be an AIS antenna (also not shown in Fig. la).
  • Fig. lb depicts a prior art device for launching surface waves which utilizes a low- profile waveguide coupler (LPWG) which protrudes from the antenna surface.
  • LPWG low- profile waveguide coupler
  • Fig 2a is a plan view very similar to Fig. la, but depicted in a larger scale and with indicia identifying certain elements and features thereof, and with the SWM and AIS depicted.
  • Fig. 2b is a section view taken through the CSWF of Fig. 2a.
  • Figs. 3a and 3b depict a simulation of the CSWF in plan (Fig. 3a) and side elevation (Fig. 3b) views.
  • the MS taper is fed by the coaxial adaptor on the right.
  • the wave propagates along the MS taper, spreading out into a broad phase front as the MS width increases.
  • a surface wave is launched into the surface-wave medium (SWM) with insertion loss ⁇ -25 dB if the wave speeds are closely matched.
  • SWM surface-wave medium
  • the surface wave is incident on the CSWF from the left.
  • the broad phase front of the surface wave is funneled through the MS taper to the narrow end of the MS taper where it is collected at a coaxial adaptor.
  • Fig. la depicts an embodiment of a CSWF 10.
  • This embodiment of CSWF 10 is integrated with a 24 GHz conformal AIS antenna 20 on a 25-mil substrate.
  • the CSWF 10 is a microstrip whose width tapers from 0.6 mm wide at a coax-to-MS adaptor (not shown in Fig. la, see element 16 in Fig. 2a) to 30 mm wide at the surface-wave medium in this embodiment.
  • the CSWF launches a surface wave with a broad phase front into a surface- wave medium (SWM) at least a portion of which may be an AIS antenna (See Fig. 2a for a representation of the AIS antenna 20) .
  • SWM surface- wave medium
  • the CSWF 10 includes a metallic microstrip 13 whose width tapers from a narrow end 11 at a transmission line 15 (typically a 50 ohm coaxial cable) to microstrip adaptor 16 (not shown in Fig. 1 a, but commercially available as model 292-04A-5 from Southwest microwave) to a wide end 12 at the surface-wave medium 22.
  • the CSWF 10 launches a surface wave with a broad phase front into the aforementioned AIS antenna.
  • the AIS 20 antenna is represented by a block in Fig. 2a.
  • the CSWF 10 need not be coupled to an AIS antenna as the CSWF 10 can be used to interface with SWMs used in devices other than AIS antennas.
  • An SWM is a "surface wave medium”. It is anything that supports surface electromagnetic waves. It is a type of artificial impedance surface (AIS). Not all AIS are SWMs as not all AIS support surface waves - on the contrary, some AIS are designed to inhibit surface waves. However, since an AIS A (an AIS antenna) works by purposefully leaking surface waves from it, it is an SWM by definition.
  • the CSWF 10 has a microstrip taper formed by a metallic layer 13 on a thin dielectric substrate 14 (typically having a thickness in the range of 25-50 mils) with relatively low relative permittivity ⁇ ⁇ ⁇ (preferably in a range of 2 - 4).
  • the relative permittivity of layer 14 is low compared to the AIS substrate's 22 relative permittivity e r2 which is typically around ⁇ 10.
  • the thickness of the substrates scale inversely to the frequency of operation. For example, 50 mil substrates 14, 22 are preferred for 8 to 14 GHz AIS, 25 mil substrates 14, 22 for 18 to 30 GHz AIS, and 1" thick substrates 14,22 for 100 to 500 MHz AIS.
  • the narrow end 11 of the taper preferably interfaces to a standard transmission line connector 30 such as the aforementioned microstrip to coaxial connector.
  • the width of the microstrip at the narrow end is chosen to match its impedance to the 50 ohm adaptor 16 according to well known technology.
  • the wider end 12 of the taper interfaces to a surface- wave medium formed by metallic patches 26 on substrate 22 that supports the desired surface wave.
  • a surface-wave impedance matching region 24 may be used if needed, which is formed by an array of metallic patches 26 on a dielectric substrate 22 whose permittivity is higher than the substrate 14 under the microstrip taper 13.
  • the CS WF 10 may be used in a number of applications , one currently preferred application is its use as a feed for an AIS antenna 20. See the application identified above for more information about AIS antennas.
  • the AIS antenna 20 typically has metallic patches similar to the metallic patches 26 and may be formed on a substrate integral with substrate 22.
  • the metallic patches of the AIS antenna 20 would typically start out with a uniform size corresponding to the smaller size patches 26 at the end of the surface wave impedance taper region 24 remote from the microstrip taper 13. Thereafter the sizes of patches in the AIS antenna 20 would be varied as discussed in the US patent application incorporated by reference to form transmission regions where the RF signal being applied via coaxial cable 15 (for example) is launched from the surface waves in the AIS antenna 20.
  • the size of the metal patches 26 varies along the direction of wave propagation denoted by arrow A with the patch size decreasing in size towards the AIS antenna 20.
  • An embodiment of disclosed CSWF 10 can be utilized, for example, to use surface waves to transmit high-rate data (>30 Mbps) or power (>1 W) in a two-dimensional surface- wave AIS antenna 20.
  • Figs, la, 2a and 2b show an exemplary embodiment of the CSWF 10 preferably used with a conformal AIS antenna 20 operating, in this embodiment, at 24 GHz.
  • the dimensions of the tapered microstrip 13 in this embodiment are 100 mm long by 30 mm maximum width at end 12 and tapering to a 0.6 mm minimum width at end 11.
  • the impedance taper in region 24 can be realized by decreasing size of patches 26, or patch period or both. Rules of thumb: 1) impedance increases with patch size for a given patch period; 2) impedance increases with patch period for a given fractional patch size (patch size/period); 3) impedance increases with substrate permittivity, and 4) impedance increases with substrate thickness. Any or all of these rules of thumb can be used to implement the impedance taper in region 24.
  • the disclosed feed will work without the impedance taper 24 (by abutting the tapered microstrip directly to an A1S antenna 20, for example). But the impedance taper 24 is highly desirable to meet specifications for most applications, especially high power applications, since the return loss tends to be unacceptably high without it.
  • the same material as substrate 22 is also preferably used as the substrate of the AIS antenna 20 and, indeed, substrate 22 is preferably shared by the AIS antenna 20 and the surface wave impedance taper 24 as an integral substrate 22.
  • Conformal artificial impedance surface antennas which are described in the US Patent Application which is incorporated by reference, modulate a surface wave and radiate its power into a designed radiation pattern.
  • any surface-wave research work the surface waves must be interfaced to external instruments that rely on conventional RF transmission line communication methods, such as coaxial cables and related connectors.
  • Artificial Impedance Surface antennas 20, whether or not they are conformal, need to be connected to transmitters and/or receivers and thus cables 15 are typically connected to such transmitters and/or receivers and those cables 15 need in turn to be connected to the AIS antenna 20.
  • the disclosed CSWF 10 facilitates that connection.
  • An important element of the CSWF 10 is its tapered microstrip 13, one end 11 of which interfaces to a conventional transmission line impedance (for example a 50 ⁇ coaxial cable 15), the other end 12 interfaces to a surface-wave medium which typically is in a surface wave impedance taper 24.
  • a very desirable element is the surface-wave impedance taper 24, which matches the wave impedance at the end of the microstrip taper 13 to the surface- wave impedance in the surface-wave medium (SWM) being fed by the CSFW 10, which may be an AIS antenna 20 as described above.
  • the SWM may comprise something other than an AIS antenna 20 since this invention is useful in launching surface waves from RF signals available in a conventional feed line, such as coaxial cable 15, into a SWM which can be used in a number of possible applications other than a AIS antenna 20.
  • the tapered microstrip 13 is designed to feed the surface wave in the SWM over a broad area, and the surface wave end 12 of the tapered microstrip 13 is therefore much wider than the coaxial end feed end 11.
  • the wave impedance changes as a function of its width according to well-known formulas governing microstrip design. The width is varied in such a way that the insertion loss between the wide and narrow ends is minimized.
  • the impedance along the taper preferably matches what is known as a "Klopfenstein" impedance taper. See Klopfenstein, R.W., "A Transmission Line of Improved Design", Proceedings of the IRE, pp. 31-35, January 1956. Other types of impedance tapers will work as well.
  • the taper shape seen in Fig. 2a is characteristic of the low-insertion loss taper formed by using a Klopfenstein impedance taper for the taper of the tapered microstrip 13.
  • the length of the tapered microstrip 13 affects the insertion loss; longer tapers lead to lower insertion loss. In practice, a length equal to approximately two wavelengths of the transmitted wave (the RF signal in coaxial cable 15) is sufficient.
  • Wave speeds should be matched between the surface wave and wave in the tapered microstrip 13 at the boundary between the impedance taper 24 and the tapered microstrip 13 in order to minimize insertion loss between the two regions.
  • the substrate 14 permittivity ⁇ for the tapered microstrip 13 is lower than the substrate 22 permittivity ⁇ 2 in the surface- wave region.
  • the wave speed in the tapered microstrip 13 is approximately cl en 1 ' 2 over a wide bandwidth, where c is the speed of light and ⁇ is the relative permittivity of substrate 14.
  • Substrate thickness and tapered microstrip 13 width affect the wave speed in a well-known, but involved way not presented here. (See: I. J. Bahl and D.
  • the surf ace- wave speed in the surface wave taper region 24 is determined by the wave's frequency, the substrate permittivity ⁇ 2 and its thickness, and the size and shape of the metallic patches 26 on the substrate 22.
  • the surface-wave speed approaches a lower limit of c/ eft 1 ' 2 as the frequency and/or the substrate thickness increase (see C. Simovskii et al, "High-impedance surfaces having stable resonance with respect to polarization and incidence angle", IEEE Trans. Antennas Prop., vol.
  • the wave speed of the microstrip - uided waves at the end of the tapered microstrip 13 is lower than desired for that application.
  • the surface-wave speed is caused to increase as the wave moves away from the tapered microstrip 13 by varying the sizes of the metallic patches in the surface-wave impedance taper region 24.
  • the shapes are varied in such a way that the surface-wave impedance is varied in a controlled fashion that minimizes insertion loss from one end of the surface-wave impedance taper region 24. In practice, this is readily accomplished with a Klopfenstein impedance taper in terms of vaiying the sizes of the patches 26 in surface- wave impedance taper region 24.
  • An impedance taper such as the Klopfenstein taper, is a mathematical entity. It describes the impedance as a function of distance along a transmission line that matches the impedances between two transmission lines with different impedances. (The SWM can be considered to be a transmission line for surface waves.) For the taper in the microstrip line 16, this is realized with a strip that gradually spreads out. For the surface-wave impedance taper in region 24, the taper is a one-dimensional change in surface- wave impedance with distance. So the patches only have to vary in size along the direction of the propagation as depicted by the arrow of region 24 in Fi . 2a.
  • the mean surface-wave impedance is relatively low - it is optimum at about 250 to 300 ohms/sq.
  • the impedance necessary to match wave speeds to an SWM at the end of the tapered microstrip 13 is much higher, approximately 500 to 800 ohms/sq. So, in this case, and pretty much for all AIS antennas 20, there has to be a transition region 24 between the AIS antenna's operational surface and the high impedance region where the microstrip 13 terminates and couples to the AIS antenna 20 if a good match is desired. In such a case, an impedance taper in region 24 is essential.
  • the AIS antenna 20 is just a SWM (like applications with power transfer or data transmission via surface waves), it is admissible to use an AIS (or SWM to be more general) with a high impedance everywhere. Then an impedance taper is not necessary. However, even in these applications, it can be desirable to taper the impedance in region 24 because for example, a lower impedance SWM is easier to make because it uses less metal or is thinner or uses a cheaper dielectric substrate with lower permittivity. These considerations are important when the SWM is very large as for a large scale SWM network. See, for example, US Patent No. 7,307,589 to Daniel Gregoire et al. entitled "Large-Scale Adaptive Surface Sensor Arrays".
  • Figs. 3a and 3b depict the results of a simulation done of the CSWF 10 of Figs. 2a and 2b.
  • the tapered microstrip 13 is fed via the coaxial adaptor 16 on the right.
  • the wave propagates along the tapered microstrip 13, spreading out into a broad phase front as the tapered microstrip 13 width increases.
  • a surface wave is launched into the surface- wave medium (SWM) region 24 with insertion loss ⁇ -25 dB if the wave speeds are closely matched.
  • SWM surface- wave medium
  • the surface wave is incident on the CSWF 10 from the left.
  • the broad phase front of the surface wave is tunneled through the tapered microstrip 13 to the narrow end 11 of the tapered microstrip 13 where it is collected at the coaxial adaptor for downstream RF to DC conversion.
  • Two possible power collection applications are (1) Broadcasting wireless power to a distributed network and (2) broadcasting wireless power from one place to another such as between a satellite and an earth station.
  • a surface-wave power and communication network distributed across a lm 2 SWM (again, see US Patent No.
  • the AIS A 20 is used as a receiving antenna in wireless power transfer. In that case, microwave power is beamed from one place to another, e.g between a satellite and the earth station.
  • the receiving antenna is an AIS A which collects the microwaves on its surface and focuses it to a single point where it is collected by the CSWF 10 and then converted to DC downstream.
  • the same system can work in reverse where the AISA 20 is the power transmitting antenna.
  • a broad surface- wave phase front is incident on the tapered microstrip 13, which then funnels the energy in the surface wave phase front down to the coaxial adaptor 16 where it can then be transmitted to an RF-to-DC converter to power devices such as CPUs, varactors, LEDs, etc.
  • Figs. 3a and 3b show the wave
  • the insertion loss for the entire device is less than -25 dB when the wave speeds are matched between the tapered microstrip 13 region and the surface wave region.
  • the overall insertion loss tends to be limited by the coax-to-microstrip adaptor 16.
  • the grey level change of the fields in Fig. 3a indicates the changing power density along the length of the taper, with a maximum power density occurring at the adaptor 16.
  • the wave energy is confined to the metallic shape of the microstrip 13. If the RF energy originates from some device (such as a transmitter) coupled to the RF cable 15, the wave energy spreads out as the width of the tapered microstrip 13 increases along the length of the taper, where it then transitions into a surface wave with a broad phase front. If the RF energy originates as surface waves (such as from an AIS antenna 20), then the wave energy concentrates as the width of the tapered microstrip 13 decreases along the length of the taper towards the adapter 16, where it then transitions into a the RF cable 15.
  • some device such as a transmitter
  • the wave energy spreads out as the width of the tapered microstrip 13 increases along the length of the taper, where it then transitions into a surface wave with a broad phase front. If the RF energy originates as surface waves (such as from an AIS antenna 20), then the wave energy concentrates as the width of the tapered microstrip 13 decreases along the length of the taper towards the adapter 16, where it
  • a transmission line feed for a surface wave medium having a dielectric substrate with an array of electrically conductive patches formed thereon comprising: a. a microstrip substrate, the microstrip substrate having a first permittivity which is lower than a second permittivity of the dielectric substrate of the surface wave medium, the microstrip substrate abutting against the dielectric substrate of the surface wave medium;
  • a tapered microstrip disposed on the microstrip substrate, the tapered microstrip tapering from a relatively narrow end to a relatively wide end, the relative wide end terminating where the microstrip substrate abuts against the surface wave substrate;
  • the transmission line feed of claim 1 coupled to an A1S antenna, the A1S antenna comprising at least a portion of said surface wave medium.
  • the transmission line feed of claim 1 coupled to an A1S antenna, the A1S antenna having a substrate which abuts against the dielectric substrate of said surface wave medium.
  • transmi ssion line feed of claim 8 wherein the metallic patches mimic a lopfenstein impedance taper in a region immediately adjacent the relative wider end of the tapered microstrip.
  • microstrip substrate having a first permittivity which is lower than a second permittivity of the dielectric substrate of the surface wave medium
  • microstrip substrate having a first permittivity which is lower than a second permittivity of the dielectric substrate of the AIS antenna
  • the tapered microstrip tapering from a relatively narrow end to a relatively wide end, the relative wide end terminating where the microstrip substrate abuts against the AIS antenna;
  • the AIS antenna has surface wave impedance taper region disposed on the dielectric substrate of the AIS antenna, the surface wave impedance taper region being disposed next to the relatively wide end of the tapered microstrip on the microstrip substrate.
  • a transmission line feed for a surface wave medium having a dielectric substrate with an array of electrically conductive patches formed thereon includes a microstrip substrate, the microstrip substrate having a first permittivity which is lower than a second permittivity of the dielectric substrate of the surface wave medium, the microstrip substrate abutting against the dielectric substrate of the surface wave medium; a tapered microstrip disposed on the microstrip substrate, the tapered microstrip tapering from a relatively narrow end to a relatively wide end, the relative wide end terminating where the microstrip substrate abuts against the surface wave substrate; and an adapter for coupling a transmission line to the relatively narrow end of the tapered microstrip.
  • the invention is a directional surface-wave coupler (SWC) that launches transverse-electric (TE) mode electromagnetic surface waves (SW) into a surface-wave medium (SWM).
  • SWC directional surface-wave coupler
  • An electromagnetic surface wave is any wave confined to propagate along the two-dimensional interface between two medium, e.g. the interface between dielectric and air is the simplest case.
  • An ideal surface wave's fields propagate along the surface with sinusoidal variation and decay exponentially away form the surface.
  • the fields have the form e' kx e 'az , (See fig. 1 ) where k is the propagation wavenumber in the propagation direction in the SWM, and a is the decay constant away form the SWM surface exponentially away form the surface SWs can be used to communicate with and/or power devices distributed on the SWM surface.
  • a TE SW is a surface wave whose electric field is parallel to the SWM surface and perpendicular to the direction of propagation.
  • An SWM is any medium that supports SW propagation. It can be composed of the interface between two dielectric materials, such as kapton and air, or it can be created out of complex engineered textures applied to a surface (an example is shown in Fig. 2).
  • the SWC is a device designed to transmit and receive SWs to and from the SWM without coupling to background radiation propagating in free space.
  • An ideal SWC when transmitting, will couple all of its energy to the SW in the SWM without radiating any power away form the SWM.
  • an ideal SWC when receiving, will receive power only form SWs in the SWM and will not couple to any external free-space radiation.
  • the SWC of this invention is formed by applying metal traces to a dielectric material.
  • a coax connector at one end allows connection to external devices.
  • the metalized patterns are designed to couple to the TE SWs in the SWM.
  • SWCPT surface wave communication and power technology
  • SWCPT can be used to communicate with an array of distributed sensors designed to monitor the structural health of a vehicle; or it could be used to allow secure reconfigurable communications or instant integration of removable devices.
  • an automobile driver could have his external electronic devices instantly integrated into the auto's control system.
  • the invention is a surface-wave coupler (SWC) that launches transverse-electric (TE) mode electromagnetic surface waves (SW) into a surface-wave medium (SWM).
  • SWC surface-wave coupler
  • An electromagnetic surface wave is any wave confined to propagate along the two-dimensional interface between two medium, e.g. the interface between dielectric and air is the simplest case.
  • An ideal surface wave's fields propagate along the surface with sinusoidal variation and decay exponentially away form the surface.
  • the fields have the form e' kx e 'az , (See fig. 1 ) where k is the propagation wavenumber in the propagation direction in the SWM, and a is the decay constant away form the SWM surface exponentially away form the surface SWs can be used to communicate with and/or power devices devices distributed on the SWM surface.
  • a TE SW is a surface wave whose electric field is parallel to the SWM surface and perpendicular to the direction of propagation.
  • An SWM is any medium that supports SW propagation. It can be composed of the interface between two dielectric materials, such as kapton and air, or it can be created out of complex engineered textures applied to a surface (an example is shown in Fig. 2).
  • the SWC is a device designed to transmit and receive SWs to and from the SWM without coupling to background radiation propagating in free space.
  • An ideal SWC when transmitting, will couple all of its energy to the SW in the SWM without radiating any power away form the SWM.
  • an ideal SWC when receiving, will receive power only form SWs in the SWM and will not couple to any external free-space radiation.
  • the SWC of this invention is formed by applying metal traces to a dielectric material.
  • a coax connector at one end allows connection to external devices.
  • the metalized patterns are designed to couple to the TE SWs in the SWM.
  • SWM supports propagation of the surface wave
  • Substrate can be made of any non conducting dielectric material. In practice, it's best to use dielectrics with low loss tangents. Typical materials used are conventional circuit board material. The material used in the pictured device is 0.050" thick Rogers 3010 duroid.
  • Coaxial coupler used to couple RF signals between the SWC and a coaxial cable. Standard off the solder-on
  • couplers can be used.
  • Groundplane a metallic coating on the bottom side of the SWC. It extends as far as via on the parasitic driver. It also helps to reflect energy in the forward direction away from the coaxial feed.
  • Striplirte feed couples RF signals between the coaxial coupler and the driver eiements. It's width is designed in conjunction with the substrate properties to match impedance with the coaxial coupler (usually 50 ⁇ )
  • Primary driver is fed RF signals by the strip line.
  • the electric fields in the primary driver are oriented so that they strongly couple to the TE SW.
  • Parasitic driver the parasitic driver forms the other half of the driver network, and works in conjunction with the primary driver to launch the TE SW.
  • Reflection plane creates an interface that prevents backward-launched TE SW from continuing in that direction; they are reflected from the edge of the ground plane and combine with the forward launched TE SW.
  • Director creates directional gain for the TE SW. directional gain is desired in some applications, particularly where power transfer is concerned. There may be one or more directors, depending on how much directional gain is desired.
  • the net effect of the TE SWC design is to convert the RF fields from the coaxial coupler to be compatible with the SW on the SWM.
  • the electric field is launched parallel to the SWM surface and perpendicular to the direction of propagation, which is identical to the SW's field configuration (see figure 5) .
  • the SWC When the SWC is close to the SWM, it efficiently couples to SWs on the SWM. Very little of the radiation is coupled to free-space radiation.
  • FIG. 4 The SWC return loss for the SWC pictured in Fig. 3.
  • the SWC has efficient coupling (return loss ⁇ -10 db) fin the frequency range form 4.5 to 7.2 GHz.

Abstract

L'invention porte sur une antenne de surface à impédance artificielle, orientable dans des angles phi et thêta comprenant un substrat de diélectrique, une pluralité de bandes métalliques sur une première surface du substrat de diélectrique, les bandes métalliques étant espacées sur une longueur du substrat de diélectrique et chaque bande métallique s'étendant sur la largeur du substrat de diélectrique, et des alimentations d'onde de surface espacées sur la largeur du substrat de diélectrique près d'un bord du substrat de diélectrique, le substrat de diélectrique étant sensiblement dans un plan X-Y défini par un axe X et un axe Y, l'angle phi étant un angle dans le plan X-Y par rapport à l'axe X, et l'angle thêta étant un angle par rapport à un axe Z orthogonal au plan X-Y.
PCT/US2013/050412 2013-07-03 2013-07-13 Antenne de surface à impédance artificielle orientable électroniquement WO2015002658A1 (fr)

Priority Applications (2)

Application Number Priority Date Filing Date Title
EP13888596.7A EP3017504B1 (fr) 2013-07-03 2013-07-13 Antenne à surface d'impédance artificielle dirigeable électroniquement
CN201380077921.1A CN105379011B (zh) 2013-07-03 2013-07-13 电子可控的人工阻抗表面天线

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US13/934,553 US9466887B2 (en) 2010-11-03 2013-07-03 Low cost, 2D, electronically-steerable, artificial-impedance-surface antenna
US13/934,553 2013-07-03

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