WO2014182000A1 - Égaliseur avant de flux robuste d'erreur de coefficient - Google Patents

Égaliseur avant de flux robuste d'erreur de coefficient Download PDF

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WO2014182000A1
WO2014182000A1 PCT/KR2014/003841 KR2014003841W WO2014182000A1 WO 2014182000 A1 WO2014182000 A1 WO 2014182000A1 KR 2014003841 W KR2014003841 W KR 2014003841W WO 2014182000 A1 WO2014182000 A1 WO 2014182000A1
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ffe
equation
equalizer
data
coefficient
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PCT/KR2014/003841
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English (en)
Korean (ko)
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김병섭
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포항공과대학교 산학협력단
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Priority to US14/889,814 priority Critical patent/US9503293B2/en
Priority claimed from KR20140052097A external-priority patent/KR20140132277A/ko
Publication of WO2014182000A1 publication Critical patent/WO2014182000A1/fr

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
    • H04L25/03006—Arrangements for removing intersymbol interference
    • H04L25/03012—Arrangements for removing intersymbol interference operating in the time domain
    • H04L25/03114—Arrangements for removing intersymbol interference operating in the time domain non-adaptive, i.e. not adjustable, manually adjustable, or adjustable only during the reception of special signals
    • H04L25/03133—Arrangements for removing intersymbol interference operating in the time domain non-adaptive, i.e. not adjustable, manually adjustable, or adjustable only during the reception of special signals with a non-recursive structure
    • H04L25/0314—Arrangements for removing intersymbol interference operating in the time domain non-adaptive, i.e. not adjustable, manually adjustable, or adjustable only during the reception of special signals with a non-recursive structure using fractionally spaced delay lines or combinations of fractionally integrally spaced taps

Definitions

  • the present invention relates to a coefficient error robust feedforward equalizer, and more particularly, to a feedband equalizer for baseband wired communication that prevents the influence of coefficient errors.
  • Feed Forward Equalizer is a channel compensation scheme widely used in baseband high-speed interconnects.
  • CMOS Complementary metal-oxide semiconductor
  • FFE transmitters can achieve high data rates with limited bandwidth.
  • CMOS technology advances to nanoscale, due to process distortions, random variables, temperature fluctuations, and aging, Since the variation of the device is increased and a large coefficitent error occurs due to the generation of nano device dispersion, there is a problem of degradation of performance and communication interference of the feedforward equalizer circuit due to the counting error.
  • the device needs to secure robustness against counting errors in a situation where the device is constantly decreasing and a high data rate is required.
  • the present invention has been proposed to solve the above problems, and provides a feedforward equalizer that more robustly responds to coefficient errors while operating like a feedforward equalizer of a general structure in a steady state. There is a purpose.
  • the count error robust feedforward equalizer includes a receiver 130 for receiving input data x according to an integer time index n, and N number of serially connected to the receiver 130. Outputting a data transition value (b) based on the change of the delay unit (D), the first operator (110) summing the tap signals output from the N delay units (D), and the input data (x), respectively. It includes a data change detection filter 120.
  • the data change detection filter 120 may be disposed between the receiving end and the delay unit D of the uppermost end adjacent to the receiving end, between any two delay units D adjacent to each other among the N delay units D, and The first operator 110 and the first operator 110 is disposed between any one of the last delay unit (D) adjacently connected.
  • the data change detection filter 120 includes one delay unit 122 and a second operator 121 connected to the one delay unit 122, and the second operator 121 includes the one delay.
  • the data transition value b [nm] based on the previous value x [n-m + 1] input to the unit 122 and the current value x [nm] output from the one delay unit 122. ]) Is preferable.
  • the data change detection filter 120 preferably calculates the data transition value based on the following equation.
  • n-m 0.5x [n-m + 1] -0.5x [n-m], where n and m are integers and n> m
  • the data transition value b [n-m] is preferably calculated through a combination of logic circuits of a plurality of consecutive digital bit values in the data stream.
  • the data change detection filter 120 is preferably a high pass filter (HPF).
  • HPF high pass filter
  • the tap signal includes a feed forward equalizer coefficient (a), and the feed forward equalizer coefficient (a) is adjustable by a user.
  • the count error robust feedforward equalizer according to the present invention operates in the same way as a general feed forward equalizer (FFE) when there is no count error, and it is robust against count errors at a small additional cost including a simple logic circuit. It has the effect of improving robustness.
  • FFE feed forward equalizer
  • the high-pass transition detection filter installed in the coefficient error robust feedforward equalizer according to the present invention reduces the perturbation of the signal caused by the coefficient error, and improves the robustness of the interconnector. It is effective to improve the eye diagram sensitivity which shows robustness by 7 to 17 times.
  • the coefficient error robust feedforward equalizer according to the present invention can be easily applied to a high-speed interconnect.
  • FIG. 1 is a conceptual diagram of a high speed interconnect system including a feed forward equalizer according to the prior art.
  • FIG 2 is an exemplary view of a feed forward equalizer according to the prior art.
  • 3 is a conceptual diagram of a three-tap feedforward equalizer according to the prior art.
  • 4A is a block diagram of a high speed interconnect system including a coefficient error robust feedforward equalizer according to an embodiment of the present invention.
  • 4B is a block diagram specifically illustrating a coefficient error robust feedforward equalizer according to an embodiment of the present invention.
  • FIG. 5 is an illustration of a three-tap equalizer of the coefficient error robust feedforward equalizer according to the present invention.
  • FIG. 7 is a graph of the coefficient error effect simulation result of the 2-tap equalizer of the coefficient error robust feedforward equalizer according to the present invention.
  • FIG. 8 is a graph comparing spectra of a feed forward equalizer according to the related art and a coefficient error robust feed forward equalizer according to the present invention.
  • FIG. 9 is a graph showing the sensitivity of the conventional 2-tap feedforward equalizer for the primary RC channel and the coefficient error robust feedforward equalizer according to the present invention.
  • FIG. 10 is a diagram illustrating an embodiment of a lost transmission line channel model.
  • FIG. 11 is a graph showing eye sensitivity and channel loss at the Nyquist frequency of a prior art 5-tap feedforward equalizer and a feedforward equalizer according to the present invention in a 40 cm PCB channel.
  • FIG. 12 is a graph illustrating an eye diagram of data rates according to FIG. 11.
  • FIG. 13 is a graph showing the sensitivity of a three-tap feedforward equalizer and a feedforward equalizer according to the present invention in a 3.5 cm silicon interposer package.
  • FIG. 14 is a graph illustrating an eye diagram of data rates according to FIG. 13.
  • a typical feedforward equalizer according to the prior art for high speed interconnects comprises a channel, a feedforward equalization transmitter (FFE Tx) and a 1-bit quantization receiver (Rx) as shown in FIG. It includes cables, backplanes, PCBs, packages, and on-chip wires. Channels range from a few centimeters to tens of meters.
  • the channel is characterized as a low-pass filter (LPF) and inter-symbol interference (ISI) Is generated.
  • LPF low-pass filter
  • ISI inter-symbol interference
  • the channel loss is large at the Nyquist frequency, the inter-signal interference seriously interferes with the communication.
  • the loss is small when the channel loss is 0 to 10 dB, the loss is large when the 10 to 20 dB, and the loss is very large when the 20 to 30 dB. Therefore, channel loss is rarely over 30dB, and it is known as an extremely difficult channel.
  • FFE feedforward equalization
  • the feedforward equalization transmitter (FFE Tx) has been used as a means for compensating for channel loss to secure a data rate.
  • a feedforward equalizer having an appropriate tap coefficient w value has a 2-level signal y [n] arriving at the receiver Rx. It works as a high pass filter (HPF) that compensates for channel loss to make a pulse amplitude modulation (PAM2) signal.
  • y [n] corresponds to the value where x [nm] arrives at the receiver, where x [n] is the transmitted data sequence whose signal level is 1 (bit '1') or -1 (bit '0') Have In this case, m means a delayed time until x [n] passes through the channel and arrives at the receiver.
  • the 1-bit quantizer of the receiver samples y [n] at periodic T intervals and then zeros Determines the value of, where y [n] is greater than 0 Is a value of 1, The value of is -1.
  • FFE feedforward equalizer
  • ISI inter-signal interference
  • 2 (a) is an exemplary diagram of the simplest and most common two-tap feedforward equalizer (2-tap FFE) structure.
  • the wire is modeled as a primary RC circuit with a time constant ⁇ .
  • the continuous-time pulse response of the system h (t) is h (t) is u (t) (1-e -t / ⁇ ) -u (tT) (1-e- (tT) / ⁇ ), where u (t) means unit step function.
  • h (t) is u (t) (1-e -t / ⁇ ) -u (tT) (1-e- (tT) / ⁇ ), where u (t) means unit step function.
  • FIG. 2 (b) the continuous time pulse response of this system is shown in a circular and without a pulse response without FFE.
  • Equation 1 The discrete time pulse response h [n] of this channel is equal to h (nT), and can be expressed by Equation 1 below.
  • n integer time index
  • the impulse response h (t) of the system without the feed forward equalizer (FFE) is the feed forward equalizer (FFE). It can be seen that the inter-signal interference tail (ISI tail) is longer than the impulse response of the system.
  • Equation 1 when n ⁇ 1, h [n] is exponentially reduced to h [n-1] e -T / ⁇ , and thus this channel is optimized as shown in Equation 2 below.
  • ISI Inter-signal interference
  • the feedforward equalizer shows the transmitted pulse in blue in FIG. 2 (b) to remove the inter-signal interference tail (ISI tail). Deform undershoot like a long dotted line.
  • the discrete time pulse response g [n] of the system equipped with the feed forward equalizer (FFE) is not subjected to signal-to-signal interference, as shown by the diamond and solid lines in FIG. As shown in (d), the eye diagram is completely opened.
  • the eye diagram is closed, which indicates that the feed forward equalizer (FFE) can improve the eye diagram. have.
  • LSE Least-squares method
  • C-FFE feed forward equalizer
  • w lse and h are vectors and truncated vector representations of w lse [n] and h [n], respectively.
  • h k and ⁇ -m are column vectors with h [n + kl] and ⁇ [nm-1] as the nth element, respectively.
  • Equation 6 the maximum value
  • of the signal transmitted from the feed forward equalizer (FFE) transmitter illustrated in FIG. 1 has a constraint as shown in Equation 6 below. Therefore, w lse [n] was normalized as shown in [Equation 7] to satisfy Equation [Equation 5].
  • Equation 6 w lse [n] can be normalized as shown in Equation 7 below.
  • a conventional feedforward equalizer (FFE) is referred to as C-FFE, and a robust-FFE to prevent the coefficient error effect of the present invention is referred to as B-FFE. Let's do it.
  • FIG. 3 illustrates a three-tap feedforward equalizer transmitter structure according to the prior art, and a flip-flop or latch shown in FIG. 3 (b) is a delay unit of FIG. 1. D).
  • the output of the latch Represents x [ni], and its value is determined according to the truth table of FIG.
  • V Tx between Txp and Txn is Thevenin-equivalent differential voltage corresponding to v [n] of FIG. 1.
  • the transmitter of the feedforward equalizer is vulnerable to coefficient errors that occur mainly due to variations in nanodevices.
  • shown in FIG. 3 (c) are sensitive to dispersion generated by the nanoscale technique, and such dispersion is not easy to control in the sub-micrometer technique. There is a problem that does not secure the robustness (roburstness).
  • the influence of the scattering of the nanodevices may be modeled as a constant random variable ⁇ w added to the coefficient w .
  • ⁇ w the coefficient error that is obtained is suitable for random number modeling such as ⁇ w.
  • FIGS. 4A and 4B are block diagram of a fast interconnect system including a count error robust feedforward equalizer 100 (B-FFE) according to an embodiment of the present invention
  • FIG. 4B is a count error according to an embodiment of the present invention.
  • a fast interconnect system including a coefficient error robust feedforward equalizer 100 (B-FFE) includes a feedforward equalizer 100 (B-FFE).
  • the receiver 300 receiving the output of the feed forward equalizer 100 and B-FFE, and the channel 200 for communication between the feed forward equalizer 100 and B-FFE and the receiver 300. It is configured to include.
  • the count error robust feedforward equalizer 100 includes a receiving end 130, N delay units D, and a first operator 110 as shown in FIG. 4B. And a data change detection filter 120.
  • the receiving end 130 is configured to receive the input data x according to the integer time index n, where x [n] is a transmitted data sequence and the signal level is 1 (bit '1') or -1 (bit ' 0 ') value.
  • m means a delayed time until x [n] passes through the channel and arrives at the receiver.
  • the first operator 110 performs a function of summing tap signals respectively output from the N delay units (D).
  • the data change detection filter 120 performs a function of outputting a data transition value b based on the change of the input data x.
  • the data change detection filter 120 will be described.
  • the count error robust feedforward equalizer 100 (B-FFE) according to an embodiment will be described.
  • the data change detection filter 120 in the coefficient error robust feedforward equalizer 100 (B-FFE) is one of the N delay units D as shown in FIG. 4B. It may be arranged between two neighboring delay units (D), and a delay unit (D) or the first operator (110) and the first stage (110) connected to the receiving end (130) and the receiving end (130) adjacently. It is also possible to be disposed between the first operation unit 110 and the delay unit (D) of the last end connected to the neighbor.
  • the data change detection filter 120 is specifically composed of one delay unit 122 and a second operator 121 connected to the one delay unit 122, and in particular, the second operator 121 is the one.
  • the data transition value b is based on a previous value (x [n-m + 1]) input to the delay unit 122 of and a current value (x [nm]) output from the one delay unit 122. [nm]).
  • x [n] is input data
  • b [n-m] is a value indicating a data transition of x [n].
  • C [n] is a high pass filter (HPF) that detects changes in data.
  • C [m-1] is 0.5
  • c [m] is -0.5
  • c [n] is any integer other than m-1 or m.
  • n] is zero. That is, as shown in Equation 9 below, c [n] is defined as a value for converting x [n-m + 1] to b [n-m].
  • b [nm] may have a value of -1, 0, 1, and when b [nm] is -1, it means that x [n] is changed from 1 to -1, and b [nm] is 1 In the case that x [n] is changed from -1 to 1, and when b [nm] is 0, it means that there is no change in x [n]. That is, b [n-m] includes information in which x [n] changes.
  • the data transition value calculated and output by Equation 9 becomes -1 when the data input to the data change detection filter 120 changes from 1 to -1, and when the input data changes from -1 to 1 It is 1, and if there is no change in the input data, it is 0.
  • the coefficient error robust feedforward equalizer may calculate the data transition value b [nm] through a logic circuit combination of a plurality of consecutive digital bit values in the data stream. .
  • Equation 9 of the 3-tap coefficient error robust feedforward equalizer (B-FFE) according to an embodiment of the present invention, and includes two digital bits D pn ⁇ . i-1 , D nn-i-1 ) is a table showing b [ni-1] values. At this time, two digital bits D pn-i-1 and D nn-i-1 are configured as shown in FIG. 5 (b) using an AND gate.
  • B-FFE 3-tap coefficient error robust feedforward equalizer
  • FIG. 5 (c) is a configuration diagram of a circuit for calculating addition and coefficient products of a 3-tap coefficient error robust feedforward equalizer (B-FFE) according to an embodiment of the present invention.
  • Equalizer (C-FFE) A circuit of the same type as a CML circuit. Since the cost of the AND gate in nanoscale CMOS technology is very low, there is an economic effect in constructing the B-FFE according to the present invention.
  • tap signals are output from N delay units D, respectively. It includes a feed forward equalizer coefficient (a) and a constant random coefficient error ⁇ a, in particular the feed forward equalizer coefficient (a) is preferably configured to be adjusted by the user, which is a steady state
  • the feed forward equalizer coefficient (a) is preferably configured to be adjusted by the user, which is a steady state.
  • the feed forward equalizer coefficient a of the feed forward equalizer 100 and B-FFE according to an embodiment of the present invention is a conventional feed forward equalizer (a). Since it is possible to map the feed forward equalizer coefficient ( w ) of the C-FFE), the feed forward equalizer 100 (B-FFE) according to an embodiment of the present invention is a conventional feed forward equalizer (C-FFE). In addition to the effect of being strong against the coefficient error, it is possible to derive the effect that the functions of all conventional feedforward equalizers (C-FFE) can be implemented as they are.
  • the scattering effect of the device in the coefficient error robust feedforward equalizer (B-FFE) according to the present invention is that the random random coefficient error ⁇ a is equal to the feedforward equalizer coefficient a .
  • feed forward equalizer is an LTI system, it is possible to analyze the effects of counting errors from the perburbation of the pulse response.
  • Equation 13 By substituting 0 into the nominal coefficient of the feedforward equalizer FFE, ⁇ y [n], which is a perturbation of y [n], can be obtained. That is, by replacing w [n] shown in FIG. 1 with 0, Equation 13 regarding the feed forward equalizer C-FFE according to the prior art can be derived.
  • H ⁇ w [m] [n] which is defined as ⁇ w [m] h [nm] in Equation 13, affects the coefficient error ⁇ w [m] on ⁇ y [n] when the input data x [n] is transmitted. Means.
  • Equation 13 ⁇ y [n] is the sum of all influences of the coefficient error ⁇ w [m], so that the perturbation of the feedforward equalizer (FFE) system is equal to all h ⁇ w as in the general pulse response. [m] can be expressed using [n]. Therefore, h ⁇ w [m] [n] can be defined as a coefficient error pulse due to ⁇ w [m].
  • the pulse response change ⁇ y [n] of the coefficient error robust feedforward equalizer B-FFE according to the present invention can be derived as shown in Equation 14 below.
  • Equation 14 h ⁇ a [m] [n] is a coefficient error pulse response due to ⁇ a [m].
  • FIG. 6 is a diagram illustrating a result of simulating the influence of ⁇ w [0] in a 2-tap feedforward equalizer (2-tap C-FFE) according to the related art mounted to compensate for loss of a primary RC channel.
  • 2-tap C-FFE 2-tap feedforward equalizer
  • the pulse and response to which the counting error ⁇ w [0] is transmitted are slightly changed from the normal state as shown in 2 or 4 of FIG. 6 (a)
  • the upper and lower portions of the eye diagram as shown in FIG. 6 (b) are spread. Phenomenon occurs. That is, the maximum value of the eye diagram spread generated in FIG. 6 (b) is ⁇ y ⁇ w [0] [n] represented by ⁇ y [n] when there is only a ⁇ w [0] error as shown in FIG. 6 (d). Is equal to the maximum of the eye diagram of.
  • ⁇ w [0] in Fig. 6D is -0.12, which is -20% of w [0] which is 0.6. Therefore, when T is 0.4 ⁇ , max
  • FIG. 7 illustrates a case in which the coefficient error robust feedforward equalizer B-FFE according to the present invention is mounted in place of the feedforward equalizer C-FFE according to the related art of FIG. 6.
  • the operation of the feedforward equalizer C-FFE according to the prior art and the coefficient error robust feedforward equalizer B-FFE according to the present invention are the same.
  • a pulse response without inter-interference (ISI) can be obtained, and referring to the solid line of FIG. 7 (b), it can be seen that there is no spread of the eye diagram.
  • ISI inter-interference
  • ⁇ a [1] changes the pulse response and spreads the eye diagram as shown by the dotted line in FIG. 7 (b).
  • c [n] does not modify h ⁇ a [0] [n]. Therefore, h ⁇ a [0] [n ] is h ⁇ w [0] [n] , h ⁇ w [1] [n] the same decreases exponentially with and perturbation occurring in h ⁇ a [1] [n] There is no reduction effect. However, as in Equation 19, the normal value a [0] has a smaller value, so
  • LPF low pass filter
  • Equation 19 a [0] value can be predicted within an error of about 20%, and it can be seen from Equation 19 that a [0] is much smaller than other coefficients.
  • is 0.04, while
  • a method of improving the robustness by the coefficient error robust feedforward equalizer (B-FFE) according to the present invention is as follows.
  • the continuous time perturbation function p ⁇ w [m] (t) of the pulse transmitted by ⁇ w [m] is derived as shown in Equation 22 below, feed forward equalizer (C) according to the prior art
  • the spectrum P ⁇ w [m] (f) of p ⁇ w [m] (t) is derived as shown in Equation 23 below.
  • [Delta] w [m], T (t) and [Delta] w [m] are defined as coefficient error pulses of the input stage reference by [Delta] w [m] and the spectrum thereof.
  • Equation 26 the continuous time transmission error pulse p ⁇ a [m] (t) and its spectrum by ⁇ a [m] in the coefficient error robust feedforward equalizer (B-FFE) according to the present invention are shown in Equation 26 below. It is derived as shown in [Equation 27].
  • c (t) 0.5 ⁇ (t) -0.5 ⁇ (tT) is the continuous time impulse response of c [n]
  • C (f) jsin ( ⁇ fT) e (-j ⁇ fT) is c (t) Means the spectrum.
  • the continuous time coefficient error pulse response h ⁇ a [m] (t) by ⁇ a [m] is (h * p ⁇ a [m] ) (t), so from H (f) and P ⁇ a [m] (f)
  • the spectrum H ⁇ a [m] (f) of h ⁇ a [m] (t) can be derived as shown in Equation 28 below.
  • T (f) for the feedforward equalizer C-FFE are coefficients according to the present invention, as compared with only H (f) as shown in Equation 25 above.
  • ⁇ ⁇ a [n] T (f) (n ⁇ 0) is a high pass filter (HPF) C (f) and a low pass filter (LPF) as shown in Equation 28 above. ) Is filtered by channel H (f).
  • the coefficient error robust feed-forward equalizer (B-FFE) according to the invention H ⁇ a [n] (f) (n ⁇ 0) is a feed forward equalizer (C-FFE) in accordance with the prior art H ⁇ w [n] is much weaker than (f), and when n is 0, ⁇ ⁇ a [0], T (f) has a value much smaller than other values, as expressed in Equation 19 above.
  • FIGS. 8A to 8C show a two-tap feedforward equalizer (C-FFE) according to the prior art and a coefficient error robustness according to the present invention in a primary RC channel where a loss of 18 dB occurs at f N. This is a graph comparing various spectra of feed forward equalizer (B-FFE).
  • the coefficient error pulse spectrum ⁇ of the input terminal of the feed forward equalizer C-FFE according to the prior art and the coefficient error robust feed forward equalizer B-FFE according to the present invention ⁇ w [0], T (f), ⁇ ⁇ w [1], T (f), ⁇ ⁇ a [0], T (f), and ⁇ ⁇ a [1], T (f) It has a concentrated sinc function form.
  • 8 (b) shows the spectrum of C (f) which is a high pass filter (HPF).
  • C (f) attenuates the magnitude of P ⁇ a [1] (f) at low frequencies so that it is smaller than P ⁇ w [0] (f) and P ⁇ w [1] (f) as shown in FIG.
  • H (f) H (f) has a small magnitude in the entire frequency range, and as shown in FIG. 8, H ⁇ a [1] (f) is H ⁇ w [0] (f) or Much smaller than H ⁇ w [1] (f). H ⁇ a [0] (f) not filtered with the high pass filter (HPF) C (f) is also less than H ⁇ w [0] (f) or H ⁇ w [1] (f) by Equation 19 above. .
  • FIG. 8 (d) shows the results of simulating the spectrum of FIG. 8 (c) in a 36 dB channel with higher loss.
  • H ⁇ a [0] (f) and H ⁇ a [1] (f) are compared to H ⁇ w [0] (f) and H ⁇ w [1] (f) as compared to FIG. 8 (c).
  • the larger difference in d) shows that the value is small. Therefore, the coefficient error robust feedforward equalizer (B-FFE) according to the present invention provides a much better improvement in high loss channels.
  • Eye diagrams are a widely used way to measure the quality of communication.
  • the eye sensitivity S for the n th feedforward equalizer (FFE) coefficient ⁇ [n] with respect to how many coefficient errors the eye diagram can tolerate.
  • S k [n] is divided by the error rate of ⁇ [n] by dividing the reduction rate for the height reduction value ⁇ v eye, ⁇ [n] by ⁇ [n] from the optimal eye height v eye .
  • the eye sensitivity of the feed forward equalizer (FFE) is large, the eye diagram of the feed forward equalizer (FFE) is more sensitive to counting errors. Thus, eye sensitivity is useful as a measure of robustness.
  • the 2-tap feedforward equalizer (FFE) for the primary RC channel has a theoretically perfect eye when there is no perturbation, and the eye height (by g opt [n] according to Eq. v eye ) is determined by 2g opt [1] as shown in Equation 30 below.
  • Equations 33 and 34 are similar to Equations 31 and 32, respectively.
  • the eye sensitivity is also very large, and the high eye sensitivity means that the amplification error is amplified.
  • the smallest coefficient error can cause the eye diagram to close, requiring designers to tightly control the variation of the coefficients, even if they consume excessive hardware area and power.
  • the coefficient error robust feedforward equalizer (B-FFE) according to the present invention has an effect of improving eye sensitivity in a channel having a very high loss.
  • the coefficient error robust feedforward equalizer (B-FFE) coefficient a opt according to the present invention optimal for the primary RC channel is expressed by Equation 35 below using Equation 2 and Equation 12 below. Induced.
  • the eye sensitivity of the coefficient error robust feedforward equalizer (B-FFE) according to the present invention is derived from h ⁇ a [m] [n] of Equations 17 and 18 above. It can be seen from the above Equations 14 and 17 that max
  • , and ⁇ v eye -2max
  • ⁇ v eye -2
  • the eye sensitivity of ⁇ a [0] (S a [0] ) can be derived from Equation 36 below.
  • Equation (37) S a [1] is simplified as shown in Equation (37) below.
  • FIG. 9 is a graph showing eye sensitivity of a feed forward equalizer (C-FFE) according to the prior art according to the data rate and a coefficient error robust feed forward equalizer (B-FFE) according to the present invention.
  • the eye sensitivity of the feed forward equalizer (C-FFE) according to the prior art increases toward infinity, and the sensitivity of the coefficient error robust feed forward equalizer (B-FFE) according to the present invention converges to 1 or 2. Able to know.
  • the difference in eye sensitivity which is a degree of robustness improvement of the feedforward equalizer (C-FFE) according to the prior art and the coefficient error robust feedforward equalizer (B-FFE) according to the present invention, increases indefinitely with increasing data. Done.
  • the channel actually used may be modeled as a lossy transmission line as shown in FIG. 10.
  • the transfer function of a lossy transmission line is expressed by using the telegrapher's equation, including the telegrapher equation and the frequency-dependent channel secondary impact. , And [Equation 41].
  • H (f) is a frequency response of a channel of length l
  • Z c (f) is a characteristic impedance of the channel
  • Z Tx (f) and Z Rx (f) are the terminal impedances of the transmitter and receiver, respectively
  • R 0 , L 0 , G 0 , and C 0 are the RLGC variables of the channel at DC
  • R s and G d are the skin effect and dielectric Variables modeled for each loss.
  • the frequency response of the PCB or package wire may be mathematically calculated from the above Equations 38, 39, 40 and 41.
  • the eye sensitivity of the coefficient error robust feedforward equalizer (B-FFE) according to the present invention is much smaller than the feedforward equalizer (C-FFE) according to the prior art.
  • the worst-case eye sensitivity (S a [2] ) of the coefficient error robust feedforward equalizer (B-FFE) according to the present invention is 3.55 and The eye sensitivity Sw [1] of the feed forward equalizer C-FFE according to the related art is 25.7.
  • FIG. 12 is an eye diagram at 5Gb / s, 7Gb / s and 8.5Gb / s selected from FIG. 11, the coefficients all assuming a 10% error. Nyquist channel losses are 20dB, 27.9dB, and 33.6dB, respectively.
  • B-FFE coefficient error robust feedforward equalizer
  • C-FFE feedforward equalizer
  • FIG. 13 shows the channel loss at the Nyquist frequency and the coefficient forward robust equalizer (B-FFE) according to the present invention and the feedforward equalizer (C-FFE) according to the prior art.
  • This graph shows the eye sensitivity according to the data rate.
  • the channel is short at 3.5cm, but because of the narrow channel, the channel loss is high at 38.8dB at 10Gb / s, causing serious robustness problems.
  • the coefficient error robust feedforward equalizer (B-FFE) improves eye sensitivity more than 15 times compared to the feedforward equalizer (C-FFE) according to the prior art.
  • FIG. 14 is a diagram showing an eye diagram of a count error robust feedforward equalizer (B-FFE) and a feedforward equalizer (C-FFE) according to the prior art in a steady state, 1Gb when a count error of 10% occurs; It is a graph comparing at / s, 5Gb / s and 10Gb / s.
  • the Nyquist channel losses are 9 dB, 27.5 dB, and 38.8 dB at 1 Gb / s, 5 Gb / s, and 10 Gb / s, respectively.
  • the eye diagram of the feed forward equalizer (C-FFE) according to the prior art closes quickly, while the eye of the coefficient error robust feed forward equalizer (B-FFE) according to the present invention has a data rate Keep the eye diagram open even though it increases. 13 and 14, according to the present invention, there is an effect of improving the robustness against the coefficient error, in particular, it is effective in securing robustness in a situation where the robustness problem is serious due to the high data rate.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)

Abstract

La présente invention porte sur un égaliseur avant de flux robuste d'erreur de coefficient et, plus spécifiquement, sur un égaliseur avant de flux pour la communication câblée en bande de base permettant de pallier l'influence d'une erreur de coefficient générée par la distribution de nanoéléments. L'égaliseur avant de flux robuste d'erreur de coefficient selon un mode de réalisation de la présente invention comprend : un terminal récepteur (130) permettant de recevoir des données d'entrée (x) selon un indice de temps (n) entier ; un nombre N d'unités de retard (D) connectées au terminal récepteur (130) en série ; un premier calculateur (110) permettant d'additionner des signaux de prise émis respectivement par le nombre N d'unités de retard ; et un filtre de détection de changement de données (120) permettant de produire une valeur de transition de données (b) sur la base du changement des données d'entrée (x).
PCT/KR2014/003841 2013-05-07 2014-04-30 Égaliseur avant de flux robuste d'erreur de coefficient WO2014182000A1 (fr)

Priority Applications (1)

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US14/889,814 US9503293B2 (en) 2013-05-07 2014-04-30 Coefficient error robust feed forward equalizer

Applications Claiming Priority (4)

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KR10-2013-0051427 2013-05-07
KR20130051427 2013-05-07
KR20140052097A KR20140132277A (ko) 2013-05-07 2014-04-30 계수 오류 로버스트 피드포워드등화기
KR10-2014-0052097 2014-04-30

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Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2019039309A (ja) * 2017-08-22 2019-03-14 株式会社Ihi 可変圧縮装置及びエンジンシステム

Citations (5)

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Publication number Priority date Publication date Assignee Title
KR20050049304A (ko) * 2003-11-20 2005-05-25 한국전자통신연구원 지상파 디지털 방송 수신 시스템에서의 판정 궤환 등화장치 및 그 방법과, 그의 심볼 검출 방법
US20060067542A1 (en) * 2004-09-27 2006-03-30 Benny Christensen Feed forward equalizer
US20070025436A1 (en) * 2005-07-28 2007-02-01 Altera Corporation High-speed data reception circuitry and methods
US20070104265A1 (en) * 2005-11-04 2007-05-10 Hou-Wei Lin Equalizer and Equalizing Method thereof
US20110182347A1 (en) * 2010-01-25 2011-07-28 Fujitsu Limited Adaptive equalizer and adaptive equalizing method

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
KR20050049304A (ko) * 2003-11-20 2005-05-25 한국전자통신연구원 지상파 디지털 방송 수신 시스템에서의 판정 궤환 등화장치 및 그 방법과, 그의 심볼 검출 방법
US20060067542A1 (en) * 2004-09-27 2006-03-30 Benny Christensen Feed forward equalizer
US20070025436A1 (en) * 2005-07-28 2007-02-01 Altera Corporation High-speed data reception circuitry and methods
US20070104265A1 (en) * 2005-11-04 2007-05-10 Hou-Wei Lin Equalizer and Equalizing Method thereof
US20110182347A1 (en) * 2010-01-25 2011-07-28 Fujitsu Limited Adaptive equalizer and adaptive equalizing method

Cited By (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2019039309A (ja) * 2017-08-22 2019-03-14 株式会社Ihi 可変圧縮装置及びエンジンシステム

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