WO2013014168A1

WO2013014168A1 - System and method for checking position and/or dimensions of mechanical pieces

Info

Publication number: WO2013014168A1
Application number: PCT/EP2012/064524
Authority: WO
Inventors: Carlo Carli; Davide Castaldini; Andrea Ferrari; Paolo Lombardo; Sylvia Procurato
Original assignee: Marposs Societa' Per Azioni
Priority date: 2011-07-28
Filing date: 2012-07-24
Publication date: 2013-01-31

Abstract

A system for checking position and/or dimensions of a mechanical piece (2) includes one probe (6) with a detecting device (9a) and a movable transceiver(10) connected to the detecting device to wirelessly transmit to a remote transceiver (12) an optical signal indicative of the status of said probe, and to receive optical control signals from the remote transceiver. The remote transceiver includes a receiver stage (51) of the homodyne type. The signals transmitted by the movable and remote transceivers have a carrier that can be modulated with a binary message. The modulated signal is transmitted through one communication channel out of a plurality of communication channels, each communication channel being defined by the value of one selected frequency of the carrier. The bits of the modulating binary message can be suitably shaped in such a way to attenuate the spectral components that are far form the carrier frequency.

Description

DESCRIPTION

«SYSTEM AND METHOD FOR CHECKING POSITION AND/OR DIMENSIONS OF MECHANICAL PIECES»

Technical field

The present invention relates to a system for checking position and/or dimensions of a mechanical piece and to corresponding checking methods.

In particular, the present invention can be advantageously, but not exclusively, applied to numerical control machine tools to determine the position and/or the dimensions of machined mechanical pieces, to which reference will be explicitly made in the following description without loss of generality.

Background art As it is known, a checking system for checking the position and/or measuring the dimensions of a machined mechanical piece includes a probe, which is provided with a contact sensor and is movably mounted on the machine tool that machines or has machined the mechanical piece, and a remote base station, which is fixed to the frame of the machine tool and receives from the movable probe radio frequency signals or infrared signals including information about position and/or dimensions of the mechanical piece or about the exact instant of time at which the movable probe touches the mechanical piece. Generally, the probe is fed by batteries and thus its electric consumption is crucial for determining the commercial success of the checking and measuring system.

The checking systems employing infrared beams have had a greater commercial success because the propagation modes help the insulation of systems outside the line of sight, contrary to what happens with radio frequency signals. Moreover, the installation of infrared checking systems has not to meet, generally, strict requirements of electromagnetic compatibility which requires onerous standardization procedures.

In the infrared checking and measuring systems the probe is provided with an infrared transmitter, for example at least one infrared LED, and the base station is provided with at least one infrared receiver, for example an infrared photodiode. The wireless transmission of the useful infrared signals between the probe and the base station can be corrupted by the lightning of the environment where the machine tool and the relative checking and measuring system are placed. Indeed, the fluorescent lamps and the incandescent lamps can emit, in an unforeseeable way, spectral components in the infrared band.

There are different circuital solutions enabling a certain immunity to optical noise signals. For example, the US patent No. 7,350,307 B2 discloses a checking and measuring system in which the base station includes a comparator circuit which reconstructs the digital signal by comparing the optical signal received with a suitable threshold, and a generation and control circuit of the threshold which is configured to allow the threshold vary as a function of the peak value of the useful optical signal and as a function of the features of the optical noise signals.

However, the solution disclosed in the US patent No. 7,350,307 B2 is not completely immune to optical noise signals when fluorescent lamps are intensively used.

There are checking and measuring systems in which the probe transmits optical signals consisting of an infrared carrier, for instance a pulse carrier, that is on-off modulated based on the bits of the messages to be transmitted to the base station. In particular, the infrared pulse carrier consists of a series of infrared pulses that follow one another with a frequency higher than the frequencies which are usually the most affected by the optical noise signals. The receiver of the base station includes a suitable band-pass filter centred on the frequency of the pulse carrier to heavily attenuate the noise components outside the filter band. Anyway, solutions employing on-off modulated carriers imply, all conditions being equal, a greater current consumption and thus a shorter length of the charge of the probe batteries.

Disclosure of the invention Object of the present invention is to provide a system for checking the position and/or the dimensions of a mechanical piece that overcomes the inconveniences above-described and concurrently is easily and cheaply implemented.

The present invention provides a system and checking methods for checking the position and/or the dimensions of a mechanical piece according to what is claimed in the attached claims.

Brief description of the drawings

The present invention is hereinafter described with reference to the attached sheets of drawings given by way of non-limiting examples, wherein:

figure 1 shows, in a schematic way, a numerical control machine tool provided with the checking system according to the present invention;

figure 2 shows a detailed block diagram of the probe of the checking system of figure 1 ;

figure 3 shows an electrical functional diagram of an input band-pass filter of the receiving section of the probe of figure 2 ;

figure 4 shows a flow chart of a gain control algorithm of the amplifier means of the receiving section of the probe of figure 2 ;

- figure 5 shows a detailed block diagram of the base station of the checking system of figure 1 ;

figure 6 shows a block diagram of the receiving section of the base station of the checking system of figure 1 according to another embodiment of the invention ;

figure 7 shows an electrical functional diagram of one part of the receiving section of figure 6;

figure 8 shows more detailed block diagram of a final stage of the receiving section of figure 6;

figures from 9 to 12 show more detailed block diagrams of the final stage of the receiving section of figure 6 according to further embodiments of the invention; figure 13 shows two examples of waveforms generated by the transmission section of the probe of figure 2; and figures from 14 to 17 shows the trend of the waveforms of signals in the final stage of the receiving section of figure 6, according to some of the embodiments of figures 9 to 12.

Best mode of carrying out the invention Figure 1 very schematically shows a numerical control machine tool 1, a mechanical piece 2 machined by the machine tool 1 and positioned in the machine tool 1 itself and a checking system for checking the position and/or the dimensions of the mechanical piece 2. The machine tool 1 includes a numerical control unit 4. The checking system 3 includes a probe 6 which is mounted on the machine tool 1 in such a way as to move, for example by means of slides 7, in the area where the piece 2 is placed, and a remote base station 8 which is fixed to the bed of the machine tool 1 at a distance from the probe 6, is connected with the numerical control unit 4 by means of suitable interface means 5 and is able to communicate with the probe 6 via infrared signals. According to a not illustrated embodiment, the interface means 5 are integrated with the base station 8.

According to the present invention, the probe 6 consists of, for example, a touch probe and includes, in particular, a movable arm with one end carrying a feeler 9, a detecting device 9a which is adapted to generate a suitable electrical signal as soon as the feeler 9 touches the mechanical piece 2, and a movable transceiver 10 to transmit information about the state of the probe 6 to the base stations 8 via infrared signals, and to receive control information from the base station 8 via infrared signals. The probe 6 includes, moreover, a microcontroller 11 connected to the detecting device 9a to receive the electrical signal generated by the detecting device 9a and to the movable transceiver 10 to control the operation of the latter. The base station 8 includes a remote transceiver 12 to communicate via infrared signals with the probe 6 and a microcontroller 13 to control the operation of the remote transceiver 12 and to interact with the numerical control unit 4 by means of the interface means 5. The transceivers 10 and 12 exchange, in use, infrared signals obtained by demodulating a carrier, for instance a pulse carrier, with suitable binary coded messages. The pulse carrier consists for instance of a series of infrared pulses that follow one another with a frequency between lOOkHZ and 10MHZ, and in particular between 300kHz and 600kHz. The pulse carrier is amplitude modulated by the binary messages. Advantageously, the amplitude modulation is an on-off modulation. Moreover, the frequency of the pulse carrier can be selected within a set of values including, for example, 350kHZ, 450kHz and 570kHz, so as to define as many communication channels that do not interfere with each other, as it will hereinafter explained in detail.

With reference to figure 2, the movable transceiver 10 of the probe 6 includes a receiving section 14, a transmission section 15 and a digital regulating device 16 consisting of, for example, a FPGA (Field Programmable Gate Array) suitable configured to regulate in real time some parameters of the receiving section 14 and transmission section 15. The digital regulating device 16 is, in turn, controlled by the microcontroller 11 and a power supply unit 32 is connected to the detecting device 9a, the digital regulating device 16, the microcontroller 11 and other components of the probe 6.

The receiving section 14 includes receiving devices with one or more infrared photodiodes 17, in particular four photodiodes 17 (only one is shown in figure 2, for the sake of simplicity) , which are oriented so as to ensure an omnidirectional reception, are mutually connected in parallel and are operated in photovoltaic mode, which means that they are not polarized, and a band-pass filter 18 which can be tuned to the wanted frequency of the pulse carrier and is connected immediately downstream from the photodiodes 17. The photodiodes 17 are operated in photovoltaic mode in order to reduce the electric current consumption of the probe 6 that is fed by a battery.

The receiving section 14 further includes variable gain amplifier means 19, to amplify the signal provided by the band-pass filter 18 and to provide a corresponding amplified signal VA, and a comparing section comprising a low-pass filter 20 to filter the amplified signal VA and a comparator 21 to square the amplified signal VA, that is to generate a corresponding processed signal, in particular a binary signal, VD on the basis of a comparison between the filtered amplified signal VF and a sum signal VTS consisting of the sum of the amplified signal VA and an adjustable reference signal or comparison threshold VTH. The low-pass filter 20 is connected to the inverting input of the comparator 21. The presence of the low-pass filter 20 in the comparing section enables to increase the immunity of the binary signal VD to the variations of the average value of the amplified signal VA. The filtered amplified signal VF represents, in substance, the average value of the amplified signal VA.

According to another, not illustrated, embodiment of the invention, the inverting input of the comparator 21 is connected to a fixed reference potential and the receiving section 14 does not include the low-pass filter 20 of figure 2 and comprises a high-pass filter to eliminate the continuous component from the amplified signal VA which consists, for example, of a capacitor in series with a resistor and is connected between the output of the amplifier means 19 and the summing junction providing the sum signal VTS.

With reference with figure 2, the digital regulation device 16 receives the binary signal VD, that is still a high frequency signal, and is configured to reconstruct, starting from the binary signal VD, the baseband envelope of the signal received by the probe 6, that is the received binary message originally transmitted by the base station 8.

There are several reconstruction techniques of the envelope. According to a preferred solution, which has provided the best results, the digital regulating device 16 is configured in such a way as to implement an IIR (Infinite Impulse Response) low-pass filter (not shown) which receives the binary signal VD, and a comparator with hysteresis (also not shown) connected to the output of the IIR filter, which receives the filtered binary signal and provides the baseband envelope of the received signal.

The digital regulating device 16 is configured to automatically control the gain of the amplifier means 19 as a function of the processed signal VD. The digital regulating device 16 is also configured to automatically adjust the threshold VTH as a function of the processed signal VD by means of a DAC (Digital Analog Converter) 22. Moreover, the regulating device 16 is configured to adjust the center band frequency of the band-pass filter 18.

The amplifier means 19 include a plurality of amplifier stages 23a-23c (three of them being show as an example in figure 2), which are cascade connected and have each a constant gain G, and selector means 24, which comprise a plurality of selection inputs, each one connected to the output of one of the amplifier stages 23a-23c. The selector means 24 are configured to be controlled by the regulating device 16 so as to select the output of one of the amplifier stages 23a-23c and thus select a different amplification level. The gain G of each amplifier stage 23a-23c is equal to about 25dB and as a consequence, in the example of figure 2, the amplification level can be selected in a set of values including 25dB, 50dB and 75dB. The figure 3 shows an electrical functional diagram of the band-pass filter 18. The band-pass filter 18 includes a plurality of reactive components, for instance inductors 25, which are mutually connected, for instance they are connected in parallel to one another, and a respective plurality of switches 26, each of them being coupled to, for instance arranged in series with, a respective reactive component/inductor 25. The switches 26 are controlled, as regards their opening and closure, by the regulating device 16 to select the center and frequency of the band-pass filter 18 and thus to select the communication channel. The inductors 25, when the respective switches are closed, are connected in parallel to the photodiodes 17. The photodiodes 17 feature respective capacitances that add up and give rise to a not negligible, overall capacitance in parallel to the inductors 25, that is shown in dashed line in figure 3. The center band frequency can be selected in a group of frequencies defined by at least one part of all the possible on/off combinations of the switches 26, the combination of all the open switches excluded. The embodiment of figure 3 showing, as an example, two inductors 25, enables to select three different center band frequencies, that is three different communication channels, by closing just one or the other of the switches 26 or both the switches 26. It is essential that at least one of the inductors 25 is connected in parallel to the photodiode 17 to provide a low impedance load for the continuous and low frequency components of the electric current generated by the photodiode 17 in the presence of environmental light. Indeed, as the photodiodes 17 are not polarized, if the voltage at their ends exceeded a few hundreds millivolts, the junctions of the photodiodes 17 would be polarized directly and this would unacceptably reduce their optical sensitivity.

The band-pass filter includes, moreover, a fine regulating circuit 27, which is provided with one or more varicap diodes, is connected in parallel to the inductors 25 and is controlled by the regulating device 16.

The fine regulating circuit 27 provide an additional capacitance that can be regulated during the calibration phase of the probe 6 by means of the regulating device 16 to compensate the tolerances of the inductors 25 and of the capacitances of the photodiodes 17 and to center with better accuracy the center band frequency on the frequency of the pulsed signal transmitted by the base station 8. According to another, not illustrated, embodiment of the band-pass filter 18, the reactive components in series with the switches 26 are capacitors that replace the inductors 25 of figure 3, and that are in parallel to a fixed inductance. Other embodiments, also not shown, include series connected reactive components, each coupled with a switch adapted to selectively exclude the corresponding reactive component.

With reference to figure 2, the transmission section 15 of the movable transceiver 10 of the probe 6 includes one or more infrared LEDs 28 adapted to transmit a modulated optical signal having a carrier, indicative of the status of the probe 6, a voltage doubler circuit 29 coupled to the power supply unit 32 and controlled by the regulating device 16 to double the power supply voltage of the LEDs, and drive means 30 of the LEDs 28 including a DAC controlled by the regulating device 16 and a voltage-to- current converter circuit to transform the drive voltage of the LEDS in output from the DAC into a drive current for the LEDs 28. The drive voltage of the LEDs consists of a pulse carrier modulated by a binary message to be transmitted to the base station 8. The voltage doubler circuit 29 enables to drive more LEDs 28 connected in series .

In particular, the voltage doubler circuit 29 includes a plurality of capacitor banks which are connected in parallel to a power supply voltage (for example 3.6 V) for most of the time and are connected in series on a command of the regulating device 16 only when it is necessary to perform the transmission and to have a double voltage in order to drive the LEDs correctly.

The structure of the transmission section 15, as described above, enables to dynamically regulate the amplitude of the envelope of the drive voltage of the LEDs 28 and thus the intensity of the infrared beams emitted by the LEDs 28, in order to obtain at least two advantageous results, that is to optimize the electric current consumption of the probe 6 during the different operation phases, and to properly shape the bits of the binary message that modulate the pulse carrier, i. e. the envelope of the drive voltage of the LEDs, in such a way to attenuate the spectral components that are far from the carrier frequency and that could interfere with other probes of the same type operating concurrently and featuring different carrier frequencies. Figure 13 shows two examples of shaping of the envelope of the drive voltage of the LEDs. The envelope marked with VLq has a rectangular shape while the envelope marked with VLs is shaped as a sinusoid portion.

The spectral components on the bordering channels of the envelope VLs are more attenuated than the

corresponding components of the envelope Vlq.

According to another shaping example, not shown in the figures, the binary signal is modulated with a carrier with a lower harmonic distortion, for example with a sinusoidal shape. In this way, the multi-frequency spectral components of the carrier, and thus the interferences on relatively far channels, are reduced.

The power supply unit 32 of the probe 6 includes an electric battery and a switching-mode power supply to fed all the electric and electronic components of the probe 6. The regulating device 16, following the instructions of the microcontroller 11, can disable the power supply unit 32 letting the probe 6 operate for a short time period powered just by the charge of the filtering and equalization electric capacitors of the power supply unit 32. Such an operation mode is useful during the receiving phase to reduce the electromagnetic noises generated by the switching-mode power supply on the receiving circuits and thus improving the performances of the probe 6.

Finally, the probe 6 includes acoustic and/or visual warning devices 33 controlled by the microcontroller 11 to provide the operator of the machine-tool 1 with information about the state of the probe 6.

As previously stated, the regulating device 16 is configured to automatically control the gain of the amplifier means 19 and automatically adjust the threshold VTH as a function of the binary signal VD, in order to compensate the great variability of the intensity of the received signals and to optimally reconstruct the baseband envelope of the signal received by the probe 6, that is the binary message received. These automatic control and adjustment take place according to respective algorithms, hereinafter described.

With reference to figure 4 showing a flow chart of the algorithm relating to the automatic control of the gain of the amplifier means 19, first of all the threshold VTH is set at a proper initial value VI (block 100 of figure 4) and the output of the first amplifier stage 23a is selected (block 101) . The value VI is defined as follows. The amplified signal VA oscillates in a substantially symmetrical way about its average value that is represented by the filtered amplified signal VF, and has a peak value, with respect to the average value, limited to a value VAM equal to less than half the power supply voltage of the amplifier stages 23a-23c. The value VI is almost equal to the value VAM divided by the typical gain of the amplifier stages 23a-23c. For example, if the power supply voltage is equal to 3V, the typical gain to 25dB and the value VAM to 1.25V, the value VI is of about 75 mV. In this way, when the peak value of the signal at the output of one of the amplifier stages 23a-23b is higher than the value VI, the subsequent amplifier stage 23b-23c is saturated and thus can not be used. This condition occurs when the values of the amplified signal VA are lower than its average value, and generates values of the sum signal VTS lower than the corresponding values of the filtered amplified signal VF and thus pulses at the low level of the binary signal VD. If the above mentioned condition does not occur, the binary signal remains constantly at the high level.

Therefore, if the binary signal VD doesn't remain constantly at the high level, that is it has pulses at the low level, which means that the peak value of the amplified signal VA is higher than the threshold VTH (output NO of the block 102), then the selection of the first amplifier stage 23a is correct and the algorithm ends. Otherwise (output YES of the block 102), the output of the second amplifier stage 23b is selected (block 103) . The same check is performed on the binary signal VD (block 104) and if it is negative (output NO of the block 104), then the selection of the second amplifier stage 23b is correct and the algorithm ends. Otherwise (output YES of the block 104) the output of the third amplifier stage 23c is selected (block 105) and then the algorithm ends.

The algorithm of figure 4 refers to an example with three amplifier stages 23a-23c. It is obvious that if there are more than three amplifier stages, the end part of the algorithm is repeated for each additional stage.

When the gain control of the amplifier means 19 ends, the threshold VTH is automatically adjusted, according to the algorithm hereinafter described, in order to find an optimal value of the threshold VTH. At the beginning, the threshold is set at a value equal to half an interval defined by the maximum dynamics of the amplified signal VA, that is half the value VAM, so as to define an upper subinterval and a lower subinterval . If the sum signal VTS has values lower than the corresponding values of the filtered amplified signal VF, which means that the peak value of the amplified signal VA is higher than the threshold VTH, the threshold VTH is set at a new value corresponding to the center of the upper subinterval. Otherwise, the threshold VTH is set at a new value corresponding to the center of the lower subinterval. This operation is repeated until the threshold VTH is close enough, taking into account the DAC 22 resolution, to the peak value of the amplified signal VA. The threshold value VTH that is used to reconstruct the envelope of the received signal is set at half the peak value that has been so identified.

To carry out the operations of automatic gain control of the amplifier means 19 and regulation of the comparison threshold VTH it is assumed that the optical signal is always present at the input of the probe 6. In order to satisfy this assumption, the base station 8 transmits, before each message, a preamble of sufficient length so that the probe 6 can do the proper adjustments, in a substantial continuous way, in its receiving section 14, that is automatically control the gain of the amplifier means 19 and regulate the comparison threshold VTH at optimal values. In this way, the probe 6 can adjust its own optical sensitivity to the actual quality of the optical channel. Moreover, the microcontroller 11 can use the adjustments of the receiving section 14 performed during the preamble to estimate the quality of the optical channel, the estimate being used to regulate the power supply of the transmission section 15, in particular the drive voltage of the transmission LEDs 28. Hence, if the optical channel is particularly reliable, for example when the probe 6 is very close to the base station 8 or, anyway, is placed in a direction ensuring a very good line of sight communication, the transmission section can reduce the electric current feeding the LEDs 28 in order to reduce the intensity of the transmitted infrared beams, and thus to enable the probe 6 to save electric power.

With reference to figure 5, the remote transceiver 12 of the base station 8 includes a receiving section 34 and a transmission section 35. The receiving section 34 and the transmission section 35 have a substantially digital structure and communicates directly with the microcontroller 13 of the base station 8. In particular, as shown in the example of figure 5, the remote transceiver 12 includes a digital, integrated circuit, in particular a FPGA (Field Programmable Gate Array) device 36 implementing a part of the functional blocks of both the receiving section 34 and the transmission section 35.

Moreover, the base station 8 includes a power supply unit 37 which can be connected to an external electric power supply, for example the electric network or the onboard, electric power supply of the machine tool 1, to feed all the electric and electronic components of the base station 8.

The power supply voltages of the different applications are very different from each other and sometimes prevent to connect the frame of the receiver to the ground reference of the electronics. Such a situation is particularly deleterious to the resulting susceptibility to electrical noises, because the unavoidable stray capacitances between the frame and the electronic circuits introduces in the electronic circuits noises, when they consist of quick variation of the potential difference between the frame and the ground reference of the electronic circuits. The power supply unit 37 includes an insulated converter, in particular a DC/DC flyback converter with an electronic control unit which can provide the electronic components of the base station 8 with the difference, necessary power supply voltages. The DC/DC flyback converter enables to electrically insulate the onboard, electric power supply of the machine tool 1 from all the electronic components of the base station 8. It is thus possible to connect the frame of the base station 8 to the ground of the electronics. In this way, the potential difference between the external, metallic frame of the base station 8 and the electronic components is steadily set to zero and thus the electrical noises introduced in the electronic components by the stray capacitances between the metallic frame and the components are drastically reduced.

The transmission section 35 includes one or more infrared LEDs 38 transmitting a modulated optical signal having a carrier, a transmission block 40 which receives, from the microcontroller 13, a message to be transmitted wirelessly to the probe 6 and transforms such message into a corresponding, digital control signal of the LEDs 38, a DAC 39 controlled by the transmission block 40 to provide a drive voltage of the LEDs 38 and drive means 39a of the LEDs 38, which comprise a voltage-to-current converter circuit to transform the drive voltage into a drive current for the LEDs 38. The transmission block 40 is implemented in the FPGA device 36. Unlike the probe 6, the transmission section 35 does not include any voltage doubler circuit because, thanks to the power supply unit 37 connected to the external, electric power supply, the power supply voltage of the LEDs is high enough. A part from that difference, the structure of the transmission section 35 is the same as that one of the transmission section 15 of the probe 6 shown in figure 2 to enable a dynamic regulation of the amplitude of the envelope of the drive voltage of the LEDs 38 in order to optimize the electric current consumption of the base station 8 in the different operation phases and attenuate the spectral components far from the carrier frequency.

The receiving section 34 includes receiving devices with one or more infrared photodiodes 41 (only one is shown in figure 5, for the sake of simplicity) , which are mutually connected in parallel and inversely polarized, and processing means with amplifier means 42 to amplify the signal provided by the photodiodes 41 and provide a received amplified signal VAB, and a conversion and filtering unit 43 connected to the output of the amplifier means 42 to provide a corresponding digital, amplified signal VDB coded with a high number of bits, for example 24 bits. The conversion and filtering unit 43 consists of an integrated component including, connected in cascade, a sigma-delta ADC 44 and band-pass filter 45 that can be tuned to the wanted frequency of the pulse carrier. The sigma-delta converter 44 has a digital output coded with a high number of bits and the band-pass filter is a FIR (Finite Impulse Response) filter. According to the example shown in figure 5, the output of the sigma-delta converter 44 is coded at 24 bits and the band-pass filter 45 is a filter of the 95^th order (96 taps) . The response of the band-pass filter 45 can be configured by changing the value of the coefficients of the filter. The selection of the frequency of the pulse carrier, and thus of the communication channel between the base station 8 and the probe 6, can be performed by writing into internal registers of the band-pass filter 45. In particular, the FPGA device 36 is adapted to tune the band-pass filter 45 to the wanted communication channel under the control of the microcontroller 13.

Moreover, the receiving section 34 includes a rectifier block 46, which is connected to the output of the conversion and filtering unit 43 to rectify the digital signal VDB by an absolute value operation and to provide a corresponding rectified signal VRB, a low-pass filter 47 which consists, for example of a IIR filter connected to the output of the rectifier block 46 to provide a corresponding rectified filtered signal VFB, a threshold generator block 48 to calculate and provide a comparison threshold VTB as a function of the rectified filtered signal VFB, and a comparison block 49 to determine, as a function of the comparison between the rectified filtered signal VFB and the threshold VTB, an output signal corresponding to the baseband envelope of the signal received from the base station 8, that is the received signal without the pulse carrier. The baseband envelope provided by the comparison block 49, or output signal, is marked in figure 5 with VIB. Finally, the receiving section 34 includes, a receiving block 50 to transform the baseband envelope VIB into a corresponding received binary message MSG, which is communicated to the microcontroller 13. In particular, the block 50 includes registers, which are read by the microcontroller 13, and implements one or more algorithms to write the received binary message MSG on said registers and to check the validity of both the baseband envelope VIB and the received binary message MSG. The rectifier block 46, the low-pass filter 47, the threshold generator block 48, the comparison block 49 and the receiving block 50 are implemented in the device FPGA 36. The threshold generator block 48 implements, and thus comprises, a peak detector (not illustrated) with rise times shorter than fall times in order to generate a peak signal that follows the peak of the rectified filtered signal VFB and an attenuator block (not illustrated) to generate the threshold VTB by reducing the amplitude of said peak signal by 50%. The so obtained threshold VTS follows the peak of the rectified filtered signal VFB and thus adjusts to the power of the optical signal received from the receiving section 34.

According to another embodiment of the invention, the infrared pulse carrier transmitted in the message exchange between the probe 6 and the base station 8 is modulated in frequency by binary messages.

For example, a frequency of 450kHz corresponds to each bit at high logic level and a frequency of 100kHz corresponds to each bit at low logic level. If there are more communication channels, the frequency of the pulse carrier for the transmission of one bit at high logic level can be selected within a set of a values including, for example, 350kHz, 450kHz and 570kHz, while the frequency of 100kHz corresponds to each bit at low logic level. This embodiment differs from those shown in figures 2 and 5 essentially in the operational logic of the part of the FPGA device that generates the drive voltage of the LEDs .

According to another embodiment of the invention, the transmitted infrared pulse carrier is phase modulated, for example by associating a phase difference of 0° of the pulse carrier to each bit at high logic level and a phase difference of 180° to each bit at low logic level.

According to a further aspect of the system, per se known, a communications protocol between the probe 6 and the base station 8, that is a method to exchange messages between the probe and the base station 8, is provided.

According to the communications protocol, there is an initial programming phase of the probe 6 including the bidirectional and not simultaneous exchange of messages between the base station 8 and the probe 6 in order to configure a series of parameters of the probe 6, for example the frequency of the communication channel, a power-off timer, etc.

When the programming phase ends, the probe 6 enters a standby state in which it transmits periodically a beacon signal to signal to the base station 8 that it can be activated .

The base station 8, when decides to activate the probe 6, responds to the beacon signal by transmitting an activation acknowledge message. The probe 6, in turn, responds by transmitting an identification message including the ID of the probe 6. At this point, the base station 8 sends a proper message of operating control to the probe 6 to commutate the latter to the operating state, and starts listening to messages transmitted by the probe 6.

The messages transmitted by the probe 6 in the operating state includes state messages and variation messages. The state messages include information on the state of the detecting device 9a and/or information on the state of the battery of the power supply unit 32. In particular, when the feeler 9 does not touch the mechanical piece 2, the probe periodically transmits state messages, for example in a sequence including four subsequent feeler state messages and one battery state message. The base station 8, immediately after having received a battery state message, transmits an acknowledge message of receipt. Otherwise, when a touch and a disengagement occurs, that is when the feeler 9 starts touching the mechanical piece 2 and release the contact with the mechanical piece 2, respectively, the probe 6 transmits, as soon as possible, or anyway with a repeatable delay, a sequence of two variation messages. The base station 8, immediately after having received the two variation messages, transmits the information to the numerical control unit 4. The delays with which the variation messages are repeated can differ depending on the communication channel (pulse carrier) used. Such an expedient enables to minimize the chance that several variation messages are transmitted simultaneously by different probes 6 that works simultaneously and have different pulse carriers.

Indeed, the interference between variation messages transmitted by several probes 6 could cause serious malfunctions in very unfavourable situations, for example when the useful signal is received very weakly and interfering signals transmitted in adjacent channels are received with high intensity.

Moreover, as previously stated, a new feature of the communications protocol between the probe 6 and the base station 8 according to the present invention is that the base station 8 transmits, immediately before each message, a respective preamble of sufficient length so that the probe 6 can calibrate its receiving section 14.

Figure 6 shows the receiving section of the remote transceiver 12 of the base station 8 according to another embodiment, in which the corresponding elements are marked with the same reference numbers and abbreviations as figure 5. According to such embodiment, the receiving section of the remote transceiver 12 has a substantially analog structure and includes processing means with a receiver stage of the homodyne type connected to the output of the receiving devices, in particular of the amplifier means 42, to provide the rectified filtered signal VFB representing a processed input signal, and a digital pulse shaper processing stage 52 which is connected to the output of the receiver stage 51 to provide the baseband envelope VIB of the signal received by the base station 8.

With reference to figure 6, the receiver stage 51 of the frequency conversion type includes a local oscillator 53 for generating a first reference signal VOl which has a nominal frequency equal to the one of the carrier of the received signal VAB, a phase shifter 54 fed by the local oscillator 53 for generating a second reference signal V02 in quadrature with the first reference signal VOl, two non^¬ linear mixer 55 consisting of, for example, respective multipliers to mix separately each of the reference signals VOl and V02 with the received signal VAB, and two identical low-pass filters 56 connected, each one, to the output of a respective mixer 55 for providing two respective baseband signals VBB1 and VBB2. The amplitude of the signals VBB1 and VBB2 is proportional to the product between the baseband envelope of the received signal and the sine and the cosine, respectively, of a same amount that is a function of the unavoidable frequency and phase difference between the pulse carrier of the received signal and the reference signal VOl that are not synchronized to each other .

Moreover, the receiver stage 51 includes a detecting circuit 57 which receives the two baseband signals VBB1 and VBB2 and provide the rectified filtered signal VFB. The receiver stage 51 enables an accurate tuning to the different pulse carriers without the need to change the filter parameters thanks to the use of the local oscillator 53 the frequency of which can be accurately programmed by the logic. At the same time, the receiver stage 51 is a simpler and cheaper solution than the digital solution of figure 5. Indeed, the reference signal VOl is already available in any case, because it is used by the transmission section of the base station 8 to generate the signals to transmit to the probe 6 and there is no need of increasing the circuit layout complexity to generate the reference signal VOl. In addition, the receiver stage 51 provides a band-pass filtering centered on the pulse carrier with performance standards equivalent to those of the digital solution of figure 5, even thought it uses low- pass filters 56 of not very high order compared to the complexity of the band-pass filter 45 of figure 5.

Figure 7 shows the circuit functional diagram of the detecting circuit 57 which comprises two active full wave rectifiers 58 which receive the baseband signals VBB1 and VBB2 respectively and have the outputs connected, by means of two respective identical resistors 59, to a single output 60 to provide the rectified filtered signal VFB.

Each active rectifier 58 includes a respective first operational amplifier 61a configured as an inverting half wave rectifier and marked with the reference number 61 and a respective second operational amplifier 62a configured as non-inverting half-wave rectifier and marked with the reference number 62. Each half-wave rectifier 61, 62 includes a respective first diode 63a, 64a and a respective second diode 63b, 64b. Each of the first diodes 63a, 64a is feedback coupled to the respective operational amplifier 61a, 62a, that is to say has the anode and the cathode connected to the inverting input and to the output of the respective operational amplifier 61a, 62a respectively. Each of the second diodes 63b, 64b is connected at the output of the respective operational amplifier 61a, 62a, that is to say has the anode and the cathode connected to the output of the respective operational amplifier 61a, 62a and to the corresponding resistor 59, respectively. Each inverting half-wave rectifier 61 includes a first resistor 65a connected in series to the inverting input of the respective operational amplifier 61a, and a second resistor 65b feedback coupled to the respective operational amplifier 61a, and more specifically between the inverting input of the operational amplifier 61a and the cathode of the corresponding diode 63b.

The resistors 65a and 65b have the same ohmic value Rl . Each non-inverting half-wave rectifier 62 includes a respective resistor 66 which is feedback coupled to the respective operational amplifier 62a, and more specifically between the inverting input of the operational amplifier 62a and the cathode of the corresponding diode 64b. The resistors 66 have a value R2 much higher than the value Rl, for example higher or equal to Rl*10. Advantageously, the value R2 is equal to Rl*100. The resistors 59 have an ohmic value equal to R1/V2: it can be demonstrated that this condition optimizes the circuit performances.

The above described detecting circuit 57 is a simpler circuit solution than the known theoretic solution which consists in extracting the square root of the sum of the squares of the signals VBB1 and VBB2. The latter would be onerous also because of the great variability of the intensity of the received signals. The downside of the simple circuitry is that the rectified filtered signal VFB provided by the detecting circuit 57 is affected by a residual oscillation with an amplitude equal to about 8% of the whole amplitude of the signal and with a frequency eight times higher than the frequency difference between the pulse carrier of the received signal and the reference signal VOl . However, the residual oscillation can be tolerated in return for simplicity of the detecting circuit 57, thanks also to the presence of the subsequent digital pulse shaper processing stage 52.

Figure 8 illustrates a block diagram of the digital pulse shaper processing stage 52 which comprises an amplifier 67, of the inverting type, with high input impedance (in order not to affect the operation of the preceding detecting circuit 57) and with a frequency response of the high-pass type to amplify the rectified filtered signal VFB and provide a corresponding amplified signal VFAB which represents a processed input signal. The digital pulse shaper processing stage 52 also comprises a control section with a threshold generator and control block 68 to generate and define, as a function of the amplified signal VFAB, a reference signal or comparison threshold VTHB, and a comparison section or comparison block 69 to determine the baseband envelope , or output signal, VIB on the basis of a comparison between the amplified signal VFAB and the comparison threshold VTHB.

As an alternative to the amplified signal VFAB, and by properly dimensioning the detecting circuit 57 and/or the circuits of the threshold generator and control block 68, the rectified filtered signal VFB can be sent directly to the input of the comparison section 69 and of the threshold generator and control block 68.

The threshold generator and control block 68 includes a threshold generating circuit 72 and circuits 70, 71 for automatically controlling the amplitude difference between the processed input signal VFB or VFAB and the reference signal VTHB, in particular a detector circuit 70 for generating , as a function of the processed input signal VFAB, a signal VU indicative of peak values of the processed input signal VFAB, and a discriminator circuit 71 for generating, as a function of the amplified signal VFAB, a signal VN indicative of the noise signal overlapping the useful signal transmitted by the movable transceiver (10) of the probe 6. The detector circuit 70 is connected to the threshold generating circuit 72, that generates the threshold VTHB, and automatically adjusts such threshold VTHB dynamically and temporarily as a function of the signals VU and VN. The threshold generator and control block 68 includes, moreover, a programmable module 73 to memorize information about the particular application in which the checking system 3 is used. The threshold generating circuit 72 determines the comparison threshold VTHB also depending on the information included in the programmable module 73. The circuits 70, 71, 72 and the module 73 are automatic sensitivity control circuits, since they perform an automatic control of the receiving sensitivity of the remote transceiver 12, by reducing automatically such receiving sensitivity on the basis of features of the processed input signal VFAB (o VFB) , in particular by automatically adjusting the comparison threshold VTHB.

The comparison block 69, the circuits 70, 71, 72 and the module 73 are similar to those described in the US patent No. 7,650,307 B2 assigned to the applicant of the present patent application. However, the present invention differs from the solution described in the US patent No. 7,650,307 B2 in that the digital pulse shaper processing stage 52 includes a inhibitor circuit 74, connected to the discriminator circuit 71 and to the threshold generating circuit 72, that prevents the signal VFAB from being applied to the discriminator circuit 71 when the output VIB of the comparison block 69 is at the high logic level, that is when the amplified signal VFAB exceeds, in absolute value, the threshold VTHB. The inhibitor circuit 74 includes an analog transmission port, with logic command, which is represented in simplified way in figure 8 by a controlled switch and as a first input receiving the signal VFAB , a second input receiving the output signal of the comparison block (VIB) and an output providing the discriminator circuit 71 with the corresponding result, that is it provides the signal VFAB only when the output of the comparison block 39 is a at low logic level.

The aim of the inhibitor circuit 74 is to inhibit the effect of the discriminator circuit 71 on the control of the threshold VTHB on the basis of features of the output signal VIB, in particular in the presence of the useful signal. Indeed, to improve the immunity to noises, the band of the useful signal transmitted, in this case by the probe 6, must be quite narrow around the pulse carrier to enable the low-pass filters 56 of the receiver stage 51 to eliminate the noise signals. As the bandwidth of the useful signal is inversely proportional to the length of the sequences of pulses, or pulse bunches, corresponding to one bit of the binary messages (useful signal) , the length of the pulse bunches can not decrease under a certain value. Hence, in case of use of a pulse carrier the bits of the binary messages must have a length greater than that one in absence of a pulse carrier. A grater length of the bits would activate the automatic sensitivity control part generated by the discriminator circuit 71 even on absence of noises, causing an unwanted, though slight, sensitivity reduction. This is the reason why the discriminator circuit 71 is inhibited in the presence of the useful signal, in order to optimize the performances.

Figure 9 shows another embodiment of the digital pulse shaper processing stage 52, with automatic sensitivity control circuits, which comprises processing means with a phase inverter 112 generating the rectified filtered inverted signal VFBN (which is here represented for the sake of congruence with the signals of the other figures, but the function of which can be obtained by simply inverting the polarity of all the diodes of the detecting circuit 57), a high-pass filter 75 composed, for example, of a capacitor in series to a resistor, that has the function of attenuating the low frequency components of the noises, a variable gain amplifier stage 76, at the output of which the signal VFAB is available, a non-linear amplifier stage 77 connected to the output of the amplifier stage 76 to amplify the weaker signals and compress the signal with more intensity, a comparison section with a comparator with hysteresis 78 to determine the baseband envelope VIB on the basis of a comparison between a processed input signal VAN generated in the non-linear amplifier stage 77 and a fixed comparison threshold VTH1 which represents reference signals, and a control section 79 with automatic sensitivity control circuits, to generate, as a function of the processed input signal VAN, a control signal VC that can control the gain of the amplifier stage 76. The comparison threshold VTH1 is provided by means of a DC voltage source having the negative terminal connected to the non-inverting input of the comparator 78 and the positive terminal connected to the ground. The comparator 78 features a hysteresis width of some tens millivolts, that substantially doubles the absolute value of the comparison threshold VTH1. The non^¬ linear amplifier stage 77 amplifies the low signals so that they can properly exceed the offset mistakes of the comparators that follow, and compress the strongest signals in order to avoid any saturation problems. The non-linear amplifier stage 77 includes an operational amplifier 101, set as an inverting amplifier, receiving a signal through the resistor 105 that is connected between the output of 80 and the inverting input of the operational amplifier 101. A feedback network of the operational amplifier 101 includes three parallel connected paths: the first path include a resistor 102; the second path includes a diode 103 having the anode and the cathode connected to the output and to the inverting input, respectively, of the operational amplifier 101; and the third path includes series connected diode 104 and resistor 106. More specifically, the diode 104 has the cathode connected to the output of the operational amplifier and the anode connected to one of the terminals of the resistor 106. The feedback network further includes a resistor 107 and a capacitor 108 that are series connected between the anode of the diode 104 and the ground .

The operation of the digital pulse shaper processing stage 52 of figure 9 is described with reference to figure 14 and 15, showing waveforms relevant to conditions in which the received useful signal is weak and strong, respectively. In use, the rectified filtered inverted signal VFBN features a series of negative pulses that are followed each by a respective "train" of transient amplitude decreasing oscillations, that are undesired but unavoidable, since they are caused by the rather steep filtering of the low- pass filters 56 of the receiver stage 51 (figure 6) . The high-pass filter 75 filters the signal VFBN to remove possible low-frequency noise components. The high-pass filter transforms each pulse of the signal VFBN in a pair of two pulses, in succession, having opposed polarities and followed by a relevant train of oscillations.

When the amplitude of signal VFB is sufficiently small, that is it cannot cause the diodes 103 and 104 to conduct, the amplifier stage 77 inverts the phase of and linearly amplifies the signal VFAB by an amplification factor defined by ratio between the resistances of resistors 102 and 105. As a consequence, the signal VAN (figure 14) includes two pulses in succession, the first one having negative polarity, the second one having positive polarity, and transient oscillations trains follow. This amplifying factor is such that the smaller receivable amplitude values of the signal VFB are caused to be sufficiently greater than the offset errors of comparators 78 and 86, so that the latter can work in an optimal way. Under these conditions, the comparator 78 can properly reconstruct the baseband envelope VIB, and the amplitude decreasing oscillations trains cannot cause commutations able to generate unwanted pulses.

As the amplitude values of the signal VFB increase, the forward conduction of diode 104 takes place at the positive peaks of the signal VFAB and the third path of the feedback network is closed through the resistor 106, so decreasing the amplification factor to, for instance, one tenth of the previous value. In this way, it is possible to preclude any possible inconveniences that might be due to a sudden saturation of the operational amplifier 101. Moreover, when the diode 104 is caused to conduct, the capacitor 108 charges, with a negative charge, through resistor 107 so that, when the useful pulse of the signal VFB ends, a proper current flows for a certain period of time from the capacitor 108 to the inverting input of the operational amplifier 101. As a result, the level of the signal VAN is pushed to higher values for a certain period of time when transient trains of signal VFB are present. However, as soon as the potential of the signal VAN reaches the threshold voltage of the diode 103, the latter enters into conduction causing a strong reduction or a substantial zeroing, of the amplifying factor of the amplifier stage 77, so preventing the signal VAN to further increase. As a consequence, the oscillations trains that follow each pulse of the signal VFB are substantially cancelled. In this case, too, the comparator 78 properly reconstructs the baseband envelope VIB without any unwanted pulses be generated by the oscillations trains.

The amplifier stage 76 comprises an operational amplifier 80, set as an inverting amplifier, with a feedback network including a resistor 81 connected between the inverting input and the output of the operational amplifier 80, and a n-channel FET 82 the source and drain terminals of which are parallel connected to the resistor 81. The operational amplifier 80 is fed by a dual voltage and its non-inverting input is connected to ground. The gate polarizing network includes a pair of resistors 83 and 85 having the same ohmic value of resistance R. The resistor 83 is connected between the gate of the FET 82 and the ground connected non-inverting input of the operational amplifier 80, while the resistor 85 is arranged between the gate of the FET 82 and the output of the operational amplifier 80. The amplifier stage 76 further includes a current sink 84, connected between the gate of the FET 82 and the ground, and controlled by the control signal VC . The current sink 84 is adapted to absorb a current I having an intensity that decreases when the amplitude of the control signal VC increases. More specifically, the current I has a maximum value 10 when the control signal VC is null, and its value progressively decreases until reaching zero when the amplitude of the control signal VC increases until reaching its maximum. Since the resistances of the two resistors 83 and 85 have the same ohmic value R, and the voltage between the inverting and non-inverting inputs of the operational amplifier 80 is substantially null, the FET 82 features a gate-source voltage that is substantially one half of the drain-source voltage, and consequently a channel resistance that is linear, at least in the "triode region", as can be easily demonstrated. In other words, the ohmic value R and the maximum value 10 of the current I are properly chosen in order to obtain the following result: when the control signal VC is null, the maximum value 10 of the current I causes a voltage drop across the resistors 83 and 85, so that the gate-source voltage takes a value that is sufficiently negative to cause the cut-off of the FET 82. In this situation the variable gain amplifier stage 76 features the maximum gain. When the amplitude of the control signal VC increases, the gate voltage consequently increases, causing the FET 82 gradually turning on and the relative channel resistance gradually decreasing. As a consequence, the gain of the amplifier stage 76 gradually decreases. Since, thanks to the action of the operational amplifier 80, the source voltage of the FET 82 can be considered grounded, the gate can be driven as provided for, and part of the control signal VC has not to be infused with the signal to be processed, contrary to what would take place in case that the resistor 83 is not grounded but connected to the source of the FET 82 and to the inverting input of the operational amplifier 80.

The control section 79 includes a comparator 86 for providing a pulse signal VP on the basis of a comparison between the processed input signal VAN and a second comparison threshold VTH2. The second comparison threshold VTH2 is determined as a fraction of the threshold VTH1 - typically 1/2 or 1/3 of VTH1 - by means of a voltage divider 87 fed by the voltage of VTH1. The control section 79 includes a low-pass filter 88 providing a signal VPM indicative of the average value of the pulse signal VP, and a comparator-amplifier 89 that is followed by a diode 90, providing the control signal VC on the basis of a comparison between the signal VPM and a third comparison threshold VTH3, provided by a proper generator arranged between the inverting input of the comparator-amplifier 89 and the ground.

The low-pass filter 88 is performed, for instance, by means of a first-order filter having a time constant (e.g. 100 ms) that is relatively greater with respect to the useful signals. The assembly including the comparator-amplifier 89, the third comparison threshold VTH3, the diode 90 and the current sink 84 is embodied, for example, by a single BJT transistor operating in a common-emitter configuration, where the threshold base-emitter voltage defines the third comparison threshold VTH3.

The control section 79 further includes an inhibitor circuit 91 that can inhibit at least partially the automatic sensitivity control circuits. The inhibitor circuit 91 includes an AND gate 92 having the impulsive signal VP as a first input, the output (VIB) of the comparator with hysteresis 78, through a NOT gate 93, as a second input, and the output connected to the low-pass filter 88. The logic AND and NOT gates 92 and 93 are embodied, for instance, by means of simple bipolar transistor stages operating at proper configurations. The operation of the circuits of the control section 79 depends on a particular feature of the useful signal, featuring a duty-cycle (e.g. about 1%) that is by far lower than the duty-cycle of the ground/environmental noise. As a consequence, the contribution of the useful signal to the duty-cycle of the signal VP is small as compared to the contribution provided by the noise. Moreover, if the useful signal is strong enough to cause commutation of the comparator 78, the inhibitor circuit 91 has the effect that such useful signal cannot provide any contribution to the optical sensitivity reduction. Furthermore the noise, that in view of its generally irregular amplitude distribution very likely includes components having an amplitude able to induce commutation of comparator 86, but not of comparator 78, can cause the control signal VC to increase and, consequently, the gain of the amplifier stage 76 to decrease. This takes place until the peak amplitude of the signal VAN that is due to noise, slightly overcomes the second threshold VTH2 without reaching the threshold VTH1, so carrying out a proper automatic control of the optical sensitivity of the base station 8, in order to conform at the best with the intensity of the received noise, and properly reconstruct the useful signal, of course when the latter as received has a sufficient amplitude.

According to the above description, the circuits of the control section 79 perform an automatic sensitivity by automatically reducing the receiving sensitivity on the basis of features of the processed input signal VAN, more specifically by varying the gain of the amplifier stage 76, wherein the ground/environmental noise is not amplified enough to overcome threshold VTH1, and so generation of spurious logical signals in the baseband envelope VIB is prevented. If the inhibitor circuit 91, that prevents the automatic sensitivity control on the basis of features of the output signal VIB, were not present, the useful signal, too, could cause a sensitivity reduction (decreasing the gain of the amplifier stage 76), and such reduction could in its turn cause a mistaken reconstruction of the baseband envelope of the received signal, mainly when the useful signal as received is weak. However, as stated above, the duty-cycle of the useful signal is normally by far lower with respect to the noise duty-cycle. So, the useful signal would cause a very small sensitivity reduction, not preventing it from overcoming the threshold VTH1 and the baseband envelope be properly reconstructed. As a consequence, a further - not shown - embodiment of the present invention, the control section 79 might not include the inhibitor circuit 91, and the pulse signal VP can be directly sent to the low-pass filter 88.

A different embodiment is shown in figure 10, where the same or corresponding elements that are present in figure 9 are marked with the same reference numbers and letters, and where the non-linear amplifier stage 77 is shown, for the sake of simplicity, as a single block. The inhibitor circuit 91 further includes a resistor 94, connected between the output of the comparator 86 and the input of the low-pass filter 88, and another resistor 95 connected between the output of the AND gate 92 and the input of the low-pass filter 88. The ratio between the resistances of the resistors 94 and 95 is properly chosen so that just a partial reduction of the gain of the amplifier stage 76 takes place when the peak amplitude of the signal VAN overcomes the threshold VTH1. It is pointed out that particular noise signals might be received, such noise signals featuring a substantially regular amplitude distribution (i.e. without any intermediate amplitude values, in substance) , and amplitude values that are sufficiently high so as to always cause commutation of both comparators 78 and 86. Even though such kind of noise signals are not so frequent, they might suddenly appear. The above-mentioned partial inhibition allows to avoid that, if and when such noise signals do appear, the desired gain reduction of the amplifier stage 76 does not take place, since this would prevent the proper reconstruction of the baseband envelope of the received signal.

Another embodiment is shown in figure 11, where the same or corresponding elements that are present in figures 9 and/or 10 are marked with the same reference numbers and letters, and the digital pulse shaper processing stage 52 includes the following different features with respect to the embodiment of figure 10.

The digital pulse shaper processing stage 52 does not include the variable gain amplifier stage 76 of figure 10, and the high-pass filter 75 is directly connected to the input of the non-linear amplifier stage 77. Figures 16 and 17 show the main waveforms relevant to conditions in which the received useful signal is weak and strong, respectively. The digital pulse shaper processing stage 52 includes a capacitor 96 connected between the output of the non-linear amplifier stage 77 and the inverting input of the comparator 78, in order to de-couple the continuous signal components between the non-linear amplifier stage 77 and the comparators 78 and 86, and a capacitor 97 connected between the inverting input of the comparator 78 and the inverting input of the comparator 86. The capacitors 96 and 97 feature a negligible electrical reactance to the frequencies of the signal VAN provided by the non-linear amplifier stage 77. As a consequence, signals VANS1 and VANS2 that are present at the inverting inputs of comparators 78 and 86, respectively, include a component corresponding to the signal VAN without substantial amplitude variations. The digital pulse shaper processing stage 52 further includes a voltage divider with a resistor 98 connected between the inverting inputs of the comparators 78 and 86, and a resistor 99 series connected to the resistor 98, between the inverting input of the comparator 86 and the ground. In the control section 79, the current sink 84 of figure 10 is replaced by a current source 100, that is connected between the inverting input of the comparator 78 and the ground, and is controlled by the control signal VC to produce an adjustable voltage offset that is added to the signal VAN, so generating the signals VANS1 and VANS2. The comparators 78 and 86 have the non-inverting inputs directly connected to the ground, that is the comparison thresholds VTH1 and VTH2 are both equal to zero.

The current I generated by the current source 100 can substantially circulate in the voltage divider 98, 99, only, so generating a corresponding voltage drop across the voltage divider 98, 99 that charges the capacitors 96 and 97, since the current I, modulated by the control signal VC, is constant or slowly variable owing to the filtering action of the low-pass filter 88. The voltage drop across the whole voltage divider 98, 99 defines a first voltage offset VOFF1 that is added to the amplified signal VAN and provides the signal VANS1 at the inverting input of the comparator 78. The voltage drop across the resistor 99 defines a second voltage offset VOFF2 that is a fraction of the first voltage offset VOFF1 and that is added to the amplified signal VAN, the latter having the original amplitude, thanks to the capacitor 97 opposing negligible reactance, and provides the signal VANS2 at the inverting input of the comparator 86. For instance, the resistors 98 and 99 can have resistances with the same ohmic value, so that the voltage offset VOFF2 is the half of the voltage offset VOFF1.

The greater is the current I, the greater is the voltage offset VOFF1, and the greater is the amplitude of the amplified signal VAN that is needed to cause the commutation of the comparator 78 and the generation of the signal VIB. In other words, the greater is the current I, the less is the optical sensitivity of the base station 8. The reduction of the optical sensitivity aims to prevent the comparator 78 from outputting spurious signals caused by ground/environmental noise, while allowing the useful signals, having greater amplitude with respect to the noise, be properly reconstructed.

The regulation of the current I is explained hereinbelow making reference to the particular, extreme circumstance wherein the effect of the inhibitor circuit 91 is null, thet is in case that the resistor 94 has zero resistance (short circuit) and the resistor 95 has infinite resistance (oper circuit) . The VPM signal at the output of the low- pass filter 88 is an analog signal that can slowly vary and is proportional to the average value of the pulse signal VP, that is it is proportional to the time percentage when the sum of the signal VAN with the offset voltage VOFF2 is lower than the threshold VTH2 (that is it is negative) . In other words, the signal VPM is proportional to the time percentage when the module of the negative peaks of the amplified signal VAN overcome the voltage offset VOFF2. When the signal VPM overcomes the threshold VTH3 the feedback loop including the control section 79 controlling the current source 100, closes, so that the voltage offset VOFF1 is generated. The loop gain of the control section 79 has a value such that the voltage offset VOFF2 approximately corresponds to the negative peaks module of the signal VAN, and consequently the voltage offset VOFF1 approximately corresponds to twice the negative peaks module of the signal VAN, in the hypothesis that the resistance of each of the resistors 98 and 99 be the same. In this way, the component of the signal VAN that is due to ground/environmental noise does not cause the comparator 78 to improperly commutate, while the component of the signal VAN that is due to the useful signal, if it is sufficiently stronger than the noise component, cause the comparator 78 to properly commutate and, thanks to its low duty-cycle, gives a limited contribution to the increase of the control signal VC and the consequent sensitivity reduction.

By employing resistors 94 and 95 having resistances with ohmic finite, non-zero values, the inhibitor circuit 91 has the same effect as the one that has been described above with reference to the embodiments of figure 9 and 10, that is it allows to limit the contribution given by the useful signal to the sensitivity reduction, meanwhile preventing the sudden interference of particular noise signals from improperly intruding and altering the needed sensitivity reduction.

The embodiment according to figure 11 operates as the embodiments of figure 9 and 10 do operate, and in view of the presence of a minor number of electronic components, allows the further advantage to get a digital pulse shaper processing stage 52 having reduced dimensions.

Another embodiment is shown in figure 12, where the same or corresponding elements that are present in figure 9 are marked with the same reference numbers and letters, and the inhibitor circuit 91 includes a further low-pass filter 109 to provide a signal VIBM that is indicative of, or proportional to, the average value of the baseband envelope VIB. The inhibitor circuit 91 also include a further comparator 110 comparing the signal VIBM with a fourth, positive comparison threshold VTH4, generated between the non-inverting input of the comparator 110 and the ground, and an AND gate 111 having the inputs connected to receive the baseband envelope VIB and the output of the comparator 110, and connected at its output to the NOT gate 93. In some cases, the low-pass filter 109 can advantageously have asymmetrical operating features, that is feature a rising time constant lower that the falling time constant. The assembly including the comparator 110 and the fourth, positive comparison threshold VTH4 is embodied, for example, by a single BJT transistor operating in a common- emitter configuration, where the threshold base-emitter voltage (typical value: 0.6 V) defines the fourth, positive comparison threshold VTH4. The logic function of the network including the gates 111, 92 and 93 are performed, for instance, by simple BJT transistor circuits, having a proper configuration.

The operation of the digital pulse shaper processing stage 52 according to the embodiment of figure 12 is hereinbelow described. When, in use, a useful signal is received, owing to the very low duty-cycle of the latter, the signal VIBM does not overcome the threshold VTH4 and, as a consequence, the output of the comparator 110 is at high logic level, while the AND gate 111 "opens". This allows the baseband envelope VIB, if it is present, to "close" the AND gate 92, so preventing the pulse signal VP from being transmitted to the low-pass filter 88 and, as a consequence, preventing the useful signal from causing an unwanted - even though small - sensitivity reduction. The noise signals, if any, having generally a rather irregular amplitude distribution, very likely include amplitude values able to cause the comparator 86, but not the comparator 78, to commutate, and consequently obtain, also thanks to a properly high loop gain, a correct sensitivity reduction. In case that particular optical noise signals suddenly interfere, such particular noise signal featuring a substantially regular amplitude distribution and including amplitude values high enough to cause both comparators 78 and 86 to commutate, it is very likely that such noise signals have a duty-cycle that is higher than the useful signal duty-cycle, so that the output of the comparator 110 is brought to a low logic level. Such low logic output of the comparator 110 "close" the AND gate 111 and consequently the AND gate 92 is "opened", so allowing the pulse signal VP to be transmitted to the low-pass filter 88, and the proper working of the automatic sensitivity control. So, as noise signals having a regular amplitude distribution do appear, the baseband envelope VIB is affected by spurious signals just for a short time interval, that is substantially equal to the sum of the response times of the low-pass filter 109 and of the feedback loop of the control section 79 (a typical value is of about some hundreds of milliseconds) . A similar behaviour takes place in the embodiments according to figures 10 and 11, too.

The parameters of the digital pulse shaper processing stage 52 according to the embodiments of figures 9 to 12 can be properly chosen so that the automatic sensitivity control circuits of the control section 79 be always operating, even when no environmental noise is present. More specifically, the working of such automatic sensitivity control circuits of the control section 79 can be caused by a ground noise component of the signal VAN that is mostly due to noise generated by the photodiodes 41 and the amplifier means 42. So, the optical sensitivity of the digital pulse shaper processing stage 52 is always kept as high as possible, and does not rely upon possible variations of the circuital parameters and/or upon the specific noise in the preceding stages, such specific noise being generally determined by the specific carrier frequency to which the apparatus is tuned.

According to an additional embodiment, not shown in the drawings, the digital pulse shaper processing stage 52 differs from the one of figure 11 in that the inhibitor circuit 91 has the features show in figure 12, that is it does not include the resistors 94 and 95 and comprises instead the low-pass filter 109, the comparator 110, a generator of the threshold VTH4 and the AND gate 111.

According to another, different, not shown embodiment of the invention all the electronics and circuits of the remote transceiver 12 of the base station 8 are fed by a not dual voltage. In this case, the connections to ground shown in the figures 7, 9, 10, 11 and 12 are replaced by connections to a fixed reference voltage having a lower value with respect to the supply voltage, for instance half the supply voltage.

Claims

1. System for checking position and/or dimensions of a mechanical piece (2) with

· at least one probe (6) comprising a detecting device (9a) and a movable transceiver (10) with a transmission section (15) that is connected to the detecting device (9a) and adapted to wirelessly transmit a signal indicative of the status of said at least one probe (6), the signal having a carrier of a known frequency, and

• a remote transceiver (12) with a receiving section (34) with

• receiving devices (41) adapted to wirelessly receive said signal, and

· processing means adapted to process the received signal to obtain an output signal (VIB) and including a receiver stage (51) of the frequency conversion type adapted to provide a processed input signal (VFB) ,

characterized in that the receiver stage (51) is of the homodyne type.

2. System according to claim 1, wherein the receiver stage (51) includes a local oscillator (53) adapted to generate a first reference signal (VOl) that has a nominal frequency equal to the one of the carrier of the received signal (VAB) , a phase shifter (54) fed by the local oscillator (53) for generating a second reference signal (V02) in quadrature with the first reference signal (VOl), two non-linear mixers (55) adapted to separately mix said first reference signal (VOl) and second reference signal (V02) with the received signal (VAB), two low-pass filters (56) each one connected to the output of a respective non^¬ linear mixer (55) for providing two respective baseband signals (VBB1, VBB2), and a detecting circuit (57) adapted to receive the two baseband signals (VBB1,VBB2) and to provide said processed input signal as a rectified filtered signal (VFB)

3. System according to claim 2, wherein the detecting circuit (57) includes two active full-wave rectifiers (58) that receive the two baseband signals (VBB1, VBB2), respectively, and have outputs connected to a single output (60) to provide said rectified filtered signal (VFB) .

4. System according to any one of the preceding claims, wherein the processing means include a digital pulse shaper processing stage (52) and is connected to the output of the receiver stage (51) to provide the output signal as baseband envelope (VIB) of the received signal (VAB) .

5. System according to any one of the preceding claims, wherein the processing means further include

• a comparison section (69; 78) that is connected to the receiver stage (51) and adapted to provide said output signal (VIB) in response to the results of comparisons carried out between the processed input signal

(VFB, VFAB; VAN) and reference signals (VTHB ; VTH1 ) , and

• a control section (68;79) connected to the receiver stage (51) and including

• automatic sensitivity control circuits adapted to automatically reduce the receiving sensitivity of the receiving section of the remote transceiver (12) depending on the characteristics of the processed input signal (VFB, VFAB; VAN) , and

• an inhibitor circuit (74; 91) adapted to inhibit at least partially said automatic sensitivity control circuits depending on characteristics of the output signal (VIB) .

6. System according to claim 5, wherein the control section includes a threshold generating and control block (68) adapted to generate and define the reference signals (VTHB) with a threshold generating circuit (72), said automatic sensitivity control circuits being included in the threshold generating and control block (68) and comprising a discriminator circuit (71) connected to the threshold generating circuit (72) in order to generate, as a function of the processed input signal (VFB,VFAB), a signal (VN) indicative of noise signals overlapping the signal transmitted by the movable transceiver (10) of the probe (6), and to automatically adjust the reference signals (VTHB) , said inhibitor circuit (74) being connected to the discriminator circuit (71) and to the threshold generating circuit (72) .

7. System according to claim 6, wherein the threshold generating and control block (68) comprises a detector circuit (70) for generating, as a function of the processed input signal (VFB,VFAB), a signal (VU) indicative of peak values of the processed input signal (VFB,VFAB), the detector circuit (70) being connected to the threshold generating circuit (72) to dynamically and temporarily adjust said reference signals (VTHB) .

8. System according to claim 5, wherein the processing means include a variable gain amplifier stage (76), the control section including control circuit (79) adapted to automatically reduce said receiving sensitivity by varying the gain of the variable gain amplifier stage (76) .

9. System according to any one of the preceding claims, wherein said carrier of a known frequency is a pulse carrier.

10. Method for checking position and/or dimensions of a mechanical piece (2), by means of a system including at least one probe (6) with a detecting device (9a) and a movable transceiver (10) with a receiving section (14) and a transmission section (15), the transmission section (15) being connected to the detecting device (9a) to transmit a modulated signal indicative of the status of said at least one probe (6), the modulated signal having a carrier, and a remote transceiver (12) with a receiving section (34) and a transmission section (35) , the method being characterized in that the modulated signal is transmitted through one communication channel out of a plurality of communication channels, each communication channel being defined by the value of one selected frequency of the carrier.

11. Method according to claim 10, wherein at least one of the said receiving sections (14, 34) of said movable transceiver (10) and of said remote transceiver (12) includes a band pass filter (18, 45), the method including tuning the band pass filter (18, 45) on said selected frequency of the carrier.

12. Method according to claim 11, wherein the band pass filter (18) of the receiving section (34) of the remote transceiver (12) includes a plurality of mutually connected reactive components (25) , and a plurality of corresponding switches (26), each of said switches (25) being coupled to one of said reactive components (25) , the method including the step of controlling the opening and closure of the switches (26) to select said communication channel by setting a band central frequency of the band pass filter (18) .

13. Method according to claim 12, wherein said reactive components (25) are parallel-connected to one another, each of said switches (25) being arranged in series with one of said reactive components (25) .

14. Method according to claim 12 or claim 13, wherein said reactive components (25) include a plurality of inductors, said band pass filter (18) further including a fine tuning circuit (27), connected to said plurality of inductors and providing an adjustable capacitance.

15. Method according to any one of claims 10 to 14, wherein the receiving section (14) of the movable transceiver (10) includes receiving devices (17) and variable gain amplifier means (19), the method comprising the further steps of:

• wirelessly transmitting via the transmission section (35) of the remote transceiver (12) a control signal comprising a message,

· receiving said control signal by means of the receiving devices (17) of the movable transceiver (10) of the probe ( 6) ,

• processing the received signal in the movable transceiver (10) of the probe (6) to obtain a processed signal (VD) comprising said message, the processing including an amplification by means of the variable gain amplifier means (19), and

• automatically controlling the gain of the variable gain amplifier means (19) depending on said processed signal (VD) ,

said control signal that is wirelessly transmitted by the transmission section (35) of the remote transceiver (12) comprising a preamble of sufficient length so that said step of automatically controlling the gain of the variable gain amplifier means (19) depending on said processed signal (VD) can be performed in a substantially continuous way .

16. Method according to claim 15, by means of said system wherein the receiving section (14) of the movable transceiver (10) further includes a comparing section, the step of processing the received signal in the movable transceiver (10) of the probe (6) to obtain said processed signal (VD) comprising a comparison between an amplified signal (VA) outputted from the variable gain amplifier means (19) and an adjustable reference signal (VTH) , the method comprising the further step of automatically adjusting the reference signal (VTH) depending on said processed signal (VD) .

17. Method according to claim 16, by means of said system wherein the movable transceiver (10) includes a digital adjusting device (16) adapted to provide said steps of automatically controlling the gain of the variable gain amplifier means (19) and adjusting the reference signal (VTH) depending on said processed signal (VD) .

18. Method according to any one of claims 15 to 17, wherein the message contained in the control signal transmitted by the transmission section (35) of the remote transceiver (12) is of the binary type.

19. Method according to any one of claims 15 to 18, wherein the modulated signal transmitted by the transmission section (15) of the movable transceiver (10) and the control signal wirelessly transmitted by the transmission section (35) of the remote transceiver (12) are optical signals in the infrared band.

20. Method according to any one of claims 15 to 19, by means of said system wherein said at least one probe (6) includes a power supply unit (32) and a microcontroller (11) that are connected to the movable transceiver (10), the microcontroller (11) being adapted to estimate, at least on the basis of the automatic control of the gain of the variable gain amplifier means (19), the transmission quality, and to consequently regulate the power supply of the transmission section (15) .

21. Method for checking position and/or dimensions of a mechanical piece (2), by means of a system including at least one probe (6) with a detecting device (9a) and a movable transceiver (10) with a receiving section (14) and a transmission section (15), the transmission section (15) being connected to the detecting device (9a) and including optical devices (28) adapted to transmit a modulated optical signal indicative of the status of said at least one probe (6), the modulated optical signal having a carrier, and a remote transceiver (12) with a receiving section (34) and a transmission section (35), the method including the steps of:

• modulating said carrier with a binary message,

• suitably shaping the bits of said binary message in such a way to attenuate the spectral components that are far form the carrier frequency.

22. Method according to claim 21, wherein the transmission section (15) of the movable transceiver (10) includes at least one infrared LED (28) adapted to emit infrared beams, the intensity of the infrared beams emitted by said at least one infrared LED (28) being dynamically regulated to perform said step of suitably shaping the bits of said binary message.

23. Method according to claim 22, wherein said at least one infrared LED (28) is driven by means of a drive voltage featuring an envelope (VLs, VLq) having dynamically regulable amplitude.

24. Method according to claim 23, wherein said step of suitably shaping the bits of said binary message includes dynamically regulating the amplitude of the envelope (VLs) of said drive voltage according to a sinusoid portion.

25. Method according to any one of claim 21 to 24, wherein the transmission section (35) of the remote transceiver (12) includes at least one infrared LED (38) adapted to transmit a modulated optical signal having a carrier, wherein the amplitude of the envelope of a drive voltage of said at least one LED 38 is dynamically regulated in order to attenuate the spectral components of the modulated optical signal far from the carrier frequency .

26. Method according to any one of claims 21 to 25, wherein the modulated optical signal is transmitted through one communication channel out of a plurality of communication channels, each communication channel being defined by the value of one selected frequency of the carrier .

27. Method according to any one of claims 10 to 26, wherein said carrier is a pulse carrier.