WO2010047455A1 - Systeme onduleur et son procede d'exploitation - Google Patents

Systeme onduleur et son procede d'exploitation Download PDF

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Publication number
WO2010047455A1
WO2010047455A1 PCT/KR2009/002890 KR2009002890W WO2010047455A1 WO 2010047455 A1 WO2010047455 A1 WO 2010047455A1 KR 2009002890 W KR2009002890 W KR 2009002890W WO 2010047455 A1 WO2010047455 A1 WO 2010047455A1
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Prior art keywords
switch
controller
diode
signal
inverter system
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PCT/KR2009/002890
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English (en)
Korean (ko)
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정동열
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주식회사 디엠비테크놀로지
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Publication of WO2010047455A1 publication Critical patent/WO2010047455A1/fr

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/42—Conversion of DC power input into AC power output without possibility of reversal
    • H02M7/44—Conversion of DC power input into AC power output without possibility of reversal by static converters
    • H02M7/48—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/538—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a push-pull configuration
    • H02M7/53803—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a push-pull configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/42—Conversion of DC power input into AC power output without possibility of reversal
    • H02M7/44—Conversion of DC power input into AC power output without possibility of reversal by static converters
    • H02M7/48—Conversion of DC power input into AC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • G—PHYSICS
    • G02—OPTICS
    • G02F—OPTICAL DEVICES OR ARRANGEMENTS FOR THE CONTROL OF LIGHT BY MODIFICATION OF THE OPTICAL PROPERTIES OF THE MEDIA OF THE ELEMENTS INVOLVED THEREIN; NON-LINEAR OPTICS; FREQUENCY-CHANGING OF LIGHT; OPTICAL LOGIC ELEMENTS; OPTICAL ANALOGUE/DIGITAL CONVERTERS
    • G02F1/00—Devices or arrangements for the control of the intensity, colour, phase, polarisation or direction of light arriving from an independent light source, e.g. switching, gating or modulating; Non-linear optics
    • G02F1/29—Devices or arrangements for the control of the intensity, colour, phase, polarisation or direction of light arriving from an independent light source, e.g. switching, gating or modulating; Non-linear optics for the control of the position or the direction of light beams, i.e. deflection
    • G02F1/33—Acousto-optical deflection devices
    • G02F1/335—Acousto-optical deflection devices having an optical waveguide structure
    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters
    • H05B41/282—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters with semiconductor devices
    • H05B41/2825—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters with semiconductor devices by means of a bridge converter in the final stage
    • H05B41/2828—Circuit arrangements in which the lamp is fed by power derived from DC by means of a converter, e.g. by high-voltage DC using static converters with semiconductor devices by means of a bridge converter in the final stage using control circuits for the switching elements
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B20/00—Energy efficient lighting technologies, e.g. halogen lamps or gas discharge lamps

Definitions

  • the present invention relates to an inverter system of a display device and a method of operating the same. More particularly, in an inverter system for generating an alternating voltage, a control is performed to reduce power consumption generated by a freewheeling current and improve efficiency. An inverter system and a method of operating the same.
  • LCD Liquid Crystal Display
  • LCD Liquid Crystal Display
  • a cold cathode tube (CCFL) is used as the back light.
  • the cold cathode tube is a fluorescent lamp that is turned on at a low temperature without heating the filament.
  • the cold cathode tube includes a glass tube containing a predetermined amount of a mixed gas such as mercury, argon, and neon, and an electrode disposed at both ends of the glass tube. Electrons are emitted by the high-voltage electric field applied to the two electrodes, mercury is excited by the emitted electrons, and ultraviolet rays are emitted. The emitted ultraviolet rays collide with the phosphor on the inner wall of the glass tube to emit visible light.
  • a cold cathode tube requires an AC voltage of several hundred V or more, and an inverter system is required.
  • Such an inverter system includes a half bridge, a full bridge, and a push pull method.
  • the half-bridge or push-pull method uses only two power semiconductors as switch elements, so it is used for LCD monitors with lower output power than large LCDs such as TVs.
  • FIG. 1 is a diagram illustrating a configuration of an inverter system implemented by a general half-bridge method, assuming an inverter system when driving one lamp.
  • a typical half-bridge inverter system uses a pulse width modulation (PWM) control method to control the lamp current to a desired value while converting an input DC voltage into an alternating voltage of several hundred V to drive a CCFL lamp.
  • PWM pulse width modulation
  • the inverter system compares the smoothed voltage VF with the reference voltage Vref to output an error voltage Verr and an error amplifier output Verr by comparing a triangular wave with a lamp current.
  • PWM comparator 12 for outputting the PWMO signal for controlling the signal
  • oscillator 30 for generating the triangular wave for PWM comparison
  • Logic Controller 14 for making the PWMO signal to drive the switch
  • Switch A switch block 16 for outputting an input voltage VIN or ground in response to an on / off control signal
  • an AC amplifier 18 for receiving an output OUT of the switch block 16 as an input and amplifying the signal to a bipolar voltage for driving a lamp
  • a CCFL lamp load 20 for outputting the PWMO signal for controlling the signal
  • oscillator 30 for generating the triangular wave for PWM comparison
  • Logic Controller 14 for making the PWMO signal to drive the switch
  • Switch A switch block 16 for outputting an input voltage VIN or ground in response to an on / off control signal
  • the gate driver 15 for amplifying the output signals DrvH and DrvL of the logic controller 14 and delivering them to the switch block 16, and the rectifier 22 for converting the lamp current into a voltage and smoothing the AC voltage to a DC voltage. It includes.
  • FIG. 2 is a circuit diagram showing a power stage circuit in FIG. 1
  • FIG. 2 (a) is a diagram showing the configuration of a switch block and an AC amplifier
  • FIG. 2 (b) is a circuit diagram of FIG. Equivalent circuit equivalently converted to the primary side of a transformer
  • 3 is a diagram illustrating waveforms of voltage and current according to the operation of the switch block in FIG. 2.
  • the two switch elements M1 and M2 constituting the switch block may be provided with a semiconductor such as a MOSFET, and as a result, the parasitic capacitances C1 and C2 may be included in a circuit as shown in FIG. do.
  • Cb the primary capacitor of the transformer
  • Cb the primary capacitor of the transformer
  • the load Ro connected to the secondary side is a transformer turn ratio. Converted to the primary side as a function of.
  • the equivalent inductance Leq is approximately equal to the leakage inductance of the transformer primary side
  • the equivalent capacitance Ceq is equal to the secondary side capacitance capacitance Cr multiplied by the squared n 2 of the turn ratio
  • the equivalent load resistance Req is the secondary side load resistance Ro Is divided by n 2 .
  • the inductor current Ip becomes zero and thus the body diode D2 of the second switch element is turned off.
  • the second switch element M2 is not yet turned on, and thus resonates due to parasitic capacitances C1 and C2 and the inductor Leq of each switch element.
  • Equivalent capacitance (Ceq) is relatively large compared to the parasitic capacitance (C1, C2), so the resonant frequency is much higher frequency than the resonance caused by the inductor (Leq) and equivalent capacitance (Ceq) and also equivalent load (Req) Since the resonance is small, the resonance disappears quickly.
  • the body diodes D1 and D2 of the switch element are automatically turned on during a freewheeling period in which the inductor current Ip direction is maintained as it is. Since the current flows to the body diodes D1 and D2 and consumes power at the body diode side, the efficiency of the inverter system is lowered and the heat generation of the switch element is increased.
  • On-resistance of MOSFET switches used in inverter systems, such as LCD monitors, is typically on the order of tens of milliohms, and the on-voltage of the body diode is about 0.5V.
  • the maximum current flowing through the switch of the inverter increases in proportion to the size of the LCD panel and the number of CCFL lamps used. At this time, when comparing the conduction loss caused by the MOSFET switch-on resistance with the diode loss, which is the power consumed when the diode is turned on, the power consumed by the diode cannot be ignored.
  • the maximum current of a switch with an on resistance of 20 m ⁇ and a body diode on voltage of 0.5 V is 7 A and freewheeling with the diode at this current
  • the maximum instantaneous power consumption by the switch on resistance is 0.98 W.
  • the diode is very large, 3.5W.
  • the present invention is to improve the above-mentioned problems, by controlling the freewheeling current to flow to the switch during the period of flow to the body diode of the switch element to improve the power efficiency of the inverter system and reduce heat generation of the switch element to improve durability
  • An inverter system and a method of operating the same are provided.
  • the inverter system of the display device a period in which the freewheeling current flows by automatically turning on the body diode of each switch while each switch performs an inverting operation.
  • the operating method of the inverter system comprising a second step of the switch to perform an on / off operation to generate an AC voltage, and applying the generated AC voltage to the lamp to emit light. And a third step of generating a control signal for temporarily turning on each switch in a period during which the freewheeling current flows to the body diode of the switch, thereby improving the power efficiency of the inverter system.
  • FIG. 1 is a view showing the configuration of an inverter system implemented in a general half-bridge method
  • FIG. 2 is a circuit diagram showing a power stage circuit in FIG. 1;
  • FIG. 3 is a diagram illustrating waveforms of voltage and current according to a switch operation in FIG. 2;
  • FIG. 5 is a diagram showing the configuration of an inverter system implemented according to the first embodiment of the present invention.
  • FIG. 6 is an exemplary view showing the configuration of a switch controller in FIG. 5;
  • FIG. 7 shows an operating waveform of a circuit constructed in accordance with FIG. 6;
  • FIG. 8 is an exemplary view showing the configuration of a switch controller implemented according to a second embodiment of the present invention.
  • FIG. 9 is an exemplary diagram showing an internal configuration of a switch controller implemented according to the third embodiment of the present invention.
  • FIG. 10 is a diagram illustrating a diode freewheeling period in an inverter system according to the present invention.
  • FIG. 11 is a circuit diagram illustrating an example of implementing a variable pulse width generator in FIG. 10.
  • FIG. 4 is a diagram illustrating an operation concept of the present invention.
  • Inverter system is a method for reducing the power loss consumed by a diode in a general inverter control method, by turning on the corresponding switch element during the period in which the body diode is on, the inductor current (Ip) is not a diode switch Control to flow.
  • Ip inductor current
  • the second switch is turned on in accordance with the freewheeling period in which the body diode of the second switch is turned on so that the inductor current maintains the direction of the current, and the second switch in a similar manner.
  • the first switch is turned on in accordance with the freewheeling period during which the body diode of the first switch is turned on.
  • FIG. 5 is a diagram illustrating a configuration of an inverter system implemented according to the first embodiment of the present invention.
  • the smoothed voltage VF is compared with the reference voltage Vref.
  • Error amplifier 10 for outputting error voltage Verr
  • PWM comparator 12 for outputting signal PWMO for PWM control of lamp current by comparing output voltage Verr of error amplifier with triangle wave CT
  • An oscillator 30 for generating a triangular wave for the switch controller 50 and a switch controller 50 for generating a control signal for turning on a corresponding switch instead of a diode in the freewheeling period in addition to the general PWM duty control signal of the switch.
  • a gate driver 60 that amplifies the switch so that it can be driven by the output signal.
  • the CCFL lamp is received by receiving the switch block 16 which connects the output signal OUT to the input voltage VIN or the ground, and OUT which is the output signal of the switch block 16.
  • It is an inverter system which consists of an AC amplifier 18 which amplifies to AC of more than several hundred V to drive, a CCFL lamp 20 as a load, and a rectifier 22 which converts lamp current into voltage and smoothes AC voltage into DC voltage. .
  • the switch controller 50 has a signal corresponding to the freewheeling period so that the corresponding switches M1 and M2 can be turned on in accordance with the freewheeling period in which the body diodes D1 and D2 are turned on. It includes a diode controller (Diode On-Time Controller) 100 for outputting the logic controller 150 to make a separate signal for the PWM control of the switches.
  • Diode controller Diode On-Time Controller
  • the diode controller 100 output signal PWMAd corresponding to the period in which the body diode (D1, D2) of each switch element (M1, M2) constituting the switch block 16 is turned on You can output
  • the logic controller 14 of the general inverter system as shown in FIG. 1 receives the PWMO which is the output signal of the PWM comparator 12 and the charge / discharge control signal of the oscillator 30 to control the first switch for PWM control.
  • the logic controller 150 included in the switch controller 50 is a PWM control signal from the PWM comparator 12.
  • the PWMO and the charge / discharge control signal CK of the oscillation controller 32 and the control signal PWMAd corresponding to the freewheeling period are input from the diode controller 100 to perform PWM control and simultaneously switch the switch according to the freewheeling period.
  • the control signals DrvH and DrvN are controlled to be turned on and supplied to the gate driver 60.
  • the error amplifier 10 amplifies the difference between the output voltage VF and the reference voltage Vref of the rectifier 22 to output the output signal Verr so that the PWM duty can be increased or decreased, and the capacitor Cc is Stabilize the inverter system.
  • the error amplifier 10 is illustrated as a transconductance amplifier that outputs a difference in input voltage as an output current, but is not limited thereto, and may also be configured as a voltage amplifier.
  • Oscillator 30 is a circuit for making a triangular wave for PWM control is composed of an oscillation controller 32 and an oscillation capacitor (Cosc).
  • the oscillation controller 32 outputs the charge / discharge control signal CK, and the triangular wave is generated by charging or discharging the oscillation current source Iosc with the oscillation capacitor Cosc as this signal becomes high or low.
  • the gate driver 60 drives the gate-source voltage of the switch elements M1 and M2 constituting the switch block 16 to a sufficiently large voltage so that the switch-on resistance is reduced while simultaneously switching It is composed of a first gate driver (not shown) and a second gate driver (not shown) for driving with a large current to charge and discharge the input capacitors of the elements M1 and M2 sufficiently fast.
  • the first gate driver amplifies the first driving control signal DrvH of the logic controller 150 to output the GP signal
  • the second gate driver amplifies the second driving control signal DrvL of the logic controller 150 to GN. Output the signal.
  • the switch block 16 alternately connects the output signal OUT to the input voltage VIN and ground.
  • the switch block 16 includes a first switch element M1 located above and a second switch element M2 located below.
  • the switch element includes body diodes D1 and D2, respectively.
  • the first switch device M1 is described with an example in which a P-type MOSFET is applied, but is not limited thereto.
  • An N-type MOSFET may be used.
  • the first switch element M1 may be turned on / off according to the first gate driving signal GP of the gate driver 60, and when turned on, the first switch element M1 may connect OUT to an input voltage VIN.
  • the second switch element M2 operates on / off according to the second gate driving signal GN and, when on, connects OUT to ground. Therefore, OUT becomes a square wave AC voltage having a voltage of VIN or ground (0V).
  • the AC amplifier 18 amplifies the input signal into an AC signal of several hundred volts or more so as to drive the CCFL lamp 20, and more specifically, OUT, which is an output signal output from the square wave AC voltage of the switch block 16, is output.
  • the LC resonance converts the output to a sine wave output VAC with positive and negative voltages.
  • the AC amplifier may include a DC blocking capacitor Cb, a transformer T1 and a resonant capacitor Cr that amplify at a turn ratio of 1: n.
  • the DC blocking capacitor Cb In the DC blocking capacitor Cb, a large value capacitor is usually used for DC blocking. Therefore, when the symmetric PWM control method is used, the DC blocking capacitor Cb is charged with a DC voltage of VIN / 2. Accordingly, the input voltage Vx of the transformer T1 becomes an AC voltage having voltages of + VIN / 2 and -VIN / 2.
  • the rectifier 22 senses and rectifies the lamp current and performs a filtering operation to enable PWM control.
  • the rectifier 22 includes a sense resistor (Rs), a half-wave rectifier diode (D3), and a smoothing resistor ( Rf) and a capacitor Cf.
  • the rectifier 22 may be a circuit configuration in a variety of ways according to the designer as an example shown when one CCFL lamp 20 is provided, and also in the case of a plurality of lamps are connected in various ways depending on the designer The configuration of the circuit may vary.
  • FIG. 6 is an exemplary diagram showing a configuration of a switch controller in FIG. 5, and FIG. 7 is a diagram showing an operation waveform of a circuit constructed according to FIG. 6.
  • the diode controller 100 of the switch controller 50 includes at least one rising-edge-delay-circuit. 102, 108 and logic circuits 104, 106, 110.
  • the PWMO_Ax signal output from the first rising-edge-delay-circuit 102 and the PWMOB signal output as the PWMO passes through the inverter 104 and the charge / discharge of the oscillation controller are delayed by the PWMO signal by a freewheeling period.
  • the control signal CK is input to the NOR gate 106, a PWMOA signal corresponding to the freewheeling period can be obtained.
  • the second rising-edge-delay- that delays the PWMOA from the rising edge by a certain time, i.e., dead time (Td), should not occur at the same time even when the corresponding switching element is turned on instead of the diode during the freewheeling period.
  • Td dead time
  • the logic controller 150 is a diode in the PWM duty control signal PWMO for the PWM duty control from the PWM comparator 12 to implement an inverter system of high efficiency. Instead, the control signal PWMOAd corresponding to the freewheeling period and the charge / discharge control signal CK for generating a signal capable of driving the gate driver separately are received from the diode controller 100 to turn on the corresponding switch. . That is, when the charge / discharge control signal CK passes through the T-type flip-flop (TFF) 152, the frequency is 1/2 of the charge / discharge control signal CK, and at the oscillation maximum voltage VH of the oscillation waveform CT.
  • T-type flip-flop T-type flip-flop
  • the transitional complementary square wave signals VFD and BVFD can be obtained, and these signals can pass through a logic gate to output switch control signals for symmetric PWM control.
  • the PWM duty control signal PWMO and the square wave signal BVFD are input terminals of the NOR gate 154, a DrvP signal corresponding to the PWM duty control period of the first switch element M1 can be obtained.
  • the output signal PWMAd and the square wave signal VFD are input terminals of the NOR gate 156, the DrvPA signal corresponding to the freewheeling period of the body diode D1 of the first switch element M1 can be obtained.
  • DrvP signal and the DrvPA signal are input to the NOR gate 162, the first switch element M1 is turned on during the PWM duty control period of the first switch element M1 and the freewheeling period of the body diode D1. DrvH can be obtained.
  • the oscillator 30 is a circuit which generates or oscillates the waveform CT by charging or discharging the oscillation capacitor Cosc with the charge / discharge current source Iosc according to the charge / discharge control signal CK of the oscillation controller 32.
  • the oscillation controller 32 causes the charge / discharge control signal CK to go low when the oscillation waveform CT becomes smaller than the oscillation minimum voltage VL, thereby driving the oscillation capacitor Cosc to the current source Iosc.
  • the charge / discharge control signal CK becomes high at the moment when the oscillation waveform CT becomes larger than the oscillation maximum voltage VH so that the oscillation capacitance Cosc is discharged. Therefore, the frequency of the CK waveform is equal to the frequency of the CT waveform.
  • FIG. 8 is an exemplary view showing the configuration of a switch controller implemented according to a second embodiment of the present invention.
  • the switch controller implemented according to the second embodiment of the present invention further includes a delay time controller 202 capable of changing the diode on period by an external signal in the diode controller 200.
  • the logic controller 250 further includes a duty limiter 270 that limits only the specific PWM duty to be switched on instead of the diode in the freewheeling period.
  • the diode on period depends on the design of the power stage circuit, such as the inductance of the transformer T1. Therefore, it is advantageous for system design to be able to vary by external signal rather than fixing diode on period.
  • the delay time controller 202 is preferably for varying the freewheeling period to achieve this characteristic, so that the delay time of the first rising-edge-delay-circuit 204 varies according to the FWT pin state.
  • the switch controller according to the second embodiment of the present invention is configured to ignore the output of the diode controller 200 when a certain PWM duty is exceeded.
  • any signal of PWMO and PWMAd is a PWM duty control signal. It is not known which signal is the diode on time control signal. Therefore, as a solution to this problem, the switch controller can be configured to ignore the output of the diode controller 200 when the PWM duty is lowered below.
  • the switch controller 50 ′ provides a duty limiter 270 for operating the output PWMAd of the diode controller 200 only in any region of the PWM duty. It may be further included in the logic controller 250.
  • Duty limiter 270 may comprise at least one comparator 272, 274 and a NAND gate 276. Since the PWM duty is determined by the output voltage Verr of the error amplifier 10, the output voltage Verr and the duty limit voltage Vmax and Vmin of the error amplifier may be compared to limit the maximum and minimum of the PWM duty. Where Vmax is the reference voltage that sets the maximum PWM duty and Vmin is the reference voltage that sets the minimum PWM duty.
  • the output FWD goes low only when the output voltage Verr of the error amplifier is greater than Vmin and less than Vmax so that the output PWMAd of the diode controller 200 operates.
  • the error amplifier output voltage (Verr) is less than Vmin or greater than Vmax, FWD is high and DrvPA and DrvNA are always low regardless of PWMAd. Therefore, the output PWMAd of the diode controller 200 is ignored and the switch controller 50 'performs only the PWM duty control.
  • the diode controller implemented according to the first and second embodiments of the present invention controls the diode on time to be constant regardless of the PWM duty.
  • the diode controller turns on the switch instead of the diode in the freewheeling period longer than the actual freewheeling period, the distortion of the current waveform is generated, and the distortion increases as the period for turning on the switch is longer.
  • the diode freewheeling period becomes shorter as the PWM duty increases. Therefore, as the PWM duty increases, shortening the diode-on time generated by the diode controller will improve the efficiency when the PWM duty is small and improve the current waveform when the PWM duty is high. .
  • FIG. 9 is an exemplary diagram showing an internal configuration of a switch controller implemented according to the third embodiment of the present invention.
  • the switch controller implemented according to the third embodiment of the present invention is to achieve the above-described object, and is implemented such that the output PWMAd of the diode controller 300 changes according to the PWM duty. More specifically, as shown in FIG. 9, when the variable pulse width generator 302 outputs a signal PWMO_Ax1 whose pulse width is changed according to the PWM duty, the output signal passes through a logic circuit to the variable diode on time. It is output with the corresponding PWMAd.
  • the minimum duty limiter instead of the duty limiter in the configuration of the logic controller implemented according to the second embodiment of the present invention in order to ignore the output of the diode controller 300 and to operate as simple PWM duty control. 370. Therefore, if the output voltage Verr of the error amplifier is less than Vmin, FWDm becomes high so that DrvPA and DrvNA always go low regardless of PWMAd, and the output PWMAd of diode controller 300 is ignored and the switch controller 50 ′′. Will only control the PWM duty.
  • variable pulse width generator In the switch controller implemented according to the third embodiment of the present invention, the configuration of the variable pulse width generator is as follows.
  • FIG. 10 is a diagram illustrating a diode freewheeling period in an inverter system according to the present invention.
  • FIG. 10 (a) illustrates a model of the freewheeling period as a linear function according to PWM duty. The relationship between the PWM duty and the diode freewheeling period T FW is shown.
  • T FW may be represented by the following equation.
  • the diode controller 300 may operate as shown in Equation 1 below.
  • the error amplifier output voltage (Verr) represents the current PWM duty. Therefore, if the voltage V FW higher than the error amplifier output voltage Verr is compared with the triangular CT, the diode freewheeling period can be obtained and if the V FW voltage is decreased as the PWM duty increases, the diode controller 300 increases the PWM duty.
  • the equation (1) can be made a signal that reduces the diode freewheeling period.
  • Equation 1 Substituting Equation 1 into Equation 2 derived as described above may obtain the following equation.
  • Equation 3 K 1 , V FW, OFFSET are derivable values, and Verr is an error amplifier output voltage, so a circuit satisfying Equation 3 can be designed and applied to the variable pulse width generator.
  • FIG. 11 is a circuit diagram illustrating an example of implementing a variable pulse width generator in FIG. 10.
  • the variable pulse width generator may include a variable voltage generator 410 and a comparator 420.
  • the variable voltage generator 410 is implemented as a circuit to satisfy Equation 3 so that the variable voltage V FW gradually decreases as the PWM duty increases.
  • V FW is always greater than the error amplifier output voltage Verr, when V FW is compared with the triangular CT in the comparator 420, a PWMO_Ax1 signal in which the pulse width gradually decreases as the PWM duty increases.
  • the variable voltage generator 410 receives a voltage-current converter 412 that outputs an output current I 1 proportional to an error amplifier output voltage Verr, and receives the output as an input sourcing current.
  • the current mirror 416 includes a voltage-current converter 414 for outputting an output current I 2 proportional to an offset voltage Voff, and a current mirror 418 for receiving the output as an input and outputting a sourcing current.
  • it consists of the resistor R3 which adds the output current of the current mirrors 416 and 418, and converts it into a voltage.
  • the gains of the current mirrors 416 and 418 are 1, the output V FW of the variable voltage generator 410 may be represented by the following equation.

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  • Physics & Mathematics (AREA)
  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Nonlinear Science (AREA)
  • General Physics & Mathematics (AREA)
  • Optics & Photonics (AREA)
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  • Dc-Dc Converters (AREA)

Abstract

L'invention concerne un système onduleur régulé pour améliorer son efficacité, et son procédé d'exploitation. Ledit système onduleur comporte une unité de commande de commutation qui génère un signal de commande de modulation d'impulsions en durée (PWM) pour chaque commutateur pour exécuter une opération d'ondulation. De plus, l'unité de commande de commutation enclenche un commutateur correspondant sur une diode de corps temporairement pendant une période de roue libre lorsque le courant circule par mise en marche de la diode de corps du commutateur de façon que la direction du courant circulant sur un inducteur peut être maintenue après arrêt de chaque commutateur. En conséquence, la consommation de puissance du commutateur peut être réduite et l'efficacité du système onduleur améliorée tant que le courant de roue libre circule vers le commutateur plutôt que vers la diode de corps.
PCT/KR2009/002890 2008-10-23 2009-05-29 Systeme onduleur et son procede d'exploitation WO2010047455A1 (fr)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
KR10-2008-0104243 2008-10-23
KR1020080104243A KR100997397B1 (ko) 2008-10-23 2008-10-23 인버터 시스템 및 그의 동작방법

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WO2010047455A1 true WO2010047455A1 (fr) 2010-04-29

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PCT/KR2009/002890 WO2010047455A1 (fr) 2008-10-23 2009-05-29 Systeme onduleur et son procede d'exploitation

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WO (1) WO2010047455A1 (fr)

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US9197146B2 (en) 2012-07-26 2015-11-24 Milwaukee Electric Tool Corporation Brushless direct-current motor and control for power tool
US10821591B2 (en) 2012-11-13 2020-11-03 Milwaukee Electric Tool Corporation High-power cordless, hand-held power tool including a brushless direct current motor

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KR102504096B1 (ko) * 2015-12-01 2023-02-27 엘지이노텍 주식회사 Bldc 전동기 pwm 제어 장치 및 그 방법
KR20230103975A (ko) 2021-12-30 2023-07-07 주식회사 엘엑스세미콘 전력 효율을 향상시킨 파워 모듈 및 2차측 스위치의 턴 오프 제어 방법

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Cited By (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US9197146B2 (en) 2012-07-26 2015-11-24 Milwaukee Electric Tool Corporation Brushless direct-current motor and control for power tool
US9647585B2 (en) 2012-07-26 2017-05-09 Milwaukee Electric Tool Corporation Brushless direct-current motor and control for power tool
US10821591B2 (en) 2012-11-13 2020-11-03 Milwaukee Electric Tool Corporation High-power cordless, hand-held power tool including a brushless direct current motor
US11141851B2 (en) 2012-11-13 2021-10-12 Milwaukee Electric Tool Corporation High-power cordless, hand-held power tool including a brushless direct current motor
US11370099B2 (en) 2012-11-13 2022-06-28 Milwaukee Electric Tool Corporation High-power cordless, hand-held power tool including a brushless direct current motor
US11673248B2 (en) 2012-11-13 2023-06-13 Milwaukee Electric Tool Corporation High-power cordless, hand-held power tool including a brushless direct current motor
US12011812B2 (en) 2012-11-13 2024-06-18 Milwaukee Electric Tool Corporation High-power cordless, hand-held power tool including a brushless direct current motor

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KR20100045170A (ko) 2010-05-03
KR100997397B1 (ko) 2010-11-30

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