WO2010004586A2 - Procédé et système de transmission et de réception de signaux - Google Patents

Procédé et système de transmission et de réception de signaux Download PDF

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Publication number
WO2010004586A2
WO2010004586A2 PCT/IN2009/000391 IN2009000391W WO2010004586A2 WO 2010004586 A2 WO2010004586 A2 WO 2010004586A2 IN 2009000391 W IN2009000391 W IN 2009000391W WO 2010004586 A2 WO2010004586 A2 WO 2010004586A2
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Prior art keywords
data sequence
signal
frequency domain
create
sequence
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PCT/IN2009/000391
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English (en)
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WO2010004586A3 (fr
Inventor
Kiran Kumar Kuchi
Deviraj Klutto Milleth Jeniston
Vinod Ramaswamy
Baskaran Dhivagar
Krishnamurthi Giridhar
Bhaskar Ramamurthi
Padmanabhan Madampu Suryasarman
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Centre Of Excellence In Wireless Technology
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Publication of WO2010004586A2 publication Critical patent/WO2010004586A2/fr
Publication of WO2010004586A3 publication Critical patent/WO2010004586A3/fr

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2614—Peak power aspects
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
    • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
    • H04L27/3405—Modifications of the signal space to increase the efficiency of transmission, e.g. reduction of the bit error rate, bandwidth, or average power
    • H04L27/3411—Modifications of the signal space to increase the efficiency of transmission, e.g. reduction of the bit error rate, bandwidth, or average power reducing the peak to average power ratio or the mean power of the constellation; Arrangements for increasing the shape gain of a signal set

Definitions

  • the embodiments herein relate to communication and, more particularly, to design and use of power efficient modulation techniques with low peak-to-average power ratio in wireless communication, and interference suppression in wireless systems
  • OFDMA Orthogonal Frequency-Division Multiple Access
  • OFDMA systems use multiple carriers to modulate and transmit data.
  • An IDFT is applied on the modulated data tone to generate the transmitted time domain signal.
  • the transmitted signal is a sum of several sinusoids modulated by random modulation symbols. Due to the summation of different sinusoids, the transmitted signal exhibits high peaks and nulls resulting in a high Peak- to-Average Power Ratio (PAPR).
  • PAPR Peak- to-Average Power Ratio
  • PA Power Amplifier
  • High power back-off has to be applied to operate the PA in the linear range if OFDM signals having high PAPR are fed as input to non-linear PA' s. With insufficient power back-off, the PA output signals exhibits significant distortion and spectral regrowth occurs and out-of band emissions become significant.
  • Portable mobile terminals have limited power to transmit and receive signals. In cells with large cell radius, the uplink generally limits the coverage and data rate due to limited transmitted signal power available in portable mobile terminals.
  • Embodiments herein disclose a method and system for precoding in a communication network, the method comprising steps of applying a constellation rotation to an input data sequence to create a constellation rotated data sequence; performing convolution on the constellation rotated data sequence using a polynomial precoder to create a precoded data sequence; transforming the precoded data sequence into frequency domain using an M-point DFT (Discrete Fourier Transform) to create a DFT output data sequence; performing mapping on the DFT output data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on the mapped data sequence.
  • M-point DFT Discrete Fourier Transform
  • N-point IDFT Inverse Discrete Fourier Transform
  • Embodiments herein disclose a method and system for precoding in a communication network, the method comprising steps of applying a constellation rotation to an input data sequence to create a constellation rotated data sequence; transforming the constellation rotated data sequence into frequency domain using an M-point DFT (Discrete Fourier Transform) to create a DFT output data sequence; multiplying the DFT output data sequence with DFT of a polynomial precoder to create a precoded data sequence; performing mapping on the precoded data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on the mapped data sequence.
  • M-point DFT Discrete Fourier Transform
  • Embodiments herein disclose a method and system for precoding in a communication network, the method comprising steps of transforming an input data sequence into frequency domain using an M-point DFT (Discrete Fourier Transform) to create a DFT output data sequence; shifting the DFT output data sequence by no samples to create a shifted data sequence; multiplying the shifted data sequence with DFT of a polynomial precoder to create a precoded data sequence; performing mapping on the precoded data sequence to create a mapped data sequence; and performing N-point IDFT (Inverse Discrete Fourier Transform) on the mapped data sequence.
  • M-point DFT Discrete Fourier Transform
  • Embodiments herein disclose a method and system for transmitting a plurality of ASK (Amplitude Shift Keying) signals in a communication scheme, wherein the plurality of ASK signals are transmitted simultaneously in a fixed block, wherein the block is fixed in frequency and time.
  • ASK Amplitude Shift Keying
  • Embodiments herein disclose a method and a receiver for processing a received communication signal, the method comprising steps of applying an N-point DFT to the received signal to create a frequency domain signal; de-mapping the frequency domain signal to create a de-mapped frequency domain signal; de-shifting the de-mapped frequency domain signal to create a de-shifted frequency domain signal; taking complex conjugate and frequency reversal of the de-shifted frequency domain signal to create a modified frequency domain signal; filtering the de-shifted frequency domain signal and the modified frequency domain signal with a plurality of filter weights to obtain a filtered signal; and applying L-point IDFT to the filtered signal.
  • Embodiments herein disclose a method and a receiver for processing a received communication signal, the method comprising steps of de-rotating the received signal to create a de-rotated signal; applying an N-point DFT to the de-rotated signal to create a frequency domain signal; de-mapping the frequency domain signal to create a de-mapped frequency domain signal; taking complex conjugate and frequency reversal of the de-mapped frequency domain signal to create a modified frequency domain signal; filtering the de-mapped frequency domain signal and the modified frequency domain signal with a plurality of filter weights to obtain a filtered signal; and applying L-point IDFT to the filtered signal.
  • Embodiments herein disclose a method and receiver for processing pilot data in a communication signal, the method comprising steps of performing circular shifting on a first pilot data to create a circular shifted first pilot data; taking conjugate of the first pilot data to create a conjugate pilot data; frequency reversing the conjugate pilot data to create a frequency reversed pilot data; performing circular shifting of the frequency reversed pilot data to create a circular shifted second pilot data; and transmitting the circular shifted first pilot data and the circular shifted second pilot data.
  • Embodiments herein disclose a method and receiver for processing received pilot data in a received communication signal, the method comprising steps of performing circular de- shifting on a first received pilot data and a second received pilot data to create a first de-shifted received pilot data and a de-shifted second pilot data; performing channel estimation using the first received pilot data and the second received pilot data to obtain an estimated channel; taking conjugate of the de-shifted second pilot data to create a conjugate received pilot data; frequency reversing the conjugate received pilot data to create a frequency reversed received pilot data; estimating a first Noise and Interference Co- variance Matrix (NICM) of background noise and interference from the received pilot data using the de-shifted first pilot data and the frequency reversed received pilot data; and multiplying elements of the NICM using a frequency dependent weight.
  • NVM Noise and Interference Co- variance Matrix
  • FIG. 1 illustrates a block diagram of an SC-FDMA transmitter with discrete data precoding, according to an embodiment herein;
  • FIG. 2 is a flowchart depicting a method to precode data in an SC-FDMA transmitter, according to an embodiment herein;
  • FIG. 3 illustrates a block diagram of an SC-FDMA transmitter with discrete data precoding done in frequency domain, according to an embodiment herein;
  • FIG. 4 is a flowchart depicting a method to precode data in the frequency domain, in an SC-FDMA transmitter, according to an embodiment herein;
  • FIG. 5 illustrates a block diagram of a receiver for precoded SC-FDMA signals, according to an embodiment herein;
  • FIG. 6 is a flowchart depicting a method to de-modulate precoded SC-FDMA data, according to an embodiment herein;
  • FIG. 7 illustrates a block diagram of a receiver with signal de-rotation done in time domain for precoded SC-FDMA signals, according to an embodiment herein;
  • FIG. 8 illustrates an SC-FDMA pilot slot structure, according to an embodiment herein;
  • FIG. 9 illustrates a block diagram of a receiver used for channel estimation, according to an embodiment herein;
  • FIG. 10 is a flowchart depicting a method to estimate the channel from pilot sequences, according to an embodiment herein.
  • the embodiments herein disclose a digitally precoded Single Carrier-Frequency Division Multiple Access (SC-FDMA) scheme with low Peak-to-Average Power Ratio (PAPR) by a constellation rotation of the input data sequence and circularly convolving the constellation rotated data.
  • SC-FDMA Single Carrier-Frequency Division Multiple Access
  • PAPR Peak-to-Average Power Ratio
  • FIG. 1 illustrates a block diagram of an SC-FDMA transmitter with discrete data precoding.
  • the input data sequence a k 101 is applied to a constellation rotation block 102.
  • the input data sequence may be Binary Phase Shift Keying (BPSK) data, Q-ary Pulse Amplitude Modulation (PAM)/Amplitude Shift Keying (ASK) data or complex-valued Quadrature Amplitude Modulation (QAM) data.
  • BPSK Binary Phase Shift Keying
  • PAM Q-ary Pulse Amplitude Modulation
  • ASK Amplitude Shift Keying
  • QAM Complex-valued Quadrature Amplitude Modulation
  • a k may take values from the set b k e [- (2Q-l),..-,3,-lX3,..,(2Q-l)] where Q is the constellation size and the constellation rotation block 102 may rotate the signal constellation by multiplying a k 101 by j k , where k is
  • the constellation rotated data sequence is then circularly convolved using a polynomial precoder 103.
  • the coefficients of the precoder 103 may take real or complex values.
  • the precoder 103 may have two taps and the tap weights may be given by:
  • the tap weights of the precoder 103 may be given by:
  • the precoded data is then transformed to frequency domain by taking an M-point Discrete Fourier Transform (DFT) 104 of the precoded data, where M is the data length.
  • DFT Discrete Fourier Transform
  • d is the M-point DFT 104 output.
  • the M-point DFT 104 output data is then mapped to specific subcarriers in the subcarrier mapping block 105.
  • the DFT output may be mapped to a contiguous set of subcarriers and if there are any subcarriers remaining, then zero padding would be used to fill the remaining subcarriers.
  • the DFT output may also be mapped to subcarriers distributed in the entire frequency domain with zeros filled in the unused subcarriers.
  • an N-Point Inverse DFT (IDFT) of the subcarriers is taken using an N-Point IDFT block 106.
  • the N-Point IDFT 106 output can be represented as
  • ⁇ / is the subcarrier width, and the signal spans over the time interval "T". ⁇ may also be chosen to be greater then M. 5(0 is the transmitted signal that is sent to the receiver. A cyclic prefix may also be added to -s(t) before being transmitted.
  • FIG. 2 is a flowchart depicting a method to precode data in an SC-FDMA transmitter.
  • the input data sequence a k 101 is applied to the constellation rotation block 102.
  • the input data sequence may be BPSK data, Q-ary-PAM/ASK data or complex- valued QAM data.
  • the constellation rotation block 102 rotates (201) the signal constellation by a specific degree.
  • the constellation rotated data sequence is then circularly convolved (202) using the polynomial precoder 103.
  • the precoded data is then transformed to frequency domain by taking (203) an M- point DFT 104 of the precoded data, where M is the data length.
  • the M-point Discrete Fourier Transform (DFT) 104 of the precoded data can be represented as
  • the DFT output may be mapped to a contiguous set of subcarriers and if there are any subcarriers remaining, then zero padding would be used to fill the remaining subcarriers.
  • the DFT output may also be mapped to subcarriers distributed in the entire frequency domain with zeros filled in the unused subcarriers.
  • an N-Point Inverse DFT (IDFT) of the subcarriers is taken (205).
  • the N-Point IDFT 106 output can be represented as
  • FIG. 3 illustrates a block diagram of an SC-FDMA transmitter with discrete data precoding done in frequency domain.
  • the input data sequence a k 101 is applied to a constellation rotation block 102.
  • the input data sequence may be BPSK data, Q-ary- PAM/ASK data or complex- valued QAM data.
  • the constellation rotated data sequence is then transformed to frequency domain by taking an M-point DFT 104 of the constellation rotated data, where M is the data length.
  • the M-point DFT 104 of the constellation data can be represented as
  • d is the output of M-point DFT 104.
  • the M-point DFT 104 output data is then precoded using a frequency domain pulse shaping block 301.
  • the frequency domain pulse shaping block 301 multiplies the elements of the M-point DFT 104 output using the coefficients of the DFT of the precoder.
  • the coefficients of the frequency domain pulse shaping block 301 may take real or complex values.
  • the frequency domain pulse shaping block 301 coefficients may be represented as:
  • the frequency domain pulse shaping is represented as:
  • the frequency domain pulse shaped data d t is then mapped to specific subcarriers in the subcarrier mapping block 105.
  • the pulse shaped data may be mapped to a contiguous set of subcarriers and if there are any subcarriers remaining, then zero padding would be used to fill the remaining subcarriers.
  • the pulse shaped data may also be mapped to subcarriers distributed in the entire frequency domain with zeros filled in the unused subcarriers.
  • an N-Point IDFT of the subcarriers is taken using an N-Point IDFT block 106.
  • the N-Point IDFT 106 signal is the transmitted signal that is sent to the receiver.
  • a cyclic prefix may also be added to N-Point IDFT 106 signal before being transmitted.
  • FTG. 4 is a flowchart depicting a method to precode data in the frequency domain, in an SC-FDMA transmitter.
  • the input data sequence a k 101 is applied to the constellation rotation block 102.
  • the input data sequence may be BPSK data, Q-ary-PAM/ASK data or complex- valued QAM data.
  • the constellation rotation block 102 rotates (401) the signal constellation by a specific degree.
  • the constellation rotated data sequence is then transformed to frequency domain by taking (402) an M-point DFT 104 of the constellation rotated data, where M is the data length.
  • the M-point DFT 104 of the precoded data can be represented as
  • d is the M-point DFT 104 output.
  • the M-point DFT 104 output data is then precoded using a frequency domain pulse shaping block 301.
  • the frequency domain pulse shaping block 301 multiplies the elements of the M-point DFT 104 output using the coefficients of the DFT of the precoder.
  • the coefficients of the frequency domain pulse shaping block 301 may take real or complex values.
  • the frequency domain pulse shaping is represented as:
  • the frequency domain pulse shaped data d t is then mapped (404) to specific subcarriers in the subcarrier mapping block 105.
  • the pulse shaped data may be mapped to a contiguous set of subcarriers and if there are any subcarriers remaining, then zero padding would be used to fill the remaining subcarriers.
  • the pulse shaped data may also be mapped to subcarriers distributed in the entire frequency domain with zeros filled in the unused subcarriers.
  • an N-Point IDFT of the subcarriers is taken (405) using the N- Point IDFT block 106.
  • the N-Point IDFT 106 signal is the transmitted signal that is sent to the receiver.
  • a cyclic prefix may also be added to N-Point IDFT 106 signal before being transmitted.
  • the various actions in the method may be performed in the order presented, in a different order, or simultaneously. Further, in some embodiments, some actions listed in FIG. 4 may be omitted.
  • FIG. 5 illustrates a block diagram of a receiver for precoded DFT-S-OFDMA signals.
  • the transmitted DFT-S-OFDMA signals would be received at the receiver and the transmitted data can be obtained from the received signal.
  • the desired signal component of the sampled time domain signal received at the receiver can be represented as:
  • N-point DFT of the received signal is taken using an N-point DFT block 501 at the receiver. N may be chosen to be greater than the number of sub- carriers used at the transmitter. The N-point DFT signal is then passed to a sub-carrier de- mapping block 502 to retrieve the data that was mapped onto the subcarriers at the transmitter.
  • n is the discrete frequency index ranging from (0, M-I)
  • p(n) is the precoding done at the transmitter
  • h(n) is the frequency domain propagation channel coefficients
  • x(ra) is the DFT of the constellation rotated data. The constellation rotation of the data was done at the transmitter.
  • x(n) can be represented as
  • x(n) can also be represented as
  • a k is the input data sequence at the transmitter and the DFT of the input data sequence can be represented as
  • the frequency shifted precoder p(n) can be defined as
  • the de-mapped low pass signal contains the desired signal transmitted from the transmitter, the Co-channel interference components which are generated by other signals and the noise components.
  • the de-mapped low pass signal can be represented in terms of the desired signal, the Co-channel interference components and the noise components as:
  • X 1 (H) denotes the DFT of /-th pi/2 rotated interference component
  • h,(n) denotes the propagation channel vectors of the CCI components
  • h(n) denotes the channel vector of the desired signal
  • x(n) is the DFT of the desired signal. If the receiver has 'Nr' antennae then the vectors y(n) , h(n) , jc(n) , p(n) ,h,(n) , jc,(n) and n(n) would be of size equal to 'Nr'. [0031]
  • the de-mapped signal is then applied to a constellation de-rotation block 503.
  • the constellation de-rotation block 503 de-rotates the signal constellation by a specific degree.
  • de-rotation can be done by applying a circular frequency de-shifting operation. If the frequency is de-shifted by H 0 tones, the frequency de-shifted signal can be represented as
  • M y(n) h(n)a(n)p(n) + ⁇ h, (n) ⁇ , (n)p(n) + ⁇ (n) (8)
  • the frequency de-shifted signal is then applied to a conjugation and frequency reversal block 504.
  • a complex-conjugation operation is performed on the frequency de-shifted signal and a frequency reversal is then performed on the signal.
  • Frequency reversal is performed by changing the sign of the frequency value in the signal. Positive values of the frequency are changed to negative values and negative values are changed to positive values under moduIo-M operation.
  • the frequency reversed and complex conjugated signal is sent to a filter 505. Also, a copy of the constellation de-rotated signal, without frequency reversal and complex conjugation, is applied to the filter 505.
  • the constellation de-rotated signal and the frequency reversed and complex conjugated signal can be represented in vector form as:
  • the signal can be written in compact form as:
  • y(n) is the signal sent to the filter 505.
  • the elements of y(n) would then be weighed by the filter 505 weights and then all the elements of the signal would be added.
  • the filter weights are obtained by estimating the channel and noise-plus-interference covariance (NICM) by the use of pilots. Pilot sequences may be transmitted for supervisory, control, equalization, continuity, synchronization or reference purposes.
  • a channel and Noise-plus Interference-Covariance- Matrix (NICM) estimation block 506 estimates the channel from the pilots received at the receiver from the transmitter. After the channel and NICM estimation block 506 estimates the channel, the filter weights would be determined.
  • the filter 505 used may be a Minimum Mean Square Error (MMSE) filter and the filter weights may be represented as:
  • R 1 ., (n) is the estimated NICM.
  • the NICM is a matrix whose elements are a measure of how much the noise and interference variables in the communication system change with respect to each other.
  • MSE Minimum Mean- Square-Error
  • the filtered signal is then applied to an M-Point IDFT block 507 and an M-Point IDFT of the filtered signal is taken.
  • the M-Point IDFT gives the decision variable which is used to demodulate the original transmitted data.
  • the bias introduced by the MMSE filter may also be corrected before demodulation.
  • the receiver is able to suppress interference.
  • the number of receiver antennas mentioned here is exemplary, and do not restrict the embodiments herein. It is obvious to a person of ordinary skill in the art that the embodiments disclosed above can be extended to a receiver with any number of antennas.
  • FIG. 6 is a flowchart depicting a method to de-modulate precoded SC-FDMA data.
  • the transmitted SC-FDMA signals would be received at the receiver and the transmitted data can be obtained from the received signal.
  • the desired signal component of the sampled time domain signal received at the receiver can be represented as:
  • N is the discrete time index.
  • An N-point DFT of the received signal is taken (601) using an N-point DFT block 501 at the receiver. N may be chosen to be greater than the number of sub-carriers used at the transmitter.
  • the N-point DFT signal is then passed (602) to a sub- carrier de-mapping block 502 to retrieve the data that was mapped onto the subcarriers at the transmitter.
  • the de-mapped low pass signal can be represented in frequency domain as: Where n is the discrete frequency index ranging from (O, M-I), p(n) is the precoding done at the transmitter, /i(n) is the frequency domain propagation channel coefficients andjc(n) is the DFT of the constellation rotated data. The constellation rotation of the data was done at the transmitter. x ⁇ n) can be represented as:
  • a k is the input data sequence at the transmitter and the DFT of the input data sequence taken along with the received signal at the receiver can be represented as:
  • the de-mapped low pass signal contains the desired signal transmitted from the transmitter, the Co-channel interference components which are generated by other signals and the noise components.
  • the de-mapped low pass signal can be represented in terms of the desired signal, the Co-Channel Interference components and the noise components as:
  • M 1 y (n) h(n)x(ri)p( ⁇ )+ ⁇ h t ( ⁇ )x, in)p ⁇ n) + n(n)
  • X 1 (Ji) denotes the DFT of /-th pi/2 rotated interference component
  • h,( «) denotes the propagation channel vectors of the CCI components
  • h(/z) denotes the channel vector of the desired signal
  • x ⁇ n) is the DFT of the desired signal. If the receiver has 'Nr' antennae then the vectors y (n) , h(/z) , x(n) , p( ⁇ ) , h, (n) , x, (n) and n(n) would be of size equal to 'Nr' .
  • the de-mapped signal is then de-rotated using a constellation de-rotation block 503.
  • the constellation de-rotation block 503 de-rotates (603) the signal constellation by a specific degree.
  • the de-rotation (603) can be done by applying a circular frequency de-shifting operation. If the frequency is de-shifted by H 0 tones, the frequency de-shifted signal can be represented as:
  • y (n) h(n)a(n)p( ⁇ ) + ⁇ h, (n)a, (n)p(n) + ⁇ (n)
  • the frequency de-shifted signal is then conjugated and frequency reversed (604) using a conjugation and frequency reversal block 504.
  • a complex-conjugation operation is performed (604) on the frequency de-shifted signal and a frequency reversal is then performed (604) on the signal.
  • Frequency reversal is performed by changing the sign of the frequency value in the signal. Positive values of the frequency are changed to negative values and negative values are changed to positive values.
  • the frequency reversed and complex conjugated signal is then filtered (605) using a filter 505. Also, a copy of the constellation de-rotated signal, without frequency reversal and complex conjugation, is applied to the filter 505.
  • the constellation de- rotated signal and the frequency reversed and complex conjugated signal can be represented in vector form as:
  • the signal can be written in compact form as:
  • M 1 y (n) h( ⁇ )a( ⁇ ) + ⁇ h, (K)O 1 (n) + n(n)
  • y(n) ,h(n) ,h j (n) and n(n) denote corresponding vectors in equation (14).
  • y(n) is the signal sent to the filter 505.
  • the elements of y(n) would then be weighed by the filter 505 weights and then all the elements of the signal would be added.
  • the filter weights are obtained by estimating the channel by the use of pilots.
  • a channel estimation block 506 estimates the channel from the pilots received at the receiver from the transmitter. After the channel and NICM estimation block 506 estimates the channel, the filter 505 weights would be determined.
  • the filter 505 used may be a Minimum Mean Square Error (MMSE) filter and the filter weights may be represented as:
  • MMSE Minimum Mean Square Error
  • the filtered signal is then applied to an M-Point IDFT block 507 and an M-Point IDFT of the filtered signal is taken (606).
  • the M-Point IDFT gives the decision variable which is used to demodulate the original transmitted data.
  • the bias introduced by the MMSE filter may be corrected before demodulation.
  • the various actions in the method may be performed in the order presented, in a different order, or simultaneously. Further, in some embodiments, some actions listed in FIG. 6 may be omitted.
  • FIG. 7 illustrates a block diagram of a receiver with signal de-rotation done in time domain for precoded SC-FDMA signals.
  • the transmitted SC-FDMA signals would be received at the receiver and the transmitted data can be obtained from the received signal.
  • the received data signal is applied to a constellation de-rotation block 503.
  • the de-rotated signal is collected and applied to an N-point DFT block 701.
  • the output of the N-point DFT block 701 is passed to a sub-carrier de-mapping block 502 to retrieve the data that was mapped onto the subcarriers at the transmitter.
  • the de-mapped low pass signal contains the desired signal transmitted from the transmitter, the Co-Channel Interference components and the noise components.
  • the de-mapped signal is then applied to a conjugation and frequency reversal block 504.
  • a complex-conjugation operation is performed on the de-mapped signal and a frequency reversal is then performed on the signal. Frequency reversal is performed by changing the sign of the frequency value in the signal. Positive values of the frequency are changed to negative values using a Module-M operation. The negative values of the frequency are changed to positive values using a Module-M operation.
  • the frequency reversed and complex conjugated signal is sent to a filter 505.
  • a copy of the constellation de-rotated signal, without frequency reversal and complex conjugation is applied to the filter 505.
  • the signal components would be weighed by the filter weights and then all the elements of the signal would be added.
  • the filter weights are obtained by estimating the channel by the use of pilots.
  • a channel estimation block 506 estimates the channel from the pilots received at the receiver from the transmitter. After the channel estimation block 506 estimates the channel, the filter 505 weights would be determined.
  • the filtered signal is then applied to an M-Point IDFT block 507 and an M-Point IDFT of the filtered signal is taken.
  • the M-Point IDFT gives the decision variable which is used to demodulate the original transmitted data.
  • FIG. 8 illustrates an SC-FDMA pilot slot structure.
  • Slot one and slot two show two slot structures used for transmitting the pilot sequence. Pilot sequences are transmitted for estimating the channel. Pilot sequences may also be transmitted for supervisory, control, equalization, continuity, synchronization or reference purposes.
  • Pl is the pilot sequence transmitted in the specific slot and Dl, D2, D3, D4, D5 and D6 are the data signals that are transmitted in the specific slots.
  • Pl is a frequency domain pilot sequence and Pl can be represented as b, ( ⁇ ) where / is the signal index.
  • the frequency domain pilot sequence used may also be a CAZAC type pilot sequence.
  • b, (n) is transmitted as the first pilot symbol.
  • P2 is the pilot sequence transmitted in the specific slot and D7, D8, D9, DlO, DIl and D12 are the data signals that are transmitted in the specific slots.
  • the signal is then circularly shifted by n 0 tones to produce another sequence
  • the circularly shifted signal is then transmitted in the second pilot.
  • the pilot sequence may also be transmitted without circularly shifting the conjugated and frequency reversed signal.
  • a slot structure can be used for DFT-S-OFDMA signals wherein the same modulator is used for the pilots and the data.
  • Time domain pilot sequences would be used as training sequences and the pilot sequences would have good auto correlation, cross correlation and low PAPR.
  • the pilots and data would be transmitted using DFT-S- OFDMA systems and low PAPR would be maintained across all OFDM symbols including the pilots and the data. If the receiver jointly filters the complex and complex-conjugate parts of the baseband received signal, then efficient interference suppression could be achieved using the slot structure.
  • FTG. 9 illustrates a block diagram of a receiver used for channel estimation. Pilot sequences are transmitted for estimating the channel.
  • Pilot sequences may also be transmitted for supervisory, control, equalization, continuity, synchronization or reference purposes.
  • the received pilot sequence is applied to an N-point DFT block 501 and an N-point DFT of the received pilot sequence is obtained.
  • the signal is then applied to a sub-carrier de- mapping block 901 to retrieve the pilot codes that were mapped onto the subcarriers at the transmitter.
  • the pilot codes are then applied to a channel estimation block 902.
  • the channel estimation block 902 estimates the propagation channel vectors from the pilots. If there is no CCI present in the channel, the propagation channel vectors in matrix form can be represented as:
  • the two pilots Pl and P2 applied to the NICM estimation block 903 can be represented as:
  • y b2 (n) h(n)b * (M - n - n 0 ) + £ h ⁇ (n)b * (M - n - n 0 ) + n b2 (n)
  • the second pilot sequence is then conjugated and frequency reversed and the conjugated and frequency reversed pilot sequence can be represented as:
  • pilot sequences can be represented in vector form as:
  • NICM can then be estimated by collecting the interference components from the pilot sequence.
  • Interference components can be obtained by subtracting the desired signal components from the pilot signals.
  • the interference components can be represented as:
  • ⁇ is the error caused by the imperfect interference sample estimation.
  • the NICM is can be estimated from the interference samples. If the channel has the same frequency response for all the sub-carriers spanning a resource block, then an estimate of R 11 ( «) can be represented as:
  • Equation 17 The averaging operation in equation 17 is done by using samples collected from the entire resource block.
  • An estimate of R 1 , (n) can be obtained by multiplying the two matrices R ( .,(n)andP(n) .
  • R ⁇ (n) can then be represented as:
  • FIG. 10 is a flowchart depicting a method to estimate the channel from pilot sequences. Pilot sequences are transmitted for estimating the channel. Pilot sequences may also be transmitted for supervisory, control, equalization, continuity, synchronization or reference purposes.
  • an N-point DFT of the received pilot sequence is taken (1001) using the N-point DFT block 501.
  • the signal is then applied to a sub-carrier de-mapping block 901 to retrieve (1002) the pilot codes that were mapped onto the subcarriers at the transmitter.
  • the pilot codes are then applied to a channel estimation block 902.
  • the channel estimation block 902 estimates (1003) the propagation channel vectors from the pilots. If there is no CCI present in the channel, the propagation channel vectors in matrix form can be represented as:
  • h(n) is the propagation channel vector. If CCI is present in the channel, then, to estimate the NICM the two pilots are sent to the NICM estimation block 903.
  • the two pilots Pl and P2 applied to the NICM estimation block 903 can be represented as:
  • y b2 (n) ( ⁇ )b](M -n -n ⁇ ) + n b2 ( ⁇ ) y fcl (n) and y fc2 (n) are then circularly de-shifted (1004) by n o , and the circularly de-shifted signals can be represented as:
  • the second pilot sequence is then conjugated and frequency reversed (1005) and the conjugated and frequency reversed pilot sequence can be represented as:
  • pilot sequences can be represented in vector form as:
  • NICM can then be estimated
  • Interference components can be obtained by subtracting the desired signal components from the pilot signals.
  • the interference components can be represented as:
  • ⁇ is the error caused by the imperfect interference sample estimation.
  • the NICM can be estimated (1007) from the interference samples. If the channel has the same frequency response for all the sub-carriers spanning a resource block, then an estimate of R n . ( «) can be represented as:
  • Equation 20 The averaging operation in equation 20 is done by using samples collected from the entire resource block.
  • An estimate of R 1 , (n) can be obtained by multiplying the two matrices
  • R 11 (n) can then be represented as:
  • Low PAPR signaling and interference cancellation (IC) features can be implemented in SC-FDMA, DFT-S-OFDMA or OFDMA networks in an IC region allocated to serve cell edge users and/or control channel transmission.
  • the communication network assigns a pre-defined IC region used exclusively for SC-FDMA, Q-ary PAM/ASK, DFT-S-OFDMA or OFDMA transmission either in Down Link (DL) or in the Up Link (UL).
  • IC region may be composed of a predefined set of resource units.
  • the IC region may be a predefined set of PRUs or slots.
  • the basic IC resource unit may be composed of a pair of slots which may be contiguous or distributed in time-frequency plane.
  • the communication network assigns a pre specified time-frequency resource in the IC region.
  • the co-channel base stations and mobile stations transmit signals in a synchronous manner. Different base stations and mobile stations may use different modulation sizes but the signal constellation used would have rectilinear signals.
  • the transmitter may use pi/2 rotated Q-ary PAM/ASK data and digital precoding.
  • Information about the IC region may be communicated to each mobile station in a broadcast control channel.
  • the receiver uses IC receivers.
  • the receiver uses interference suppression MMSE algorithms to increase the data rate and reliability of reception.
  • the slot structure which is depicted in FIG. 8 can be used for Q-ary PAM/ASK DFT-S-OFDMA or OFDMA transmission.
  • the embodiments herein reduce the PAPR and allow the PA to be used near the saturation region.
  • PA' s can be operated in the linear range without the need to have a high power back-off to operate the PA.
  • Transmitted signal power can thus be increased in communication terminals having limited reserve power.
  • Signals with greater power can be used in the Up-link and Down Link.
  • the data rate of signals transmitted can also be increased.
  • Embodiments herein can be implemented without using DFT and DDFT blocks. Also, the disclosed modulation and de-modulation schemes can be used for DFT-S-OFDMA systems.
  • the embodiments disclosed herein can be implemented through at least one software program running on at least one hardware device and performing network management functions to control the network elements.
  • the network elements shown in Fig. 1 include blocks which can be at least one of a hardware device, or a combination of hardware device and software module.
  • the embodiment disclosed herein specifies a system for a digitally precoded SC- FDMA scheme with low PAPR.
  • the mechanism allows transmission and reception of signals having low PAPR providing a system thereof. Therefore, it is understood that the scope of the protection is extended to such a program and in addition to a computer readable means having a message therein, such computer readable storage means contain program code means for implementation of one or more steps of the method, when the program runs on a server or mobile device or any suitable programmable device.
  • the method is implemented in a preferred embodiment through or together with a software program written in e.g. Very high speed integrated circuit Hardware Description Language (VHDL) another programming language, or implemented by one or more VHDL or several software modules being executed on at least one hardware device.
  • VHDL Very high speed integrated circuit Hardware Description Language
  • the hardware device can be any kind of device which can be programmed including e.g. any kind of computer like a server or a personal computer, or the like, or any combination thereof, e.g. one processor and two FPGAs.
  • the device may also include means which could be e.g. hardware means like e.g. an ASIC, or a combination of hardware and software means, e.g. an ASIC and an FPGA, or at least one microprocessor and at least one memory with software modules located therein.
  • the means are at least one hardware means and/or at least one software means.
  • the method embodiments described herein could be implemented in pure hardware or partly in hardware and partly in software.
  • the device may also include only software means.
  • the invention may be implemented on different hardware devices, e.g. using a plurality of CPUs.

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  • Radio Transmission System (AREA)

Abstract

L’invention concerne dans des modes de réalisation un procédé SC-FDMA numériquement précodé avec un faible PAPR, le procédé impliquant l’application d'une rotation de constellation à la séquence de données d’entrée de fréquence M, avant la formation de convolutions circulaires de la séquence de données pivotée avec un décodeur polynomial. Les données sont ensuite transformées selon le domaine fréquentiel, puis mappées. Un N-point IIFT est alors appliqué aux données pour produire des échantillons de domaine temporel.
PCT/IN2009/000391 2008-07-10 2009-07-09 Procédé et système de transmission et de réception de signaux WO2010004586A2 (fr)

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KR20150101837A (ko) * 2014-02-27 2015-09-04 삼성전자주식회사 이동 통신 시스템에서 신호 송수신을 위한 변조 방법 및 장치
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US11824694B2 (en) 2015-09-02 2023-11-21 Astrapi Corporation Systems, devices, and methods employing instantaneous spectral analysis in the transmission of signals
US11411785B2 (en) 2015-09-02 2022-08-09 Astrapi Corporation Spiral polynomial division multiplexing
WO2017178871A1 (fr) 2016-04-15 2017-10-19 Indian Institute Of Technology Hyderabad Procédé et émetteur pour générer une forme d'onde avec un rapport papr optimisé
EP3443714A4 (fr) * 2016-04-15 2019-11-20 Indian Institute Of Technology Hyderabad Procédé et émetteur pour générer une forme d'onde avec un rapport papr optimisé
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EP3497828A4 (fr) * 2016-08-12 2020-04-15 Indian Institute Of Technology Hyderabad Système et procédé pour générer une forme d'onde dans un réseau de communication
WO2018130973A1 (fr) * 2017-01-13 2018-07-19 Wisig Networks Private Limited Récepteur de réception d'un ofdm étalé à transformée de fourier discrète avec précodage de domaine de fréquence
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US11283657B2 (en) 2017-08-03 2022-03-22 Samsung Electronics Co., Ltd. Device and method for processing received signal in wireless communication system
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WO2019027284A1 (fr) * 2017-08-03 2019-02-07 삼성전자주식회사 Dispositif et procédé permettant de traiter un signal reçu dans un système de communication sans fil
CN111406389B (zh) * 2017-11-29 2023-05-23 高通股份有限公司 使用低峰均功率比基序列的参考信号生成
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