WO2008082367A1 - Procédé de détermination d'un décalage de fréquence porteuse entre un émetteur et un récepteur - Google Patents

Procédé de détermination d'un décalage de fréquence porteuse entre un émetteur et un récepteur Download PDF

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Publication number
WO2008082367A1
WO2008082367A1 PCT/SG2008/000001 SG2008000001W WO2008082367A1 WO 2008082367 A1 WO2008082367 A1 WO 2008082367A1 SG 2008000001 W SG2008000001 W SG 2008000001W WO 2008082367 A1 WO2008082367 A1 WO 2008082367A1
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WIPO (PCT)
Prior art keywords
offset
carrier frequency
sampling interval
determined
frequency offset
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PCT/SG2008/000001
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English (en)
Inventor
Xiaoming Peng
Khiam Boon Png
Po Shin Francois Chin
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Agency For Science, Technology And Research
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Application filed by Agency For Science, Technology And Research filed Critical Agency For Science, Technology And Research
Priority to CN200880007350A priority Critical patent/CN101689953A/zh
Publication of WO2008082367A1 publication Critical patent/WO2008082367A1/fr

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2657—Carrier synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2668—Details of algorithms
    • H04L27/2673—Details of algorithms characterised by synchronisation parameters
    • H04L27/2676—Blind, i.e. without using known symbols

Definitions

  • Embodiments of the invention relate to the field of wireless communications.
  • embodiments of the invention relate to a method of determining a first carrier frequency offset between a transmitter and a receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band, a method of compensating a first carrier frequency offset between a transmitter and a receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band, and a method of compensating a sampling interval offset between a transmitter and a receiver, a first carrier frequency offset between the transmitter and the receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band, as well as corresponding communication devices.
  • carrier frequency offset refers to the frequency difference between the carrier of a received signal at a receiver and a demodulation signal generated at the receiver (from its own clock crystal oscillator) in order to demodulate the received signal.
  • a first conventional method [1] would be to first estimate the carrier frequency offset using the transmitted pilot symbols, and then to carry out the compensation of carrier frequency offset based on the estimated carrier frequency offset, for example.
  • the angle of the phase ramp is estimated using the four pilot tones embedded in each OFDM symbol.
  • the effect of the sampling interval offset is then compensated by multiplying the OFDM symbol with the inverse of the phase ramp.
  • a second conventional method [2] would be to use a best fit method to estimate the slope of the phase angle change with respect to symbol time.
  • the best fit method used is a weighted least squares best fit straight line.
  • a third conventional method [3] would be to jointly estimate the carrier frequency offset and the sampling interval offset.
  • This method involves using a specially designed preamble to estimate the carrier frequency offset and then compensate the said preamble in the time domain first. Following which, the compensated preamble is transformed into the frequency domain and then used to estimate the sampling interval offset.
  • the first conventional method in the case when the packet size of the transmitted data large, the solution suffers from the phase wrapping effect.
  • a method of determining a first carrier frequency offset between a transmitter and a receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band includes determining an estimate of the first carrier frequency offset, determining an estimate of the second carrier frequency offset, determining an estimate of a first sampling interval offset between the transmitter and the receiver for the first frequency band based on the determined estimate of the first carrier frequency offset, and determining an estimate of a second sampling interval offset between the transmitter and the receiver for the second frequency band based on the determined estimate of the second carrier frequency offset.
  • the method provided further includes determining an averaged sampling interval offset based on the calculation of an average of the determined estimate of the first sampling interval offset and the determined estimate of the second sampling interval offset, and determining the first carrier frequency offset and the second carrier frequency offset based on the determined averaged sampling interval offset.
  • Figure 1 shows a communication device according to one embodiment of the invention.
  • FIG 2 shows an illustration of how frequency ranges may be used by the communication system shown in Figure 1 according to one embodiment of the invention.
  • FIG 3 shows an illustration of the preamble structure used by the communication device shown in Figure 1 according to one embodiment of the invention.
  • Figure 4 shows a block diagram illustrating the implementation of one embodiment of the invention, where the sampling interval compensation is performed in the frequency domain.
  • Figure 5 shows a block diagram illustrating the implementation of one embodiment of the invention, where the sampling interval compensation is performed in the time domain.
  • Figure 6 shows the packet error rate (PER) performance comparison for the data rate mode of 53.3 Mbps in CM4, between conventional methods of estimating and compensating carrier frequency offset and embodiments of the invention.
  • PER packet error rate
  • Figure 7 shows an illustration of how the wrapping effects in the measured phase when the number of OFDM symbols is large will result in significant errors in the gradient estimation, in the first conventional method.
  • Figure 8 shows the packet error rate (PER) performance comparison for the data rate mode of 106.7 Mbps in CM4, between conventional methods of estimating and compensating carrier frequency offset and embodiments of the invention.
  • PER packet error rate
  • FIG. 9 shows the mean squared error (MSE) of the sampling interval offset estimation observed for the different data rate modes, according to one embodiment of the invention.
  • MSE mean squared error
  • Figure 10 shows the packet error rate (PER) performance comparison for the data rate mode of 200 Mbps in CM2, between conventional methods of estimating and compensating carrier frequency offset and embodiments of the invention.
  • PER packet error rate
  • Figure 11 shows the packet error rate (PER) performance comparison for the data rate mode of 480 Mbps in CM1 , between conventional methods of estimating and compensating carrier frequency offset and embodiments of the invention.
  • PER packet error rate
  • Figure 12 shows the packet error rate (PER) performance comparison for the data rate mode of 53.3 Mbps in CM4 (with shadowing), between a conventional method of estimating and compensating carrier frequency offset and one embodiment of the invention.
  • PER packet error rate
  • Figure 13 shows the packet error rate (PER) performance comparison for the data rate mode of 106.7 Mbps in CM4 (with shadowing), between a conventional method of estimating and compensating carrier frequency offset and one embodiment of the invention.
  • Figure 14 shows the packet error rate (PER) performance comparison for the data rate mode of 200 Mbps in CM2 (with shadowing), between a conventional method of estimating and compensating carrier frequency offset and one embodiment of the invention.
  • Figure 15 shows the packet error rate (PER) performance comparison for the data rate mode of 480 Mbps in CM1 (with shadowing), between a conventional method of estimating and compensating carrier frequency offset and one embodiment of the invention.
  • PER packet error rate
  • the frequency offset between the clock crystal oscillators in the transmitter and the receiver may be scaled based on the processing involved. For example, if the signal generated from the clock crystal oscillator has a higher frequency than the oscillating frequency of the clock crystal oscillator, the frequency offset between the signals generated in the transmitter and in the receiver is higher than the frequency offset between the clock crystal oscillators in the transmitter and the receiver.
  • a method of determining the carrier frequency offset between the transmitter and the receiver is developed by appropriately combining all the above discussed observations.
  • the carrier frequency offset is estimated in the time domain.
  • the estimated carrier frequency offset is scaled (or translated) accordingly to the estimated sampling interval offset.
  • the estimated sampling interval offsets are then averaged across the frequency bands, since the sampling interval offset for each frequency band is primarily caused by the frequency difference between the clock crystal oscillators in the transmitter and in the receiver. By so doing, a more accurate estimation of the sampling interval offset can be obtained.
  • sampling interval offset refers to the time difference between the sampling interval used in the sampling process of a transmitter in order to generate digital data carried by a signal to be transmitted, and the sampling interval used in the sampling process of a receiver of the transmitted signal in order to recover the digital data from the transmitted signal.
  • a sampling interval offset may be considered as corresponding to a sampling frequency offset.
  • the respective carrier frequency offsets for each frequency band are obtained by scaling the estimated sampling interval offset accordingly.
  • the respective carrier frequency offsets for each frequency band are then used in the process of carrier frequency offset compensation for each frequency band.
  • the process of carrier frequency offset compensation is typically carried out in the frequency domain.
  • the process of sampling interval offset compensation is carried out based on the estimated sampling interval offset. It should be noted that the process of sampling interval offset compensation may be carried out either in the frequency domain (using a frequency domain phase compensator, for example) or in the time domain (using a low complexity minimum mean square error (MMSE) interpolator, for example).
  • MMSE low complexity minimum mean square error
  • a method of determining a first carrier frequency offset between a transmitter and a receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band includes determining an estimate of the first carrier frequency offset, determining an estimate of the second carrier frequency offset, determining an estimate of a first sampling interval offset between the transmitter and the receiver for the first frequency band based on the determined estimate of the first carrier frequency offset, and determining an estimate of a second sampling interval offset between the transmitter and the receiver for the second frequency band based on the determined estimate of the second carrier frequency offset.
  • the method provided further includes determining an averaged sampling interval offset based on the calculation of an average of the determined estimate of the first sampling interval offset and the determined estimate of the second sampling interval offset, and determining the first carrier frequency offset and the second carrier frequency offset based on the determined averaged sampling interval offset.
  • the method provided includes determining an estimate of the carrier frequency offset between the transmitter and the receiver for each frequency band of a multiplicity of frequency bands, and determining an estimate of the sampling interval offset between the transmitter and the receiver for each frequency band in the multiplicity of frequency bands based on the determined estimate of the carrier frequency offset for each frequency band.
  • the method provided further includes determining an averaged sampling interval offset based on the calculation of an average of the determined estimates of the sampling interval offsets for each frequency band of the multiplicity of frequency bands, and determining a carrier frequency offset for each frequency band of the multiplicity of frequency bands based on the determined averaged sampling interval offset.
  • multiplicity refers to "three or more".
  • a multiplicity of frequency bands refers to three or more frequency bands.
  • the method provided may be extended to determine the carrier frequency offset for a multiplicity of frequency bands. In this case, an estimate of the carrier frequency offset between the transmitter and the receiver for each frequency band of a multiplicity of frequency bands is first determined. This means that if there are N frequency bands in the multiplicity of frequency bands, then there would be N estimates of the carrier frequency offset in total (one estimate of the carrier frequency offset for each frequency band).
  • the averaged sampling interval offset is then determined by averaging the N estimates of the carrier frequency offset. Finally, the respective carrier frequency offsets for each frequency band are determined based on the averaged sampling interval offset.
  • the first carrier frequency offset is determined by multiplying the determined averaged sampling interval offset with a first coefficient and the inverse of the sampling interval
  • the second carrier frequency offset is determined by multiplying the determined averaged sampling interval offset with a second coefficient and the inverse of the sampling interval
  • the estimate of the first sampling interval offset is determined by multiplying the determined estimate of the first carrier frequency offset with the sampling interval and the inverse of the first coefficient
  • the estimate of the second sampling interval offset is determined by multiplying the determined estimate of the second carrier frequency offset with the sampling interval and the inverse of the second coefficient
  • the ratio between the first coefficient and the second coefficient is about the same as the ratio between the carrier frequency of the first frequency band and the carrier frequency of the second frequency band.
  • the first coefficient and the second coefficient are respectively selected from a group of numbers in the range from 0 to 255.
  • the averaged sampling interval offset is calculated as an arithmetic average of the determined estimates of the sampling interval offsets for each frequency band.
  • a method of compensating a first carrier frequency offset between a transmitter and a receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band includes determining an estimate of the first carrier frequency offset, determining an estimate of the second carrier frequency offset, determining an estimate of a first sampling interval offset for the first frequency band based on the determined estimate of the first carrier frequency offset, and determining an estimate of a second sampling interval offset for the second frequency band based on the determined estimate of the second carrier frequency offset.
  • the method provided further includes determining an averaged sampling interval offset based on the calculation of an average of the determined estimate of the first sampling interval offset and the determined estimate of the second sampling interval offset, determining the first carrier frequency offset and the second carrier frequency offset based on the determined averaged sampling interval offset, and compensating the first carrier frequency offset and the second carrier frequency offset based on the determined first carrier frequency offset and the determined second carrier frequency offset.
  • compensating the first carrier frequency offset and the second carrier frequency offset includes determining a first phase rotation based on the determined first carrier frequency offset, determining a second phase rotation based on the determined second carrier frequency offset, compensating the determined first phase rotation, and compensating the determined second phase rotation.
  • compensating the first carrier frequency offset and the second carrier frequency offset further includes determining a third coefficient iteratively, determining the residual first carrier frequency offset based on the determined third coefficient, determining the residual second carrier frequency offset based on the determined third coefficient, and compensating the determined residual first carrier frequency offset and the determined residual second carrier frequency offset.
  • a method of compensating a sampling interval offset between a transmitter and a receiver, a first carrier frequency offset between the transmitter and the receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band includes determining an estimate of the first carrier frequency offset, determining an estimate of the second carrier frequency offset, determining an estimate of a first sampling interval offset for the first frequency band based on the determined estimate of the first carrier frequency offset, and determining an estimate of a second sampling interval offset for the second frequency band based on the determined estimate of the second carrier frequency offset.
  • the method provided further includes determining an averaged sampling interval offset based on the calculation of an average of the determined estimate of the first sampling interval offset and the determined estimate of the second sampling interval offset, determining the first carrier frequency offset and the second carrier frequency offset based on the determined averaged sampling interval offset, compensating the first carrier frequency offset and the second carrier frequency offset based on the determined first carrier frequency offset and the determined second carrier frequency offset, and compensating the sampling interval offset based on the determined averaged sampling interval offset.
  • compensating the first carrier frequency offset and the second carrier frequency offset includes determining a first phase rotation based on the determined first carrier frequency offset, determining a second phase rotation based on the determined second carrier frequency offset, compensating the determined first phase rotation, and compensating the determined second phase rotation.
  • compensating the first carrier frequency offset and the second carrier frequency offset further includes determining a third coefficient iteratively, determining the residual first carrier frequency offset based on the determined third coefficient, determining the residual second carrier frequency offset based on the determined third coefficient, and compensating the determined residual first carrier frequency offset and the determined residual second carrier frequency offset.
  • compensating the sampling interval offset is carried out after the processed received signal has been further processed according to a transform.
  • the transform may be selected from a group including Fast Fourier Transform (FFT), Discrete Fourier Transform (DFT) and Discrete Cosine Transform (DCT).
  • compensating the sampling interval offset is carried out using a frequency domain compensator.
  • compensating the sampling interval offset includes determining a third phase rotation based on the determined averaged sampling interval offset, and compensating the determined third phase rotation.
  • compensating the sampling interval offset further includes determining the residual sampling interval offset, and compensating the determined residual first carrier frequency offset, the determined residual second carrier frequency offset and the residual sampling interval offset.
  • compensating the determined residual first carrier frequency offset, the determined residual second carrier frequency offset and the residual sampling interval offset includes determining a fourth phase rotation due to the residual first carrier frequency offset, the determined residual second carrier frequency offset and the residual sampling interval offset, and compensating the determined fourth phase rotation.
  • determining the fourth phase rotation due to the residual first carrier frequency offset, the determined residual second carrier frequency offset and the residual sampling interval offset is carried out using pilot symbols, preamble symbols or synchronization symbols.
  • compensating the sampling interval offset is carried out before the processed received signal is further processed according to the transform.
  • compensating the sampling interval offset is carried out using a time domain compensator.
  • the time domain compensator is a finite impulse response (FIR) filter.
  • the coefficients of the FIR filter are determined based on one algorithm selected from the group including the minimum mean square error (MMSE) algorithm, the linear interpolation algorithm, the polynomial- based interpolation algorithm, the Lagrange algorithm, the piece-wise polynomial interpolation algorithm, the spline interpolation algorithm, the trigonometric interpolation algorithm and the Discrete Fourier Transform (DFT) interpolation algorithm.
  • MMSE minimum mean square error
  • the linear interpolation algorithm the polynomial- based interpolation algorithm
  • the Lagrange algorithm the piece-wise polynomial interpolation algorithm
  • the spline interpolation algorithm the trigonometric interpolation algorithm
  • DFT Discrete Fourier Transform
  • a communication device includes a first determining unit for determining an estimate of a first carrier frequency offset between a transmitter and a receiver for a first frequency band, a second determining unit for determining an estimate of a second carrier frequency offset between the transmitter and the receiver for a second frequency band, a third determining unit for determining an estimate of a first sampling interval offset between the transmitter and the receiver for the first frequency band based on the determined estimate of the first carrier frequency offset, and a fourth determining unit for determining an estimate of a second sampling interval offset between the transmitter and the receiver for the second frequency band based on the determined estimate of the second carrier frequency offset.
  • the communication device further includes a fifth determining unit for determining an averaged sampling interval offset based on the calculation of an average of the determined estimate of the first sampling interval offset and the determined estimate of the second sampling interval offset, and a sixth determining unit for determining the first carrier frequency offset and the second carrier frequency offset based on the determined averaged sampling interval offset.
  • a communication device includes a first determining unit for determining an estimate of a first carrier frequency offset between a transmitter and a receiver for a first frequency band, a second determining unit for determining an estimate of a second carrier frequency offset between the transmitter and the receiver for a second frequency band, a third determining unit for determining an estimate of a first sampling interval offset between the transmitter and the receiver for the first frequency band based on the determined estimate of the first carrier frequency offset, and a fourth determining unit for determining an estimate of a second sampling interval offset between the transmitter and the receiver for the second frequency band based on the determined estimate of the second carrier frequency offset.
  • the communication device further includes a fifth determining unit for determining an averaged sampling interval offset based on the calculation of an average of the determined estimate of the first sampling interval offset and the determined estimate of the second sampling interval offset, a sixth determining unit for determining the first carrier frequency offset and the second carrier frequency offset based on the determined averaged sampling interval offset, and a compensating unit for compensating the first carrier frequency offset and the second carrier frequency offset based on the determined first carrier frequency offset and the determined second carrier frequency offset.
  • a communication device includes a first determining unit for determining an estimate of a first carrier frequency offset between a transmitter and a receiver for a first frequency band, a second determining unit for determining an estimate of a second carrier frequency offset between the transmitter and the receiver for a second frequency band, a third determining unit for determining an estimate of a first sampling interval offset between the transmitter and the receiver for the first frequency band based on the determined estimate of the first carrier frequency offset, and a fourth determining unit for determining an estimate of a second sampling interval offset between the transmitter and the receiver for the second frequency band based on the determined estimate of the second carrier frequency offset.
  • the communication device further includes a fifth determining unit for determining an averaged sampling interval offset based on the calculation of an average of the determined estimate of the first sampling interval offset and the determined estimate of the second sampling interval offset, a sixth determining unit for determining the first carrier frequency offset and the second carrier frequency offset based on the determined averaged sampling interval offset, a first compensating unit for compensating the first carrier frequency offset and the second carrier frequency offset based on the determined first carrier frequency offset and the determined second carrier frequency offset, and a second compensating unit for compensating the sampling interval offset based on the determined averaged sampling interval offset.
  • the communication device may be, but is not limited to, a wireless communication device, a high speed wireless communication device, an orthogonal frequency division multiplexing (OFDM) communication device, a multi-band (MB) OFDM communication device, a ultra-wideband (UWB) communication device, a WiMedia communication device or a Bluetooth communication device.
  • OFDM orthogonal frequency division multiplexing
  • MB multi-band OFDM
  • UWB ultra-wideband
  • WiMedia communication device or a Bluetooth communication device.
  • the embodiments which are described in the context of the method of determining a first carrier frequency offset between a transmitter and a receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band are analogously valid for the method of compensating a first carrier frequency offset between a transmitter and a receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band, the method of compensating a sampling interval offset between a transmitter and a receiver, a first carrier frequency offset between the transmitter and the receiver for a first frequency band and a second carrier frequency offset between the transmitter and the receiver for a second frequency band, and the respective communication devices, and vice versa.
  • Fig. 1 shows a communication device 100 according to one embodiment of the invention.
  • the communication device 100 includes a transmitter 101 and a receiver 103.
  • the transmitter 101 includes a transmit data source 105, a baseband processing block (TX) 107 and a radio frequency (RF) processing block (TX) 109.
  • the receiver 101 includes a RF processing block (RX) 113, a baseband processing block (RX) 115 and a receive data block 117.
  • the data generated by the transmit data source 105 is first processed by the baseband processing block (TX) 107.
  • the baseband processing block (TX) 107 may include, but is not limited to, a channel encoder, an interleaver, a data scrambler, a code spreader and a modulator, for example.
  • the processed data is then processed by the RF processing block (TX) 109 before being transmitted over a channel 111.
  • the RF processing block (TX) 109 may include, but is not limited to, a mixer, a frequency up-converter, a filter, an amplifier and an antenna system, for example.
  • the transmitted signal passes through the channel 111 before arriving at the receiver 103 side, as the received signal.
  • the transmitted signal may suffer degradation, such as additive white Gaussian noise (AWGN), fading, shadowing, reflection and refraction, for example.
  • AWGN additive white Gaussian noise
  • the received signal is first processed by the RF processing block (RX) 113, and then followed by the baseband processing block (RX) 115, in order to retrieve the data transmitted for the receive data block 117.
  • the RF processing block (RX) 113 may include, but is not limited to, an antenna system, a mixer, a frequency down-converter, a filter and an equalizer, for example.
  • the baseband processing block (RX) 115 may include, but is not limited to, a demodulator, a code de-spreader, a data de- scrambler, an de-interleaver and a channel decoder, for example.
  • each block on the transmitter has a corresponding block on the receiver.
  • the RF processing block (TX) 109 on the transmitter 101 has a corresponding block on the receiver 103, namely, the RF processing block (RX) 113.
  • the communication device 100 may represent an ultra-wideband (UWB) communication device (which operates based on the multi-band (MB) OFDM communication technology), for example.
  • UWB ultra-wideband
  • MB multi-band OFDM communication technology
  • Fig. 2 shows an illustration of the frequency bands used by the communication device 100 shown in Fig. 1 according to one embodiment of the invention.
  • Fig. 2 shows 3 frequency bands, the first frequency band 201 being from 3168 MHz to 3696 MHz, the second frequency band 203 being from 3696 MHz to 4224 MHz, and the third frequency band 205 being from 4224 MHz to 4752 MHz.
  • the ultra-wideband (UWB) communication device is based on the multi-band (MB) OFDM technology.
  • the ultra- wideband (UWB) communication device may be considered as a 3-band OFDM communication device.
  • Fig. 2 also shows a transmission period 207, being of 937.5 nanoseconds (ns), which consists of 3 OFDM symbol transmission periods 209 (each being of 312.5 ns).
  • Each OFDM symbol transmission period 209 comprises a cyclic prefix 211 (being of 60.6 ns) and a guard interval 213 (being of 9.5 ns).
  • the arrangement of the 3 symbol transmission periods 209 in the transmission period 207 is determined according to the preamble structure used in the ultra-wideband (UWB) system.
  • UWB ultra-wideband
  • An example of the preamble structure used is shown in Fig. 3.
  • Fig. 3 shows an illustration of the preamble structure 300 used by the communication device 100 shown in Fig. 1 according to one embodiment of the invention.
  • the preamble structure 300 is designed to assist receiver side algorithms to perform specific functions, such as synchronization, carrier frequency offset estimation and channel estimation, for example.
  • the preamble structure typically consists of two portions, a time domain portion and a frequency domain portion.
  • the time domain portion includes a packet synchronization sequence 301 and a frame synchronization sequence 303, while the frequency domain portion includes a channel estimation sequence 305.
  • the packet synchronization sequence 301 may be used for packet detection and acquisition, coarse carrier frequency estimation and coarse symbol timing estimation, for example.
  • the frame synchronization sequence 303 may be used for frame synchronization and for synchronizing the relevant receiver algorithms within the preamble transmission period, for example.
  • the channel estimation sequence 305 may be used for channel estimation, for example. It can be seen that the packet synchronization sequence 301 , the frame synchronization sequence 303 and the channel estimation sequence 305 consists of 21 OFDM symbols, 3 OFDM symbols and 6 OFDM symbols, respectively. Further, it can be seen that the preamble structure 300 requires a transmission period 307 of 9.375 microseconds ( ⁇ s), which is equivalent to 10 of the transmission periods 207 shown in Fig. 2.
  • the preamble structure shown in Fig. 3 is for Time Frequency Code (TFC) 2, which is used to determine the "hopping" sequence shown in Fig. 2 (where the first OFDM symbol is transmitted in the third frequency band 205, the second OFDM symbol is transmitted in the second frequency band 203 and the third OFDM symbol is transmitted in the first frequency band 201 ).
  • TFC Time Frequency Code
  • Fig. 4 shows a block diagram 400 illustrating the implementation of one embodiment of the invention, where the sampling interval compensation is performed in the frequency domain.
  • a signal transmitted from transmitter 401 passes through the wireless channel 403 before arriving at the receiver 405 (as the received signal).
  • the received signal may be considered to include additive white Gaussian noise (AWGN) 407.
  • AWGN additive white Gaussian noise
  • the receiver 405 includes a local oscillator unit 409, a low pass filter (LPF) and analog-to-digital (A/D) converter unit 411 , a carrier frequency offset (CFO) processing block 413, a sampling interval offset (SIO) processing block 415, a Fast Fourier Transform (FFT) unit 417, an equalizer 419, a phase compensation block 421 and a baseband processing block 423.
  • LPF low pass filter
  • A/D analog-to-digital
  • CFO carrier frequency offset
  • SIO sampling interval offset
  • FFT Fast Fourier Transform
  • the carrier frequency offset (CFO) processing block 413 includes a carrier frequency offset (CFO) estimator unit 425 and a carrier frequency offset (CFO) phase compensator unit 427.
  • the sampling interval offset (SIO) processing block 415 includes a sampling interval offset (SIO) estimator unit 429 and a sampling interval offset (SIO) phase compensator unit 431.
  • phase compensation block 421 includes a summation and phase extractor unit 433, a residual phase averaging unit 435 and a data phase compensator unit 437.
  • the baseband processing block 423 includes functional units such as an interleaver and a Viterbi decoder, for example.
  • the local oscillator unit 409 From the point of view on how the received signal is processed in the receiver 405, first, the local oscillator unit 409 generates and provides the carrier clock signal (of frequency f c ) in order to perform frequency down-conversion of the received signal. After being processed by the low pass filter and analog-to- digital converter unit 411 , parts of the preamble are then used for synchronization and for automatic gain control (AGC), for example.
  • AGC automatic gain control
  • carrier frequency offset estimator unit 425 for a ultra-wideband (UWB) system, 6 OFDM symbols (2 in each band) are typically used for carrier frequency offset estimation.
  • a conventional method of estimating the carrier frequency offset may be used, such as estimating an auto-correlation function based on the packet synchronization sequence in the preamble, for example.
  • the symbol timing is synchronized first in one embodiment.
  • symbol timing recovery is performed prior to carrier frequency offset estimation in one embodiment.
  • r ⁇ n represents the n th received sample within the /* OFDM symbol in band m
  • T is the sample duration.
  • the auto-correlation 127 function refers to the term r * n r ⁇ +U ⁇ in Equation (1 ).
  • Equation (1 ) refers to the number of samples which separates the current timing synchronization sequence to the previous timing synchronization sequence within the same band for Time Frequency Code (TFC) 1.
  • Equation (1 ) an estimate of the carrier frequency offset should be obtained for each frequency band.
  • the next step of processing is performed in the sampling interval offset (SIO) estimator unit 429.
  • the estimated carrier frequency offset obtained from the carrier frequency offset (CFO) estimator unit 425) is scaled (or translated) accordingly to an estimated sampling interval offset for each frequency band.
  • An average of the estimated sampling interval offset for each frequency band is then obtained. This is done based on the observation that the estimated sampling interval offset for each frequency band is caused by the same frequency offset between the clock crystal oscillators in the transmitter 401 and the receiver 405.
  • the averaged sampling interval offset is then fed back to the carrier frequency offset (CFO) estimator unit 425.
  • the respective carrier frequency offsets for each frequency band are then obtained by scaling the averaged sampling interval offset accordingly.
  • the carrier frequency offset for each frequency band is then used to perform carrier frequency offset compensation for each frequency band, for the preamble portions used for channel estimation and for data. This processing step is performed in the carrier frequency offset (CFO) phase compensator unit 427.
  • CFO carrier frequency offset
  • the averaged sampling interval offset (obtained from the sampling interval offset (SIO) estimator unit 429) is also used by the sampling interval offset (SIO) phase compensator unit 431 , to perform sampling interval offset compensation for the preamble portions used for channel estimation and for data (in the frequency domain).
  • the estimate of the sampling interval offset in the frequency domain is obtained in this embodiment of the invention by scaling the estimated carrier frequency offset and then performing an averaging of the results of the scaling.
  • the residual carrier frequency offset and the residual sampling interval offset may estimated and then compensated, using conventional methods based on the pilot signals transmitted in predetermined sub-carriers within each OFDM symbol, for example.
  • An example of a method for estimating and compensating the residual carrier frequency offset and the residual sampling interval offset may be the following.
  • the residual carrier frequency offset and the residual sampling interval offset cause a phase rotation in the received signal samples.
  • This phase rotation increases with time.
  • this phase rotation is tracked and compensated. Since pilot signals are transmitted in predetermined sub-carriers within each OFDM symbol, these pilot signals may be used for tracking and compensating the said phase rotation.
  • the transmitted signal, the received signal and the estimated channel response at pilot sub-carrier k are** , , y k l , and h k , respectively, for the f h
  • the estimated phase rotation may then be used to compensate for the phase rotation of the current OFDM symbol.
  • the estimated residual carrier frequency offset and the estimated residual sampling interval offset may also be averaged as well. This can be done because the residual carrier frequency offset and the residual sampling interval offset are correlated from one frequency band to another, based on the earlier discussed observations. Finally, after compensating the residual carrier frequency offset and the residual sampling interval offset in the data phase compensator unit 437 and further processing in the baseband processing block 423, the received data is obtained.
  • the effects of the carrier frequency offset and the sampling interval offset on each sub-carrier in an ultra-wideband (UWB) system may be represented mathematically as follows.
  • the time domain baseband received signal may be written as
  • a l k is the transmitted symbol at the /c" 1 sub-carrier of the f h OFDM symbol
  • H k is the frequency-domain channel response for the k? h sub-carrier
  • T is the sampling interval at the transmitter
  • N is the number of inverse Fast Fourier Transform (IFFT) samples
  • M • T is the total duration of the OFDM symbol including the zero prefix duration
  • ⁇ (t) is the additive white
  • AWGN Gaussian noise
  • the sampling interval offset, ⁇ T may be a positive number or a negative number, depending on whether the sampling clock signal at the receiver 405 has a higher frequency or a lower frequency compared to that of the transmitter 401.
  • Equation (4) it can be seen from Equation (4) that the term A corresponds to the phase distortion due to the sampling interval offset, the term B corresponds to the phase distortion due to the carrier frequency offset and the term C corresponds to the phase distortion due to the combined effect of the carrier frequency offset and the sampling interval offset.
  • the preamble is used to estimate the carrier frequency offset, Af 0 .
  • MB OFDM systems including ultra-wideband (UWB) systems
  • UWB ultra-wideband
  • the respective estimates of the carrier frequency offset ⁇ / c obtained for each frequency band be ⁇ / c) , Af c2 and ⁇ / c3 .
  • the carrier frequency offset for each frequency band is also different.
  • the sampling interval offset, AT is approximately a constant value for all frequency bands, since the sampling interval offset, ⁇ , is caused by the frequency offset between the clock crystal oscillators in the
  • the quantity — may be denoted as ⁇ ,
  • the quantity ⁇ is typically specified as ⁇ ⁇ 40ppm.
  • the relationship between the carrier frequency offset and the sampling interval offset may be written as follows
  • Af, ⁇ f, .
  • Af ci , ⁇ / c2 , ⁇ / c3 are the carrier frequency offset for the first frequency band (1 ), the second frequency band (2) and the third frequency band (3) respectively
  • / cl , f c2 , / c3 are the carrier frequency for the first frequency band (1 ), the second frequency band (2) and the third frequency band (3), respectively.
  • the estimated carrier frequency offset ⁇ / el , ⁇ / c2 and ⁇ / c3 may be obtained for each frequency band, for example, using either Equation (5) or Equation (6).
  • the carrier frequency offset ⁇ / c and the sampling interval offset AT may be obtained as follows.
  • the estimated sampling interval offset for the first frequency band AT 1 , the estimated sampling interval offset for the second frequency band AT 2 and the estimated sampling interval offset for the third frequency band ⁇ 3 may be obtained as
  • AT ⁇ Afc l T
  • AT 2 ⁇ Afc L T 1
  • AT ⁇ Afc l T (7)
  • Equation (7) may not be easily implemented. Additionally, since Af c « f c , and ⁇ f « T , Equation (7) may be simplified as follows A f 1 - A/c3 j > (8)
  • the coefficients Qr 1 - Qf 3 may be used to replace f c - ⁇ f 0 and such a
  • the coefficients a x - a ⁇ may be selected to be approximately in the same order as f c - ⁇ f 0 .
  • the second coefficient ⁇ 2 and the third coefficient a 3 may be selected to be approximately the same as the ratio between the carrier frequency of the first frequency band / cl , the carrier frequency of the second frequency band f c2 and the carrier frequency of the third frequency band / c3 (/ d : / c2 : / e3 ).
  • the ratio between the first coefficient a x , the second coefficient ⁇ and the third coefficient ⁇ 3 , ( ⁇ , : a 2 : ⁇ 3 ) may be selected as
  • Equation (9) is not used directly.
  • the simplification used to obtain Equation (9) from Equation (8) may be applied to the averaging of the sampling interval offsets for each frequency band, as shown below.
  • the averaged sampling interval offset AT may be obtained by averaging the estimated sampling interval offset for all three frequency bands, namely, ⁇ 7; , AT 2 and AT 3 as where N 5 is the number of frequency bands.
  • Equation (10) may then be reduced to
  • the carrier frequency offset for each frequency band may be obtained as follows
  • Equation (12) may be reduced to
  • the received signal in the time domain may be written as
  • sampling interval offset compensator there are two approaches to implement the sampling interval offset compensator, either as a frequency domain compensator or as a time domain compensator. Since the block diagram 400 is also an illustration of the implementation of a frequency domain compensator for the sampling interval offset, the frequency domain compensator is discussed here. The time domain compensator will be discussed subsequently in relation to Fig. 5 later.
  • the received time domain samples r ⁇ zier are then passed through the
  • the received signal at the ft" 1 sub-carrier of the t h OFDM symbol may then be described as *U - l2 ⁇ —
  • ⁇ i m is assumed to be an independently and identically distributed (i.i.d.)
  • ICI inter-carrier interference
  • AWGN additive white Gaussian noise
  • the combined effects due to the residual carrier frequency offset and the sampling interval offset may be considered as a phase rotation, which is dependent on the sub-carrier index k and the OFDM symbol index / ' .
  • the received data in frequency domain in the presence of the sampling interval offset and the residual carrier frequency offset may then be approximated as 2 —- M
  • V 1 k is the additive white Gaussian noise (AWGN)
  • F corresponds to the phase distortion due to the sampling interval offset
  • G corresponds to the phase distortion due to the combined effect of the residual frequency offset and the sampling interval offset.
  • AWGN additive white Gaussian noise
  • the received signal in the frequency domain may be written as
  • V is same as the term G of Equation (18), because the sampling interval offset compensation does not affect this term. This means that ⁇ r in term V is not compensated for, since it is already part of the phase information.
  • phase distortion in the pilot of each OFDM symbol is only due to the residual sampling interval offset and the residual carrier frequency offset.
  • the phase distortion in the terms U and V may be compensated by using the conventional phase compensation algorithms based on the pilot symbols in each OFDM symbol.
  • An example of such a conventional phase compensation algorithm is described in earlier in relation to Equation (2).
  • the carrier frequency offset obtained for each frequency band is also correlated.
  • the carrier frequency offset between the transmitter 401 and the receiver 405 is mainly due to the frequency difference between the respective clock crystal oscillators of the transmitter 401 and the receiver 405.
  • the residual carrier frequency offset and the residual sampling interval offset are 5 correlated from one frequency band to another.
  • the residual phase distortion is considered to be random and independent from one frequency band to another. As such, the phase compensation is performed on each frequency 10 band as well.
  • the average of the estimated carrier frequency offset and the sampling interval offset across frequency bands is used.
  • a new method to estimate and compensate the phase 15 distortion due to the residual carrier frequency offset and the residual sampling interval offset may be used, as shown in the following.
  • the respective residual carrier frequency offset in each frequency band may be given by
  • Equation (20) may be rewritten as
  • ⁇ v the index of sub-carrier.
  • 2 varies from between 0 to 15%.
  • ⁇ ⁇ the absolute values for both ⁇ ⁇ and ⁇ v are relative small.
  • f c is the carrier frequency for the corresponding frequency band.
  • ⁇ is a value which is independent from the index of frequency bands and the index of OFDM symbols.
  • a weighted average of the estimates of ⁇ may be calculated from the preceding OFDM symbols across the frequency bands, and then used to compensate for the residual phased.
  • ⁇ may be iteratively updated in order to obtain a more accurate estimate, since the number of OFDM symbols received has also increased.
  • ⁇ represent the estimate of ⁇ for the t h OFDM symbol derived from ⁇ i (which may be estimated using the conventional method discussed earlier in relation to Equation (2), for example).
  • the weighted average ⁇ i may be given by
  • Equation (22) is now represented in Equation (23) as an iteratively updated estimate, with the initial value
  • Fig. 5 shows a block diagram 500 illustrating the implementation of one embodiment of the invention, where the sampling interval compensation is performed in the time domain.
  • the block diagram 500 has a number of blocks and units which are the same as those as the block diagram 400 shown in Fig. 4. As such, the labels used for such blocks and units would have the same two rightmost digits.
  • the carrier frequency offset (CFO) processing block is labeled as 425 in Fig. 4 and as 525 in Fig. 5.
  • the block diagram 500 When compared to the block diagram 400, the block diagram 500 does not include a phase compensation block (421 ) or a sampling interval offset (SIO) phase compensator unit (431 ). Instead, the block diagram 500 has a minimum mean square error (MMSE) interpolator unit 539. In this context, the minimum mean square error (MMSE) interpolator unit 539 is used for compensating the averaged sampling interval offset.
  • MMSE minimum mean square error
  • block diagram 500 the entire process of estimating and compensating the carrier frequency offset and the averaged sampling interval offset is performed before the Fast Fourier Transform (FFT) unit. As such, the sampling interval compensation is considered as being performed in the time domain.
  • FFT Fast Fourier Transform
  • any time domain compensator may be used in place of the minimum mean square error (MMSE) interpolator unit 539.
  • MMSE minimum mean square error
  • the time domain compensator may be a finite impulse response (FIR) filter.
  • the coefficients of the FIR filter are determined based on one algorithm selected from the group including the minimum mean square error (MMSE) algorithm, the linear interpolation algorithm, the polynomial-based interpolation algorithm, the Lagrange algorithm, the piece-wise polynomial interpolation algorithm, the spline interpolation algorithm, the trigonometric interpolation algorithm and the Discrete Fourier Transform (DFT) interpolation algorithm.
  • MMSE minimum mean square error
  • DFT Discrete Fourier Transform
  • the minimum mean square error (MMSE) interpolator unit 539 is simply a FIR filter, which employs the minimum mean square error (MMSE) algorithm in order to determine the FIR filter coefficients.
  • MMSE minimum mean square error
  • the received time domain samples r ⁇ n are passed through a digital finite impulse response (FIR) filter.
  • FIR digital finite impulse response
  • a digital interpolation is then carried out based on the criterion of minimizing the mean squared error between the impulse response of an ideal filter and the (approximate) interpolated signal.
  • the digital interpolator is simply a variable ⁇ / / r //r tap FIR filter which has been designed to interpolate the correct sampled values from the incorrectly sampled data.
  • the digital interpolator may be designed to interpolate for a fixed resolution of sampling offset, i.e., the digital interpolator only works at the predetermined resolutions. For example, if the digital interpolator's predetermined resolution is 1/8 of the sampling period, T, then the digital interpolator would only be able to
  • a counter may be used to calculate the estimated accumulated timing drift
  • the digital interpolator would be activated using the predetermined filter coefficients for the said amount of timing drift.
  • the digital interpolation will be carried out using predetermined filter coefficients, until the counter informs the digital interpolator that sufficient samples have been digitally interpolated, and that it is time to switch over to a new set of
  • T predetermined filter coefficients to compensate for a timing drift of— .
  • sampling interval offset compensation process may be described as follows.
  • timing drift of — may be calculated as — ⁇ l ⁇ 6 .
  • 8 4 counter may be used to keep track of the number of samples which have
  • control signal is sent to the FIR filter to load the next set of filter
  • the interpolated samples are then passed to the baseband processing block 523 for subsequent processing.
  • System B uses the earlier described second conventional method [2], with some adaptations made so that it may be applied to an ultra-wideband (UWB) system (or a multi-band (MB) OFDM system).
  • UWB ultra-wideband
  • MB multi-band
  • the number of OFDM symbols used for carrier frequency offset estimation is limited to only 60 (20 per frequency band) in the simulations.
  • System C uses the earlier described third conventional method [3], again with the appropriate modifications made so that it may be applied to a ultra- wideband (UWB) system (or a multi-band (MB) OFDM system).
  • the number of preambles used in System C for carrier frequency offset estimation is limited to only 30 (10 per band). This is comparable to existing ultra-wideband (UWB) systems, where 24 preambles are typically used.
  • the channel models (CM) used in the simulations are all taken from IEEE 802.15.3a [4]. In this context, 100 channel realizations are generated for each simulation. Simulations are then carried out for 50 data packets (each of 1 Kbytes in size) over the 100 channel realizations.
  • the packet error rate (PER) is then calculated using the remaining 90 channel realizations.
  • the calculated packet error rate (PER) is used for the simulation results shown in Figs. 6 to 15.
  • the different data rate modes (such as 480Mbps, 200Mbps, 106.7Mbps and 53.3 Mbps, for example) have been simulated in the likely operating channel models, CM1 to CM4 [4].
  • Figs. 6, 8, 10 and 11 show the simulation results for the channel models without shadowing
  • Figs. 12 to 15 show the simulation results for the channel models with shadowing.
  • the performance results for System A, System B, System C and the ideal system have been included in Figs. 8, 10 and 11.
  • the performance results for System C and the ideal system have also been included in Figs. 12 to 15 for the same reason.
  • the performance results for System A and the ideal system have been included in Fig. 6 as well.
  • the respective performance results for System A, System B, System C and the ideal system would be labeled as n01 , nO3, nO5 and nO7, where n corresponds to the Figure number.
  • Fig. 6 shows the packet error rate (PER) performance comparison for the data rate mode of 53.3 Mbps in CM4, between conventional methods of estimating and compensating carrier frequency offset and embodiments of the invention.
  • PER packet error rate
  • an averaging of the residual carrier frequency offset and the residual sampling interval offset may be carried out across the frequency bands.
  • the performance results of the embodiment of the invention which carry out such an averaging is labeled as 611 in Fig. 6. Additionally, in order to demonstrate the effects of such an averaging, the performance results for the embodiment where the said averaging is not carried out for the residual carrier frequency offset and the residual sampling interval offset (609) is also included in Fig. 6.
  • the embodiment without carrying out the averaging shows a performance loss of about 2 dB, when compared to that for the ideal system.
  • the embodiment with the averaging being carried out only shows a performance loss of about 0.5 dB, when compared to that for the ideal system.
  • there is an about 1.5dB performance improvement which may be obtained by carrying out the averaging of the residual carrier frequency offset and the residual sampling interval offset across the frequency bands.
  • the embodiments of the invention show a performance improvement over System A.
  • Fig. 7 shows an illustration of how the wrapping effects in the measured phase when the number of OFDM symbols is large will result in significant errors in the gradient estimation, in the first conventional method.
  • the wrapping effects in the measured phase is illustrated by the labeled lines 701 , 703 and 705.
  • the measured phase of a sub-carrier index approaches the value of -/7
  • the measured phase of the next sub-carrier index would have a value of about /7. This is referred to as the wrapping effect.
  • the best fit line determined (which may be the weighted least squares best fit straight line, for example) is the labeled line 707. It is clear that line 707 would consistently give the wrong estimate on the gradient of the measured phase.
  • Fig. 7 shows the packet error rate (PER) performance comparison for the data rate mode of 106.7 Mbps in CM4, between conventional methods of estimating and compensating carrier frequency offset and embodiments of the invention.
  • PER packet error rate
  • FIG. 8 It can be seen from Fig. 8 that the embodiments of the invention (809, 811 ) show a performance improvement compared to that of all three conventional methods (System A (801 ), System B (803) and System C (805)).
  • Fig. 9 shows the mean squared error (MSE) of the sampling interval offset estimation observed for the different data rate modes, according to one embodiment of the invention. It should be noted that each data rate mode has a different operating range based on the signal energy to noise ratio (Eb/No) parameter.
  • MSE mean squared error
  • the operating range is from about 3dB to about 6dB (for the bit energy to noise ratio (Eb/No) parameter), which corresponds to the signal energy to noise ratio (Es/No) parameter range from about -7.5dB to about -5dB.
  • Eb/No bit energy to noise ratio
  • the mean squared error (MSE) of the sampling interval offset estimation observed within the said range for the Es/No parameter is in the range of from about 2x10 12 to about
  • the operating range is from about 4dB to about 7dB (for the bit energy to noise ratio (Eb/No) parameter), which corresponds to the signal energy to noise ratio (Es/No) parameter range from about -4.OdB to about -0.5dB.
  • Eb/No bit energy to noise ratio
  • the mean squared error (MSE) of the sampling interval offset estimation observed within the said range for the Es/No parameter is in the range of from about 3x10 13 to about 10x10 13 .
  • the operating range is from about 5dB to about 8dB (for the bit energy to noise ratio (Eb/No) parameter), which corresponds to the signal energy to noise ratio (Es/No) parameter range from about OdB to about 2.5dB.
  • Eb/No bit energy to noise ratio
  • the mean squared error (MSE) of the sampling interval offset estimation observed within the said range for the Es/No parameter is in the range of from about 1x10 "13 to about 2.5x10 13 .
  • the operating range is from about 7dB to about 1OdB (for the bit energy to noise ratio (Eb/No) parameter), which corresponds to the signal energy to noise ratio (Es/No) parameter range from about 6dB to about 8.5dB.
  • Eb/No bit energy to noise ratio
  • the mean squared error (MSE) of the sampling interval offset estimation observed within the said range for the Es/No parameter is in the range of from about 0.7x10 '13 to about 0.8x10 "13 .
  • the mean squared error (MSE) of the sampling interval offset estimation observed for 53.3Mbps is about 10 times greater than the one observed for 106.7Mbps. It should be noted that with larger values of the mean squared error (MSE) of the sampling interval offset estimation, the step of averaging of the residual carrier frequency offset and the residual sampling interval offset has greater effect on the performance improvement obtained.
  • Fig. 10 shows the packet error rate (PER) performance comparison for the data rate mode of 200 Mbps in CM2, between conventional methods of estimating and compensating carrier frequency offset and embodiments of the invention.
  • PER packet error rate
  • the performance improvement due to the averaging of the residual carrier frequency offset and the residual sampling interval offset being carried out is only 0.2dB. This performance improvement is smaller than the one observed for the data rate mode of 53.3Mbps over that for the data rate mode of 106.7Mbps (OJdB). This observation may be explained as follows, based on Fig. 9.
  • the mean squared error (MSE) of the sampling interval offset estimation observed within its said operating range is in the range of from about 3x10 "13 to about 7x10 13 .
  • This MSE range is very small, and therefore, the effects of the averaging of the residual carrier frequency offset and the residual sampling interval offset being carried out on the performance improvement obtained is not significant.
  • Fig. 11 shows the packet error rate (PER) performance comparison for the data rate mode of 480 Mbps in CM1 , between conventional methods of estimating and compensating carrier frequency offset and embodiments of the invention.
  • the mean squared error (MSE) of the sampling interval offset estimation observed within its said operating range is in the range of from about 0.7x10 "13 to about 2.5x10 "13 .
  • This MSE range is very small, and therefore, the effects of the averaging of the residual carrier frequency offset and the residual sampling interval offset being carried out on the performance improvement obtained is not significant.
  • the performance improvement of the embodiments of the invention over the three systems would become less significant when a higher data rate mode is used. This is because as the data rate increases, the resultant decreased packet size causes less severe distortion due to the sampling interval offset.
  • FIGs. 12 to 15 would show the performance comparison between a conventional method of estimating and compensating carrier frequency offset (System C) and one embodiment of the invention, over a multi-path channel with shadowing. Further, it should be noted that the averaging of the residual carrier frequency offset and the residual sampling interval offset is not carried out in the embodiment of the invention.
  • Fig. 12 shows the packet error rate (PER) performance comparison for the data rate mode of 53.3 Mbps in CM4 (with shadowing), between a conventional method of estimating and compensating carrier frequency offset and one embodiment of the invention.
  • PER packet error rate
  • Fig. 13 shows the packet error rate (PER) performance comparison for the data rate mode of 106.7 Mbps in CM4 (with shadowing), between a conventional method of estimating and compensating carrier frequency offset and one embodiment of the invention.
  • PER packet error rate
  • FIG. 13 It can be seen from Fig. 13 that the embodiment of the invention show better performance compared to that of System C. In more detail, the embodiment of the invention requires about 6dB less (in terms of Eb/No) compared to System C.
  • the embodiment of the invention requires only about 0.6dB more (in terms of Eb/No) compared to the ideal system. This implies that the embodiment of the invention can almost compensate fully for the sampling interval offset.
  • Fig. 14 shows the packet error rate (PER) performance comparison for the data rate of 200 Mbps in CM2 (with shadowing), between a conventional method of estimating and compensating carrier frequency offset and one embodiment of the invention.
  • PER packet error rate
  • FIG. 14 It can be seen from Fig. 14 that the embodiment of the invention show better performance compared to that of System C. In more detail, the embodiment of the invention requires about 1.8dB less (in terms of Eb/No) compared to System C.
  • the embodiment of the invention requires only about 0.3dB more (in terms of Eb/No) compared to the ideal system.
  • Fig. 15 shows the packet error rate (PER) performance comparison for the data rate of 480 Mbps in CM1 (with shadowing), between a conventional method of estimating and compensating carrier frequency offset and one embodiment of the invention.
  • PER packet error rate
  • Embodiments of the invention have the following effects.
  • embodiments of the invention typically show better performance compared to the existing conventional methods.
  • the embodiments of the invention are easy to implement. As such, the embodiments of the invention show great promise as effective solutions for the problem of estimating and compensating the carrier frequency offset as well as the sampling interval offset.

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Abstract

La présente invention concerne un procédé de détermination d'un premier décalage de fréquence porteuse entre un émetteur et un récepteur pour une première bande de fréquence et d'un second décalage de fréquence porteuse entre un émetteur et un récepteur pour une seconde bande de fréquence. Le procédé selon l'invention comprend la détermination d'une estimation du premier décalage de fréquence porteuse, la détermination d'une estimation du second décalage de fréquence porteuse, la détermination d'une estimation d'un premier décalage d'intervalle d'échantillonnage entre l'émetteur et le récepteur pour la première bande de fréquence en fonction de l'estimation déterminée du premier décalage de fréquence porteuse, et la détermination d'une estimation d'un second décalage d'intervalle d'échantillonnage entre l'émetteur et le récepteur pour la seconde bande de fréquence en fonction de l'estimation déterminée du second décalage de fréquence porteuse. Le procédé comprend également la détermination d'un décalage d'intervalle d'échantillonnage moyenné en fonction du calcul d'une moyenne de l'estimation déterminée du premier décalage d'échantillonnage déterminé et de l'estimation déterminée du second décalage d'échantillonnage déterminé, et la détermination du premier décalage de fréquence porteuse et du second décalage de fréquence porteuse en fonction du décalage d'intervalle d'échantillonnage moyenné déterminé.
PCT/SG2008/000001 2007-01-05 2008-01-04 Procédé de détermination d'un décalage de fréquence porteuse entre un émetteur et un récepteur WO2008082367A1 (fr)

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WO2023014272A1 (fr) * 2021-08-06 2023-02-09 Telefonaktiebolaget Lm Ericsson (Publ) Test de transmission par cohérence de dispositif sans fil
CN114338325A (zh) * 2021-12-24 2022-04-12 深圳市联平半导体有限公司 载波频偏和采样频偏的确定方法及装置
CN114338325B (zh) * 2021-12-24 2023-07-18 深圳市联平半导体有限公司 载波频偏和采样频偏的确定方法及装置

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