WO2008008446A2 - Dispositif de protection de circuit de moteur électronique - Google Patents

Dispositif de protection de circuit de moteur électronique Download PDF

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Publication number
WO2008008446A2
WO2008008446A2 PCT/US2007/015914 US2007015914W WO2008008446A2 WO 2008008446 A2 WO2008008446 A2 WO 2008008446A2 US 2007015914 W US2007015914 W US 2007015914W WO 2008008446 A2 WO2008008446 A2 WO 2008008446A2
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WO
WIPO (PCT)
Prior art keywords
trip
current
circuit
voltage
threshold
Prior art date
Application number
PCT/US2007/015914
Other languages
English (en)
Other versions
WO2008008446A3 (fr
Inventor
Susan Jean Walker Colsch
Jason Robert Colsch
Ignacio Dapic
William Davison
David Joseph Dunne
Dennis W. Fleege
Kevin John Malo
Steven M. Meehleder
Ryan James Moffitt
Paul Andrew Reid
Marco Antonio Ramirez Rodriguez
Ii Richard Allen Studer
Gary Michael Stumme
Jr. James G. Tipton
Original Assignee
Square D Company
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from US11/818,679 external-priority patent/US8154373B2/en
Priority claimed from US11/824,652 external-priority patent/US7788055B2/en
Priority claimed from US11/824,683 external-priority patent/US7683586B2/en
Priority claimed from US11/824,682 external-priority patent/US7791849B2/en
Priority claimed from US11/824,684 external-priority patent/US7592888B2/en
Priority claimed from US11/824,680 external-priority patent/US7859802B2/en
Priority claimed from US11/824,681 external-priority patent/US7550939B2/en
Priority claimed from US11/824,693 external-priority patent/US7697250B2/en
Priority claimed from US11/824,651 external-priority patent/US7869169B2/en
Priority claimed from US11/824,654 external-priority patent/US7869170B2/en
Application filed by Square D Company filed Critical Square D Company
Publication of WO2008008446A2 publication Critical patent/WO2008008446A2/fr
Publication of WO2008008446A3 publication Critical patent/WO2008008446A3/fr

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Classifications

    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F38/00—Adaptations of transformers or inductances for specific applications or functions
    • H01F38/20—Instruments transformers
    • H01F38/22—Instruments transformers for single phase ac
    • H01F38/28—Current transformers
    • H01F38/30—Constructions
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01H—ELECTRIC SWITCHES; RELAYS; SELECTORS; EMERGENCY PROTECTIVE DEVICES
    • H01H71/00—Details of the protective switches or relays covered by groups H01H73/00 - H01H83/00
    • H01H71/10—Operating or release mechanisms
    • H01H71/12—Automatic release mechanisms with or without manual release
    • H01H71/123—Automatic release mechanisms with or without manual release using a solid-state trip unit
    • H01H71/125—Automatic release mechanisms with or without manual release using a solid-state trip unit characterised by sensing elements, e.g. current transformers
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02H—EMERGENCY PROTECTIVE CIRCUIT ARRANGEMENTS
    • H02H3/00—Emergency protective circuit arrangements for automatic disconnection directly responsive to an undesired change from normal electric working condition with or without subsequent reconnection ; integrated protection
    • H02H3/02—Details
    • H02H3/04—Details with warning or supervision in addition to disconnection, e.g. for indicating that protective apparatus has functioned
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F27/00—Details of transformers or inductances, in general
    • H01F27/02—Casings
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F27/00—Details of transformers or inductances, in general
    • H01F27/28—Coils; Windings; Conductive connections
    • H01F27/30—Fastening or clamping coils, windings, or parts thereof together; Fastening or mounting coils or windings on core, casing, or other support
    • H01F27/306—Fastening or mounting coils or windings on core, casing or other support
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F3/00—Cores, Yokes, or armatures
    • H01F3/02—Cores, Yokes, or armatures made from sheets
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01F—MAGNETS; INDUCTANCES; TRANSFORMERS; SELECTION OF MATERIALS FOR THEIR MAGNETIC PROPERTIES
    • H01F3/00—Cores, Yokes, or armatures
    • H01F3/10—Composite arrangements of magnetic circuits
    • H01F3/14—Constrictions; Gaps, e.g. air-gaps
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01H—ELECTRIC SWITCHES; RELAYS; SELECTORS; EMERGENCY PROTECTIVE DEVICES
    • H01H71/00—Details of the protective switches or relays covered by groups H01H73/00 - H01H83/00
    • H01H71/74—Means for adjusting the conditions under which the device will function to provide protection
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02H—EMERGENCY PROTECTIVE CIRCUIT ARRANGEMENTS
    • H02H3/00—Emergency protective circuit arrangements for automatic disconnection directly responsive to an undesired change from normal electric working condition with or without subsequent reconnection ; integrated protection
    • H02H3/20—Emergency protective circuit arrangements for automatic disconnection directly responsive to an undesired change from normal electric working condition with or without subsequent reconnection ; integrated protection responsive to excess voltage

Definitions

  • a circuit breaker is an automatically operated electro-mechanical device designed to protect a load from damage caused by an overload or a short circuit.
  • a circuit breaker may be tripped by an overload or short circuit, which causes an interruption of power to the load.
  • a circuit breaker can be reset (either manually or automatically) to resume current flow to the load.
  • One application of circuit breakers is to protect motors as part of a motor control center (“MCC”).
  • MCC motor control center
  • a typical MCC includes a temperature triggered overload relay, a contactor and a motor circuit protector (“MCP").
  • MCP motor circuit protector
  • the MCP is a specialized circuit breaker that provides instantaneous protection against instantaneous short-circuit events. In the United States, these motor circuit protector devices must meet National Electric Code (“NEC”) requirements when installed as part of a UL-listed MCC to provide instantaneous short-circuit protection.
  • NEC National Electric Code
  • Mechanical circuit breakers energize an electro-magnetic device such as a solenoid to trip instantaneously in response to a rapid surge in current such as a short circuit.
  • MCPs protect only a limited range of motors, but should avoid tripping in response to in-rush motor currents that occur during motor start-up.
  • MCPs that sense relatively low currents may not be suitable for motors having a relatively low in-rush current because tripping will occur during normal operation of the motor.
  • MCPs that sense relatively high currents may not trip on relatively low current levels such as those corresponding to locked-rotor current levels. Because of their limited operating range, some existing MCPs cannot protect for both relatively low current levels and relatively high current levels.
  • Other existing MCPs that can protect against a wider range of fault currents are very large and their current transformers require large volumes of steel to remain in their linear range of operation.
  • Some circuit breakers include a current transformer, along with other electrical components, to make up the breaker system.
  • current transformers used in existing circuit breaker devices are designed to supply power to trip unit electronics, or to sense low current ranges, or to sense high current ranges, and have a limited operating range.
  • current transformer devices designed to sense low fault currents cannot effectively sense high fault currents.
  • An additional current transformer specifically designed for supplying power to the trip unit electronics must be incorporated into the circuit breaker, increasing its size, complexity, and cost.
  • current transformer devices designed to sense high fault currents cannot effectively sense low fault currents.
  • MCP device with user-adjustable and automatically configurable trip point settings over a broad range of current ratings.
  • circuit protection device that couples a mechanical adjustment button and a potentiometer for adjusting trip levels of an electrical circuit.
  • Fault currents are sensed by one or more current transformers that inductively couple a primary current into a secondary current according to a transfer function that defines a linear and saturation operating region of the current transformer.
  • the transfer function of a current transformer shifts with temperature such that a higher secondary current output is produced for the same primary current input as temperature increases.
  • the higher secondary current output has the effect of causing the MCP to trip sooner.
  • trip curves should be adjusted upwards or downwards. What is needed is a temperature compensation algorithm that automatically adjusts trip curve settings to compensate for changes in temperature.
  • Mechanical circuit breakers energize an electro-magnetic device such as a solenoid to trip instantaneously in response to a rapid surge in current such as a short circuit.
  • An electro-magnetic device such as a solenoid to trip instantaneously in response to a rapid surge in current such as a short circuit.
  • Existing MCPs protect only a limited range of motors, but should avoid tripping in response to in-rush motor currents that occur during motor start-up while tripping on a range of fault currents including instantaneous short-circuit currents. What is needed is a circuit breaker system with improved trip protection.
  • Circuit breakers may have current transformers that serve a dual function of measuring fault currents and supplying stored or instant energy to trip the breaker.
  • the range of the circuit breaker is limited by the linear region of operation of the current transformer. What is needed is to provide a calibration process to use the saturation region of current transformers to increase the operating range of a circuit breaker. There is also a need for a calibration process that may be adjusted via programming without altering the basic test process.
  • a current transformer that extends the range of a circuit breaker, such as a motor circuit protector, includes both solid and gapped laminations that are staked and stacked together to form a single core.
  • the solid laminations produce secondary current sufficient to power electronic components of the circuit breaker and sense relatively low currents.
  • the gapped laminations produce secondary current sufficient to power the electronic components and sense relatively high currents, thereby extending the range of sensed currents for the MCP.
  • the gapped laminations decrease the amount of remnant flux or saturation in the current transformer compared to solid cores.
  • the number of solid laminations and gapped laminations as well as the size of the gap in the gapped laminations are selected to fault power the MCP electronic components and sense a range of currents corresponding to locked-rotor or in-rush motor currents as well as high instantaneous short-circuit currents.
  • the saturation knee threshold region of the core's transfer function is pushed higher, resulting in saturation at a higher peak current.
  • Gapped laminations are added for higher current sensing based on remnant flux requirements. As each gapped lamination is. added, the core's saturation region shifts to a higher peak current value.
  • the linear region of the current transformer can be extended by increasing the ratio of solid-to-gapped laminations and/or by varying the number of turns wound on the primary coil of the current transformer, resulting in more accurate approximation of the primary current.
  • the core includes sixteen solid laminations and eight gapped laminations, resulting in a current transformer that can sense locked-rotor currents in the range of 1OA as well as high fault currents in the range of 3000A.
  • a current transformer for fault powering trip unit electronics and sensing low currents and high currents includes a core with solid laminations and gapped laminations to sense a wide range of currents .from locked-rotor currents to high, instantaneous short-circuit currents in a single current transformer.
  • the current transformer can also fault power trip unit electronics without requiring an additional current transformer.
  • the operating range of the circuit breaker is significantly enhanced compared to existing breakers that can sense only a limited range of current levels.
  • Aspects of the present invention improve conventional techniques of translating user-adjustable trip unit settings to pickup levels. These aspects enable a fail-safe operation mode where user adjustments can revert to greater or any other predetermined protective levels. Overall system performance is improved with lower-cost components without requiring switch calibration. Switch performance is verified during the production test process with quantitative techniques.
  • the MCP includes a user adjustment assembly for adjusting the tripping levels of the MCP.
  • the user adjustment assembly includes a mechanical button with switch-like stop and detent features corresponding to mechanical orientation angles that are translated to a potentiometer mechanical orientation via a user adjustment circuit.
  • the user adjustment circuit may include a potentiometer and is configured to present a percentage of an A/D's full-scale voltage to an A/D input pin, which converts the scaled voltage to a corresponding digital value that determines the button position.
  • the user adjustment circuit is a cheaper alternative to existing mechanical solutions by substantially eliminating the number of mechanical parts required to translate mechanical switch positions to meaningful data.
  • Software embedded in the MCP and executed by a controller in the MCP implements a switch detection algorithm that includes a failure mode detection.
  • Mechanical button positions are determined via the controller's A/D converter, and changes to the mechanical button positions are sensed by the A/D converter and the MCP 's trip levels are automatically adjusted based upon the new position.
  • the failure mode detection reverts to predetermined protective levels.
  • Position thresholds are determined by producing a statistical distribution of data corresponding to the switch settings, and as each user adjustment assembly is produced, the position thresholds and user adjustment assembly performance are monitored and stored.
  • a user adjustment assembly for adjusting tripping levels of an electrical trip unit includes a potentiometer and an adjustment button.
  • the potentiometer is positioned inside a protective cover of the electrical trip unit and has a top surface.
  • the adjustment button is coupled to the potentiometer for mechanically adjusting the potentiometer and has an insulation disc for increasing resistance to electrostatic discharge.
  • the adjustment button is dimensioned and located so that it covers the potentiometer.
  • an electrical circuit breaker has adjustable tripping levels and includes an enclosing cover, a potentiometer, and an adjustment button.
  • the enclosing cover has a button hole.
  • the potentiometer is coupled to a voltage source and is mounted to a printed wire assembly in an interior area of the enclosing cover.
  • the adjustment button has an insulation disc for protecting the potentiometer from electrostatic discharge. The adjustment button is dimensioned and located so that it covers the potentiometer.
  • the potentiometer is positioned inside a cover of the trip unit and includes a potentiometer button.
  • the adjustment button is coupled to the potentiometer for mechanically adjusting it and includes an insulation disc for increasing resistance to electrostatic discharge, preventing contaminants from entering the printed wire assembly components, and preventing application of downward force to the potentiometer button.
  • the insulation disc has a bottom surface that is dimensioned to be larger than the potentiometer button.
  • the adjustment button includes one or more stops that trigger a fail safe operation mode where the tripping levels are automatically adjusted to higher or predetermined protective levels when the adjustment button is moved to those stop positions. Switch calibration is obviated and the simplified design reduces overall cost.
  • aspects of the present invention improve conventional techniques of translating user-adjustable trip unit settings to pickup levels. These aspects enable a fail-safe operation mode where user adjustments can revert to greater or any other predetermined protective levels. Overall system performance is improved with lower-cost components without requiring switch calibration. Switch performance is verified during the production test process with quantitative techniques.
  • the MCP includes .a user adjustment assembly for adjusting the tripping levels of the MCP.
  • the user adjustment assembly includes a mechanical button with switch-like stop and detent features corresponding to mechanical orientation angles that are translated to a potentiometer mechanical orientation via a user adjustment circuit. There is a continuity or lack of discontinuity between switch position ranges.
  • the user adjustment circuit may include a potentiometer and is configured to present a percentage of an A/D's full-scale voltage to an A/D input pin, which converts the scaled voltage to a corresponding digital value that determines the button position.
  • the user adjustment circuit is a cheaper alternative to existing mechanical solutions by substantially eliminating the number of mechanical parts required to translate mechanical switch positions to meaningful data.
  • Position thresholds are determined by producing a statistical distribution of data corresponding to the switch settings, and as each user adjustment assembly is produced, the position thresholds and user adjustment assembly performance are monitored and stored.
  • aspects of various embodiments described herein provide for methods for translating trip unit switch positions to trip point settings for embedded software-controlled trip unit systems. These aspects offer an improvement over a simple lookup table search for accessing stored calibrated trip data.
  • the algorithms that implement these aspects can be extended to any trip unit system that requires access to calibrated trip pickup data.
  • the switch-to-trip point translation algorithms involve data compression of trip point data, diagnostic checksums, switch-to-trip point memory mapping, and the extension of data settings to elevated temperatures. This solution allows trip unit devices to be updated easily and securely, independent of embedded software product design. Normalized templates including normalized trip point data are used as a starting point for calibrating the embedded software.
  • a method for translating mechanical positions to trip curves of an electrical tripping system.
  • the method includes operatively coupling a mechanical button to a potentiometer of an electrical tripping system, the electrical tripping system being operable to trip at an operating trip curve.
  • a first signal indicative of a trip curve of a plurality of trip curves is received from the potentiometer.
  • the first signal is associated with one of the plurality of trip curves to produce a first trip curve.
  • An operating trip curve is set to be the first trip curve.
  • a method for adjusting a circuit breaker operating trip curve in response to selecting a mechanical position.
  • the method includes providing a mechanical button for selecting any of a plurality of mechanical positions.
  • a potentiometer is mounted to a printed wire assembly of a circuit breaker, the potentiometer having a plurality of potentiometer positions indicative of respective ones of a plurality of trip curves.
  • Each of the plurality of mechanical positions is operatively coupled to a corresponding one of the plurality of potentiometer positions.
  • a potentiometer first signal is sent.
  • the potentiometer first signal is indicative of a first curve of the plurality of trip curves.
  • An operative trip curve of the circuit breaker is set to be the first curve.
  • a potentiometer second signal is sent.
  • the potentiometer second signal is indicative of a second curve of the plurality of trip curves.
  • the operative trip curve is changed to be the second trip curve.
  • an electrical tripping system is operable at an operating trip curve and includes a mechanical button, a potentiometer, and a controller.
  • the mechanical button has a plurality of mechanical positions.
  • the potentiometer is operatively coupled to the mechanical button and produces a first data signal in response to selection of a first position of the plurality of mechanical positions.
  • the controller is communicatively coupled to the potentiometer and is programmed to associate the first data signal with one of a plurality of trip curves to produce a first trip curve and to set the operating trip curve to be the first trip curve.
  • a translation technique for translating mechanical button positions of a circuit breaker to trip point settings stored in a memory of the circuit breaker.
  • a turn of a mechanical button turns a potentiometer button, whose output is converted to scaled voltages and converted to corresponding digital values. These digital values are checked against a range of thresholds (minimum/maximum) corresponding to mechanical orientation positions of the mechanical button.
  • a trip curve lookup table stored in memory is accessed to determine which trip point setting should be set for the circuit breaker based upon the button position.
  • the circuit breaker's trip curve settings can be changed easily via the mechanical button. They can also be changed easily by modifying the trip curve lookup table without having to recalibrate the circuit breaker or the switch settings.
  • aspects of the various embodiments disclosed herein relate to a temperature compensation algorithm that uses low-cost sensor technology and a low-cost microcontroller to achieve real-time temperature compensation.
  • a sensor equation transformation relationship is disclosed to efficiently convert temperature sensor readings directly to burden resistance percentage values.
  • the burden resistance percentage values are used to efficiently adjust trip point thresholds.
  • the temperature compensation techniques disclosed herein can be applied to a wide variety of industrial sensor detection applications that incorporate copper sensing resistors. In general, aspects disclosed herein can be extended to other sensor technologies where temperature sensor equations are deliberately matched with compensated sensors.
  • the temperature compensation algorithm takes advantage of the following two sensor relationships: first, the base-emitter voltage equation of a PNP transistor as a function of temperature; second, the temperature versus resistance relationship of a copper burden resistor. These sensor relationships are deliberately matched to enable a simple transformation from sensor temperature to burden resistor percentage values. Data values from the negative-sloping temperature sensor are transformed to the positive sloping copper relationship of the burden resistor. The transformation is designed about the raw sensor data intercepts with adjusted offset compensation. [00038] The data output of the temperature compensation algorithm is the percentage operation point of the burden resistor relative to 100% at 25 degrees C (ambient temperature). Operation range is designed for the intended temperature operating range of the compensated burden resistor sensor.
  • an automatic temperature compensation method that automatically adjusts trip point thresholds of a motor circuit protector in response to changes in temperature.
  • the relationship between two curves is exploited to match temperature sensor readings from a temperature sensor circuit with burden resistor percentage values derived from a burden resistor circuit.
  • a temperature inflection point is determined from the intersection of (1) the temperature sensor curve plotting the voltage output of the temperature sensor versus temperature and (2) the burden resistance curve plotting burden resistance versus temperature.
  • a temperature value along the temperature sensor curve is transformed into the corresponding burden resistance on the burden resistance curve.
  • the burden resistance is expressed as a percentage variance from a burden resistance at an ambient temperature.
  • An adjusted trip point threshold is calculated from the computed burden resistor percentage, and the adjusted trip point threshold is stored in a memory.
  • a Trip Activation Sequence for stored energy tripping includes the following:
  • Output ports of the microcontroller are configured to "Charge" mode and the Backup Trip Circuitry will be activated if the power supply reaches its voltage trip threshold.
  • Output ports of the microcontroller are configured to set a trip circuit output that is received by a trip circuit.
  • the trip solenoid will normally be activated via this signal path.
  • the backup trip output will active the trip circuit.
  • the first signal path is to a voltage regulation module that charges a power supply, which supplies power to the trip solenoid and trips the solenoid when the appropriate voltage level is reached.
  • the second signal path is to a trip circuit that also activates the trip solenoid.
  • a redundant trip activation scheme whereby at least one pre-trip condition must be satisfied before a trip signal is sent to a trip solenoid of a motor circuit protector.
  • a trip module sets a trip variable upon detection of a trip signal. Instead of tripping the MCP, the trip module requires at least an additional trip signal to be detected. The next time a trip signal is detected, the trip module checks whether the trip variable has been set, and, if so, sends a trip signal to a trip circuit, causing the solenoid to be activated. To ensure that a trip actually occurs, the solenoid can be tripped from the microcontroller via the trip circuit or from an over-voltage protection circuit in the event of a system failure, which operates independently of the trip circuit.
  • aspects of embodiments disclosed herein implement a fault-powered voltage regulation control algorithm with a wide operating range, such as 9 A ⁇ through to 2500 Arms-
  • the embedded software control algorithm utilizes re-configurable microcontroller technology to minimize power supply peak overshoot, minimize voltage regulation ripple, and maintain stored energy trip voltages.
  • Numerous advantages are realized, including at least the following: improved system level performance while reducing the requirements for expensive external hardware components; reduced risk of nuisance tripping of the trip unit system while the system is transitioning between powered and unpowered states; providing a robust fault tolerant backup trip detection system by cooperating with external backup trip circuitry; the power supply control algorithm can be applied to a variety of breaker products having different operating ranges and can. be extended to other similar trip unit design platforms.
  • the algorithm cooperates with a power-supply activated, backup trip system. During normal operation, the algorithm maintains voltage regulation below backup trip setpoints. A variety of software and system failure modes will activate the backup trip detection circuitry.
  • the voltage regulation control algorithm includes the following:
  • the voltage regulation control algorithm configures the power supply for a brief charge-only state, so as to achieve power supply startup stability. This configuration reduces the expense of external stability hardware components. Then, the voltage regulation control algorithm configures the power supply voltage sense inputs to connect to fast reacting microcontroller comparator circuitry. This reduces peak overshoot during high instantaneous startup scenarios, while charging the power supply to stored energy trip voltages more efficiently. After stored energy trip levels have been reached, the microcontroller's internal circuitry is reconfigured to connect the power supply voltage sense inputs to more accurate internal analog-to-digital ("A/D") converters. This reconfiguration improves power supply ripple performance after startup. A variable number of charge pulses are generated each time the voltage regulator routine is serviced.
  • A/D analog-to-digital
  • a method and system are provided for supply voltage regulation in a motor circuit protector (MCP) that includes a current transformer coupled to a rectifier and a stored energy circuit. A solenoid is actuated by that circuit" when a sufficient voltage is present. A controller having a configurable input is coupled to the stored energy circuit.
  • MCP motor circuit protector
  • the controller Upon startup of the motor circuit protector, the controller causes the stored energy circuit to be charged to a startup voltage level via secondary current from the current transformer. The controller periodically interrupts the charging to measure the secondary current to detect fault levels.
  • the configurable input is set to a comparator input for rapid current measurements.
  • the configurable input is set to an A/D input for accurate measurements. The controller measures the voltage of the stored energy circuit while charging it to a power level sufficient to actuate the solenoid.
  • the circuit breaker is self-protected against faults in parallel with the trip unit's voltage regulation cycle. Improved performance over conventional microprocessor sampling trip systems is achieved by utilizing reconfigurable microprocessor technology. As a result of the reconfiguration aspects disclosed herein, the overall cost of the trip unit system is substantially reduced.
  • the time-synchronized self-protection embedded software includes the following: [00055] The microcontroller wakes up, and the embedded software stabilizes its power supply. The power supply is charged for several microseconds before proceeding to the first self protection sensing window.
  • the microcontroller is then configured for first half-cycle high instantaneous faults with minimal initializing overhead.
  • An on-board comparator circuit is configured for the trip unit's self-protection level. Exceeding the set threshold will direct drive a trip solenoid, rather than waiting for stored energy voltage detection.
  • the embedded software reconfigures the trip sense comparator circuit to sense voltage and charge the power supply. High instantaneous faults in this mode will charge the power supply quickly. A fast charge proceeds to the next time synchronized step and thus self protection is not comprised.
  • the trip unit After the stored energy trip voltage (sufficient to actuate the solenoid) is reached, the trip unit reconfigures its resources for an interrupt-based self protective mode. This is achieved by again reconfiguring the onboard comparator circuit for self-protection trip levels. In this mode and in the time immediately following, because stored energy trip voltages are reached, stored energy trip activation occurs.
  • a time-synchronized trip implementation is provided for a motor circuit protector (MCP) having a reconfigurable microcontroller.
  • MCP motor circuit protector
  • the microcontroller causes a power supply to be charged for an initial time period during a charging mode.
  • An onboard comparator is configured for a predetermined self-protection level of the MCP, and fault currents that exceed the comparator's threshold will directly drive a solenoid to trip the MCP.
  • the microcontroller reconfigures the comparator's threshold to both measure and charge the power supply toward a stored energy trip voltage, which will charge quickly when high fault currents are present. As a result, self-protection is not compromised.
  • the microcontroller reconfigures the onboard comparator's threshold for self-protection trip levels. When a trip event occurs in this mode, stored energy trip activation occurs.
  • the MCP includes user-selectable trip settings, and the microcontroller reconfigures the comparator threshold levels for user-selectable self- protection levels.
  • a motor circuit protector trips in accordance with an instantaneous trip curve that is split into three protection regions (in no particular order), a self-protection region, an in-rush avoidance region, and a locked-rotor avoidance region.
  • Software modules for detecting whether primary current exceeds each region is run in parallel or simultaneously, providing redundant instantaneous trip detection, and these redundant protection regions remain active throughout the startup and steady-state modes of operation of the motor circuit protector. This redundancy provides improved time current trip performance for a wide variety of short circuit conditions and improved system safety properties are realized.
  • the current threshold for the self-protection region can be adjusted toward the in-rush avoidance region once steady-state operation is achieved.
  • the redundant trip detection of the present invention improves short-circuit fault detection for both "Close on Fault” and "Fault in Steady State” scenarios. Higher protection levels remain active throughout startup and steady-state modes of operation.
  • Aspects of the present invention advantageously improve self-protection detection of the motor circuit protector and provide a fast response to high instantaneous faults. Additional advantages include improved instantaneous short-circuit protection with simultaneous implementation of locked-rotor and in-rush avoidance protection. As a result, the motor circuit protector is less prone to nuisance tripping while having high availability for enhanced flinctions.
  • the redundant trip detection aspects of the present invention also improve system safety properties.
  • software aspects of the various embodiments described herein will be compliant with UL-1998 and IEC-61508.
  • a motor circuit protector trips in accordance with an instantaneous trip curve that is split into three protection regions, a self-protection region, an in-rush avoidance region, and a locked-rotor avoidance region.
  • Software modules for detecting whether primary current exceeds each region are run in parallel or simultaneously, providing redundant instantaneous trip detection, and these redundant protection regions remain active throughout the startup and steady-state modes of operation of the motor circuit protector. This redundancy provides improved time current trip performance for a wide variety of short circuit conditions and improved system safety properties are realized.
  • the current threshold for the self-protection region can be adjusted toward the in-rush avoidance region once steady-state operation is achieved. In startup mode, only the self-protection region may be detected, but can remain active during steady-state mode.
  • the product package may be a motor circuit protector, a circuit breaker generally, or any industrial control device.
  • transfer functions are derived for low-cost current transformers to characterize their performance over a wide range of primary operating currents extending from the linear region of the current transformers well into their saturation region. These transfer functions are modeled with mathematical characteristic equations to show the typical properties of each current transformer design. Specific calibration points are chosen for each design based on transition points in the characteristic equation. During production testing, well-defined primary currents are injected into the product at the calibration points. The actual outputs are measured simultaneously for these injected currents.
  • a method and system are provided to detect currents in the saturation region of a current transformer for a circuit breaker.
  • An example method is sensing a fault condition with a current transformer in t a circuit breaker.
  • the characteristic curve of the current transformer in a saturation mode is determined based on peak current.
  • a current is received on the transformer.
  • a secondary current is output from the transformer. It is determined whether the secondary current is indicative of a fault current in the saturation mode of the transformer.
  • the breaker is tripped if the secondary current is indicative of a fault current.
  • Various aspects of the embodiments disclosed herein are directed to calibration of variable components of a low-cost current measurement signal chain in a circuit breaker, such as a motor circuit protector, to achieve accurate current measurement.
  • the signal chain includes one or more current transformers, a serpentine copper resistor, the R ds,o n of a FET, a microcontroller, a voltage regulator for an AID reference, and a temperature sensor.
  • the current transformers have a characteristic V ou t to Vj n over the range of the product under calibration. The product's range is in the saturated and linear region of the characteristic curve of the current transformer.
  • the characteristic output of the current transformer is provided to the functional tester prior to calibration.
  • the calibration of the product is extended to the design process rather than just to the manufacturing process. Calibration responsibility can be seamlessly integrated between the manufacturing and design functions.
  • the calibration techniques disclosed herein store the nominal templates during the design process at high temperatures, such as 90 0 C, and scaling is performed on this elevated nominal calibration template.
  • the temperature sensor measures temperature based on the voltage across the p-n junction of a BJT as it varies with temperature.
  • the BJT reacts quickly to shifts in temperature.
  • the temperature sensor is calibrated to a reference temperature on the functional tester.
  • the temperature of the circuit board is important because the burden resistance includes the resistance of the serpentine copper resistor and the R ⁇ i s , on of the FET.
  • the resistance of the FET and the copper resistor combination changes at a rate of 0.393 percent per degree C.
  • a test current is independently injected into each of the three current transformers from the functional tester and the response of the current transformers is read from the microcontroller.
  • the responses of the current transformers to the injected currents, and the temperature of the circuit board are used to scale the characteristic curves of the transformer to provide a curve that will fit the system as a whole, i.e., the current transformers and the circuit board. This process eliminates error from the voltage reference, -some of the A/D error, and error associated with the burden resistor and FET R ds . on -
  • An example method calibrates a signal chain of a circuit breaker.
  • the signal chain includes a current transformer, a burden resistor, a stored energy circuit and a controller.
  • the circuit breaker includes a memory coupled to the controller.
  • a calibration instruction routine is written in a first location of the memory.
  • a test current is injected in the circuit breaker signal chain.
  • the test current peak of the test current in the circuit breaker signal chain is measured.
  • Data indicative of the test current peak is stored in a second location of the memory.
  • the test current peak data is read from the second location of the memory.
  • the test current peak data is compared with nominal current data related to the signal chain remotely from the circuit breaker.
  • a calibration factor is determined based on the comparison.
  • FIG. 1 is perspective view of a motor circuit protector according to certain embodiments of the present disclosure
  • FIG. 2 is a functional block diagram of the motor circuit protector in FIG. 1 according to certain embodiments of the present disclosure
  • FIG. 3 is a functional block diagram of the operating components of a control algorithm of the motor circuit protector in FIG. 1 according to certain embodiments of the present disclosure
  • FIG. 4 is a circuit diagram of the stored energy circuit of the motor circuit protector in FIG. 1 according to certain embodiments of the present disclosure.
  • FIG. 5 is an isometric view of a current transformer according to certain embodiments of the present disclosure.
  • FIG. 6 is an exploded isometric view of the current transformer in FIG. 5 according to certain embodiments of the present disclosure
  • FIG. 7 is an isometric view of a solid lamination from the current transformer shown in FIG. 6 according to certain embodiments of the present disclosure
  • FIG. 8 is an isometric view of a current transformer core according to certain embodiments of the present disclosure.
  • FIG. 9 is an isometric view of a current transformer core according to certain embodiments of the present disclosure.
  • FIG. 10 is a perspective view of the interior of a current transformer according to certain embodiments of the present disclosure.
  • FIG. 1 IA is a functional electrical schematic of an user adjustment switch for use with the motor circuit protector of FIG. 1 ;
  • FIG. HB is an illustration of an electro-mechanical orientation for adjustment in accordance with the diagram of FIG. 1 IA;
  • FIG. HC is a flowchart diagram for setting an operating trip curve of the motor circuit protector of FIG. 1 by adjusting a mechanical switch
  • FIG. 12A is a perspective view of a trip unit assembly according to an alternative implementation of the present application.
  • FIG. 12B is an enlarged view of a top portion of the trip unit assembly of FIG.
  • FIG. 13 is a cross-sectional view showing a portion of the trip unit assembly of
  • FIG. 12A at a rotational center of an adjustment switch
  • FIG. 14A is a top perspective view of the adjustment switch of FIG. 13;
  • FIG. 14B is a bottom perspective view of the adjustment switch of FIG. 13;
  • FIG. 15A is a perspective view of a printed wire assembly including two potentiometers according to another alternative implementation of the present application;
  • FIG. 15B is a perspective view of the printed wire assembly of FIG. 15A including two adjustment switches coupled to the two potentiometers;
  • FIG. 16A is an enlarged view showing the adjustment switch inserted into a cover of the trip unit assembly of FIG. 12 A;
  • FIG. 16B is an enlarged bottom perspective view illustrating a hole in the cover of the trip unit assembly of FIG. 12 A;
  • FIG. 16C illustrates a cross-sectioned portion of the adjustment switch of FIG. 13 inserted into the hole of FIG. 16B;
  • FIG. 17 illustrates another cross-sectioned portion of the adjustment switch of FIG.
  • FIG. 18A illustrates a top perspective view of an adjustment switch having a insulative skirt according to yet another alternative implementation of the present application
  • FIG. 18B illustrates a bottom perspective view of the adjustment switch of FIG.
  • FIG. 19A is a functional block diagram of a temperature compensation system according to aspects of the various embodiments disclosed herein;
  • FIG. 19B is a functional circuit schematic for a temperature sensor coupled to the microcontroller shown in FIG. 2;
  • FIG. 20 is a flow chart diagram of an automatic temperature compensation method according to some aspects of the various embodiments disclosed herein;
  • FIG. 21 is a Unified Modeling Language (UML) diagram of a trip curve adjustment sequence
  • FIG. 22A is a UML sequence diagram of a trip curve initialization sequence shown in FIG. 21;
  • FIG. 22B is a UML diagram of an update trip curve settings sequence shown in
  • FIG. 21 is a diagrammatic representation of FIG. 21.
  • FIG. 23 A is a UML diagram of a get burden resistor data sequence shown in FIG.
  • 23B is a UML diagram of a get scaled temperature sensor data sequence shown in
  • FIG. 23A [000108]
  • FIG. 23C is a UML diagram of a range limit, scaled temperature sensor data sequence shown in FIG. 23B;
  • FIG. 23D is a UML diagram of a read raw temperature sensor data sequence shown in FIG. 23B;
  • FIG. 24A is a diagram of a trip state according to embodiments of the present invention.
  • FIG. 24B is a diagram of a direct drive trip state and a stored energy trip state according to embodiments of the present invention.
  • FIG. 24C is a flow diagram of the trip state shown in FIG. 24a according to embodiments of the present invention.
  • FIG. 25 is a UML diagram illustrating a state diagram for the INST trip regions A
  • FIG. 26A is a diagram expressed in a Unified Modeling Language (UML) illustrating a power-up activity diagram according to an implementation of the present invention
  • FIG. 26B is a UML diagram showing the run/main loop states according to aspects of the various embodiments disclosed herein;
  • FIG. 26C is a UML diagram for a start regulator activity according to aspects of the various embodiments disclosed herein;
  • FIG. 27 is a flow chart diagram of a process of the voltage regulation module that is part of the control algorithm of the motor circuit protector in FIG. 1;
  • FIG. 28 is an exemplary trip curve having three protection regions, according to an implementation of the present invention.
  • FIG. 29 is a diagram expressed in a Unified Modeling Language (UML) illustrating a power-up activity diagram according to an implementation of the present invention
  • FIG. 3OA is a UML diagram illustrating a run-mode state diagram according to an implementation of the present invention.
  • FIG. 30B is a UML diagram illustrating a state diagram for run-mode auxiliary tasks
  • FIG. 31 is a UML diagram illustrating a state diagram for instantaneous trip regions A (locked-rotor avoidance region) and B (i ⁇ -rush avoidance region);
  • FIG. 32 is a UML diagram illustrating a state diagram for high instantaneous self- protection tripping for region C (self-protection region);
  • FIG. 33 is a block diagram of a calibration system used to calibrate the operating components of the motor circuit protector in FIG. 1;
  • FIGs. 34A and 34B are current waveforms of the primary and secondary currents from current transformers of the motor circuit protector in FIG. 1 in the non-saturated region;
  • FIG. 35 is a current waveform of the primary and secondary currents from a current transformer of the motor circuit protector in FIG. 1 in the saturated region;
  • FIG. 36 is a graph of a transfer function of the current transformers in the motor circuit protector in FIG. 1;
  • FIG. 37 is a functional block diagram of the operating components of the calibration software of the calibration system in FIG. 33;
  • FIG. 38 is a flow chart diagram of the calibration process that is employed by the calibration system in FIG. 33.
  • FIG. 39 is calibration state diagram in Unified Modeling Language (UML) according to aspects of various embodiments disclosed herein.
  • UML Unified Modeling Language
  • the motor circuit protector 100 includes a durable housing 102 including a line end 104 having line terminals 106 and a load end 108 having load lugs or terminals 110.
  • the line terminals 106 allow the motor circuit protector 100 to be coupled to a power source and the load terminals 110 allow the motor circuit protector 100 to be coupled to an electrical load such as a motor as part of a motor control center ("MCC").
  • MCC motor control center
  • the motor circuit protector 100 includes a three-phase circuit breaker with three poles, although the concepts described below may be used with circuit protectors with different numbers of poles, including a single pole.
  • the motor circuit protector 100 includes a control panel 112 with a full load ampere (“FLA”) dial 114 and an instantaneous trip point (“I m ”) dial 116 which allows the user to configure the motor circuit protector 100 for a particular type of motor to be protected with the rated current range of the motor circuit protector 100.
  • the full load ampere dial 114 allows a user to adjust the full load which may be protected by the motor circuit protector 100.
  • the instantaneous trip point dial 116 has settings for automatic protection (three levels in this example) and for traditional motor protection of a trip point from -8 to 13 times the selected full load amperes on the full load ampere dial 114.
  • the dials 114 and 116 are located next to an instruction graphic 118 giving guidance to a user on the proper settings for the dials 114 and 116.
  • the instruction graphic 118 relates to NEC recommended settings for the dials 114 and 116 for a range of standard motors.
  • the motor circuit protector 100 includes a breaker handle 120 that is moveable between a TRIPPED position 122 (shown in FIG. 1), an ON position 124 and an OFF position 126.
  • the position of the breaker handle 120 indicates the status of the motor circuit protector 100. For example, in order for the motor circuit protector 100 to allow power to flow to the load, the breaker handle 120 must be in the ON position 124 allowing power to flow through the motor circuit protector 100.
  • FIG. 2 is a functional block diagram of the motor circuit protector 100 in FIG. 1 as part of a typical MCC configuration 200 coupled between a power source 202 and an electrical load such as a motor 204.
  • the MCC configuration 200 also includes a contactor 206 and an overload relay 208 downstream from the power source 202.
  • the motor circuit protector 100 protects the motor 204 from a short circuit condition by actuating the trip mechanism, which causes the breaker handle 120 to move to the TRIPPED position when instantaneous short-circuit conditions are detected.
  • the power source 202 in this example is connected to the three line te ⁇ ninals 106, which are respectively coupled to the primary windings of three current transformers 210, 212 and 214. Each of the current transformers 210, 212 and 214 has a phase line input and a phase load output on the primary winding.
  • the current transformers 210, 212 and 214 correspond to phases A, B and C from the power source 202.
  • the current transformers 210, 212 and 214 in this example are iron-core transformers and function to sense a wide range of currents.
  • the motor circuit protector 100 provides instantaneous short- circuit protection for the motor 204.
  • the motor circuit protector 100 includes a power supply circuit 216, a trip circuit 218, an over-voltage trip circuit 220, a temperature sensor circuit 222, a user adjustments circuit 224, and a microcontroller 226.
  • the microcontroller 226 is a PIC16F684-E/ST programmable microcontroller, available from Microchip Technology, Inc. based in Chandler, Arizona, although any suitable programmable controller, microprocessor, processor, etc. may be used.
  • the microcontroller 226 includes current measurement circuitry 241 that includes a comparator and an analog-to-digital converter.
  • the trip circuit 218 sends a trip signal to an electro-mechanical trip solenoid 228, which actuates a trip mechanism, causing the breaker handle 120 in FIG. 1 to move from the ON position 124 to the TRIPPED position 122, thereby interrupting power flow to the motor 204.
  • the electromechanical trip solenoid 228 is a magnetic latching solenoid that is actuated by either stored energy from a discharging capacitor in the power supply circuit 216 or directly from secondary current from the current transformers 210, 212 and 214.
  • the signals from the three current transformers 210, 212 and 214 are rectified by a conventional three-phase rectifier circuit (not shown in FIG. 2), which produces a peak secondary current with a nominally sinusoidal input.
  • the peak secondary current either fault powers the circuits 216, 218, 220, 222, and 224 and the microcontroller 226, or is monitored to sense peak fault currents.
  • the default operational mode for current sensing is interlocked with fault powering as will be explained below.
  • a control algorithm 230 is responsible for, inter alia, charging or measuring the data via analog signals representing the stored energy voltage and peak current presented to configurable inputs on the microcontroller 226.
  • the control algorithm 230 is stored in a memory that can be located in the microcontroller 226 or in a separate memory device 272, such as a flash memory.
  • the control algorithm 230 includes machine instructions that are executed by the microcontroller 226. All software executed by the microcontroller 226 including the control algorithm 230 complies with the soflware safety standard set forth in UL-489 SE and can also be written to comply with IEC- 61508. The software requirements comply with UL-1998.
  • the configurable inputs may be configured as analog-to-digital (“A/D") converter inputs for more accurate comparisons or as an input to an internal comparator in the current measurement circuitry 241 for faster comparisons.
  • A/D analog-to-digital
  • the A/D converter in the current measurement circuitry 241 has a resolution of 8/10 bits, but more accurate A/D converters may be used and may be separate and coupled to the microcontroller 226.
  • the output of the temperature sensor circuit 222 may be presented to the A/D converter inputs of the microcontroller 226.
  • the configurable inputs of the microcontroller 226 include a power supply capacitor input 232, a reference voltage input 234, a reset input 236, a secondary current input 238, and a scaled secondary current input 240, all of which are coupled to the power supply circuit 216.
  • the microcontroller 226 also includes a temperature input 242 coupled to the temperature sensor circuit 222, and a full load ampere input 244 and an instantaneous trip point input 246 coupled to the user adjustments circuit 224.
  • the user adjustments circuit 224 receives inputs for a full load ampere setting from the full load ampere dial 114 and either a manual or automatic setting for the instantaneous trip point from the instantaneous trip point dial 116.
  • the microcontroller 226 also has a trip output 250 that is coupled to the trip circuit 218.
  • the trip output 250 outputs a trip signal to cause the trip circuit 218 to actuate the trip solenoid 228 to trip the breaker handle 120 based on the conditions determined by the control algorithm 230.
  • the microcontroller 226 also has a burden resistor control output 252 that is coupled to the power supply circuit 216 to activate current flow across a burden resistor (not shown in FIG. 2) and maintain regulated voltage from the power supply circuit 216 during normal operation.
  • the breaker handle 120 controls manual disconnect operations allowing a user to manually move the breaker handle 120 to the OFF position 126 (see FIG. 1).
  • the trip circuit 218 can cause a trip to occur based on sensed short circuit conditions from either the microcontroller 226, the over-voltage trip circuit 220 or by installed accessory trip devices, if any.
  • the microcontroller 226 makes adjustment of short-circuit pickup levels and trip-curve characteristics according to user settings for motors with different current ratings.
  • the current path from the secondary output of the current transformers 210, 212, 214 to the trip solenoid 228 has a self protection mechanism against high instantaneous fault currents, which actuates the breaker handle 120 at high current levels according to the control algorithm 230.
  • the over- voltage trip circuit 220 is coupled to the trip circuit 218 to detect an over- voltage condition from the power supply circuit 216 to cause the trip circuit 218 to trip the breaker handle 120 independently of a signal from the trip output 250 of the microcontroller 226.
  • the temperature sensor circuit 222 is mounted on a circuit board proximate to a copper burden resistor (not shown in FIG. 2) together with other electronic components of the motor circuit protector 100.
  • the temperature sensor circuit 222 and the burden resistor are located proximate each other to allow temperature coupling between the copper traces of the burden resistor and the temperature sensor.
  • the temperature sensor circuit 222 is thermally coupled to the power supply circuit 216 to monitor the temperature of the burden resistor.
  • the internal breaker temperature is influenced by factors such as the load current and the ambient temperatures of the motor circuit protector 100.
  • the temperature sensor 222 provides temperature data to the microcontroller 226 to cause the trip circuit 218 to actuate the trip solenoid 228 if excessive heat is detected.
  • the output of the temperature sensor circuit 222 is coupled to the microcontroller 226, which automatically compensates for operation temperature variances by automatically adjusting trip curves upwards or downwards.
  • the microcontroller 226 first operates the power supply circuit 216 in a startup mode when a reset input signal is received on the reset input 236.
  • a charge mode provides voltage to be stored for actuating the trip solenoid 228.
  • the microcontroller 226 shifts to a normal operation mode and monitors the power supply circuit 216 to insure that sufficient energy exists to power the electro-mechanical trip solenoid 228 to actuate the breaker handle 120. During each of these modes, the microcontroller 226 and other components monitor for trip conditions.
  • the control algorithm 230 running on the microcontroller 226 includes a number of modules or subroutines, namely, a voltage regulation module 260, an instantaneous trip module 262, a self protection trip module 264, an over temperature trip module 266 and a trip curves module 268.
  • the modules 260, 262, 264, 266 and 268 generally control the microcontroller 226 and other electronics of the motor circuit protector 100 to perform functions such as governing the startup power, establishing and monitoring the trip conditions for the motor circuit protector 100, and self protecting the motor circuit protector 100.
  • a storage device 270 which in this example is an electrically erasable programmable read only memory (EEPROM), is coupled to the microcontroller 226 and stores data accessed by the control algorithm 230 such as trip curve data and calibration data as well as the control algorithm 230 itself.
  • the EEPROM may be internal to the microcontroller 226.
  • FIG. 3 is a functional block diagram 300 of the interrelation between the hardware components shown in FIG. 2 and software/firmware modules 260, 262, 264, 266 and 268 of the control algorithm 230 run by the microcontroller 226.
  • the secondary current signals from the current transformers 210, 212 and 214 are coupled to a three-phase rectifier 302 in the power supply circuit 216.
  • the secondary current from the three-phase rectifier 302 charges a stored energy circuit 304 that supplies sufficient power to activate the trip solenoid 228 when the trip circuit 218 is activated.
  • the voltage regulation module 260 ensures that the stored energy circuit 304 maintains sufficient power to activate the trip solenoid 228 in normal operation of the motor circuit protector 100.
  • the trip circuit 218 may be activated in a number of different ways. As explained above, the over- voltage trip circuit 220 may activate the trip circuit 218 independently of a signal from the trip output 250 of the microcontroller 226. The microcontroller 226 may also activate the trip circuit 218 via a signal from the trip output 250, which may be initiated by the instantaneous trip module 262, the self protection trip module 264, or the over temperature trip module 266. For example, the instantaneous trip module 262 of the control algorithm 230 sends a signal from the trip output 250 to cause the trip circuit 218 to activate the trip solenoid 228 when one of several regions of a trip curve are exceeded.
  • a first trip region A is set just above a current level corresponding to a motor locked rotor.
  • a second trip region B is set just above a current level corresponding to an in-rush current of a motor.
  • the temperature sensor circuit 222 outputs a signal indicative of the temperature, which is affected by load current and ambient temperature, to the over temperature trip module 266.
  • the over temperature trip module 266 will trigger the trip circuit 218 if the sensed temperature exceeds a specific threshold. For example, load current generates heat internally by flowing through the current path components, including the burden resistor, and external heat is conducted from the breaker lug connections. A high fault current may cause the over temperature trip module 266 to output a trip signal 250 (FIG.
  • the over temperature trip module 266 protects the printed wire assembly from excessive temperature buildup that can damage the printed wire assembly and its components. Alternately, a loose lug connection may also cause the over temperature trip module 266 to output a trip signal 250 if sufficient ambient heat is sensed by the temperature sensor circuit 222.
  • the trip signal 250 is sent to the trip circuit 218 to actuate the solenoid 228 by the microcontroller 226.
  • the trip circuit 218 may actuate the solenoid 228 via a signal from the over-voltage trip circuit 220.
  • the requirements for "Voltage Regulation,” ensure a minimum power supply voltage for "Stored Energy Tripping.”
  • the trip circuit 218 is operated by the microcontroller 226 either by a "Direct Drive” implementation during high instantaneous short circuits or by the control algorithm 230 first ensuring that a sufficient power supply voltage is present for the "Stored Energy Trip.” In the case where the "Stored Energy" power supply voltage has been developed, sending a trip signal 250 to the trip circuit 218 will ensure trip activation.
  • the control for Direct Drive tripping requires a software comparator output sense mode of-operation. When the comparator trip threshold has been detected, the power supply charging current is applied to directly trip the trip solenoid 228, rather than waiting for full power supply voltage.
  • the over-voltage trip circuit 220 can act as a backup trip when the system 200 is in "Charge Mode.”
  • the control algorithm 230 must ensure "Voltage Regulation,” so that the over-voltage trip circuit 220 is not inadvertently activated.
  • the default configuration state of the microcontroller 226 is to charge the power supply 216. In microcontroller control fault scenarios where the power supply voltage exceeds the over voltage trip threshold, the trip circuit 218 will be activated. Backup Trip Levels and trip times are set by the hardware design.
  • the user adjustments circuit 224 accepts inputs from the user adjustment dials 114 and 116 to adjust the motor circuit protector 100 for different rated motors and instantaneous trip levels.
  • the dial settings are converted by a potentiometer to distinct voltages, which are read by the trip curves module 268 along with temperature data from the temperature sensor circuit 222.
  • the trip curves module 268 adjusts the trip curves that determine the thresholds to trigger the trip circuit 218.
  • a burden circuit 306 in the power supply circuit 216 allows measurement of the secondary current signal, which is read by the instantaneous trip module
  • the self-protection trip module 264 also receives a scaled current (scaled by a scale factor of the internal comparator in the current measurement circuitry 241) from the burden resistor in the burden circuit 306 to determine whether the trip circuit 218 should be tripped for self protection of the motor circuit protector
  • a trip module 265 is coupled between the trip circuit
  • the self protection trip module 264, and the over temperature trip module 266 are received by the trip module 265.
  • OVER-VOLTAGE TRIP BACKUP - A trip sequence that uses the over-voltage trip circuit 220 to trip the breaker. This sequence is a backup for the normal "trip circuit” method. This sequence can be activated later in time due to a higher V C A P 304 activation voltage.
  • FIG. 4 is a detailed circuit diagram of various circuits of the motor circuit protector
  • the power supply circuit 216 derives the secondary current from the secondary windings of the three current transformers 210, 212 and 214, which are rectified by the three-phase rectifier 302.
  • the output of the three-phrase rectifier 302 is coupled to the burden circuit 306, which is coupled in parallel to the stored energy circuit 304.
  • the power supply circuit 216 also includes a peak current input circuit 402 that is provided to the microcontroller 226, a scaled current comparator input circuit 404 that is provided to the comparator of the current measurement circuitry 241 of the microcontroller 226 via the scaled secondary current input 240, a stored energy capacitor voltage input circuit 406 and a voltage regulator circuit 408.
  • the stored energy capacitor input 232 of the microcontroller 226 is coupled to the stored energy capacitor input circuit 406, the reference voltage input 234 is coupled to the voltage regulator circuit 408, the secondary current input 238 is coupled to the peak current input circuit 402, and the scaled secondary current input 240 is coupled to the scaled current comparator input circuit 404.
  • the burden circuit 306 includes a burden resistor 410 connected in series with a burden resistor control field effect transistor (FET) 412.
  • the gate of the burden resistor control FET 412 is coupled to the burden resistor control output 252 of the microcontroller 226. Turning on the burden resistor control FET 412 creates a voltage drop across the burden resistor 410 and the burden resistor control FET 412 allowing measurement of the secondary current for fault detection purposes. The voltage drop may also provide an indication of current available to charge the stored energy circuit 304.
  • the secondary current from the rectifier 302 is measured by the peak current input circuit 402 and the scaled current comparator input circuit 404.
  • the stored energy circuit 304 includes two energy storage capacitors 420 and 422. The energy storage capacitors 420 and 422 are charged by the secondary current when the burden resistor control FET 412 is switched off and are discharged by the trip circuit 218 to actuate the trip solenoid 228 in FIG. 2.
  • the scaled current comparator input circuit 404 has an input that is coupled to the rectifier 302.
  • the scaled current comparator input circuit 404 includes a voltage divider to scale down the signal from the rectifier 302 and is coupled to the scaled secondary current input 240 of the microcontroller 226.
  • the voltage regulator circuit 408 provides a component power supply (in this example, 5 volts nominal) to the electronic components such as the microcontroller 226 in the motor circuit protector 100.
  • the microcontroller 226 includes an internal comparator in the current measurement circuitry 241 that may be switched to compare the input 232 or the input 240 with a reference voltage that is received from the voltage regulator circuit 408 to the reference voltage input 234.
  • the reference voltage is also a reference voltage level when the inputs 232 and 240 are configured to be coupled to analog- to-digital converters.
  • the internal comparator is switched to receive the input 240 to the self protection trip module 264, the peak current is scaled for the comparator input by external hardware such as the scaled current comparator input circuit 404.
  • An internal comparator reference is set by the microcontroller 226 to control the comparator trip thresholds.
  • the stored energy capacitor voltage input circuit 406 includes the parallel- connected capacitors 420 and 422 and measures the voltage level of the stored energy circuit 304, which is indicative of the stored energy in the capacitors 420 and 422.
  • the stored energy capacitor voltage input circuit 406 provides a signal indicative of the voltage on the capacitors 420 and 422 to the stored energy capacitor input 232 of the microcontroller 226 to monitor the voltage of the stored energy circuit 304.
  • the voltage regulator circuit 408 and the microcontroller 226 receive a reset signal from the power supply circuit 216 and the rectifier 302 begins to charge the capacitors 420 and 422.
  • a start-up delay time including a hardware time delay and a fixed software time delay elapses.
  • the hardware time delay is dependent on the time it takes the secondary current to charge the stored energy circuit 304 to a voltage sufficient to operate the voltage regulator circuit 408.
  • the voltage regulator circuit 408 needs a minimum of 5 volts (nominal) to operate.
  • the fixed software time delay is the time required for stabilization of the regulated component voltage from the voltage regulator circuit 408 to drive the electronic components of the motor circuit protector 100.
  • the software delay time is regulated by an internal timer on the microcontroller 226.
  • the overall start-up delay time typically covers the first half-cycle of the current.
  • the microcontroller 226 executes the control algorithm 230, which is optionally stored in the internal memory of the microcontroller 226, and enters a "Self Protection" measurement mode, which relies upon the internal comparator of the microcontroller 226 for rapid detection of fault currents.
  • the microcontroller 226 turns on the burden resistor control FET 412 allowing measurement of the secondary current.
  • the burden resistor control FET 412 is turned on for a fixed period of time regulated by the internal timer on the microcontroller 226.
  • the voltage regulation module 260 configures the microcontroller 226 to couple the scaled secondary current input 240 to an input to the internal comparator of the microcontroller 226.
  • the scaled secondary current input 240 reads the signal from the scaled peak current input circuit 404, which measures the secondary current from the rectifier 302 and requires minimal initializing overhead.
  • the peak current from the secondary current is predicted via the secondary current detected by the scaled current comparator input circuit 404.
  • the internal comparator in the microcontroller 226 is a relatively fast device (compared to, for example, an A/D converter, which may be more accurate but operates more slowly) and thus can detect fault currents quickly while in this mode. If the peak current exceeds a threshold level, indicating a fault current, the burden resistor control FET 412 is turned off by a signal from the burden resistor control output 252 of the microcontroller 226.
  • the threshold level is set depending on the desired self-protection model of the range of currents protected by the particular type of motor circuit protector 100.
  • the disconnection of the FET 412 causes the fault current to rapidly charge the capacitors 420 and 422 of the stored energy circuit 304 and actuate the trip solenoid 228 to trip the trip mechanism of the motor circuit protector 100, which is visually indicated by the breaker handle 120.
  • the control algorithm 230 enters into a charge only mode of operation in order to charge the capacitors 420 and 422 of the stored energy circuit 304.
  • the control algorithm 230 sends a signal to turn off the burden resistor control FET 412, causing the capacitors 420 and 422 to be charged.
  • the control algorithm 230 remains in the charge only mode until sufficient energy is stored in the stored energy circuit 304 to actuate the trip solenoid 228 in the event of a detected fault condition.
  • the voltage regulation module 260 configures the microcontroller 226 to take a voltage input from the peak current input circuit 402 to the secondary current input 238 which is configured for an analog to digital converter. The signal from the secondary current input 238 analog to digital conversion is more accurate then the internal comparator but relatively slower.
  • the stored energy circuit 304 is charged quickly and the fault current actuates the trip solenoid 228 therefore providing self protection.
  • control algorithm 230 can be programmed to multiplex current measurement for self-protection sensing and power-supply charging for minimum stored-energy tripping.
  • the voltage regulation module 260 also configures the internal comparator in the current measurement circuitry 241 to be connected to the stored energy capacitor voltage input circuit 406 via the capacitor voltage input 232 to detect voltage levels from the stored energy circuit 304.
  • the voltage regulation module 260 thus maintains real time monitoring over the regulated voltage output from the stored energy circuit 304 while performing other software tasks such as monitoring fault currents.
  • the control algorithm 230 charges the stored energy circuit 304 from the minimum voltage regulation level (5 volts in this example from the hardware startup period) to a voltage level (15 volts in this example) indicative of sufficient energy to actuate the trip solenoid 228.
  • the charging of the capacitors 420 and 422 is regulated by the voltage regulation module 260, which keeps the burden resistor control FET 412 off via the burden resistor control output 252 causing the capacitors 420 and 422 to charge.
  • the voltage regulation module 260 holds the stored energy circuit, 304 in the charge mode until a start voltage threshold level (15 volts in this example) is reached for the supply voltage from the stored energy circuit 304 and is thus sensed through the stored energy capacitor voltage input circuit 406.
  • the timing of when the start voltage threshold level is reached depends on the secondary current from the rectifier 302 to the stored energy circuit 304.
  • the ability of the voltage regulation module 260 to hold the charge mode allows designers to avoid external stability hardware components. This process reduces peak overshoot during high instantaneous startup scenarios while charging the capacitors 420 and 422 to the start voltage threshold level more efficiently.
  • the control algorithm 230 proceeds to a steady state or run mode.
  • the control algorithm 230 maintains control of the voltage from the stored energy circuit 304 ' with the voltage regulation module 260 after the sufficient energy has been stored for tripping purposes.
  • the voltage regulation module 260 maintains a voltage above the stored energy trip voltage by monitoring the voltage from the stored energy circuit 304 from the stored energy capacitor voltage input circuit 406 to the stored energy capacitor input 232.
  • the stored energy capacitor input 232 is internally configured as an AID converter input for more accurate voltage level sensing for the run mode.
  • the voltage regulation module 260 also regulates the stored energy circuit 304 and avoids unintended activation of the over-voltage trip circuit 220.
  • the power supply regulation task is serviced in the run mode on a periodic basis to maintain the necessary energy in the stored energy circuit 304.
  • the regulation task may be pre-empted to service higher priority tasks such as the trip modules 262 and 264.
  • the voltage regulation module 260 monitors the voltage from the stored energy circuit 304.
  • the voltage regulation module 260 maintains the voltage output from the stored energy circuit 304 above the backup trip set points, which include a high set point voltage and a low set point voltage.
  • the voltage regulation module 260 initiates fixed width charge pulses, by sending control signals via the burden resistor control output 252 to the burden resistor control FET 412 to turn on and off until a high voltage set point for the power supply voltage is reached.
  • the width of the pulse corresponds with the maximum allowable voltage ripple at the maximum charge rate of the stored energy circuit 304.
  • the number of fixed width charge pulses is dependent on the voltage level from the stored energy circuit 304. If the energy is above the high set point voltage, the voltage regulation module 260 will not initiate fixed width charge pulse in order to avoid unintended activation of the over-voltage trip circuit 220.
  • a threshold voltage low set point (13.5 volts in this example) for the stored energy circuit 304 is reached and the control algorithm 230 will charge the stored energy circuit 304 to reach a minimum voltage necessary for trip activation of the trip solenoid 228.
  • the microcontroller 226 will restart the charge mode to recharge the capacitors 420 and 422 in the stored energy circuit 304.
  • fault current measurement is disabled, however if a fault current of significant magnitude occurs, the fault current will rapidly charge the capacitors 420 and 422 of the measured stored energy circuit 304 and thus overall trip performance is not affected.
  • the application will also restart when the watchdog timer in the microcontroller 226 resets.
  • the microcontroller 226 In the run mode, the microcontroller 226 is in measurement mode by keeping the burden resistor control FET 412 on.
  • the microcontroller 226 monitors the secondary current via the secondary current input 238, which is configured as an analog-to-digital converter for more accurate measurements.
  • the instantaneous trip module 262 sends an interrupt signal from the trip output 250 of the microcontroller 226 to cause the trip circuit 218 to activate the trip solenoid 228 for conditions such as a motor in-rush current or a locked motor rotor (trip conditions A and B), which cause a trip curve to be exceeded based on the secondary current.
  • the internal comparator of the microcontroller 226 is configured to accept an input from the scaled secondary current input 240, which is read by the self protection trip module 264 to determine whether the trip circuit 218 should be tripped for self protection of the motor circuit protector 100 in the case of high instantaneous current (trip condition C) detected from the faster measurement of the comparator.
  • the trip conditions for self protection are a function of the user settings from the dials 114 and 116.
  • the solenoid 228 is triggered by the over voltage trip circuit 220 (shown schematically in FIG. 4).
  • the over voltage trip circuit 220 includes a voltage divider 430, which steps down the voltage level. In this example, pull up transistors cause the over voltage trip circuit 220 to send a discrete trip signal 280 to the trip circuit 218, causing the trip circuit 218 to actuate the trip solenoid 228 to trip the breaker handle 120.
  • FIG. 5 is an isometric view of a current transformer ("CT") 500 according to certain embodiments of the present disclosure, and is suitable for use as the current transformers 210, 212, or 214 shown in FIG. 2.
  • the current transformer 500 is enclosed within a housing 510.
  • the housing can be configured in any of a number of ways, the housing 510 can, for example, comprise two housing elements 512, 514 formed to fit in manner that encloses the various current transformer components. In certain embodiments, the housing can be configured to partially enclose the current transformer components.
  • a first lead pin 522 and a second lead pin 524 extend from the enclosed current transformer components through the housing 510.
  • the lead pins 522, 524 from current transformer 500 are connected to power supply circuit 216 as illustrated, for example, in FIGs. 2-4.
  • the housing can be constructed with nonconductive materials such as, for example, plastics or ceramics.
  • the current transformer 500 is mounted to a printed wire board (PWB) (not shown).
  • PWB printed wire board
  • FIG. 6 is an exploded isometric view of the current transformer 500 in FIG. 5 according to certain embodiments of the present disclosure.
  • the current transformer 500 includes two housing elements 512, 514 that enclose a number of current transformer components.
  • Housing element 514 is configured with a joint edge 516 that overlaps joint edge 518 of the housing element 512.
  • the overlapping joint arrangement between housing elements 512, 514 allows for dielectric integrity between current transformer 500 and uninsulated current path sections of the current transformer 500 in contact with current transformer housing 510.
  • Housing 510 can be configured to substantially enclose or to partially enclose the current transformer components 630, 632, 642, 644, 650. In the implementation illustrated in FIG.
  • the components 630, 632, 642, 644, 650 of the current transformer are substantially enclosed upon the joining of housing elements 512, 514 with the exception of a tunnel 620 through which first lead pin 522 and second lead pin 524 extend from the interior to the exterior of housing 510.
  • Current transformer 500 includes a core 630 that includes gapped laminations 632 and solid laminations 734 (an exemplary solid lamination 734 is shown in FIG. 7) combined together to form a single core element.
  • the solid laminations 734 in the current transformer core 630 provide secondary current output sufficient to power the electronic components of the system 200 and further sense relatively low currents (for example, current in the range of 10 amperes).
  • the gapped laminations 632 in current transformer core 630 decrease the amount of remnant flux or saturation in the current transformer 500, while providing secondary current sufficient to power the electronic components of the system 200 and sensing relatively high currents (for example, currents in the range of 3,000 amperes).
  • the combination of the solid and gapped laminations increases the range of primary currents that can be sensed by the current transformer 500 while also providing a sufficient amount of secondary current available for powering the electronic components of the system 200, including in particular the trip solenoid 228 and the power supply circuit 216.
  • Both power supply and current sensing are accomplished in a single current transformer that also senses current over a very wide range of currents, e.g., motor locked-rotor ("LRA") currents (on the order of 1OA for a lower threshold) to motor in-rush currents to high instantaneous short-circuit currents (as high as 3000A for an upper threshold for in-rush motor currents).
  • LRA motor locked-rotor
  • the ratio of the upper current threshold to the lower current threshold exceeds 100: 1 and can be as high as 300: 1.
  • the gapped laminations 632 and the solid laminations 734 are combined in a single stacked core 630 having a central opening 660.
  • Some benefits of a single stack core include that a higher lamination factor is achieved and post-annealing stresses are minimized in the current transformer core 630. Another benefit simplifies the manufacturing process, e.g., the gapped laminations 632 and the solid laminations 734 can be punched from the same die. A retractable insert can be used to punch out the gap 636 in the gapped lamination 632. Because both laminations are made from the same die, the consistency between individual laminations is increased.
  • the current transformer 500 can be assembled efficiently with the single stacked core 630 according to aspects of the present invention.
  • the stacked gapped laminations and solid laminations are staked together to form the single stacked core 630.
  • Bobbin halves 642, 644 circumscribe the core 630 when the two halves 642, 644 are joined together.
  • the two bobbin halves 642, 644 are held together by a layer of tape 650 after the two bobbin halves are joined.
  • Bobbin halves 642, 644 function as an insulator while holding the secondary windings in place.
  • the number of gapped laminations 632 and solid laminations 734 in a current transformer core 630 can be adjusted depending upon the range of current values that need to be sensed by the motor circuit protector 100.
  • the ratio of gapped-to-solid laminations of the current transformer 500 can be adjusted to sense currents ranging from 9 amperes to 3,000 amperes or any ranges in between. The particular range may depend upon the particular locked-rotor or in-rush current specifications provided by the motor manufacturer.
  • FIG. 8 illustrates an isometric view of a current transformer core 800 according to certain embodiments of the present disclosure with a gapped-to-solid laminations ratio of approximately 1:3. In other implementations, the gapped-to-solid laminations ratio is 1:2.
  • a current transformer, such as the one illustrated in FIG. 8 includes solid laminations 810 for relatively low current sensing, for example, in the range of 9 or 10 amperes, for current under locked-rotor conditions.
  • the gapped laminations 812 can then be utilized to decrease the amount of remnant flux so that the electronic components of the system 200 will be able to receive power from the current transformers 210, 212, 214 and while also accurately sensing high instantaneous fault currents.
  • the core 800 can be constructed with twenty- four laminations comprising eight solid laminations 810 and sixteen gapped laminations 812. Alternately, the core 800 can be constructed with sixteen solid laminations and eight gapped laminations. The twenty-four laminations are stacked and staked together as shown for core 800 so that the lamination edges 820 are substantially aligned with each other. The core 800 is assembled with the gapped laminations 812 having a cumulative thickness ranging from 0.13 inches to 0.145 inches. The solid laminations 810 are stacked to the gapped laminations 812 to achieve a total core thickness ranging from 0.39 inches to 0.44 inches.
  • the individual gapped laminations 812 and solid laminations 810 are approximately 0.016-0.019 inches thick at lamination edge 820. In certain embodiments, no lamination materials extend beyond the surface (the lamination edge 820) of the outermost and innermost laminations due to the staking process.
  • the nominal solid lamination area is approximately 0.0607 in 2
  • the nominal gapped lamination area is approximately 0.0304 in 2 .
  • the size of the gap in the gapped laminations is approximately 0.085 in.
  • FIG. 9 illustrates an isometric view of a current transformer core 900 according to certain embodiments of the present disclosure.
  • the core 900 includes seven-eighths gapped laminations and one-eighth solid laminations, representing a gapped-to-solid laminations ratio of 7:1.
  • the core 900 includes twenty-four laminations as illustrated with three solid laminations 910 and twenty-one gapped laminations 912. The twenty-four laminations are stacked and staked together as shown for the core 900 so that the lamination edges 920 are substantially aligned with each other.
  • the core 900 has a total core thickness ranging from 0.39 inches to 0.44 inches.
  • the individual solid laminations 910 and gapped laminations 912 are each approximately 0.016-0.019 inches thick at the lamination edge 920.
  • no lamination materials extend beyond the surface (formed by the lamination edge 920) of the outermost and innermost laminations due to the staking process.
  • the nominal solid lamination area is approximately 0.0114 in 2
  • the nominal gapped lamination area is approximately 0.0797 in 2 .
  • the size of the gap in the gapped laminations is approximately 0.085 in.
  • the ratio of gapped laminations 632, 812, 912 to solid laminations 734, 810, 910 in the single stacked current transformer core 630, 800, 900 can be determined by balancing output level and remnant flux parameters.
  • the power-up output levels are adjusted by the number of solid laminations, and as the number of solid laminations increases, the linear portion of the current transformer's operating range is extended, pushing the knee threshold of the core's transfer function higher (i.e., the core's saturation region begins at higher peak currents).
  • gapped laminations are added for higher fault current detection based on the remnant flux requirements. As each gapped lamination is added, the core's saturation region shifts to a higher peak current value.
  • the gapped laminations 812, 912 and the solid laminations 810, 910 in FIGs. 8 and 9 have similar dimensions.
  • the gap 830, 930 in gapped laminations 812, 912 is approximately 0.085 inches.
  • the gap can comprise air.
  • the thickness of the individual laminations can be around 0.016-0.019 inches each.
  • the width of the individual laminations from the side inner edge 835, 935 to the side outer edge 845, 945 and the top inner edge 850, 950 to the top outer edge 855, 955 can be around 0.21-0.22 inches.
  • the height of the lamination from the top outer edge 855, 950 to the bottom outer edge 865, 965 can be around 1.13-1.15 inches.
  • the height of the space define by the interior space of the laminations from top inner edge 855, 955 to the bottom inner edge 860, 960 can be around 0.70-0.72 inches.
  • the width of the laminations can vary and the laminations can taper slightly from the upper portion to the lower portion of the laminations.
  • the lamination width from the left upper outer edge 870, 970 to the right upper outer edge 875, 975 can be around 0.90-0.94 inches.
  • the lamination width from the left lower outer edge 880, 980 to the right lower outer edge 885, 985 can be around 0.86-0.90 inches.
  • the lamination width for the interior space of the laminations can also vary and taper slightly from the upper portion to the lower portion.
  • the lamination width from the left upper inner edge 872, 972 to the right upper inner edge 877, 977 can be around 0.47-0.51 inches.
  • the lamination width from the left lower inner edge 882, 982 to the right lower inner edge 887, 987 can be around 0.44-0.48 inches.
  • the gapped laminations 812, 912 are generally shown as C-shaped or reverse C-shaped.
  • Other embodiments of the present invention contemplate gapped laminations that are L-shaped or U-shaped, or variations thereof, where the gapped laminations are staked with some solid laminations at the front or back of the core.
  • the gapped laminations are partially gapped or notched instead of having a full gap.
  • Gapped laminations 812, 912 and solid laminations 810, 910 can be made of an iron alloy that, for example, comprises silicon, aluminum and iron, such as 26 gauge non- oriented Si-Al-Fe semi-processed cold rolled steel (ASTM 47Sl 75).
  • the laminations can further be heat treated for approximately one hour at a temperature of approximately 1,550 0 F in a hydrogen/nitrogen atmosphere as set forth in the American Society of Testing Material (ASTM) Standard 683.
  • alternate metallic materials can be used including, but not limited to, steel, transformer iron, or nickel.
  • the laminations can be coated with a C4-AS antistick coating available from AK Steel Corp., or an equivalent coating.
  • the coating is applied to the surface of the individual laminations in the current transformer's core prior to the punching and stacking operations.
  • the coating provides an insulating barrier between the laminations that can withstand elevated temperatures during the annealing process.
  • a primary function of the coating is to provide surface insulation between the layers of the stacked core, which prevents eddy currents from flowing from one lamination to the next. Eddy currents are undesirable, because they cause the resistive steel laminations to heat up. This heating reduces the current transformer's efficiency and requires a more expensive construction to withstand the additional heat rise.
  • Application of a coating can also inhibit rusting to a certain extent.
  • FIG. 10 is a perspective view of a current transformer 1000 without a housing according to certain embodiments of the present disclosure.
  • the current transformer 1000 includes a core 1030 comprising both solid laminations 1032 and gapped laminations 1035 combined in a single stacked core.
  • a bobbin 1040 can be secured around the stacked laminations.
  • a first lead pin 1022 and a second lead pin 1024 can be secured to the bobbin 1040 such that the pins 1022, 1024 extend vertically from the bobbin 1040.
  • the bobbin 1040 is placed around the core 1030, and a magnet wire 1050 is wrapped around the bobbin 1040.
  • the wire 1050 is first wrapped approximately six turns around first lead pin 1022, then wound around the bobbin 1040, and then finished with approximately six turns around the second lead pin 1024.
  • the wire 1050 can be wrapped around the bobbin 1040 for approximately 420 turns to achieve an approximate resistance of 12 ⁇ .
  • the magnet wire 1050 can be #32 AWG with heavy build polyurethane and a temperature requirement of 155 0 C.
  • a circuit breaker such as motor circuit protector 100
  • the current transformers can have iron cores and function to send current and to fault power the trip unit electronics.
  • Each current transformer 210, 212, 214 senses different phase currents (traditionally labeled A, B, and C, each 120 degrees apart from one another) of the motor circuit protector 100.
  • the number of secondary turns of wire 1050 about the bobbin varies and in certain embodiments ranges from 400 to 420 turns.
  • Embedded software 230 is provided for switching a trip unit, such as the motor circuit protector 100, when detecting a failure mode in the trip unit.
  • the software 230 implements switch detection algorithms that include failure mode detection.
  • the algorithm 230 can be used on any trip unit system that accesses calibrated trip pick-up data, including the motor circuit protector 100.
  • the software translates user-adjustable trip unit settings to pick-up levels by accessing stored calibrated trip data in a data table.
  • the translation technique includes data compression of trip point data, diagnostic checksums, switch to trip point memory mapping, and extension of data settings to elevated temperatures. Normalized templates including normalized trip point data are used as a starting point for calibrating the embedded software.
  • aspects of the present invention enable a fail-safe operation mode where user adjustments (such as adjustments of the full load ampere dial 114 and/or the instantaneous trip point dial 116) can revert to predetermined protective levels.
  • An electronic circuit for a potentiometer is configured to present a percentage of a microcontroller's analog/digital ("A/D") full scale to an A/D input pin, where one channel is used for each user adjustment position.
  • the user adjustment circuit 224 can be used as a switch for detecting an open contact fault, a short-to-ground fault, and/or a short to a supplied or reference voltage.
  • the potentiometer is coupled with an adjustment button, which is generally a mechanical button, that includes switch-like stop and detent features for translating mechanical orientation angles to a potentiometer mechanical orientation.
  • the user adjustment circuit 224 can be adjusted by rotating a dial similar to the full load ampere dial 114 and/or the instantaneous trip point dial 116.
  • the potentiometer's vulnerability to electrostatic discharge is decreased by increasing an over-surface distance of the adjustment button.
  • the adjustment button interacts with a cover to increase the likelihood that the adjustment button will easily rotate only to a designed switch position, not to an unintended in-between position.
  • the adjustment button interacts with the cover to have increased consistent feel to a user by incorporating, for example, three detent pressure arms (or spring elements) located symmetrically around the user adjustment button 120 degrees apart.
  • low cost components can be utilized (while achieving improved over-all system performance), eliminating need for switch- calibration, and providing the ability to use quantitative techniques to verify switch performance in a production test process.
  • Trip unit products can be easily and securely updated, independent of embedded software product design. For example, trip point changes in relation to switch settings can be made without changing product software code as long as data points are within a maximum/minimum range.
  • a statistical distribution of data corresponding to switch settings can be used to determine position thresholds.
  • the position thresholds and device performance are monitored for each trip unit.
  • automated process techniques can be used during product development to quantitatively monitor user adjustment performance. For example, mechanical torque, angular orientation, and microprocessor data have correlated profiles that can be quantitatively adapted for monitoring user-adjustment performance. This quantitative approach is an improvement over an approach that requires manual inspection of mechanical user adjustment.
  • the automated process technique involves a functional tester with two motors that can rotate the switches 114, 116 to any position. The motors are coupled to motor drivers that detect the amount of current needed to drive each switch 114, 116 to different positions.
  • a torque can be derived directly from this current, and the rotation (in degrees) can be derived from the torque or from optical decoders in the motors that detect the amount of rotation a motor shaft has turned.
  • the functional tester is coupled to communicate the switch rotation angle to the microcontroller 226.
  • the automated process technique automatically rotates the switches 114, 116 to various positions, measures the corresponding torque required to put the switches into the various positions, calculates the angle of rotation (i.e., the distance traveled by the motor) from the torque or from the optical decoders, and communicates, via the microcontroller 226, an A/D count that represents the voltage level from a potentiometer 1210. [000198] FIGs.
  • HA and HB illustrate an electrical schematic of a user-adjustment button and a plurality of electro-mechanical orientations (i.e., "P1"-"P9"), respectively.
  • Pl corresponds to a first position of the user-adjustment button
  • P2 corresponds to a second position
  • Switch position ranges, Pl Range through P9 Range correspond to respective ranges of mechanical orientation positions of the user-adjustment button. For example, if the user-adjustment button has a mechanical orientation position anywhere within Pl Range, then its position is Pl.
  • An important aspect of this implementation is that there is a lack of continuity between switch position ranges. Each position range is continuous with respect to its neighboring position range(s).
  • the electro-mechanical orientations are generally mechanical switch orientations of a user-adjustment button that are translated to corresponding analog signal levels by way of a resistive potentiometer.
  • the button and the user adjustment circuit are described in more detail below in reference to FIGs. 12A-18B.
  • the user adjustment circuit is mechanically aligned with the user-adjustment button so that button position "P5" 1103 is nominally at 50% resistance.
  • An analog/digital (“A/D”) reference voltage (“Vdd”) is presented to a switch circuit, and each analog voltage converted by the A/D converter into corresponding digital values can be expressed as a percentage of the reference voltage (i.e., "%Vdd").
  • the mechanical orientation of the switch relative to a resistive element of the potentiometer sets a signal presented to a microcontroller for measurement.
  • the mechanical design of the switch is illustrated as a nine-position switch, with a "Detent” feature in-between positions and “Stop” features at the switch extremes (i.e., "Pl” and “P9”).
  • Table 1 shows some of the electro-mechanical parameters considered in the software design. Table 1 - User Adjustment Switch Electro-Mechanical Orientation
  • the switch positions can be determined from experimental test results of voltages at the microcontroller's inputs for each of the desired mechanical positions, i.e., A/D inputs also referred to as "FLA” (full load amperes) and “Im” (instantaneous trip point current) inputs.
  • A/D inputs also referred to as "FLA” (full load amperes) and “Im” (instantaneous trip point current) inputs.
  • FLA full load amperes
  • Im instantaneous trip point current
  • Switch error detection is accomplished by implementation of a "SW_HIGH_ERR" specification, independently, for both "FLA” and “Im” switches. If a switch is oriented past a stop-feature maximum limit, then a switch error will be detected and the switch logic shall revert to a specified position, such as illustrated in Table 2. For example, when the "SW_HIGH_ERR” limit is reached, both the "FLA” and the “Im” switches default to position 1 setting, independently.
  • trip points stored in the EEPROM 270 (there are 81 in a specific aspect, which represent high temperature settings) are associated with 27 FLA and Im position combinations.
  • a diagnostic routine periodically adds up all the trip point data values and compares the summed values against a checksum. If the checksum does not match the summed values, a Diagnostics Trip will occur, eventually causing the MCP 100 to trip. Alternately, instead of causing a Diagnostics Trip, the diagnostic routine can revert to predetermined trip point settings. In an aspect, the predetermined settings are set to a low pickup level. In this manner, the integrity of trip points and trip data stored in the EEPROM 270 can be verified. When the verification fails, either tripping can occur, or the trip curve settings can be automatically reverted to predetermined low pickup settings. [000204] On start-up, switch positions should be determined before attempting instantaneous ("INST”) trip detection.
  • INHT instantaneous
  • the software 230 should read the correct switch positions at the nominal (or center) mechanical switch adjustment markings. Labels identifying the adjustment markings should be aligned to mechanical specifications.
  • a user adjusts the switch positions, either from an "Energized” or “De-energized” state.
  • the software design considers one or more of the electrical and software parameters shown below in Table 3. While the application is running, the switch settings are updated at the "Switch Change Perception” rate. A minimum “Switch Change Perception” rate may be specified to spread over time a temperature compensation calculation.
  • FSv corresponds to the full-scale voltage of the A/D converter to which the FLA and Im inputs 244, 246 are coupled.
  • FSv may correspond to 5 volts (nominal).
  • the A/D converter may be part of the measurement circuit 241 shown in FIG. 2. Note, for clarity, the measurement circuit 241 is shown coupled to inputs 232, 238, and 240. However, it is understood that the measurement circuit may also be coupled to inputs 244, 246. Alternately, the inputs 244, 246 may be presented to another A/D converter, either in the microcontroller 226 or external to the microcontroller 226.
  • Switch position settings may determine product trip curve settings. These settings are realized by implementing a switch to an EEPROM 270 trip point lookup algorithm. The same translation algorithm can be implemented in a plurality of circuit breakers. Each switch setting permutation may correspond to a specified pair of "A" and "B/C" trip points as per breaker trip settings specifications.
  • the "A” and “B/C” trip points may be implemented as 16 bit words in 8 bit EEPROM memory 270.
  • the formatting of "A” and “B” trip data can be identical and 10 bit left justified.
  • the “C” trip points are packed within the "B/C” word and 5 bit right justified. This trip data organization is convenient for implementing the switch translation algorithm, specified by the equations listed below in Table 4.
  • the trip curve profiles are stored in the EEPROM memory 270.
  • the various combinations of "FLA” 114 and “Im” 116 adjustments will cause the control algorithm 230 to point to specific pickup values stored in EEPROM memory 270.
  • the EEPROM values will represent the actual A/D pickup levels for the corresponding settings.
  • Table 5.13.1 shows the storage requirements for trip curve implementation in the EEPROM 270.
  • the software trip curve settings are dependent on the combination of "FLA” and "Im” user adjustment switches 114, 1 16. For example, in an implementation, there are nine different FLA settings, in addition to nine "Im” settings for each of the "FLA” settings. This is equivalent to eighty-one different trip curve profiles for the circuit breaker 100. Each of the eighty-one different settings correspond to a different trip profile.
  • the circuit breaker 100 may have a current rating of 3OA rms, 50A rms, etc. For each current rating, there are different FLA settings as set forth in the table below.
  • Trip Curve Adjustment "FLA" Trip Curve Adjustment
  • FIG. HC is a flowchart illustrating the coupling of a mechanical button to a user adjustment circuit for setting an operating trip curve in a circuit breaker.
  • the mechanical button is operatively coupled to the potentiometer (1110).
  • the mechanical button can be operatively coupled to the user adjustment circuit as described below in reference to FIGS. 12A-17. Accordingly, adjustment of the mechanical button results in adjustment of the user adjustment circuit.
  • the mechanical button is adjusted to a first position (1112).
  • the mechanical adjustment causes a first signal to be received from the user adjustment circuit (1114).
  • the first signal is indicative of a trip curve.
  • the first signal is associated with one of a plurality of trip curves (1116) and a first trip curve is produced in response to the association between the first signal and the plurality of trip curves (1118).
  • An operating trip curve is set to be the first trip curve (1120).
  • FIGS. 12A and 12B illustrate a trip unit assembly 1200 that generally includes one or more copper components to carry electrical current, a set of current transformers (one per phase) to measure the electrical current, and a circuit board to process information.
  • the trip unit assembly 1200 is an alternative embodiment of the motor circuit protector 100 and can generally include similar components and operate as described above in reference to FIGS. 1- 3.
  • the internal components of the trip unit assembly 1200 e.g., copper components, circuit board, etc.
  • the trip unit assembly 1200 includes one or more user adjustment buttons 1206 for controlling electrical current trip curves of the trip unit assembly 1200. These buttons 1206 may correspond to the FLA dial 114 and the instantaneous trip point dial 116 shown in FIGS. 1-3.
  • FIG. 13 illustrates a partial cross-sectional view of the trip unit assembly 1200 at a rotational center of one of the adjustment button 1206.
  • the trip unit assembly 1200 includes a printed wire assembly 1208 to which a potentiometer 1210 is attached.
  • the potentiometer 1210 has a shaped pocket 1211 at a top face of a potentiometer button 1212 for receiving snugly the corresponding adjustment button 1206.
  • the potentiometer button 1212 via the shaped pocket 1211, connects the adjustment button 1206 and the potentiometer 1210 during rotational movement of the button 1206.
  • the cover 1204 encapsulates an upper portion of the adjustment button 1206.
  • FIGs. 14A and 14B illustrate features of the adjustment button 1206.
  • the adjustment button 1206 includes a spring element 1206a, a rigid base 1206b, a flex member 1206c, a location nipple 1206d, a stop 1206e, a stopping surface 1206f, an insulation disc 1206g, a protrusion 1206h, and a shoulder 1206j.
  • the adjustment button 1206 can include any number of features in accordance with the claimed invention.
  • the illustrated adjustment button 1206 includes three spring elements 1206a and two stopping surfaces 1206f.
  • the spring element 1206a includes the rigid base 1206b, the flex member 1206c, and the location nipple 1206d.
  • the rigid base 1206b is in direct contact with the shoulder 1206j and connects two flex members 1206c of respective adjacent spring elements 1206a.
  • a gap separates the flex member 1206c and the shoulder 1206j, and the location nipple 1206d is located generally in a central location of the flex member 1206c.
  • the stop 1206e is located generally over one of the rigid bases 1206b and is in contact with the shoulder 1206j. Furthermore, the stop 1206e includes the two stopping surfaces 1206f, which are symmetrically located at opposing ends of the stop 1206e.
  • the shoulder 1206j is generally a cylinder centrally located on top of the insulation disc 1206g. The shoulder 1206j is surrounded by the spring elements 1206a and the stop 1206e. Starting on a top surface of the shoulder 1206j, an arrow-shaped blind hole 1206k is provided for receiving a tool when rotational movement of the adjustment switch 1206 is required.
  • the insulation disc 1206g is located at the bottom of the adjustment button 1206, below the shoulder 1206j.
  • the insulation disc 1206g has a diameter that is greater than the diameter of the shoulder 1206j, to increase resistance to ESD and to provide protection against pollutants entering the cavity located between the insulation disc 1206g and the printed wire assembly 1208.
  • a user such as a customer, touches a top exterior surface of the cover 1204, static electricity carried by the user may try to reach internal electronics through air or over surfaces located between the adjustment button 1206 and the cover 1204.
  • the insulation disc 1206g increases the distance that ESD needs to travel to go from a front face of the adjustment button 1206 (e.g., a top surface of the adjustment button 1206 in which the arrow-shaped hole 1206k is located) to the potentiometer 1210 and other components on the printed wire assembly 1208.
  • the insulation disc 1206g increases ESD protection by increasing through-air or over-surface distance of the adjustment button 1206.
  • the insulation disc 1206g protects against pollutants (such as environmental debris, dust, oil, and the like) from entering the cavity between the insulation disc 1206g and the printed wire assembly 1208, which may interfere with the potentiometer 1210.
  • a bottom surface of the insulation disc 1206g is greater than the bottom face of the potentiometer 1210.
  • the insulation disc 1206g has a diameter that is greater than the largest dimension of the potentiometer button 1212.
  • the bottom surface of the insulation disc 1206g is shaped and sized such that it exceeds the largest dimension of the potentiometer button 1212 to protect the potentiometer 1210 from ESD and/or pollutants.
  • the larger size of the insulation disc 1206g also prevents application of down force on the potentiometer button 1212, thereby protecting the potentiometer button 1212 from damage.
  • the protrusion 1206h is centrally located on a bottom surface of the insulation disc 1206g and has a cross-shaped profile.
  • the illustrated embodiment of the protrusion 1206h is also referred to as an "X" style protrusion.
  • FIGs. 15 A and 15B illustrate the printed wire assembly 1208 having two potentiometers 1210.
  • Each potentiometer 1210 has a rotational center with the pocket 1211 on the potentiometer button 1212 for receiving a respective protrusion 1206h.
  • the pocket 1211 is an "X" style pocket for receiving the respective "X” style protrusion 1206h.
  • the adjustment switches 1206 are assembled correspondingly on the potentiometers 1210, with the "X" style protrusion 1206h being snugly inserted into the "X" style pocket 1211 of a respective potentiometer button 1212.
  • FIGs. 16A-16C illustrate the interaction between the adjustment switch 1206 and the cover 1204 (viewing from inside the cover in FIGs. 16B and 16C) at the spring elements 1206a level.
  • the adjustment switch 1206 has been sectioned in FIG. 16C to remove the insulation disc 1206g for more clearly showing the spring elements 1206a from below.
  • the cover includes a hole 1204e through which the shoulder 1206J of the adjustment switch 1206 protrudes such that the top surface of the shoulder 1206j is generally planar with a top surface of the cover 1204 (as shown in FIG. 16A).
  • the hole 1204e of the cover 1204 includes a bearing surface 1204a, two stop limits 1204b, a plurality of position detents 1204c, a plurality of detent walls 1204d, a plurality of crests 1204f, and a plurality of troughs 1204g.
  • the bearing surface 1204a defines in part the circular hole 1204e, which locates the adjustment switch 1206 and allows rotational movement of the adjustment switch 1206.
  • the shoulder 1206j has a diameter dimensioned such that a top portion of the shoulder 1206J can protrude through the hole 1204e.
  • each stop limit 1204b is located below the bearing surface 1204a. Specifically, each stop limit 1204b is a surface formed by removing material along the depth of the hole 1204e such that a partial greater-diameter hole is formed within the hole 1204e.
  • the position detents 1204c are located below the stop limits 1204b, along the circumference and near the bottom of the hole 1204e (in the interior of the cover 1204).
  • Each detent 1204c is defined by two detent walls 1204d coupled by a trough 1204g.
  • each detent 1204c is connected to another detent 1204c by a common crest 1204f. Specifically, the crest 1204f is located at the intersection of two detent walls 1204d that are not part of the same detent 1204c and that is a point generally closest to a center axis of the hole 1204e.
  • the flex members 1206c are generally aligned with the position detents 1204c along an axial direction of the hole 1204e. Additionally, a center axis of the adjustment switch 1206 is generally collinear with the center axis of the hole 1204e. Each of the location nipples 1206d is located within a corresponding clearance formed by two detent walls 1204d between two consecutive crests 1204f.
  • the location nipples 1206d comes into contact with the detent walls 1204d.
  • the flex member 1206c of the spring elements 1206a elastically deforms towards the center axis of the adjustment switch 1206 to allow the location nipple 1206d to move over a crest 1204f of a position detent 1204c.
  • the location nipple 1206d of each spring element 1206a past a respective crest 1204f the location nipple 1206d is forced by the flex member 1206c into a centered position between two detent walls 1204d that are not joined by a crest 1204f. In the centered position the location nipple 1206d is generally aligned with the trough 1204g of a respective detent 1204c.
  • the crests 1204f are designed such that they reduce the likelihood that a location nipple 1206d of the adjustment switch 1206 will statically stop on top of any crest 1204.
  • the angles and radius sizes of the crests are selected to provide crests that are as small as possible for achieving the current invention.
  • the detent walls 1204d should have an angle that allows easy centering of the location nipples 1206d. Accordingly, the design of the position detents 1204c should reduce, or eliminate, the amount of play that the adjustment switch 1206 can move relative to the hole 1204e.
  • the feel and accuracy of the position detents 1204c movements should take into considerations other factors, such as possible tolerance stack-ups of the potentiometer 1210 relative to the printed wire assembly 1208, the "X" style protrusion 1206h relative to the "X” style pocket 1211, etc.
  • FIG. 17 illustrates the interaction between the adjustment switch 1206 and the cover 1204 (viewing from inside the cover) at the stop 1206e level, wherein the adjustment switch 1206 has been sectioned to remove features located below the stop 1206e (e.g., insulation disc 1206g, spring elements 1206a, etc.).
  • the adjustment switch 1206 can rotate in either direction (clockwise or counterclockwise) until opposing stops of the two parts make contact. Specifically, the adjustment switch 1206 can rotate until either one of its stopping surfaces 1206f makes contact with a respective stop limit 1204b of the cover 1204. The contact between the stopping surfaces 1206f and the stop limits 1204b ensures that the adjustment switch 1206 will not be rotated beyond a design rotation specification.
  • FIGs. 18 A and 18B illustrate an adjustment switch 1806 according to an alternative aspect of the present invention.
  • the adjustment switch 1806 includes an insulation disc 1806g having a skirt 1806i around its bottom surface to further increase ESD protection and/or to reduce any pollution from entering a corresponding potentiometer.
  • the skirt 1806i is designed to totally encircle the potentiometer.
  • FIG. 19A illustrates a temperature compensation system 1900 according to aspects of the various embodiments disclosed herein.
  • the temperature compensation system 1900 automatically adjusts the trip curves for regions A and B (and not region C) based on fluctuations in environmental temperature sensed by the temperature sensor circuit 222.
  • the temperature compensation system 1900 includes a burden resistor calculation module 1902 and a trip curve calculation module 1904.
  • the burden resistor calculation module 1902 receives a scaled voltage signal, Tsv, from the temperature sensor circuit 222 indicative of temperature.
  • the module 1902 calculates a percentage operation point of the burden resistor relative to 100% at 25 0 C (%BR) by exploiting two sensor relationships: (1) the base-emitter voltage equation of a bipolar transistor 1906 (shown in FIG.
  • the burden resistor 410 is a serpentine, copper burden resistor on a printed circuit board.
  • the burden resistor 410 is disposed near the temperature sensor 222 such that there is temperature coupling between the copper traces of the burden resistor 410 and the temperature sensor 222, which is a pnp transistor 1906.
  • the voltage output of the temperature sensor 222 represents the circuit board temperature and is scaled and presented to an analog-to-digital converter input of the microcontroller 226, which may be an 8-bit microcontroller such as a PIC16F684-E/ST programmable microcontroller available from Microchip Technology, Inc. based in Chandler, Arizona.
  • the trip curve calculation module 1904 receives calibrated trip point data at 90 0 C or 128% from the EEPROM 270.
  • 90 0 C represents the upper temperature range of the compensated burden resistor sensor, though it should be understood that this value is merely exemplary and other upper temperature thresholds may be selected depending upon the desired operation range.
  • the trip curve calculation module 1904 adjusts the trip points upwards or downwards depending upon whether the temperature sensor reading falls above or below the temperature inflection point.
  • burden resistance increases generally linearly with temperature with a positive-going slope.
  • the resistance is determined from the resistance of the burden resistor 410 and the turn-on resistance of the FET 412.
  • the slope of the curve depends upon the temperature coefficient for copper, which is approximately 4000 parts per million in this particular example.
  • the temperature sensor circuit 222 includes the pnp transistor 1906 having a base-emitter voltage that varies with temperature. As the temperature increases, the base-emitter voltage of the pnp transistor 1906 decreases, creating a negative-going slope.
  • the nominal temperature sensor equation can be determined experimentally.
  • the offset, 0.6504, is typical, but may have to be adjusted upwards or downwards to represent a nominal curve.
  • the nominal slope (-0.0021) does not require compensation. If the burden resistance curve is superimposed over the temperature sensor curve, the two curves intersect at an inflection point, which in this very specific and non- limiting example is 51.3 0 C. While assumptions have to be made about the symmetry of the two curves before and after the inflection point, the inflection point is useful for efficiently converting the temperature sensor readings directly to burden resistance percentage values.
  • the burden resistor calculation module 1902 calculates the percentage on the normalized burden resistance from the scaled temperature sensor voltage, Tsv.
  • the resolution and range of temperature sensor readings and burden resistor percentage readings are matched about the temperature inflection point.
  • a linear equation for temperature is converted to a linear equation for normalized burden resistance.
  • the following exemplary table illustrates the various parameters and their values for converting the temperature curve to a corresponding burden resistance curve. Of course, it should be understood that the values provided in the following table are merely exemplary.
  • the trip curve calculation module 1904 adjusts trip thresholds downwards in specified constant steps until an estimated burden resistance is determined.
  • Trip points A and B are stored in the EEPROM 270 at Ts_HIGH or BR_MAX (e.g., 90 0 C or 128%).
  • Trip curve initialization iterates a specified number of steps (27 in this non-limiting example) to match estimated burden resistance with actual burden resistance readings.
  • the following table summarizes the parameters and their values involved in the trip curve initialization:
  • the temperature sensor 222 may have an internal offset that is known at calibration time. Temperature sensors from one to another may vary, and temperature calibration values can be stored to add or subtract the internal offset so that the output of the temperature sensor 222 mimics a true nominal sensor. For example, a temperature sensor 222 outputting high readings relative to nominal can be corrected as follows. Suppose the temperature sensor 222 outputs at 25 0 C 0.3812 volts corresponding to an A/D value of 78 [dec]. A nominal sensor would read 0.5962 volts or 122 [dec]. The temperature compensation algorithm would add 0.215 volts or 44 [dec] to every sensor reading to calibrate the low-reading temperature sensor to a nominal sensor.
  • FIG. 20 is a flow chart diagram of an exemplary temperature compensation algorithm 2000 of the temperature compensation system 1900 according to various embodiments disclosed herein.
  • the algorithm 2000 is implemented as machine instructions executed by the microcontroller 226.
  • the algorithm 2000 determines the temperature inflection point, Inflection_Point, corresponding to the intersection of the temperature sensor curve and the burden resistance curve (2002).
  • the burden resistance includes the resistance of the burden resistor 410 and the resistance of the FET 412.
  • the temperature sensor 222 is read (2004).
  • the algorithm 2000 determines the scaled sensor voltage Tsv from the temperature read by the temperature sensor 222 (2006).
  • the algorithm 2000 may offset the scaled value Tsv by a temperature calibration to calibrate the scaled readings to a nominal sensor output.
  • the algorithm 2000 determines whether Tsv is above the inflection point (2008).
  • the algorithm 2000 determines the %BR for the Tsv value above the inflection point (2010); otherwise it determines the %BR for the Tsv value below the inflection point (2012).
  • the algorithm 2000 retrieves trip points A and B from the calibration EEPROM 270 (2014), which correspond to trip point settings at the upper temperature range or 90 0 C (128%). The trip points are adjusted downward until an estimated BR matches the actual BR by shifting the estimated BR in specified steps (2016). When a match is found, the new trip points A' and B' at the actual BR are stored (2016).
  • FIGS. 21, 22A-22B, and 23A-23D are activity and sequence diagrams expressed in Unified Modeling Language (UML).
  • UML Unified Modeling Language
  • FIG. 21 a trip curve adjustment activity diagram 2100 is shown.
  • the microcontroller 226 Upon power up of the MCP 100, the microcontroller 226 initializes trip curve settings (2102) and then updates the trip curve settings (2104). If there is a change in the switches 114, 116, the trip curve settings are initialized to the trip points for the switch combination pair (2102).
  • FIG. 22 A is a trip curve initialization adjustment sequence diagram of the initialize trip curve settings activity 2102 shown in FIG. 21.
  • a TCInit function is called from the main module 2202.
  • the TC (trip curve) module 2204 reads the switch positions 114, 116 from the switch module 2208 and fetches the thresholds from the EEPROM 270 (2210).
  • the BRGet function is called and the TCUpdate function is iterated 27 times to determine the actual %BR.
  • FIG. 22B expands upon the update trip curve settings activity 2104 shown in FIG. 21.
  • the Get Burden Resistor Data (%BR) activity 2220 is carried out, which is detailed in FIG. 23 A. If the estimated BR is greater than the actual BR, the trip curve data is adjusted downward (2222). If the estimated BR is less than the actual BR, the trip curve data is adjusted upward (2224).
  • FIG. 23A illustrates the Get Burden Data (%BR) activity 2220 in more detail.
  • the scaled temperature sensor data is obtained (2302) (shown in FIG. 23B), and the scaled temperature sensor readings are ranged if necessary (2304) (shown in FIG. 23C).
  • the following guards and actions are applicable to the UML diagram shown in FIG. 23A:
  • the stored sensor data which may be scaled and calibrated for an internal offset of the temperature sensor is read (2310), and the temperature sensor readings are ranged within upper and lower limits (2312) as shown in FIG. 23 C.
  • Tsv is ranged to
  • Tsv if it is less than the HIGHJTEMP. If Tsv is greater than the LOW-TEMP, it is ranged to LOWJTEMP. Otherwise, if Tsv is within the range limits, it is not adjusted.
  • FIG. 23D illustrates an activity diagram for reading temperature sensor 222 data
  • the calibrated offset magnitude is read from the EEPROM 270
  • This offset is a decimal value corresponding to the variance of the temperature sensor 222 readings compared to nominal.
  • the calibrated polarity is read from the EEPROM
  • the offset stored in the EEPROM 270 is added to the scaled raw output of the temperature sensor 222 (2324), effectively calibrating it to a nominal output. Otherwise, the offset is subtracted from the scaled raw output of the temperature sensor 222 (2326) to calibrate its readings to nominal.
  • the microcontroller 226 checks for over-temperature events (2328) and- stores the scaled temperature sensor data (2330).
  • the following exemplary source code exemplifies a TCUpdate routine for updating trip curve thresholds, which can vary as a function of the switch positions and the burden resistance (BR), which varies as a function of temperature.
  • BR_estimate BR_estimate - BRJSTEP
  • TCA. word TCA. word - TC.A_STEP;
  • TCB.word TC.B.word - TCB_STEP;
  • BR_estimate BR_estimate + BR-STEP
  • TC A. word TCA. word + TC.A_STEP;
  • TC.B.word TC.B.word + TCB_STEP;
  • the following exemplary source code exemplifies a TCInit routine for initializing the TC object.
  • BR_estimate BRJNITJESTIMATE
  • FIG. 24 An example flow diagram of an algorithm of the trip module 265 of the control algorithm 230 for redundant trip activation in the motor circuit protector 100 is shown in FIG. 24.
  • the redundant trip aspects of the present invention provide a layer of protection against inadvertent tripping that may be caused by a spurious microprocessor operation, a software error, external effects such as electromagnetic fields, and the like, which can cause the software code to jump to a trip state and trip the motor circuit protector 100 even though no instantaneous short-circuit condition has actually been satisfied.
  • the control algorithm 230 includes the trip module 265 that sets a variable the first time the trip state is reached by the algorithm 230 (here, called a "pre-trip" state).
  • the trip module 265 may require that the pre-trip state be invoked a predetermined number of times, such as two for "B" trip events and five for "A" trip events, before invoking the trip state wherein the trip solenoid 228 is activated.
  • the machine readable instructions comprise -an algorithm for execution by: (a) a processor, (b) a controller, such as the microcontroller 226, and/or (c) any other suitable processing device.
  • the algorithm may be embodied in software stored on a tangible medium such as, for example, a flash memory, a CD-ROM, a floppy disk, a hard drive, a digital versatile disk (DVD), or other memory devices, but persons of ordinary skill in the art will readily appreciate that the entire algorithm and/or parts thereof could alternatively be executed by a device other than a processor and/or embodied in firmware or dedicated hardware in a well known manner (e.g., it maybe implemented by an application specific integrated circuit (ASIC), a programmable logic device (PLD) 5 a field programmable logic device (FPLD), discrete logic, etc.). Also, some or all of the machine readable instructions represented by the flowchart of FIG. 24 may be implemented manually.
  • ASIC application specific integrated circuit
  • PLD programmable logic device
  • FPLD field programmable logic device
  • FIG. 24a is a unified modeling language diagram of a trip state 2402 according to embodiments of the present invention.
  • FIG. 24b is a unified modeling language diagram of a direct drive trip state 2404 and a stored energy trip state 2406 according to embodiments of the present invention. The following guards and actions are shown in FIGS. 24a and 24b:
  • a trip event can be detected from an instantaneous "A" or "B" trip, a diagnostic trip (caused in a diagnostic mode of the motor circuit protector 100), an over-temperature trip, or a comparator-interrupted trip for self protection of the motor circuit protector 100.
  • FIG. 24a shows the trip state 2402 in a run state mode of operation
  • FIG. 24b shows trip states in a power-up mode of operation of the motor circuit protector 100.
  • the direct drive trip state 2404 is entered upon detection of a high instantaneous self-protection fault at startup of the motor circuit protector 100.
  • the stored energy trip state 2406 is entered upon detection of a trip or self-protection trip.
  • a sample source code for the run mode trip state 2402 is reproduced below. It can be called from a pretrip state for "A” and “B” trips or from an auxiliary task state for diagnostic or over-temperature trips. It would not be called for "C” trips; rather a comparator interrupt service routine is called.
  • TripSetCode (TRIP_CODE); (set trip code variable)
  • FIG. 24c is a unified modeling language diagram the trip state 2402 shown in FIG. 24a, which is implemented as a software algorithm executed by the microcontroller 226.
  • the interrupts are stopped (2420), and the TripCode variable is set (2412).
  • decision (2414) the algorithm determines whether the code equals TRIP_CODE. If so, the TripCode variable is set to TRIP_CODE. If not, the TripCode variable is initialized to zero (2416). After the algorithm determines whether the code equals TRIP-CODE 3 the algorithm calls the Trip Module Activate (2418).
  • the guard at decision (2420) determines whether the TripCode variable equals TRIP CODE, and if so, the algorithm sends trip signals to the I/O (2422), which are received by the voltage regulation module 260 and the trip circuit 218.
  • the trip signals include a set to charge mode that is received by the voltage regulation module 260 and a trip circuit signal that is received by the trip circuit 218. In the charge mode, the burden resistor control FET 412 is turned off by a signal from the burden resistor control output 252 of the microcontroller 226. Additionally, the over-voltage trip circuit 220 (which is also referred to as backup trip circuitry) will activate the trip circuit 218 independent of the control algorithm 230, when the power supply voltage exceeds its voltage trip threshold.
  • the microcontroller 226 is configured to send the trip signal 250 (2406) to the trip circuit 218.
  • the solenoid 228 will normally be activated via this signal path.
  • the over-voltage trip circuit 220 will activate the trip circuit 218.
  • FIG. 25 is a UML diagram illustrating a state diagram 2500 for the INST trip regions A (locked-rotor avoidance region) and B (in-rush avoidance region).
  • the state diagram 2500 includes the following Guards and Actions specified below.
  • Trip B At least 2 INST Pre-trip B events AND Pre-trip B delay
  • Trip A At least 5 ESfST Pre-trip A events AND Pre-trip A delay.
  • the control algorithm 230 fetches the trip curve from EEPROM 270 (Fl).
  • the trip curve may be selected based upon the positions of the dials 114, 116 detected by the control algorithm 230.
  • the trip curve for the selected dial 114, 116 combination is read from the EEPROM 270.
  • the peak current is sensed and monitored in monitor composite state 2504. If the peak current is greater than the in-rush avoidance current threshold B, a pre-trip B event signal (F3) and a pre-trip A event signal (F4) are activated.
  • the controller 230 enters simultaneously an INST pre-trip B state 2506 and an INST pre-trip A state 2508 and polls until a pre-trip event signal is received. If the peak current is greater than the locked-rotor avoidance current threshold A and less than the in-rush avoidance current threshold B, a pre- trip A event signal (F4) is activated and the INST pre-trip A state 2508 is entered. In the INST pre-trip B state 2506, a pre-trip B timer is active (G3), and in the INST pre-trip A state 2508, a pre-trip A timer is active (G4). When a pre-trip event signal is received (F5), an INST pre-trip state 2510 is entered for the duration of a pre-trip timer (G5).
  • Trip logic (G6) is parsed before transitioning to a trip state 2512.
  • the trip logic depends upon whether a Trip B or Trip A event has been detected.
  • the Trip Logic requires at least two INST pre-trip B events to occur and a pre-trip B delay to expire.
  • the Trip Logic requires at least five INST pre-trip A events to occur and a pre- trip A delay to expire. Once these conditions are satisfied, the trip state 2512 is entered and the state machine 2500 ends.
  • the solenoid 228 may be activated by the direct-drive sequence or stored-energy trip sequence or both.
  • FIG. 26a is a diagram expressed in a Unified Modeling Language (UML) illustrating a power-up (i.e., startup mode) activity diagram 2600 according to an implementation of the present invention.
  • the activity diagram 2600 conventionally includes Guards, designated by the letter G, and Actions, designated by the letter F.
  • Guards and Actions are provided below:
  • the state diagram 2600 initializes to a PowerUp 1 state 2602, which detects a power-up or startup of the motor circuit protector 100 (e.g., primary current is applied when the handle 120 is moved to the ON position 124).
  • the control algorithm 230 is initialized for first half-cycle self-protection (region C), and half-cycle self-protection 2604 is carried out by the control algorithm 230. If a high INST self-protection fault is sensed (G2), the state diagram 2600 moves to a Direct Drive Trip state 2606, which activates a Direct Drive trip W
  • the control algorithm 230 initializes for voltage regulation start (in the power supply circuit 216) (F3), and the state diagram 2600 transitions to a start regulator state 2610. If the voltage regulator in the power supply circuit 216 reaches a Stored Energy trip voltage level, the control algorithm 230 is initialized for a run (or steady- state) mode (F4). A run mode INST self-protection state 2612 is maintained until a self- protection trip is detected (G7), and the state diagram 2600 enters a Stored Energy trip state 2614.
  • FIG. 26b is a UML diagram of the Run/Main Loop state 2616 shown in FIG. 26a.
  • the Run/Main Loop state 2616 includes a Peak Detection state 2620, a PreTrip Detection state 2622, an Auxiliary Task Execution state 2624, a Regulation state 2626, and a Trip state 2628. Secondary currents are sampled via the scaled current comparator input circuit 404 and their peaks are recorded via the peak current input circuit 402 in the Peak Detection state 2620. Pre-trip conditions are monitored periodically in the PreTrip Detection state 2622. Auxiliary tasks are carried out in the Auxiliary Task Execution state 2624, including updating trip curves based on temperature, diagnostics, or dial 114, 116 positions. The Trip state 2628 sets the trip software code and activates the trip sequence.
  • FIG. 26c is a UML activity diagram of the Start Regulator state 2610 shown in FIG. 26a. If the Start Regulator state 2610 is transitioned from the Run/Main Loop state 2616, the A/D converter is configured 2632 and the power supply is charged 2634 to a stored energy trip voltage. If the Start Regulator state 2610 is transitioned from the PowerUp 2 state 2608, A/D configuration is bypassed and the power supply is charged 2634 to a stored energy trip voltage.
  • FIG. 27 Another example flow diagram 2700 of the voltage regulation module 260 of the control algorithm 230 for voltage regulation in the motor circuit protector 100 is shown in FIG. 27.
  • the machine readable instructions comprise an algorithm 2700 for execution by: (a) a processor, (b) a controller, such as the microcontroller 226, and/or (c) any other suitable processing device.
  • the algorithm 2700 may be embodied in software stored on a tangible medium such as, for example, a flash memory, a CD-ROM, a floppy disk, a hard drive, a digital versatile disk (DVD), or other memory devices, but persons of ordinary skill in the art will readily appreciate that the entire algorithm and/or parts thereof could alternatively be executed by a device other than a processor and/or embodied in firmware or dedicated hardware in a well known manner (e.g., it may be implemented by an application specific integrated circuit (ASIC), a programmable logic device (PLD), a field programmable logic device (FPLD) 5 discrete logic, etc.). Also, some or all of the machine readable instructions represented by the flowchart of FIG. 27 may be implemented manually.
  • ASIC application specific integrated circuit
  • PLD programmable logic device
  • FPLD field programmable logic device
  • example algorithm 2700 is described with reference to the flowchart illustrated in FIG. 27, persons of ordinary skill in the art will readily appreciate that many other methods of implementing the example machine readable instructions may alternatively be used. For example, the order of execution of the blocks may be changed, and/or some of the blocks described may be changed, eliminated, or combined.
  • the motor circuit protector 100 is first activated via the reset of the voltage regulator circuit 408, which causes the stored energy circuit 304 to charge to a level sufficient to run the electronic components (2702).
  • the process enters a first power-up mode (2704), which provides time to initialize the hardware and software components.
  • the microcontroller 226 initializes the software in the first half cycle (e.g., about 4 ms) (2706).
  • the microcontroller 226 turns on the burden resistor control FET 412 for a fixed time period and measures whether an excessive instantaneous current is detected by the scaled secondary current input 238 (2708).
  • the burden resistor control FET 412 is turned off thereby coupling the secondary current to the stored energy circuit 304, and the breaker trips (2710).
  • the microcontroller 226 enters the charge only mode (2712).
  • the microcontroller 226 monitors the secondary current via the secondary current input 238 for more accurate measurement via the internal A/D converter (2714). If a fault current is detected (2716), the trip circuit 218 is activated to trigger a break (2710). It is to be understood that the current monitoring functions occurs simultaneously with the charging functions described below.
  • the microcontroller 226 charges the capacitors 420 and 422 of the stored energy circuit 304 by turning off the burden resistor control FET 412 and allowing the secondary current to flow to the stored energy circuit 304.
  • the microcontroller 226 configures the energy storage capacitor voltage input 232 to connect to the internal comparator input (2718).
  • the microcontroller 226 measures the voltage of the stored energy circuit 304 (2720) to determine whether the voltage has reached the voltage required for the stored energy circuit 304 to actuate the trip solenoid 228. If the voltage has not reached the voltage necessary to actuate the trip solenoid 228 (2722), the control algorithm 230 continues the charging process. If the requisite voltage is reached (2722), the microcontroller 226 enters the steady-state or run mode (2724).
  • the microcontroller 226 turns the burden resistor control FET 412 on to allow for measurement of the secondary current.
  • the microcontroller 226 also sets the comparator input in the measurement circuitry 241 to detect whether a high instantaneous current is detected from the scaled peak current input circuit 404, the secondary current input 238 remains operatively coupled to the analog-to-digital converter of the microcontroller 226 and the capacitor voltage input 232 remains operative coupled to the analog-to-digital converter (2726).
  • the microcontroller 226 monitors the secondary current continuously while the power cycle occurs for fault currents and high instantaneous currents (2728). If no fault or excessive instantaneous currents are detected the microcontroller 226 remains in normal operation. If an excessive instantaneous current is detected, the microcontroller 226 sends a signal to the trip circuit 218 to trip the breaker (2710). The microcontroller 226 also detects whether the voltage from the stored energy circuit 304 falls under the low set point voltage threshold (2720). If the voltage from the stored energy circuit 304 falls under the low set point voltage, the voltage regulation module 260 changes to the charge mode (2712) to recharge the stored energy circuit 304.
  • the microcontroller 226 determines if the voltage from the stored energy circuit 304 is below the high point voltage threshold (2732). If the voltage from the stored energy circuit 304 is below the high point voltage threshold, the microcontroller 226 initiates charge pulsing of the secondary current via the burden resistor control FET 412 (2734) and returns to the run mode.
  • the charge pulses are of a fixed pulse width. The number of pulses varies depending on the voltage from the stored energy circuit 304. This process continues until the sensed voltage exceeds the high set point voltage threshold.
  • the voltage regulation module 260 allows a wide operating range such as between 9A rms through 250OA rms for the motor circuit protector 100.
  • the module 260 utilizes the configurability of the microcontroller 226 to minimize power supply peak overshoot, minimize voltage regulation ripple, and maintain stored energy trip voltages.
  • the module 260 also reduces the risk of nuisance tripping of the trip circuit 218 while the motor circuit protector 100 transitions between powered and unpowered states.
  • the module 260 cooperates with other modules of the control algorithm 230 to provide a robust fault tolerant backup trip detection system.
  • the examples relate to motor circuit protectors, it is to be understood that the principles described above may be applied to all types of circuit breakers.
  • FIG. 28 shows a non-limiting software trip curve 2800.
  • the trip curve 2800 implements the motor circuit protector's full load ampere (FLA) settings input by a user via the FLA dial 114.
  • the trip curve 2800 further implements the instantaneous trip points from the instantaneous trip point (Im) dial 116 for making motor configuration adjustments at the specified full load ampere setting.
  • the trip curve 2800 illustrates current level thresholds, or pickup thresholds, for triggering the trip circuit 218. It may be desirable that the motor circuit protector 100 be designed to enable easy motor protection setup for users familiar with the NEC or other applicable code or standard.
  • the trip curve 2800 of FIG. 28 assumes that the applied fault starts at time zero on the vertical axis, which measured in seconds.
  • the magnitude of the applied fault is shown on the horizontal axis and is by convention expressed in Amps.
  • the trip curve 2800 of FIG. 28 depicts three independent time/current protection regions, namely a trip region A 2802, a trip region B 2804, and a trip region C 2806.
  • Region A 2802 is also referred to as a locked-rotor avoidance region.
  • Region B is also referred to as an in-rush avoidance region.
  • Region C is also referred to as a self protection region.
  • Software detection algorithms including the instantaneous trip subroutine 268 and the self protection trip subroutine 264 are run in parallel after steady state is reached, focusing on each of the corresponding trip regions.
  • Each trip region 2802, 2804, 2806, is defined by pickup threshold current values A, B, and C, which are stored in the EEPROM 270.
  • the current values A, B, and C form a trip curve, such as the exemplary one shown in FIG. 28.
  • region it is meant that current values corresponding to the primary current below the threshold current value A, B, or C, optionally within a predetermined tolerance (e.g., +/- 5%), will not cause a trip, whereas current values corresponding to the primary current above the threshold current value A, B, or C 3 optionally within a predetermined tolerance, will cause a trip either immediately (such as when region C is exceeded) or after an intentional delay (such as when region A is exceeded) or after an unintentional delay (such as when region B is exceeded).
  • a region may be characterized as a zone of protection, which when exceeded, will cause a trip.
  • the zone of protection may optionally include a predetermined tolerance at the threshold value (A, B, or C).
  • primary current it is meant any characteristic of the primary current, such as the peak primary current, its rms or nominal values, or any current proportional to the primary current including the secondary current (or its peak, rms, or nominal values), to name a few by way of example only.
  • the trip region A 2802 is considered a locked rotor avoidance region and is generally designed to avoid nuisance tripping at motor locked rotor current levels on specified instantaneous trip point settings.
  • a locked rotor condition generally occurs when the rotating member of the motor 204 is locked in a stationary position, causing excess current to be drawn to the locked rotor.
  • the trip region A 2802 implements a delayed trip.
  • the trip region B 2804 is considered an in-rush avoidance region, and the trip region C 2806 is considered a high current breaker self-protection region.
  • the trip regions B and C 2804, 2806 are designed to avoid motor in-rush, which occurs when, on startup of the power source, surges of current or voltage cause erroneous tripping, thereby creating a nuisance.
  • the trip regions B and C 2804, 2806 cause the motor circuit protector to trip as soon as the input signals are qualified and, thus, exhibit a "no intentional delay" characteristic.
  • the trip region G 2806 focuses on the instantaneous self protection trip feature. In an implementation neither the trip region A 2802 nor the trip region B 2804 is active until the steady-state trip region is achieved. After steady state has been reached and the switch settings have been determined, pickup thresholds for the trip region C 2806 may be moved closer to the trip region B 2804. The trip feature of the trip region C 2806 extends upward in time and may be considered a backup to the trip characteristics of both the trip regions A and B 2802, 2804. Note that the trip regions A and B 2802, 2804 can be simultaneously active, or one or both regions 2802, 2804 can be disabled. It should also be understood that regions 2802 and 2804 can be identical.
  • the trip region A 2802 and the trip region B 2804 use the peak current input 238 from the burden resistor and are based on temperature compensated analog to digital (A/D) values.
  • the trip region C 2806 uses the startup peak current comparator input 240, which is connected to a comparator input contained in the power supply 216.
  • the startup peak current comparator input 240 has a relatively fast comparator circuit such that high instantaneous short circuits and/or fault currents may be detected immediately.
  • the trip region C 2806 has a faster qualification trip time than the trip region B 2804 but is less accurate than the trip regions A and B 2802, 2804.
  • the instantaneous trip point settings of the motor circuit protector 100 have pickup thresholds of the trip regions A and B 2802, 2804 set to the same levels without disabling the function of the trip region A 2802.
  • the trip region A 2802 may be considered a backup to the trip region B 2804.
  • the motor circuit protector 100 is designed to have flexible control over the characteristics of the trip curve 2800. All trip curve settings and behavior are specified in the calibration EEPROM 270, where possible.
  • the trip curve 2800 of FIG. 4 shows four test points, Tl 2808, T2 2810, T3 2812, and T4 illustrating current levels that will activate the various trip regions 2802, 2804, 2806.
  • test point Tl 2808 is below the pickup threshold of the trip region A 2802, test point Tl 2808 will not trip the motor circuit protector 100.
  • Test point T2 2810 is above the pickup threshold of the trip region A 2802 and will, thus, pickup with a time delay associated with the trip region A 2802.
  • Test point T3 2812 is above the pickup threshold of the trip region B 2804 and will pickup at time T d (T3).
  • Test point T4 2814 is above the pickup threshold of the trip region C 2806 and will pickup at time T d (T4).
  • the trip regions A, B, and C 2802, 2804, 2806 have variable activation times that are largely dependent on the fault powered supply and trip priorities.
  • the trip region C 2806 may be broken down into a first half-cycle detection "CiHC” during the first 4mS after the processor starts, a second half-cycle detection “C 2 HC” after the power supply comes up but before the switch settings and temperature settings are known, and finally the steady-state trip region "C n HC.”
  • Table 5 below provides the relationship between the trip curve 2800 and the power-up sequence according to one embodiment.
  • the trip curve 2800 of FIG. 28 shows nominal pickup and trip time delays per trip region 2802, 2804, 2806. It is generally desirable to attempt a tripping action close to the nominal pickups and trip times of each trip region 2802, 2804, 2806. It is contemplated that the tolerance specifications for specific settings may cause overlap in the trip regions A, B, and/or C 2802, 2804, 2806, which must be considered when determining test points T2 2810, T3 2812, and T42814.
  • FIG. 29 is a diagram expressed in a Unified Modeling Language (UML) illustrating a power-up (i.e., startup mode) activity diagram 2900 according to an implementation of the present invention.
  • the activity diagram 2900 conventionally includes Guards, designated by the letter G, and Actions, designated by the letter F. A legend of the Guards and Actions is provided below:
  • the state diagram 2900 initializes to a PowerUp 1 state 2902, which detects a power-up or startup of the motor circuit protector 100 (e.g., primary current is applied when the handle 120 is moved to the ON position 124).
  • the control algorithm 230 is initialized for first half-cycle self-protection (region C 2806 of the trip curve 2800 shown in FIG. 28), and half-cycle self-protection 2904 is carried out by the control algorithm 230. If a high INST self-protection fault is sensed (G2), the state diagram 2900 moves to a Direct Drive Trip state 2906, which activates a Direct Drive trip (F2).
  • the state diagram 2900 Upon expiration of a self-protection monitor time (G3), the state diagram 2900 transitions to a PowerUp 2 state 2908.
  • the control algorithm 230 initializes for voltage regulation start (in the power supply circuit 216) (F3), and the state diagram 2900 transitions to a start regulator state 2910. If the voltage regulator in the power supply circuit 216 reaches a Stored Energy trip voltage level, the control algorithm 230 is initialized for a run (or steady-state) mode (F4).
  • a run mode INST self- protection state 2912 is maintained until a self-protection trip is detected (G7), and the state diagram 2900 enters a Stored Energy trip state 2914.
  • a run-mode main loop state 2916 that attempts to maintain the voltage regulator at the Stored Energy trip voltage level.
  • the control algorithm 230 initializes for regulation start (F3) and enters the start regulator state 2910 and maintains this loop until the voltage regulator has reached a Stored Energy trip voltage level. In this way, if a trip is detected (G6), the Stored Energy trip state 2914 has a sufficient voltage to apply to the trip solenoid 228 to trip the motor circuit protector 100.
  • FIG. 30a is a UML diagram illustrating a run-mode state diagram 3000 according to an implementation of the present invention.
  • the state diagram 3000 begins with a monitor peak current state 3002. When the monitor period is complete, the state diagram 3000 transitions to a pre-trip detection state 3004. If an instantaneous trip is detected, the state diagram 3000 moves to a Stored Energy trip state 3006, which may correspond to the Stored Energy trip state 2914 shown in FIG. 29.
  • the state diagram 3000 enters an auxiliary task state 3008 (shown in FIG. 31). If a pre-trip is detected, the state diagram 3000 returns to the monitor peak current state 3002. When the auxiliary tasks have been completed, the state diagram 3000 returns to the monitor peak current state 3002. If an over-temperature trip is detected, the state diagram 3000 moves to the Stored Energy trip state 3006.
  • FIG. 30b is a UML diagram illustrating a state diagram 3008 for run-mode (steady- state mode) auxiliary tasks according to an implementation of the present invention.
  • the Guards and Actions applicable to this state diagram 3008 are as follows.
  • the state diagram 3008 enters the auxiliary tasks state 3010.
  • Various auxiliary tasks may be carried out, incl ⁇ ding diagnostics 3012, temperature sensing 3016., voltage regulation 3014, and switch position detection 3018.
  • the switch position detection state 3018 if a switch change is detected (G2), the position(s) of the dials 114, 116 are converted to digital values representing the dial position(s), and the corresponding trip curve settings (e.g., trip threshold current values for trip regions A, B, and C) are fetched from the EEPROM 270.
  • the auxiliary tasks are terminated upon detection of an over-temperature trip or a low-voltage across the voltage regulator.
  • FIG. 31 is a UML diagram illustrating a state diagram 3100 for the INST trip regions A (locked-rotor avoidance region) and B (in-rush avoidance region).
  • the state diagram 3100 includes the following Guards and Actions specified below.
  • Trip B At least 2 INST Pre-trip B events AND Pre-trip B delay
  • Trip A At least 5 INST Pre-trip A events AND Pre-trip A delay.
  • the control algorithm 230 fetches the trip curve from EEPROM 270 (Fl).
  • the trip curve may be selected based upon the positions of the dials 114, 116 detected by the control algorithm 230.
  • the peak current is sensed and monitored in monitor module 3104. If the peak current is greater than the in-rush avoidance current thxeshold B (e.g., trip region 2804), a pre-trip B event signal (F3) and a pre-trip A event signal (F4) are activated.
  • the controller 230 enters simultaneously an INST pre-trip B state 3106 and an ESFST pre-trip A state 3108 and polls until a pre-trip event signal is received.
  • a pre-trip A event signal (F4) is activated and the INST pre-trip A state 3108 is entered.
  • a pre-trip B timer is active (G3)
  • a pre-trip A timer is active (G4).
  • G5 When a pre-trip event signal is received (F5), an INST pre-trip state 3110 is entered for the duration of a pre-trip timer (G5).
  • Trip logic (G6) is parsed before transitioning to a trip state 3112.
  • the trip logic depends upon whether a Trip B or Trip A event has been detected. In the case of a Trip B event, the Trip Logic requires at least two INST pre-trip B events to occur and a pre-trip B delay to expire. In the case of a Trip A event, the Trip Logic requires at least five INST pre-trip A events to occur and a pre- trip A delay to expire. Once these conditions are satisfied, the trip state 3112 is entered and the state diagram 3100 ends.
  • FIG. 32 is a UML diagram illustrating a state diagram 3200 for high instantaneous self-protection tripping for region C (self-protection region).
  • the Guards and Actions applicable to this state diagram 3200 are provided below.
  • a comparator software object is initialized (Fl), and a monitoring peak current state 3202 is entered.
  • An idle state 3204 is entered when a comparator interrupt is stopped (F2) and is exited when a comparator interrupt is started (F3).
  • a comparator trip is confirmed (Gl)
  • the state diagram 3200 transitions to a Stored Energy trip state 3206, which may correspond to the Stored Energy trip state 2914 shown in FIG. 29.
  • the comparator interrupt is started when a self-protection voltage signal corresponding to the secondary current from the current transformers 210, 212, 214 exceeds a calibration reference voltage value.
  • FIG. 33 is a block diagram of a calibration and testing system 3300 that calibrates the output responses in a customized calibration table prepared from a nominal template and referenced by the control algorithm 230.
  • the control algorithm 230 along with the customized calibration table with scaled values is transferred into the flash memory 272 of the motor circuit protector 100 in the production and testing process.
  • the scaled values in the customized calibration table are obtained as a result of the calibration process.
  • the calibration and testing system 3300 includes a tester unit 3302 and a motor circuit protector (also referred to as a device under test or "DUT") to be tested and calibrated such as the motor circuit protector 100 described above.
  • DUT device under test
  • the tester unit 3302 includes a communications interface 3306 that is in data communication with the EEPROM 270 of the motor circuit protector 100 in the calibration process.
  • the tester unit 3302 also includes a current output 3308 that is coupled to the current transformers 210, 212 and 214 of the motor circuit protector 100.
  • the current output 3308 injects currents to the current transformers 210, 212 and 214 for calibration purposes.
  • the tester unit 3302 also .includes a signal connector 3310 for transmitting additional test data signals to components such as the power supply capacitor input circuit 406.
  • the tester unit 3302 includes production test software 3320 that provides analysis of the data and determines scaling values for the customized calibration table eventually stored on the EEPROM 270 and accessed by the control algorithm 230.
  • the flash memory 272 is loaded with the calibration software 3330 via the communications interface 3306.
  • the calibration software 3330 implements calibration and testing routines such as current transformer characterization equation calibration, switch testing, temperature sensor testing, voltage input testing, etc.
  • the production test software 3320 records sensor readings and current peak detection data obtained by the calibration software 3330 by reading the EEPROM 270.
  • the calibration software 3330 acts as a data recorder for sensor feadings and input current peaks from the motor circuit protector 100.
  • the signal chain for the current peak injection includes the current transformers 210, 212 and 214, the serpentine copper burden resistor 410, the burden resistor control FET 412, the microcontroller 226, the voltage regulator circuit 408 (or the voltage regulation module 260) and the temperature sensor circuit 222 as shown in FIGs. 3-4.
  • the calibration software 3330 is a Java-based, signal chain simulator. Of course other types of coding language may be used to perform the same functions. Nominal calibration templates may be generated from a spreadsheet program, for example.
  • the production test software 3320 stimulates the motor circuit protector 100 with power supply, switch, and current signals.
  • the calibration software 3330 is loaded in the flash memory 272 and writes the test data to the EEPROM 270.
  • the tester unit 3302 includes normalized templates of equipment operating parameters for product calibration of different types of motor circuit protectors (e.g., having different current operating ranges).
  • the normalized templates include expected performance parameters such as trip curves for the type of motor circuit protector 100.
  • the production test software 3320 manipulates the template in a restrictive manner for calibration purposes to produce the customized calibration table.
  • critical calibration information is delivered to the EEPROM 270 in the customized calibration table written by the production test software 3320 using data from running the calibration software 3330.
  • FIG. 34A shows a set of typical balanced three-phase 60Hz secondary currents 3402, 3404 and 3406 that are fed into a three-phase rectifier such as the rectifier 302.
  • An ideal peak current output signal 3408 from the three-phase rectifier 302 is shown in FIG. 34A.
  • FIG. 34B a single-phase secondary current 3412 having a phase A, Isa, from the current transformer 210 results in a rectified output current 3414 from a rectifier .
  • FIG. 35 shows current graphs 3510, 3520, 3530, and 3540 of the transfer-function behavior of the current transformer 210 for various fault currents.
  • the current graph 3510 includes a primary current waveform 3512 at 25 A and a corresponding saturated secondary current 3514.
  • the current graph 3520 includes a primary current waveform 3522 at IOOA and a corresponding saturated secondary current 3524.
  • the current graph 3530 includes a primary current waveform 3532 at 250A and a corresponding saturated secondary current 3534.
  • the current graph 3540 includes a primary current waveform 3542 at 2000A and a corresponding saturated secondary current 3544.
  • the motor circuit protector 100 is operational for currents in the saturation ranges of the current transformers 210, 212, and 214, the secondary current waveforms are not uniform over the entire pickup range of instantaneous fault currents.
  • the secondary current signals are also sinusoidal as shown in FIGs. 34A and 34B and sampling errors can be calculated.
  • the secondary current signals are distorted due to being in the saturation region of the current transformers 210, 212, and 214 as shown in FIG. 35.
  • Experimental data determines the maximum peak detection errors. The maximum peak error due to worst case instantaneous current sampling or self protection comparator response is considered in the control algorithm 230 via the normalization template. ,
  • the peak secondary currents are predictable over the operating ranges of the motor circuit protector 100.
  • a series of typical current transformer transfer functions 3600, 3602, and 3604 are shown in FIG. 36, where secondary peak currents (y-axis) vary with known primary current signals (x-axis).
  • the transfer function 3600 represents a relatively high temperature (110 0 C in this example)
  • the transfer function 3602 represents a relatively ambient temperature (25 0 C in this example)
  • the transfer function 3604 represents a relatively low temperature (-35 °C in this example).
  • the current measurement performance of the current transformer is non-linear over both the fault current and high instantaneous current detection ranges that fall in the saturation region of the current transformer.
  • An ideal current transformer has an output predicted by the ratio of secondary turns to primary turns. It is convenient to characterize the current transformers with a parameter known as an "Effective Turns Ratio" at the interested measurement points and normalize the effective turns ratio to the ideal turns ratio. Iron-core current transformers also exhibit temperature performance.
  • the transfer functions for the current transformers in this example take both temperature performance and effective turns ratio into account. [000363]
  • the equations for the transfer functions are developed by part experimentation or by models. The equations are modified by software design to improve the system measurement accuracy where applicable. The equations are mostly for the second half cycle and beyond current signals. Expected first half cycle signal errors depend on the current transformer configuration, closing angle and current magnetization.
  • the transfer function may be expressed generally as the following equation:
  • Is is the secondary current and "Ip” is the primary current.
  • C0-C4 are determined by experimentation involving a test setup for different temperatures and varying signals to determine outputs over different current levels for a particular type of current transformer. The performance characteristics are determined experimentally for each current transformer configuration at all the fault current and high instantaneous current trip points. The magnitude performance of the current transformers is important for predicting trip pickup levels. The current sensing signal width is important for digital sampling constraints, specifically for single-phase scenarios. The following table indicates exemplary values for the coefficients at various current ratings.
  • a calibration point or points are determined for the testing and calibration process described in more detail below.
  • a single calibration current or point may be selected for a range of trip points or two or more calibration points may be selected for each different desired range of trip points.
  • a calibration current or point is selected based on different candidates of current levels. In this example, four potential candidates of current levels are tested to determine a calibration current which will meet acceptable calibration standards. The candidates are selected depending on the desired operating range of the current transformer. For example, different candidates of current levels may be selected near the transition to the saturation region of a specific current transformer if the desired current range is primarily in the linear region.
  • the calibration point or points are stored at the high temperature curve 3600 in FIG. 36 to the nominal templates.
  • the high temperatures may be temperatures that are high relative to an ambient temperature of 25 0 C such as 90 C or 110 0 C.
  • the storage of calibration points at a higher temperature level prevents nuisance tripping when errors occur in the temperature calibration system.
  • the scaling of the calibrated values is performed on the nominal templates that are derived from the elevated or relatively high temperatures.
  • the different candidates for calibration points are each calibrated via the device under test (DUT) with the tester unit 3302 in accordance with procedures detailed below to obtain a scaling factor.
  • the DUT is removed from the tester unit 3302 and the response at some or all of the current trip points are measured.
  • the corresponding customized calibration tables for each are stored and the values at the trip points from the tables are compared with actual response at some or all of the trip points from the DUT.
  • the candidate with the minimal amount of error across some or all of the trip points is selected .as the calibration point for production testing.
  • each calibration point candidate is compared with the corresponding trip points within the desired ranges.
  • the characteristic equation and average resistance for the burden resistor 412 and the on state of the burden resistor control FET 412 is used to produce a normalized table of trip points.
  • FIG. 37 is a functional block diagram of the components of the calibration software 3330 when installed in conjunction with the hardware components of the motor circuit protector 100.
  • the calibration software 3330 has a switch reading module 3702, a temperature readings module 3704, a voltage readings module 3706, a voltage regulation module 3708, a sensor readings module 3710, a peak detection module 3712 and a read/write module 3714.
  • the switch reading module 3702 receives inputs from the user adjustments circuit 224 during the testing process and provides switch data in response to test signals.
  • the temperature readings module 3704 receives inputs from the temperature sensor circuit 222 and provides temperature test data.
  • the temperature readings module 3704 records raw temperature sensor readings when triggered. These readings and tester fixture temperature data determine the temperature sensor offset sign and magnitude.
  • the temperature sensor offset is written by the read/write module 3714 to the EEPROM 270 by the production test software 3320. Given the production test software 3320 is operating within calibration temperature limits, the difference from the nominal temperature reading may be determined. If the sensor reading from the temperature readings module 3704 is greater than the nominal, the read/write module 3714 writes a positive offset to the EEPROM 270. Conversely, a negative difference will result in the read/write module 3714 writing a negative offset to the EEPROM 270.
  • the voltage readings module 3706 is coupled to the power supply capacitor input circuit 406 and provides voltage readings by injecting a test voltage from the power supply capacitor input circuit 406 to determine any needed voltage offset to the microcontroller 226.
  • the voltage regulation module 3708 may provide voltage regulation fbrjhe motor circuit protector 100 during the calibration process.
  • the sensor readings module 3710 receives switch reading data, temperature data, and voltage data from the switch, temperature and voltage modules 3704, 3706 and 3708, respectively, and sends the readings to the read/write module 3714 that writes the test data into the EEPROM 270 for retrieval by the production test software 3320.
  • the peak detection module 3712 is coupled to the burden resistor circuit 306 and reads the peak current data in response to test currents that are injected to the three current transformers 210, 212 and 214 via the current output 3308. The peak detection data is sent to the read/write subroutine 3714 for storage on the EEPROM 270.
  • the production test sequence implemented by the calibration and testing system 3300 to gather sensor information can either be initiated with an Auto Trigger or by a Primary Current Trigger mode.
  • the Auto Trigger mode is used by the sensor reading subroutine 3710 to gather sensor data that does not depend on primary current injection, such as the switch readings from the switch readings subroutine 3702.
  • the current calibration test sequences associated with the Primary Current Trigger mode of operation allows the communications interface 3306 and the signal connector 3310 to be disconnected during primary current injection to reduce signal noise.
  • the Auto Trigger mode is configured by the voltage readings subroutine 3706 of the production test software 3320, which sets a peak threshold value to 0 in the EEPROM 270 while applying a voltage to the energy storage circuit 304.
  • the applied voltage should be greater than the required product startup voltage, which in this example is 16 volts, the voltage level sufficient to start the power supply Vcap circuit 304.
  • the Primary Current Trigger mode is adjusted in order to capture the synchronized peak current and secondary current signals at the specified calibration level. This mode is initiated by setting the peak threshold value to a value on the signal chain and expected tolerances for the particular motor circuit protector 100. Once the threshold value is exceeded, the current peaks are recorded by the calibration software 3330.
  • the production test software 3320 injects a targeted primary calibration current in all three phases to the current transformers 210, 212, and 214.
  • the primary calibration current is determined by the process described above.
  • the secondary currents of the current transformers 210, 212, and 214 are rectified by the three-phase rectifier 302.
  • the calibration software 3330 is programmed in the microcontroller 226 to record the first eight peaks of the secondary current from the three-phase rectifier 302 after the secondary current exceeds the peak threshold.
  • the production test software 3320 injects an actual current into one pole of motor circuit protector 100 for a sufficient duration for the calibration software 3330 to record the eight peaks.
  • the peaks are written into the EEPROM 270 in decimal count values via the read/write subroutine 270.
  • the production test software 3320 records the peaks of the input actual current and matches those with the peaks recorded by the calibration software 3330 in the EEPROM 270. This process is repeated for the other two current transformers 212 and 214. The sensor responses are recorded in specific locations in the EEPROM 270 by the read/write module 3714.
  • the communications interface 3306 is reconnected to the EEPROM 270.
  • the responses are read by the production test software 3320 to determine whether the nominal template values need to be scaled.
  • the production test software 3320 determines the scaling factors for the normalized template to produce the customized calibration table loaded into the EEPROM 270.
  • the scaling factors are determined by calculating temperature and current magnitude scaling constants or adjustment factors.
  • the peak current scaling constants are applicable over specified current ranges set forth in the calibration specifications for the type of motor circuit protector 100.
  • the temperature scaling constants are applicable over all operating current ranges.
  • the temperature scaling constant is a function of the ambient temperature of the motor circuit protector 100 to be tested. This adjustment factor compensates for burden resistor changes with temperature.
  • These normalized codes are stored in a comparator threshold lookup table with corresponding secondary current comparator values that is referenced by the test production software 3320.
  • the overall scaling constants determined by the production test software 3320 are multiplied by the normalized secondary current comparator values and then rounded down to the nearest secondary current comparator level.
  • the new secondary current comparator values are translated back to the applicable codes.
  • the new codes are written to the customized calibration table for loading in the EEPROM 270.
  • the test production software After loading the customized calibration table in the EEPROM 270, the test production software writes the control algorithm 230 into the flash memory 272. In this example, the control algorithm 230 overwrites the space occupied by the calibration software 3330 in the flash memory 272 to conserve memory space for the production ready motor circuit protector 100.
  • the motor circuit protector 100 is now calibrated and ready for use.
  • the production test and calibration process has restrictions on manipulation of the nominal templates implemented with the calibration software 3330.
  • the trip value adjustments are made within the limits of expected burden resistances and temperatures for the particular motor circuit protector. It is to be understood that different motor circuit protectors with different operating ranges have different normalized calibration templates.
  • the nominal template is altered by the production calibration process if the data recordings of the signal chain differ from the nominal values. Sensor readings and calibration data are bounded by a maximum current error and current delta error.
  • the maximum current error is an absolute difference of the equivalent primary current from the synchronized actual primary current injected by the production test software 3320.
  • the current delta error is a difference error between the three current transformers 210, 212, 214.
  • FIG. 38 An example flow diagram 3800 of the production test software 3320 and the calibration software 3330 for testing and calibration of the motor circuit protector 100 is shown in FIG. 38.
  • the machine-readable instructions comprise an algorithm for execution by: (a) a processor, (b) a controller, and/or (c) any other suitable processing device.
  • the algorithm may be embodied in software stored on a tangible medium such as, for example, a flash memory, a CD-ROM, a floppy disk, a hard drive, a digital versatile disk (DVD), or other memory devices, but persons of ordinary skill in the art will readily appreciate that the entire algorithm and/or parts thereof could alternatively be executed by a device other than a processor and/or embodied in firmware or dedicated hardware in a well known manner (e.g., it maybe implemented by an application specific integrated circuit (ASIC), a programmable logic device (PLD), a field programmable logic device (FPLD), discrete logic, etc.). Also, some or all of the machine-readable instructions represented by the flowchart of FIG. 38 may be implemented manually.
  • ASIC application specific integrated circuit
  • PLD programmable logic device
  • FPLD field programmable logic device
  • the example test sequence is as follows.
  • the calibration software 3330 is loaded into the flash memory 272 of the motor circuit protector 100 to be tested (3802).
  • the calibration software 3330 initializes itself and waits a set delay (4ms in this example) for a startup voltage to be reached (3804).
  • the test production software 3320 configures the auto trigger mode (3806).
  • the test production software 3320 reads test data from the various sensors via the readings modules.
  • the dials 1 14 and 116 are set to their maximum and minimum settings, which are received by the user adjustments circuit 224, converted to corresponding digital values indicative of the respective maximum and minimum positions of the dials, and provided to the switch reading module 3702.
  • a test voltage is applied to the power supply capacitor input circuit 406, whose value is read by the voltage readings module 3706.
  • the temperature readings module 3704 reads temperature sensor 222, which provides a voltage indicative of the temperature.
  • the resulting test data is collected (3808) and the calibration software 3330 records the test data in the EEPROM 270 via the read/write module 3714 (3810). It is to be understood that blocks 3806, 3808 and 3810 are optional test routines and any or all of them may be carried out subsequent to the current injection or not at all depending on the desired test process.
  • the peak trigger mode is initiated that samples the input current for the trigger threshold (3812).
  • the input current peak threshold is set to a desired value by the test production software 3320 writing the desired value to the EEPROM 270 (3814).
  • the input current peak threshold is selected depending on the desired operational range of the motor circuit protector 100.
  • the inputs of the current transformers 210, 212 and 214 are stimulated with current signals (3816) one at a time or simultaneously.
  • the peak detection module 3712 detects eight half cycle peak samples for calibration purposes and sends the peak sample data to the read/write module 3714.
  • the read/write module 3714 writes the peak sample data in the EEPROM 270 (3818).
  • the production test software 3320 reads the peak sample data stored in the EEPROM 270 (3820).
  • the production test software 3320 compares the input signals with the test data (3822).
  • the production test software 3320 determines the scaling factors for the template for the motor circuit protector 100 under test (3824).
  • the scaling factors are used to modify the nominal template to create a customized calibration table for the motor circuit protector 100 under test (3826).
  • the customized calibration table is written to the EEPROM 270 (3828).
  • the control algorithm 230 then is written over the calibration software 3330 (3830) once the calibration is complete.
  • An advantage of the calibration techniques above is the employment of flexible software architecture that accommodates trip point adjustments between MCP limits without changing the source code for the MCP.
  • the use of the separate testing software and calibration software enables the calibration process to be controlled by software engineering part releases.
  • the software architecture allows the product software code to have high commonality across circuit breakers with different operational current ranges.
  • the flexible software architecture and implementations reduce product test times while maintaining product test coverage.
  • the calibration also is repeatable, which results in low variance in trip points for different calibrations of the same unit.
  • FIG. 39 is a calibration state diagram in Unified Modeling Language (UML) according to aspects of the various embodiments disclosed herein. The following guards and actions are applicable to FIG. 39:
  • UML Unified Modeling Language
  • the Calibration Initialize state initializes the calibration system and waits for the startup voltage to be reached.
  • the Read Sensors state records the A/D readings for the analog inputs, FLA, Im, Vs, and Ts.
  • the Peak Trigger state samples the input current for a trigger threshold.
  • the Peak Detection state records half-cycle peak samples for calibration purposes.
  • the Regulator Service state maintains power supply voltage until power is removed.

Abstract

Un disjoncteur de circuit muni d'un transformateur de courant pour alimenter des dispositifs électroniques d'unité de déclenchement et détecter des courants élevés et faibles comprend un cœur présentant des laminations solides et écartées. Le disjoncteur comprend un ensemble d'ajustement permettant de traduire des réglages de cadran décimal en niveau de déclenchement stockés en mémoire. Un algorithme de compensation de température ajuste des points de déclenchement en réponse à des variations de température. Un schéma d'activation de déclenchement redondant nécessite au moins qu'une condition de pré-déclenchement soit satisfaite avant que le solénoïde de déclenchement soit activé. Un système de régulation de tension d'alimentation comprend un circuit d'énergie stockée. Un microcontrôleur reconfigurable exécute un algorithme de déclenchement synchronisé temporellement. Le disjoncteur se déclenche conformément à une courbe de déclenchement divisée en trois zones de protection. Un logiciel détecte si un courant primaire dépasse chaque zone est exécuté en parallèle ou simultanément, fournissant une détection de déclenchement instantanée redondante. Un procédé d'étalonnage utilise une chaîne de signal qui comprend un transformateur de courant, une résistance de charge, un circuit d'énergie stockée et un contrôleur.
PCT/US2007/015914 2006-07-14 2007-07-12 Dispositif de protection de circuit de moteur électronique WO2008008446A2 (fr)

Applications Claiming Priority (22)

Application Number Priority Date Filing Date Title
US83100606P 2006-07-14 2006-07-14
US60/831,006 2006-07-14
US11/818,679 US8154373B2 (en) 2006-07-14 2007-06-15 Circuit breaker-like apparatus with combination current transformer
US11/818,679 2007-06-15
US11/824,683 US7683586B2 (en) 2006-07-14 2007-07-02 Method and system of fault powered supply voltage regulation
US11/824,682 US7791849B2 (en) 2006-07-14 2007-07-02 Redundant trip activation
US11/824,682 2007-07-02
US11/824,652 US7788055B2 (en) 2006-07-14 2007-07-02 Method and system of calibrating sensing components in a circuit breaker system
US11/824,683 2007-07-02
US11/824,684 US7592888B2 (en) 2006-07-14 2007-07-02 Low cost user adjustment, resistance to straying between positions, increased resistance to ESD, and consistent feel
US11/824,654 2007-07-02
US11/824,680 US7859802B2 (en) 2006-07-14 2007-07-02 Burden resistor temperature compensation algorithm
US11/824,684 2007-07-02
US11/824,681 US7550939B2 (en) 2006-07-14 2007-07-02 Redundant instantaneous trip detection
US11/824,652 2007-07-02
US11/824,693 US7697250B2 (en) 2006-07-14 2007-07-02 Switch-to-trip point translation
US11/824,651 2007-07-02
US11/824,651 US7869169B2 (en) 2006-07-14 2007-07-02 Method and system of current transformer output magnitude compensation in a circuit breaker system
US11/824,681 2007-07-02
US11/824,693 2007-07-02
US11/824,654 US7869170B2 (en) 2006-07-14 2007-07-02 Method and system for time synchronized trip algorithms for breaker self protection
US11/824,680 2007-07-02

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WO2008008446A3 WO2008008446A3 (fr) 2008-05-02

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US9184014B2 (en) 2013-02-01 2015-11-10 General Electric Company Electrical operator for circuit breaker and method thereof
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KR101916468B1 (ko) 2012-11-19 2018-11-07 현대자동차주식회사 외부 온도에 따른 부하 전류 제한 자동 보정 방법
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