WO2008004417A1

WO2008004417A1 - Sensorless control apparatus of synchronous machine

Info

Publication number: WO2008004417A1
Application number: PCT/JP2007/061894
Authority: WO
Inventors: Kazuya Yasui
Original assignee: Kabushiki Kaisha Toshiba
Priority date: 2006-07-05
Filing date: 2007-06-13
Publication date: 2008-01-10
Also published as: JP2008017608A; US20090200974A1; CN101485079B; EP2075904A1; CN101485079A; EP2075904A4; JP4928855B2

Abstract

A rotating machine sensorless control apparatus comprising an inverter (05) that converts DC and AC powers to each other; a permanent magnet synchronous machine (07) the rotator of which has a magnetic saliency and which is driven by a power supplied from the inverter; a PWM modulating means (04) that decides, based on a command for controlling the permanent magnet synchronous machine, an output voltage at the inverter; a current detecting means (06) that detects a current flowing through the permanent magnet synchronous machine; a high frequency component calculating means (10) that calculates the high frequency components of a current change caused by the voltage which is decided by the PWM modulating means and which is outputted from the inverter; and a rotation phase angle estimating means (08) that estimates the rotation phase angle of the permanent magnet synchronous machine based on a spatial distribution on a rotating coordinate axes synchronous with the rotation of the permanent magnet synchronous machine of the high frequency components; wherein the increase of loss and noise that are caused by the sensorless control of the rotating machine can be suppressed and a simple adjustment can provide a stable operation.

Description

Specification

Sensorless control device for synchronous machine

Technical field

The present invention relates to a sensorless control device for a synchronous machine.

Background art

[0002] In order to control a synchronous machine using a permanent magnet as a rotor, a sensor for detecting the rotational phase angle of the rotor is required to drive and control the synchronous machine (electric motor or generator). In the case of a control device that controls a synchronous machine using a sensor that detects the rotational phase angle by applying a force, the following problems exist.

First, the presence of a sensor that detects the rotational phase angle increases the volume of the entire drive system. This hinders expansion of the output of the synchronous machine in a limited installation space.

[0004] Secondly, the maintenance and inspection work of the sensor itself for detecting the rotational phase angle is required.

. This deteriorates the maintenance inspection efficiency.

[0005] Third, noise or the like is superimposed on the signal line of the sensor force that detects the rotational phase angle.

The detected value is disturbed and the control performance is deteriorated.

[0006] Fourthly, most of the sensors for detecting the rotational phase angle require a power source for driving the sensor, and it is necessary to install a power source different from that for driving the synchronous machine.

. This becomes a factor of increasing the burden in the power supply installation space, the power supply line, the cost and the like.

[0007] For the reasons described above, a control method has been developed in which a rotational phase angle is estimated without using a sensor, and drive control is performed based on the estimated rotational phase angle. This is called “sensorless control”.

[0008] As described in Japanese Patent No. 3168967, as a sensorless control device that performs sensorless control of such a synchronous machine and is effective particularly in a stop / low speed state, in a system that drives the synchronous machine by a PWM inverter In addition, a high frequency voltage command with a frequency sufficiently high with respect to the operating frequency of the synchronous machine is superimposed on the command for controlling the inverter, and a component corresponding to the superimposed high frequency command is detected from the high frequency current response caused by this. Process It is known that the rotational phase angle is estimated by using this to obtain the rotational phase angle error.

[0009] The conventional sensorless control device for a synchronous machine described above has an advantage that the synchronous machine can be controlled without using a rotational phase angle sensor, and maintenance is improved at low cost. However, a control method that detects a component corresponding to a high-frequency voltage command for high-frequency current response, such as the sensorless control device described in the above-mentioned patent document, basically requires a desired high-frequency current to flow. Compared to a control device using sensors, there were problems when the loss and noise increased extremely. As a secondary matter, in order to estimate the rotational phase angle stably, it is necessary to finely adjust the amplitude and frequency of the high-frequency command to be superimposed and the high-frequency superposition method. In practice, it was necessary to make complicated and time-consuming adjustments. Specifically, since the motor characteristics fluctuate due to inductance fluctuations due to saturation of the motor windings, changes to the high-frequency superposition method according to the motor torque current, fine adjustment of the high-frequency current detection method, etc. Was necessary.

Disclosure of the invention

[0010] The present invention has been made to solve the above-described problems of the prior art, and suppresses loss and noise increase caused by sensorless control, and enables stable operation with simple adjustment. It is an object of the present invention to provide a sensorless control device for a synchronous machine that enables this.

[0011] The present invention provides an inverter that mutually converts direct current power and alternating current power, a synchronous machine that has a magnetic saliency in a rotor and is driven by power supplied from the inverter, and controls the synchronous machine PWM modulation means for determining the output voltage in the inverter based on a command for the current, current detection means for detecting the current flowing in the synchronous machine, and the voltage determined in the PWM modulation means and output from the inverter A high-frequency component calculating means for calculating a high-frequency component of the generated current change, and a rotation for estimating a rotation phase angle of the synchronous machine based on a spatial distribution of the high-frequency component on a rotating coordinate axis synchronized with the rotation of the synchronous machine. It features a sensorless control device for a synchronous machine equipped with phase angle estimation means.

[0012] According to the sensorless control device for a synchronous machine of the present invention, the change in current flowing through the synchronous machine is high. Control the synchronous machine by calculating the frequency component and estimating the phase angle of the motor rotor without the rotational phase angle sensor based on the spatial distribution of the high-frequency current change in the dq axis coordinate system that is synchronized with the rotation of the synchronous machine As a result, the loss of the synchronous machine and the increase in noise caused by sensorless control can be suppressed, and stable operation can be achieved with simple adjustment.

Brief Description of Drawings

FIG. 1 is a block diagram showing a general permanent magnet synchronous machine model and coordinate definitions.

FIG. 2 is a block diagram showing a definition of a voltage vector of the permanent magnet synchronous machine.

FIG. 3 is a block diagram of the sensorless control device of the synchronous machine according to the first embodiment of the present invention.

[FIG. 4] FIG. 4 is a graph showing a high-frequency component distribution of current change on the dq coordinate axis of the permanent magnet synchronous machine.

[FIG. 5] FIG. 5 is a graph showing a high-frequency component distribution when the current change error on the dq coordinate axis of the permanent magnet synchronous machine is 30 °.

[FIG. 6] FIG. 6 is a graph showing a high-frequency component distribution in a state where the permanent magnet synchronous machine has a 100% torque output on a dq axis coordinate and a current change error of 0 °.

[FIG. 7] FIG. 7 is a graph showing a high-frequency component distribution in a state where the permanent magnet synchronous machine has a torque 100% output on a dq axis coordinate and a current change error of 30 °.

[FIG. 8] FIG. 8 is a graph showing a high-frequency component distribution in a state where a permanent magnet synchronous machine has a torque 100% output on a dq axis coordinate and a current change error + 30 °.

[Fig. 9] Fig. 9 is a graph showing the relationship between the characteristic amount and the error in the sensorless control device of the synchronous machine according to the third embodiment of the present invention.

FIG. 10 is a block diagram of a sensorless control device for a synchronous machine according to a fourth embodiment of the present invention.

BEST MODE FOR CARRYING OUT THE INVENTION

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

[0015] (First embodiment)

The sensorless control device for a synchronous machine according to the present invention calculates a high frequency component of a current change flowing through a permanent magnet synchronous machine, and synchronizes with the rotation of the synchronous machine. Based on the spatial distribution of changes, the phase angle of the motor rotor is estimated without the rotational phase angle sensor, and the permanent magnet synchronous machine is controlled.

FIG. 1 shows a configuration of a general permanent magnet synchronous machine. The stator 01 of the permanent magnet synchronous machine is composed of U, V, W three-phase wire 01U, 01V, 01W, and the rotor is composed of a rotor core 02 and a permanent magnet 03. In the sensorless control device for a synchronous machine according to the present embodiment, as a coordinate system that rotates in synchronization with the rotation of the permanent magnet synchronous machine, the direction of the magnetic flux of the permanent magnet is d-axis, and the axis orthogonal to this d-axis is Defined as q axis. Also, the U-phase direction is defined as the _α- axis, the direction perpendicular to this is defined as the β-axis, and the angle from the a-axis direction to the d-axis direction is defined as the rotational phase angle Θ of the synchronous machine. Based on this definition, the relationship between the voltage and current of the permanent magnet synchronous machine is expressed by the following equation (1).

[Number 1]

Here, Vd and Vq are d-axis voltage and q-axis voltage, respectively, Id and Iq are d-axis current and q-axis current, R is resistance, Ld is d-axis inductance, and Lq is q-axis. Inductance, Φ is permanent magnet flux, ω is rotational speed, and p is a differential operator.

However, since the sensorless control device of the synchronous machine of the present embodiment cannot detect the rotational phase angle Θ itself that the rotational phase angle sensor does not have, the phase angle estimated by the control device is Use instead of sensor output. Therefore, as shown in Fig. 1, the estimated phase angle is defined as Θ est, and the corresponding coordinate system is defined as the γ-axis and δ-axis. When the estimation error Δ 0 occurs, the γ δ axis is rotated by the dq axial force estimation error Δ 0.

FIG. 3 shows a configuration of the sensorless control device of the synchronous machine according to the first embodiment of the present invention. The inverter 05 receives the gate command for driving the inverter 05 as an input, and converts AC Z DC power to each other by switching ONZOFF of the main circuit switching element built in the inverter 05. The permanent magnet synchronous machine 07 generates a magnetic field due to the three-phase alternating current flowing in each excitation phase of the UVW, and torque is generated by magnetic interaction with the rotor. Generate

[0020] The control command calculation unit 10 calculates a control command for the torque command force by calculation described later, and outputs the control command to the PWM modulation unit 04. The PWM modulation unit 04 modulates a control command for driving the permanent magnet synchronous machine 07 by PWM (Pulse Width Modulation), and a gate command that is an ONZOFF command for each phase switching element of the inverter 05. Is output.

The current detection unit 06 detects a two-phase or three-phase current response value of the three-phase AC current flowing through the permanent magnet synchronous machine 07. In this embodiment, the two-phase currents Iu and Iw are detected. The high frequency component calculation unit 11 extracts the high frequency current component from the response values of the input currents Iu and Iw, calculates the time change rate thereof, and outputs it. It also outputs a signal indicating the timing of computation.

[0022] The rotational phase angle estimation unit 08 uses the high-frequency component of the current change calculated from the current response values Iu and Iw detected by the current detection unit 06, and uses the spatial component of the component in the y δ-axis coordinate system. Based on the distribution, the rotational phase angle Θ est of the permanent magnet synchronous machine 07 is estimated.

Next, the operation of the sensorless control apparatus for a synchronous machine according to the present embodiment configured as described above will be described. In FIG. 3, the control command that is an input to the PWM modulation unit 04 is given by the control command calculation unit 10 by the following calculation processing based on the torque command to be output by the permanent magnet synchronous machine 07.

[0024] First, a torque command is given from the host control system, and based on the torque command! /, A y-axis current command I γ ref and a δ-axis current command I δ rel ^ are calculated according to Equation 2.

[Equation 2]

I _y ' ^ef = Trq rcf ^-n -cos ()

If = Trq ' ^ef -.sin (

Here, Trqref is a torque command, k is a constant, and 0 i is a current phase angle with respect to the γ axis in the γ δ axis coordinate system.

However, the current commands I γ ref and I δ ref are tapes that can refer to the torque command as a parameter. It is also possible to prepare the file and refer to this table. The method using a table is effective in cases where it is preferable to formulate the relationship between torque and current as shown in the above equation (2).

Next, the current command I γ ref, I δ ref obtained from the torque command as described above, and the γ-axis response value I γ res, δ-axis response value I of the current flowing through the permanent magnet synchronous machine 07 Using δ res as an input, γ-axis voltage command V γ ref and δ-axis voltage command V δ re are calculated by proportional integral control as follows.

[Equation 3] j

v: ^ef = K + K ^.--

Here, ここ ρ is a proportional gain, Ki is an integral gain, and s is a Laplace operator.

[0029] Next, based on the y-axis voltage command ν γ ref, the δ-axis voltage command V δ re output as described above, and the rotational phase angle estimated value Θ est output from the rotational phase angle estimator 08. Then, coordinate conversion is performed by the following calculation, and three-phase voltage commands Vuref, Vvref, Vwrel ^ are calculated.

[Equation 4]

[0030] The control command calculation unit 10 uses the three-phase voltage command obtained by the above calculation as a control command.

, Input to PWM modulator 04.

[0031] The PWM modulation unit 04 performs PWM modulation and outputs a gate command to the inverter 05. Here, PWM modulation is a given control command, which is a three-phase voltage command, and is constant or variable in advance. Compared with a triangular wave carrier wave set to have a frequency of, the comparison result is used as a gate command.

The rotational phase angle estimation unit 08 is based on the high-frequency component of the current change flowing through the permanent magnet synchronous machine 07 obtained by the high-frequency component calculation unit 11 and based on the spatial distribution in the γδ axis coordinate system as follows. To estimate the rotational phase angle Θ est.

First, in the high frequency component calculation unit 11, the phase currents Iu and Iw detected by the current detection unit 06 are used as the rotation phase angle estimation value 0 est 〖based on the rotation phase angle estimation unit 08. Then, coordinate transformation is performed by the following calculation !, and the γ-axis current response value I y res and the δ-axis current response value I δ res are obtained.

[Equation 5]

Here, using the fact that the sum of the three-phase currents flowing through the permanent magnet synchronous machine 07 is 0, the γ-axis current response is calculated from the current values of the two phases of the three-phase current, as shown in the following equation. The value I y res and the δ-axis current response value I δ res can be obtained. In this case, it is only necessary to provide the current detection unit 06 for two phases, and the apparatus can be simplified compared to the case of detecting for three phases.

[Equation 6]

2 / '"COS ^! Cos (,

Next, a high frequency component corresponding to a change in the γδ axis current response value obtained as described above is obtained and output by the following calculation.

[0036] Here, Im is the input current I at time tm, In is the input current I at time tn, and dlbase / dt is the rate of time change of the fundamental wave component of the input current. The fundamental wave component is an electrical rotational frequency component.

[0037] As a calculation method of dlbaseZdt, there is a method of calculating the rate of change of the input current and the rate of change of the current command value in a sufficiently long time interval compared to (tm-tn), strictly speaking, the rotational frequency component However, if tm-tn is set to be sufficiently shorter than the fundamental wave component computation time interval, computation can be performed without any particular problem.

[0038] Further, as a more precise calculation means of the high-frequency component of the current change, the current change between times tm and tn is linearly approximated to obtain the change between the times, and this is expressed by Equation 7. It is also possible to use as (Im-In). As a linear approximation method, a generally used least square approximation is used. In that case, at least multiple sampling points are required between the computation times tm and tn. The number of samplings depends on the sampling frequency of the AZD converter, but this is because the general AZD variation in recent years can take a sufficient number of samplings to linearly approximate current changes that also increase the sampling frequency. It can be said that the method is also sufficiently practical. In addition, this method can eliminate the influence of noise, and can therefore be expected to improve estimation accuracy.

The rotational phase angle estimator 08 estimates the rotational phase angle from the spatial distribution of the high frequency component of the current change obtained by the high frequency component calculator 11. Here, we first explain the estimation principle. The high-frequency component of the current change flowing through the permanent magnet synchronous machine 07 can be derived from the current differential term force in Equation (2). Dividing Eq. 2 into high-frequency components and fundamental wave components as shown in Eq. 8, and extracting high-frequency components, Eq. 9 is obtained.

[Equation 8]

[0040] where Xbase is the fundamental component of waveform X, and the high frequency component of waveform X

PL I

[0041] The high-frequency voltages V to d and V to (! Can be defined as the sum of the inverter output voltage and the high-frequency component of the motor-induced voltage. However, the motor-induced voltage varies depending on the rotational speed and drive current, but the high-frequency component The high frequency component of the inverter output voltage is dominant as the high frequency voltage related to the spatial distribution of the high frequency component used in the rotational phase angle estimation unit 08. The high frequency component of the inverter output voltage is Therefore, the amplitude values of the high-frequency voltages V to d and V to q can be regarded as almost constant, so that the high-frequency currents d and I to q are as follows: , V ~ q, Ld, Lq.

[Equation 10]

[0042] When the present inventor spatially plots the high-frequency component of the current change in the dq axis coordinate system of the permanent magnet synchronous machine, which is the development facility used by the inventor, the distribution is shown in FIG. It was experimentally confirmed that the distribution was almost elliptical. This proves the principle of Equation 10. By using this distribution, rotational phase angle estimation can be realized.

[0043] As is apparent from Fig. 4, the spatial distribution of the high-frequency component has a shape having a characteristic in the dq axis direction. This distribution can be said to be the inductance distribution of the stator winding through which the drive current of the permanent magnet synchronous machine 07 flows, in terms of the structural power of the motor. However, the spatial distribution of the stator inductance is strongly influenced by the rotor inductance, and the stator current is driven by the drive current. It can be said that the rotor inductance distribution is almost equal to the torque zero to middle range operation where the ductance is not saturated. Figure 4 shows the inductance distribution obtained by superimposing a spatially uniform high-frequency voltage with zero output torque, that is, zero current. For the reasons described above, this distribution is considered to be rotor inductance. It is done.

In FIG. 4, the major axis direction of the elliptic distribution is coincident with the d axis, and the minor axis direction is coincident with the q axis.

As described above, since Fig. 4 represents the rotor inductance, this elliptical distribution rotates according to the rotation of the rotor. Therefore, if the rotation of the ellipse is extracted from the distribution by approximation, the d-axis direction can be estimated. According to this method, if the minimum high-frequency component that can extract the rotation of the elliptic distribution can be obtained, the rotational phase angle can be estimated.

[0045] On the other hand, in the conventional method, it is necessary to superimpose a high-frequency voltage and observe a high-frequency current corresponding to or synchronized with at least the superimposed high-frequency voltage in some form. For example, it was necessary to observe the amplitude of the high-frequency current matched to the superposition period of the high-frequency voltage. In this method, if a disturbance such as noise occurs at the sampling time point when the high-frequency current amplitude is observed, the accuracy of the estimated phase angle is greatly affected. In order to avoid such drawbacks, the conventional method has been adjusted so that the high frequency current amplitude is increased by increasing the amount of superimposed high frequency in order to increase the SZN ratio so that noise does not significantly affect the estimation result. . However, this is not preferable because loss and noise due to high-frequency current increase.

However, since the sensorless control device for a synchronous machine according to the present embodiment employs the phase angle estimation method described above, the current sampling time point is converted into a voltage within a range in which a correct current change high-frequency component can be measured. In addition to being able to select freely without being influenced, it is also possible to reduce the influence of noise by calculating the current change using linear approximation, and as a result, the amount of high-frequency multiplication is reduced to 0 or lower than before. Is possible.

Therefore, according to the sensorless control device of the synchronous machine according to the present embodiment, the phase angle of the rotor can be estimated without using the rotational phase angle sensor, and the sensorless configuration reduces the size. Cost reduction, easy maintenance, and loss due to high-frequency currents' Adjustment for stable operation without incurring or increasing the noise can be simplified. Become. [0048] (Second embodiment)

A sensorless control device for a synchronous machine according to a second embodiment of the present invention will be described. The configuration of the present embodiment is the same as that of the first embodiment and is shown in FIG. 3, but the calculation process performed by the rotational phase angle estimator 08 differs from the first embodiment in that it is permanent. In a high torque output state in which a current at which the inductance of the magnet synchronous machine 07 saturates flows, the rotational phase angle is estimated based on a change in the shape of the spatial distribution of the current changing high frequency component.

[0049] The estimation of the rotational phase angle performed by the rotational phase angle estimation unit 08 shown in the first embodiment is that the spatial distribution of the current change high frequency component represents the rotor inductance of the permanent magnet synchronous machine 07, which It utilized the fact that there was a rotational relationship corresponding to the synchronized dq axis coordinate system. In this case, when a high torque is output as a characteristic of the permanent magnet synchronous machine 07, the stator inductance is saturated by the drive current, and the rotational relationship of the above distribution cannot be observed.

[0050] In the present embodiment, the rotational phase angle estimation unit 08 performs effective rotational phase angle estimation even in such a case, and particularly pays attention to the shape of the above distribution, and the shape of the spatial distribution and the estimated phase. The rotational phase angle is estimated in correspondence with the angular error.

The operation of the sensorless control apparatus for a synchronous machine of the present embodiment configured as described above will be described. Figure 4 shows the high-frequency component distribution of the current change in the low torque state. If an error ΔΘ occurs in the estimated phase angle Θest, this high-frequency component distribution is a distribution rotated by -ΔΘ. Figure 5 shows the distribution of high-frequency components in a state where the error ΔΘ is intentionally generated by 30 °. From Fig. 5, the spatial correspondence between the dq-axis coordinate system and the high-frequency component distribution is clear.

[0052] The rotational phase angle estimation method employed in the first embodiment is to estimate the rotational phase angle by using such a feature. However, depending on the synchronous machine, the distribution shape of the high frequency component is not an ellipse but an ellipse close to a perfect circle as shown in Fig. 6 in a high torque operation state. In this state, an estimation error is likely to occur due to the influence of noise or the like that makes it difficult to extract the relationship corresponding to the dq axis coordinate system as described above. Furthermore, as can be seen from Fig. 6, the rotational relationship between the ellipse and the dq axis itself also varies depending on the current phase. Since the elliptic distribution and dq axis do not have a fixed rotational relationship, it is necessary to correct appropriately according to the driving situation.

Therefore, in the present embodiment, the rotational phase angle estimator 08 uses the high frequency component distribution shape itself rather than performing the rotational phase angle estimation using the rotational relationship between the distribution of the high frequency component and the dq axis coordinate system. To estimate.

[0054] In a high torque output state, inductance saturation is induced by the drive current flowing through the stator, and the salient pole ratio that can be observed in the stator is reduced. In the worst case, the salient pole ratio is 1. However, if an estimation error Δ Θ occurs in a high torque state that induces inductance saturation, the influence of that causes a phase of current Θ i controlled in the γ δ axis coordinate system and a force in the dq axis coordinate system to be only Δ Θ. It becomes a rotated phase. At this time, because the current phase changes, the saturation state of the stator inductance changes, and the shape of the high-frequency component of the current change varies greatly according to ΔΘ as shown in Figs. Therefore, even in such a high torque state, the rotational phase angle is estimated by observing the shape of the distribution and associating it with ΔΘ. At this time, the high frequency component distribution is affected by both the inductance of the stator itself and the rotor inductance. That is, for example, the shape of a substantially elliptic distribution is stored in correspondence with the torque command and the estimation error, and the rotational phase angle is estimated by comparing the obtained distributions.

As described above, in the sensorless control device for a synchronous machine according to the present embodiment, the rotor phase angle is estimated without using the rotation phase angle sensor, thereby reducing the size and cost of the device and maintaining the device. In addition, the rotation phase angle can be estimated stably even in a high torque state.

[0056] (Third embodiment)

A third embodiment of the sensorless control device for a synchronous machine of the present invention will be described. The configuration of the sensorless control device of this embodiment is the same as that of the first and second embodiments shown in FIG. 3, but the rotational phase angle estimation processing power performed by the rotational phase angle estimation unit 08 therein is the same. Unlike these embodiments, a predetermined feature amount is calculated in the spatial distribution of the high-frequency component of the current change in the dq axis coordinate system, and the rotational phase angle is estimated based on the feature amount. . In the present embodiment, the maximum value of the component detected in the predetermined angle direction from the γ-axis of the high-frequency component distribution of the current change on the γ-δ axis is selected as the feature amount. The correspondence between the selected feature quantity and the estimation error is the characteristic shown in Fig. 9, and is a linear characteristic with an offset especially near the zero point of the estimation error. In Fig. 9, the calculation is performed with the predetermined angle set to 45 °. Therefore, the rotational phase angle is estimated by performing feedback estimation so that the feature amount converges to a point with zero error.

With this configuration, it is possible to perform estimation based on an amount obtained by a simple comparison operation rather than a complicated operation. In FIG. 9, the predetermined angle is about 45 degrees. However, the feature quantity may be calculated in any way as long as the characteristics of the feature quantity are obtained. For example, the feature amount may be the area occupied by the distribution in the first quadrant on the γδ coordinate axes. The area calculated in this way also has a characteristic that varies depending on the estimation error, so it can be adopted as a feature quantity.

[0059] As described above, in the sensorless control device for a synchronous machine according to the present embodiment, the phase angle of the rotor is estimated without using the rotational phase angle sensor, thereby reducing the size and cost of the device. Maintenance can be facilitated, and stable rotation phase angle estimation can be performed without performing complicated calculations. Furthermore, by calculating the feature value based on the components detected in the predetermined angle direction from the γ-axis, it is possible to simplify the calculations required for rotational phase angle estimation, shortening the calculation time and improving the performance of the processing unit. Reduction can be achieved.

[0060] (Fourth embodiment)

A sensorless control device for a synchronous machine according to a fourth embodiment of the present invention will be described with reference to FIG. In the present embodiment, the same elements as those in the first embodiment shown in FIG. 3 are denoted by the same reference numerals.

[0061] The sensorless control device for a synchronous machine of the present embodiment is characterized in that a high frequency command superimposing unit 09 is added to the first embodiment. The high-frequency command superimposing unit 09 is configured so that a high-frequency command having a frequency sufficiently higher than the fundamental frequency for operating the permanent magnet synchronous machine 07 is superimposed on the voltage output from the inverter 05. Superimpose the high-frequency command on the command. Furthermore, a rotating high frequency command is superimposed as a high frequency command. Furthermore, as a high frequency command, it is significant in a predetermined angular direction on the γ δ coordinate axes. A high-frequency command that generates a high-frequency component of current change is superimposed.

Next, the operation of the sensorless control device of the present embodiment configured as described above will be described. The principle of estimating the rotational phase angle is based on the time change rate of the high-frequency current. In order to obtain the rate of change, the change in the high-frequency current must be observed with high accuracy. However, the voltage command is small when the voltage output from the inverter 05 is low, for example, when the engine is stopped at low speed and low torque. In addition, the state in which the high-frequency current time change rate can be calculated significantly is a state in which the voltage vector force VO, V7 other than the voltage vector force VO, V7 output from the inverter 05 as a result of modulation by the PWM modulator 04 is a so-called non-zero voltage vector Often there is. This is because large current fluctuation occurs when a non-zero voltage vector is output. The definition of the voltage vector output from inverter 05 is shown in FIG. However, in the above-mentioned stop, low speed, and low torque states, the non-zero voltage vector time interval becomes very short, and a significant high-frequency current change may not be observed depending on the sampling period of the AZD converter. In such a case, an estimation error of the rotational phase angle tends to occur as a result.

Therefore, in the present embodiment, the high frequency command superimposing unit 09 adds the high frequency command shown in Formula 11 to the current command, thereby extending the time interval of the non-zero voltage vector. The reason why the superimposed command is a high frequency command is to act as a disturbance on the torque generated by the motor force.

[Equation 11] 1 rf = ^sin , Ri

¹呵-I _hf cos m _hf tj

[0064] where Ihf is the high frequency command amplitude and ω hf is the high frequency command angular velocity.

In the present embodiment, the high frequency command can be freely selected as long as the above-described characteristics can be obtained. Therefore, it is not always necessary to set as shown in Equation 11. However, by setting the rotational high frequency command to be superimposed as shown in Equation 11, the above characteristics can be obtained and the following effects can be obtained. The rotational phase angle estimation unit 08 uses a high-frequency current change as a rotational phase angle estimation method. When the shape or area of the component is used, a uniform distribution can be obtained in the δδ coordinate system by giving a rotational high frequency, and the shape can be accurately grasped, so that the accuracy of phase angle estimation is improved. It becomes possible.

[0066] When a high-frequency command that generates a high-frequency component of a significant current change in a predetermined angular direction on the γδ coordinate axis is superimposed as a high-frequency command, the following effects can be obtained. This is significant in the predetermined angular direction when the method for estimating the rotational phase angle based on the component in the predetermined angular direction from the γ-axis of the high-frequency component of the current change described in the third embodiment is implemented. High frequency can be obtained, and the detection accuracy can be improved, and hence the accuracy of phase angle estimation can be improved.

Here, the high-frequency command that generates a high-frequency component of a significant current change in a predetermined angular direction may be selected as follows:

[Equation 12]

=, cos (sin (CT _{A /} t)

2 = _v sin (sin (

[0068] where φ is a predetermined angle.

[0069] As described above, in the sensorless control apparatus for a synchronous machine according to the present embodiment, the phase angle of the rotor is estimated without using the rotation phase angle sensor, thereby reducing the size and cost of the apparatus. Maintenance can be facilitated, and a stable rotational phase angle can be estimated even when stopped at low speed and low torque.

Claims

The scope of the claims

[1] an inverter that mutually converts DC power and AC power;

A synchronous machine having a magnetic saliency on the rotor and driven by the inverter power,

PWM modulation means for determining an output voltage in the inverter based on a command for controlling the synchronous machine;

Current detecting means for detecting a current flowing in the synchronous machine;

High-frequency component calculation means for calculating a high-frequency component of a current change caused by a voltage determined by the PWM modulation means and output from the inverter;

A rotation phase angle estimating means for estimating a rotation phase angle of the synchronous machine based on a spatial distribution of the high frequency component on a rotation coordinate axis synchronized with the rotation of the synchronous machine. Sensorless control device.

[2] The rotational phase angle estimating means estimates the rotational phase angle based on a shape of a spatial distribution of the high frequency component in a high torque output state in which a current at which an inductance of the synchronous machine is saturated flows. The sensorless control device for a synchronous machine according to claim 1,

[3] The synchronous machine according to claim 1 or 2, wherein the rotational phase angle estimator estimates a rotational phase angle of the synchronous machine based on a feature amount of the high-frequency component on a rotational coordinate axis. Sensorless control device.

[4] The rotational phase angle estimation means estimates the rotational phase angle based on a component of the high-frequency component on a rotational coordinate axis in a predetermined angular direction. A sensorless control device for a synchronous machine as described.

5. The sensorless control device for a synchronous machine according to any one of claims 1 to 4, further comprising a high frequency command superimposing unit that superimposes a high frequency command on a command for controlling the synchronous machine.

6. The sensorless control device for a synchronous machine according to claim 5, wherein the high-frequency command superimposing unit superimposes a rotating high-frequency command.

[7] The high-frequency command superimposing means has a significant height in a predetermined angular direction on the rotational coordinate axis. The sensorless control device for a synchronous machine according to claim 1, wherein a high-frequency command that causes a change in frequency current is superimposed.