JP2024083840A - Inverter control device and power conversion device - Google Patents

Abstract

[Problem] To provide an inverter control device capable of estimating with high accuracy the rotational phase angle and rotational angular velocity of a rotating synchronous machine. [Solution] The device according to the embodiment includes a current command generation unit 101 that generates a current command, a current detection unit 110 that detects the value of a current flowing between an inverter main circuit INV and a synchronous machine M, a current control unit 102 that calculates a voltage command corresponding to the current command, a modulation unit 104 that generates a drive signal for the inverter main circuit INV based on the voltage command, coordinate conversion units 103 and 105 that perform coordinate conversion of the voltage command value and the current detection value using an estimated value of the rotational phase angle of the synchronous machine M, an initial value calculation unit 107 that calculates initial values of the rotational phase angle and rotational angular velocity of the synchronous machine M using a pulsation component generated by magnetic salient poles when the estimated value of the rotational phase angle used in the coordinate conversion is set to zero, and a calculation unit 106 that calculates estimated values of the rotational phase angle and rotational angular velocity of the synchronous machine M using the initial value calculated by the initial value calculation unit 107. [Selected Figure] FIG.

Description

Embodiments of the present invention relate to an inverter control device and a power conversion device.

Conventionally, in inverter control devices that drive synchronous motors such as permanent magnet synchronous motors (PMSM) and synchronous reluctance motors (SynRM), technology has been proposed that performs sensorless control without using a rotor position (rotation angle) sensor or rotation speed sensor for the motor.

For example, when restarting an inverter from a free-running state using an inverter control device that performs sensorless control, it is necessary to synchronize the inverter output phase and frequency with the position and rotational angular velocity of the motor's rotor. For this reason, methods have been proposed that use the magnetic salient poles of the motor to estimate the rotor's rotational phase angle and rotational angular velocity when restarting the inverter, and methods that use the residual voltage of an AC motor to estimate the rotational phase angle and rotational angular velocity.

However, when the inverter is restarted from a stopped state, the former method causes the frequency of the high-frequency superposition to approach the fundamental frequency frequency, which can lead to poor high-frequency current detection accuracy during high-speed rotation and poor speed estimation accuracy. Also, with the latter method, the estimation accuracy improves as the speed increases, but in motors with very few or no magnets, no residual magnetic flux or no-load magnetic flux is generated, making it impossible to estimate the angle or speed.

Publication No. WO2020/110315

For example, in a control device that employs the former method, it has been proposed to use a filter to separate the fundamental current and pulsating current that are generated when a DC voltage is applied, and a map to correct the filter delay. However, using filters and maps makes the system more complicated.

The present invention has been made in consideration of the above circumstances, and aims to provide an inverter control device and a power conversion device that can estimate the rotational phase angle and rotational angular speed of a rotating synchronous machine with high accuracy.

The inverter control device according to the embodiment includes a current command generation unit that generates a current command, a current detection unit that detects the value of a current flowing between the inverter main circuit and the synchronous machine, a current control unit that calculates a voltage command according to the current command, a modulation unit that generates a drive signal for the inverter main circuit based on the value of the voltage command, a coordinate conversion unit that performs coordinate conversion of the voltage command value and the detected current value using an estimated value of the rotational phase angle of the synchronous machine, an initial value calculation unit that calculates initial values of the rotational phase angle and rotational angular velocity of the synchronous machine based on a pulsating component of the output current of the synchronous machine generated by the magnetic salient pole of the synchronous machine when the estimated value of the rotational phase angle of the synchronous machine used in the coordinate conversion is set to a value that does not change, and an angle/speed calculation unit that calculates the estimated values of the rotational phase angle and rotational angular velocity of the synchronous machine using the initial value calculated by the initial value calculation unit.

FIG. 1 is a diagram illustrating an example of a configuration of an inverter control device and a power conversion device according to a first embodiment. FIG. 2 is a diagram for explaining one configuration example of the synchronous machine shown in FIG. FIG. 3 is a diagram for explaining the definitions of the d-axis, the q-axis, and the estimated rotating coordinate system (dc-axis, qc-axis) in the first embodiment. FIG. 4 is a diagram illustrating an example of flags for controlling various operation modes in the inverter control device of the first embodiment. FIG. 5 is a diagram illustrating an example of the configuration of a current command generating unit of the inverter control device according to the first embodiment. FIG. 6 is a diagram illustrating an example of the configuration of the high-frequency voltage superimposing unit of the inverter control device according to the first embodiment. FIG. 7 is a diagram illustrating an example of the configuration of the angle/speed initial value calculation unit of the inverter control device according to the first embodiment. FIG. 8 is a diagram illustrating an example of the configuration of a rotation angle/speed calculation unit of the inverter control device according to the first embodiment. FIG. 9 is a diagram illustrating an example of the configuration of a current control unit of the inverter control device according to the first embodiment. FIG. 10 is a diagram for explaining an example of a transfer function when the inverter control device of the first embodiment performs PI control on a current. FIG. 11 is a diagram illustrating an example of timing for changing the current control gain in the current control unit of the inverter control device according to the first embodiment. FIG. 12 is a flowchart for explaining an example of the operation of the inverter control device of the first embodiment. FIG. 13 is a diagram for explaining an example of an effect of the inverter control device according to the first embodiment. FIG. 14 is a diagram illustrating an example of another configuration of the angle/speed initial value calculation unit of the inverter control device according to the first embodiment. In FIG.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An inverter control device and a power conversion device according to embodiments will be described below with reference to the drawings.
FIG. 1 is a diagram illustrating an example of a configuration of an inverter control device and a power conversion device according to a first embodiment.
The power conversion device of the first embodiment includes an inverter main circuit INV and an inverter control device 100.

The inverter main circuit INV converts DC power into three-phase AC power and outputs it to the synchronous machine M. The inverter main circuit INV has an upper arm switching element and a lower arm switching element in each phase.

The inverter main circuit INV is supplied with control signals (gate commands) for the upper and lower arm switching elements from the inverter control device 100. The inverter main circuit INV can convert AC power to DC power and vice versa by switching the switching elements on and off.

The synchronous machine M is, for example, a motor with magnetic salient poles, such as a permanent magnet synchronous motor (PMSM) or a synchronous reluctance motor (SynRM). In this embodiment, an example in which a SynRM is used as the synchronous machine M will be described.

FIG. 2 is a diagram for explaining one configuration example of the synchronous machine shown in FIG.
Here, the configuration of SynRM is shown as an example of a synchronous machine M.
The synchronous machine M includes a rotor 20 and a stator 10. A magnetic field is generated by three-phase AC current flowing through each excitation phase, and torque is generated by magnetic interaction with the rotor. Note that only a part of the synchronous machine M is shown here, and the stator 10 and the rotor 20 of the synchronous machine M are, for example, a combination of multiple configurations shown in FIG.

The rotor 20 has an air gap 21, an outer circumferential bridge BR1, and a center bridge BR2.
The center bridge BR2 is disposed on a line connecting the outer periphery and center of the rotor 20. The line on which the center bridges BR2 are arranged corresponds to the d-axis. The outer periphery bridge BR1 is located between the outer periphery of the rotor 20 and the air gaps 21. The synchronous machine M portion shown in FIG. 2 is provided with six air gaps 21 extending between the outer periphery and center of the rotor 20. The air gaps 21 extend between the center bridge BR2 and the outer periphery bridge BR1, symmetrically with respect to the d-axis.

FIG. 3 is a diagram for explaining the definitions of the d-axis, the q-axis, and the estimated rotating coordinate system (dc-axis, qc-axis) in the first embodiment.
In this embodiment, the d-axis is the axis where the magnetic saliency is small, and the q-axis is the axis where the magnetic saliency is large. The dc-axis is the d-axis in the estimated coordinate system, and the qc-axis is the q-axis in the estimated coordinate system.

The d-axis is a vector axis rotated by a rotational phase angle θ from the α-axis (U-phase) of the αβ fixed coordinate system, and the q-axis is a vector axis orthogonal to the d-axis in electrical angle. In contrast, the dcqc estimated rotating coordinate system corresponds to the d-axis and q-axis at the estimated position of the rotor 20. That is, the dc-axis is a vector axis rotated by the rotational phase angle estimated value θest from the α-axis, and the qc-axis is a vector axis orthogonal to the dc-axis in electrical angle. In other words, the vector axis rotated by the estimated error Δθ from the d-axis is the dc-axis, and the vector axis rotated by the estimated error Δθ from the q-axis is the qc-axis.

For example, in a method of estimating the rotational angular velocity by superimposing a high-frequency voltage in the d-axis direction, the estimated values of the rotational angular velocity and rotational phase angle of the synchronous machine M can be calculated by performing PLL (Phase Locked Loop) control for the axis where the q-axis high-frequency current is zero, that is, where the inductance to the harmonics is minimum.

The inverter control device 100 includes a calculation device having at least one processor, such as a CPU or MPU, and a memory in which a program executed by the processor is recorded. The inverter control device 100 can realize the various functions described below by software or a combination of software and hardware.

The inverter control device 100 receives a torque command T ^* and an on/off command Gst from a higher-level control device (not shown). The higher-level control device controls a plurality of components in an apparatus equipped with a synchronous machine M and an inverter main circuit INV so that they operate in a coordinated manner. The higher-level control device may include a user interface such as an operation panel, and may output the torque command T ^* and the on/off command Gst to the inverter control device 100 based on the operation of the user interface.

The inverter control device 100 includes a current command generation unit 101, a current control unit 102, a coordinate (dq/3Φ) conversion unit 103, a modulation unit 104, a coordinate (3Φ/dq) conversion unit 105, a rotation angle/speed calculation unit 106, an angle/speed initial value calculation unit 107, a flag generation unit 108, a high-frequency voltage superposition unit 109, current detectors (current detection units) 110U, 110V, 110W, and an adder A1.

The flag generation unit 108 generates flags for switching between various modes for driving the synchronous machine M. The flag generation unit 108 generates and outputs a first flag Flg1, a second flag Flg2, a third flag Flg3, and an initialization flag Flg_init.

FIG. 4 is a diagram illustrating an example of flags for controlling various operation modes in the inverter control device of the first embodiment.
The first flag Flg1 is "1" during the initial value estimation period and is "0" during the normal control period. The second flag Flg2 is "0" during the initial value estimation period and is "1" during the normal control period. The initialization flag Flg_init indicates the timing for initializing (setting the initial values) the rotational angular velocity and rotational phase angle calculated during the initial value estimation period. In this embodiment, the initialization flag Flg_init is "1" during a predetermined period (one sample period) from the end of the initial value estimation period and is "0" during other periods (initial value estimation period and normal control period).

The third flag Flg3 is a flag for controlling switching between low-speed and high-speed sensorless control during normal control. In this embodiment, the third flag Flg3 is "1" during the period during which high-speed sensorless control is performed during the normal control period, and is "0" during other periods (the period during which low-speed sensorless control is performed during the normal control period and the initial value estimation period).

The first delay flag Flg1_old is a flag obtained by delaying the first flag Flg1 by one sample period. The second delay flag Flg_old is a flag obtained by delaying the second flag Flg2 by one sample period.

The current detectors 110U, 110V, 110W detect the values of three-phase AC currents iu, iv, iw flowing to the synchronous machine M. The current detection values of the current detectors 110U, 110V, 110W are input to a coordinate (3Φ/dq) conversion unit 105 and converted into current detection values _Idc , _Iqc in a dq-axis rotating coordinate system.
The current command generating unit 101 generates and outputs a d-axis current command I _dref and a q-axis current command I _qref based on a torque command T ^* supplied from a higher-level control device.

FIG. 5 is a diagram illustrating an example of the configuration of a current command generating unit of the inverter control device according to the first embodiment.
The current command generating unit 101 includes a command generating unit 11, a limiting unit 12, a time delay unit 13, switching units 14 and 15, and logical AND operation units 16-18.

The command generating unit 11 calculates and outputs a d-axis current command value I _dref1 and a q-axis current command value I _qref1 that minimize the copper loss, for example, by using a map, an approximation formula, a theoretical formula, or the like.
The limit unit 12 calculates the absolute value of the second d-axis current command I _dref2 by setting the absolute value of the d-axis current command value I _dref1 to be equal to or greater than the lower limit value dlim, and calculates and _{outputs the second d-axis current command I dref2 so} that its sign is the same as that of the d-axis current command value I _dref1 .
The limiting section 12 includes, for example, an absolute value calculating section, a lower limiting section, a sign determining section, and a multiplying section (none of which are shown).

The absolute value calculation unit receives the d-axis current command I _dref1 from the command generation unit 11, calculates and outputs the absolute value of the d-axis current command I _dref1 .
The lower limit unit receives the absolute value of the d-axis current command I _dref1 from the absolute value calculation unit, and outputs an absolute value of a second d-axis current command I _dref2 equal to the absolute value of the d-axis current command I _dref1 when the absolute value of the d-axis current command I dref1 is equal to or greater than the lower limit value idlim. The lower limit unit LIM outputs the absolute value of the second d-axis current command I _dref2 equal to the lower limit value dlim when the absolute value of the d-axis current command I _dref1 is less than _the lower limit value idlim.

The sign determination unit receives the d-axis current command I _dref1 from the command generation unit 11 and determines whether the d-axis current command I _dref1 is greater than zero or less than or equal to zero. The sign determination unit outputs "+1" when the d-axis current command I _dref1 is greater than zero, and outputs "-1" when the d-axis current command I _dref1 is less than or equal to zero.

The multiplier multiplies the absolute value of the second d-axis current command I _dref2 output from the lower limiter by the output value of the sign determiner, and outputs the result.
As described above, by limiting the lower limit of the amplitude of the d-axis current, it becomes possible to pass a fundamental current in the d-axis direction (or −d-axis direction) equal to or greater than a predetermined threshold value through the synchronous machine M.

The time delay unit 13 outputs the on/off command Gst with a predetermined delay. The on/off command Gst is a control command for a logical AND operation unit 16 that switches the electrical connection of a path that supplies a torque command to the command generation unit 11, and a logical AND operation unit 18 that switches the electrical connection of a path that outputs the second d-axis current command I _dref2 from the limit unit 12 to the switching unit 14. The on/off command Gst is also supplied, via the time delay unit 13, to a logical AND operation unit 17 that switches the electrical connection of a path that outputs the q-axis current command I _qref1 from the command generation unit 11.

The switching units 14 and 15 switch the current command value between the initial estimation period and the normal control period.
The switching unit 14 includes a first input terminal, a second input terminal, and a first output terminal. The second d-axis current command I _dref2 output from the limit unit 12 is input to the first input terminal via the logical product calculation unit 18. The d-axis current command initial value I _{dref_init} is input to the second input terminal. When the value of the first flag Flg1 is "0", the first input terminal and the first output terminal of the switching unit 14 are electrically connected, and when the value of the first flag Flg1 is "1", the second input terminal and the first output terminal are electrically connected. The value input to the output terminal of the switching unit 14 is output as the d-axis current command I _dref .

The switching unit 15 has a third input terminal, a fourth input terminal, and a second output terminal. The second q-axis current command _iqref2 output from the logical product calculation unit 17 is input to the third input terminal. The input value of the fourth input terminal is zero. When the value of the first flag Flg1 is "0", the third input terminal and the output terminal of the switching unit 15 are electrically connected, and when the value of the first flag Flg1 is "1", the fourth input terminal and the output terminal are electrically connected. The value input to the output terminal of the switching unit 15 is output as the q-axis current command _Iqref .

The current control unit 102 is equipped with, for example, a PI (proportional integral) controller, and compares the dc-axis current value _Idc and the qc-axis current value _Iqc supplied from the coordinate conversion unit 105 with the d-axis current command _Idref and the q-axis current command _Iqref , and calculates and outputs voltage commands _{Vd_ACR} and Vq_ACR so that the dc-axis current value _Idc and the d-axis current command _Idref become zero and the difference between the qc-axis current value _{Iqc and the q-axis current command Iqref becomes zero} .

FIG. 6 is a diagram illustrating an example of the configuration of the high-frequency voltage superimposing unit of the inverter control device according to the first embodiment.
The high frequency voltage superimposing unit 109 generates a high frequency voltage of an arbitrary frequency for the dc axis or the qc axis or for both the dc axis and the qc axis in accordance with a triangular wave carrier (carrier command), and outputs the generated voltage to the adder A1, the rotation angle/speed calculation unit 106, and the rotation angle/speed calculation unit 106. In this embodiment, the high frequency voltage superimposing unit 109 outputs a high frequency voltage _Vdch for the dc axis.

The high-frequency voltage superimposing unit 109 operates during the normal control period when the synchronous machine M is under low-speed sensorless control (when the delayed second flag Flg_old is "1" and the third flag Flg3 is "0") and outputs the high-frequency voltage _Vdch .

The high frequency voltage superimposing unit 109 includes a delay unit 91 , a synchronization pulse generating unit 92 , and a high frequency voltage synchronization unit (logical AND operation unit) 93 .
The synchronization pulse generating unit 92 generates a synchronization pulse synchronized with the carrier command and outputs it to the high frequency voltage synchronization unit 93 .

The high frequency voltage synchronization unit 93 multiplies a voltage Vh, which is a DC voltage command value of a predetermined magnitude generated internally, by a synchronization pulse and outputs the result. That is, the high frequency voltage Vdch output from the high frequency voltage superimposing unit 109 is a high frequency voltage command _Vdch having a predetermined amplitude Vh and a high frequency voltage period (1/ _fdch ) synchronized with the _period of the carrier command.

The high frequency voltage command value V _dch output from the high frequency voltage superimposing unit 109 is added to the d axis voltage command V _{d_ACR} by an adder A1, and the output value of the adder A1 is supplied to the dq/3Φ conversion unit 103 as the d axis voltage command value V _dref .

The coordinate (dq/3Φ) conversion unit 103 vector-converts values in a dq rotating coordinate system synchronized with the rotational angular velocity of the rotor of the synchronous machine M into values in a three-phase fixed coordinate system, and outputs the converted values. In this embodiment, the coordinate (dq/3Φ) conversion unit 103 converts the voltage command values _Vdref , _Vqref in the dq rotating coordinate system into three-phase voltage command values Vu ^* , Vv ^* , Vw ^* in the three-phase fixed coordinate system, and outputs them.

The modulator 104 converts the three-phase voltage command values Vu ^* , Vv ^* , Vw ^* into gate commands for the inverter main circuit INV. In this embodiment, the modulator 104 generates gate commands (drive signals) by PWM modulation that compares a triangular wave carrier with the voltage command values Vu ^* , Vv ^* , Vw ^* , and outputs the gate commands to the inverter main circuit INV.

Using the rotational phase angle estimate value θest output from the rotational angle/speed calculation unit 106, the 3Φ/dq conversion unit 105 vector-converts the values in the three-phase fixed coordinate system into values in a dq rotating coordinate system synchronized with the rotational angular velocity of the rotor of the synchronous machine M, and outputs the converted values. In this embodiment, the 3Φ/dq conversion unit 105 converts the current detection values in the three-phase fixed coordinate system into current detection values _Idc , _Iqc in the dq rotating coordinate system and outputs them.

The angle/speed initial value calculation unit 107 operates when the value of the delayed first flag Flg1_old is “1”, and uses the values of the three-phase AC currents iu, iv, iw and the d-axis current command value _Idref to perform three-phase/dq conversion and PLL on a pulsating current (described later) to calculate a rotation angle initial value θest_init and a rotation angular velocity initial value ωest_init.

Here, the pulsating current used when the angle/speed initial value calculation unit 107 calculates the rotation angle initial value θest_init and the rotation angular velocity initial value ωest_init will be described.
The voltage equation for the target motor, a synchronous machine M having no magnets or few magnets, is expressed by equation (1).
R: armature resistance, ω: motor angular velocity, Ld: d-axis inductance, Lq: q-axis inductance, vd: d-axis voltage, vq: q-axis voltage, id: d-axis current, iq: q-axis current

Moreover, when equation (1) is transformed into a coordinate system having a certain angle θ, equation (2) is obtained.

Here, L _dc , L _qc , L _dqc , L ₀ , and L ₁ are as follows, and p is the differential operator (d/dt).

When the above equation (2) is solved for the current, equation (6) is obtained.

Furthermore, by converting equation (6) back into the form of a rotation angle using equations (3)-(5), we obtain equation (7).
Assuming that the estimated voltage values V _dc , V _qc are equal to the voltage command values from the inverter, if the average value (DC component) of the motor voltage can be controlled, the above equation (7) can be transformed into the following equation (8).

Furthermore, if it is assumed that a current is passed only through the d-axis, the above formula (8) can be transformed into formula (9).

Further, formula (9) above can be rearranged to obtain formula (10).
According to equation (10), a current having a sine wave component with a frequency of 2θ is generated on the d-axis, and a current having a cosine wave component with a frequency of 2θ is generated on the q-axis. Note that the pulsating currents generated on the d-axis and q-axis can be suppressed by increasing the gain in the current control.

In this embodiment, the angle/speed initial value calculation unit 107 uses the pulsating current generated by the above principle to calculate the rotation angle initial value θest_init and the rotation angular velocity initial value ωest_init.

FIG. 7 is a diagram illustrating an example of the configuration of the angle/speed initial value calculation unit of the inverter control device according to the first embodiment.
The angle/speed initial value calculation unit 107 includes a delay unit 71, a coordinate conversion unit 72, a normalization unit 73, division units 74, 77, 7A5, subtraction units 75, 7A1, a PI control unit 76, integration units 78, 79, a hold unit 70, addition units 7A2, 7A3, and a multiplication unit 7A4.

The delay unit 71 outputs a value (Flg_old) obtained by delaying the first flag Flg1 by a predetermined period (one sampling period).
The multiplication unit 7A4 calculates the product (G×I _dref ) of the d-axis current command value I _dref by the gain G, and outputs the product to the addition unit 7A1 and the division unit 7A5.
The division unit 7A5 divides the value output from the multiplication unit 7A4 by 2, and outputs the quotient ((G×I _dref )/2) to the addition units 7A2 and 7A3.

Subtraction unit 7A1 outputs the difference (iu-G×I _dref ) obtained by subtracting the output value of multiplication unit 7A4 from the value of U-phase current iu.
The adder 7A2 outputs the sum (iv+(G×I _dref )/2) of the value of the V-phase current iv and the output value of the divider 7A5.
The adder 7A3 outputs the sum (iw+(G×I _dref )/2) of the value of the W-phase current iw and the output value of the divider 7A5.

Here, when current control is performed with the -d axis as the target and the angle set to zero, a current that is sqrt (2/3) times the d-axis current command value I _dref flows through the U phase. At this time, a current that is halved and has its sign inverted flows through the V and W phases. By utilizing this, this embodiment adds a value ((G×I _dref )/2) that is the d-axis current command value I _dref multiplied by the gain to each of the V-axis current iv and the W-axis current iw, and thereby the pulsating current can be easily calculated without using a high-pass filter or a complicated map.

The coordinate conversion unit 72 receives the output value of the subtraction unit 7A1, the output values of the addition units 7A2 and 7A3, and the rotational phase angle θest2, and converts the current values in the three-phase fixed coordinate system into current values d2 and q2 in the dq-axis rotating coordinate system using the rotational phase angle θest2, and outputs the converted values.

The normalization unit 73 receives the current values d2 and q2 from the coordinate conversion unit 72, and calculates and outputs a value A for normalizing the q-axis current value q2. For example, the normalization unit 73 calculates the value A for normalizing the amplitude of the q-axis current value q2.

The division unit 74 divides the q-axis current value q2 by the value A output from the normalization unit 73, and outputs the normalized value to the subtraction unit 75. By normalizing the q-axis current value q2, the amplitude of the q-axis current value q2 is normalized (the amplitude becomes 1 or less), and it is possible to prevent the input value to the downstream PI control unit 76 from becoming excessively large.

Note that, according to the above formula (10), the d-axis current contains a sine component, and the q-axis current contains a cosine component. For this reason, the angle/speed initial value calculation unit 107 may calculate the arctangent (a tan), for example, using the d-axis current as the numerator and the q-axis current as the denominator. In this case, since the amplitude of the pulsating current is almost the same on the d and q axes, there is no need to normalize the amplitude, and the gain design of the subsequent PI control is simplified.

The subtraction unit 75 calculates a difference by subtracting the value output from the division unit 74 (the normalized q-axis current value q2) from zero, and outputs the difference to the PI control unit 76.
The PI control unit 76 calculates the rotation angular velocity ωest2 so that the value input from the subtraction unit 75 follows zero, and outputs the calculation result to the division unit 77 and the integration unit 78. Here, since the difference between the normalized q-axis current value q2 and zero is input to the PI control unit 76, the PI control unit 76 calculates the rotation angular velocity ωest2 such that the q-axis current value q2 becomes zero. This is because the inverter control device of this embodiment causes a current to flow through the d-axis during the initial value calculation period.

In the inverter control device of this embodiment, a method of passing an offset current through the d-axis during the normal control period is adopted, but a method of passing a current through the q-axis may also be adopted. This offset current is passed for the purposes of magnetically saturating the rotor bridge, generating a no-load voltage during high-speed rotation to estimate the position, and so on. If a current is passed in the d-axis direction during the initial value calculation period, there is an advantage that the transition to normal control is smooth, and the d-axis is relatively more susceptible to magnetic saturation than the q-axis, making magnetic salient poles more pronounced.

The division unit 77 outputs the value ωest1 obtained by dividing the rotational angular velocity ωest2 output from the PI control unit 76 by 2 to the integration unit 79 and the hold unit 70. Since the rotational angular velocity ωest2 is a rotational angular velocity obtained from a pulsating current having a frequency of 2θ (twice the fundamental frequency of the synchronous machine M), the division unit 77 calculates the rotational angular velocity ωest1 by dividing the rotational angular velocity ωest2 by 2.

The integrator 79 integrates the rotational angular velocity ωest1 output from the divider 77 to calculate a rotational phase angle θest1, and outputs the calculated value to the hold unit 70.
When the initialization flag Flg_init changes from "0" to "1", the hold unit 70 holds the rotational angular velocity ωest1 and the rotational phase angle θest1, and outputs the held values as the rotational angle initial value θest_init and the rotational angular velocity initial value ωest_init.
The integrator 78 integrates the rotational angular velocity ωest2 input from the PI controller 76 to calculate a rotational phase angle θest2, and outputs the result to the coordinate converter 72.

FIG. 8 is a diagram illustrating an example of the configuration of a rotation angle/speed calculation unit of the inverter control device according to the first embodiment.
The rotation angle/speed calculation unit 106 switches the position error estimation method in response to the third flag Flg3, and initializes the integral value in response to the initialization flag Flg_init. In addition, the rotation angle/speed calculation unit 106 sets the values of the rotation angle and speed to zero during the period in which the initial values of the rotation angle/speed are calculated.

The rotation angle/speed calculation unit 106 includes a bandpass filter 61, an FFT analysis unit 62, a first position error calculation unit 63, a second position error calculation unit 65, PLL units 64 and 66, an integration unit 67, a delay unit 68, and switching units SW1, SW2, and SW3.

The bandpass filter 61 receives the qc-axis response current values (output currents) _Idc , _Iqc from the 3Φ/dq conversion unit 105, and extracts and outputs the high-frequency current _Idc ', _Iqc ' components in a band including a frequency equal to the frequency of the high-frequency voltage command _Vdh (superimposed high-frequency voltage frequency) _fh .

The FFT analysis unit 62 performs FFT (Fast Fourier Transform) analysis of, for example, the high frequency current _Idc ', _Iqc ' components to detect the high frequency current amplitude _Ih . The FFT analysis unit 62 acquires the high frequency current _Idc ', _Iqc ' components and the high frequency voltage command _Vdch , samples the values of the high frequency current _Idc ', _Iqc ' components at timings of every 1/4 period of the high frequency voltage, and detects the high frequency current amplitude Ih from the difference between the sampled values. The FFT analysis unit 62 outputs the detected high frequency current amplitude _Ih to the first position error calculation unit 63.

The first position error calculation unit 63 calculates the rotational phase angle error Δθ by utilizing, for example, the following rotational angle dependent characteristic.
The rotation angle is transformed into a coordinate axis having a rotation phase angle error Δθ, and a low speed state is assumed, ignoring components including the rotation angular velocity ω and resistance voltage drop. In this case, the above formula (7) becomes formula (11).

Furthermore, if the high frequency voltage is applied only to the dc-axis, which is the estimated d-axis, then equation (11) can be rewritten as equation (12).

According to equation (12), it can be seen that the harmonic current of the qc axis changes depending on the rotational phase angle error Δθ. If the qc-axis current in equation (12) is divided by the dc axis and the amplitudes of the high-frequency currents are i _dch and i _qch , the rotational phase angle error Δθ is given by equation (13).

The first position error calculation unit 63 uses the rotational phase angle dependency characteristics of the harmonic currents of the dc and qc axes to calculate an estimated value Δθest of the rotational phase angle error Δθ, and outputs the calculation result Δθest to the PLL unit 64.

The PLL unit 64 performs PLL control so that the rotational phase angle error estimate Δθest input from the first position error calculation unit 63 converges to zero, calculates the rotational angular velocity estimate ωest, and outputs it to the switching unit SW1.

When the initialization flag Flg_init changes from "0" to "1", the PLL unit 64 sets the rotational angular velocity initial value ωest_init supplied from the angle/velocity initial value calculation unit 107 as the initial value for PLL control.

The second position error calculation unit 65 calculates the rotational phase angle error Δθ by applying a method that uses the relationship between the output of the current control unit and the feedforward voltage, for example.
The voltage equation when an error occurs in the rotational phase angle can be expressed by the above-mentioned equation (2), and the feedforward voltage command at this time is given by equation (14).

Here, _{R_set} is a resistance set value, _{Ld_set} is a d-axis inductance set value, _{Lq_set} is a q-axis inductance set value, and ωest is an angular velocity estimated value.

At this time, the difference (error) between them becomes the output of the current controller, and when the motor angular velocity ω and the estimated value ωest are almost equal and the voltage drop due to resistance can be ignored, equation (15) is obtained.

Focusing on the d-axis component of the above formula (15) gives formula (16).
In equation (16), the estimated value Δθest of the rotational phase angle error Δθ can be expressed by equation (17).

In actual calculations, ωest is calculated by integrating Δθest, so that the value of one sample period before can be used as ωest in this equation.
The second position error calculation unit 65 calculates the rotational phase angle error Δθest from the above equation (17) and outputs it to the PLL unit 66 .
The PLL unit 66 performs PLL control so that the rotational phase angle error estimate Δθest input from the second position error calculation unit 65 converges to zero, calculates a rotational angular velocity estimate ωest, and outputs it to the switch unit SW1.

When the initialization flag Flg_init changes from "0" to "1", the PLL unit 66 sets the rotational angular velocity initial value ωest_init supplied from the angle/velocity initial value calculation unit 107 as the initial value for PLL control.

For example, when the voltage of the motor model (the voltage required to pass a certain current) and the feedforward voltage command match, the PI controller no longer needs to output a voltage. Sensorless control considers this state to be the operating point to aim for, and constructs PLL control so that the voltage value of the PI control becomes zero. In other words, when the rotational angular velocity estimate ωest is changed with PLL control, the rotational phase angle estimate θest, which is its integral value, also changes, and the rotational phase angle error Δθ (= true value of rotational phase angle θ - rotational phase angle estimate θest) becomes smaller. By utilizing this fact, PLL control is constructed to achieve the above operating point.

The switching unit SW1 has a fifth input terminal, a sixth input terminal, and a third output terminal. The output value ωest of the PLL unit 64 is input to the fifth input terminal. The output value ωest of the PLL unit 66 is input to the sixth input terminal. The third output terminal is electrically connected to the fifth input terminal when the value of the third flag Flg3 is "0", and is electrically connected to the sixth input terminal when the value of the third flag Flg3 is "1". The third output terminal outputs the rotational angular velocity estimate value ωest to the integrator 67, the second position error calculator 65, and the switching unit SW3.

In the inverter control device of this embodiment, the flag generation unit 108 compares the result of the initial speed estimation (ωest) with the threshold value (ωsh) of the low-speed/high-speed sensorless control switching frequency, and switches the value of the third flag Flg3, thereby making it possible to selectively use the above-mentioned low-speed sensorless control and high-speed sensorless control.

The integrator 67 integrates the rotational angular velocity estimate ωest input from the switch SW1 to calculate a rotational phase angle estimate θest, and outputs the estimate to the switch SW2.
When the initialization flag Flg_init changes from "0" to "1", the integrator 67 sets the rotational phase angle initial value θest_init supplied from the angle/speed initial value calculator 107 as the initial value for the integral calculation.

The switching unit SW2 has a seventh input terminal, an eighth input terminal, and a fourth output terminal. The output value θest of the integrator 67 is input to the seventh input terminal. The input value of the eighth input terminal is zero. The fourth output terminal is electrically connected to the eighth input terminal when the delayed value of the second flag Flg_old is "0", and is electrically connected to the seventh input terminal when the delayed value of the second flag Flg_old is "1". The output value θest of the fourth output terminal is supplied to the coordinate conversion units 103 and 105 as the output value of the rotation angle/speed calculation unit 106.

The switching unit SW3 has a ninth input terminal, a tenth input terminal, and a fifth output terminal. The rotational angular velocity estimation value ωest is input to the ninth input terminal from the switching unit SW1. The input value of the tenth input terminal is zero. The fifth output terminal is electrically connected to the tenth input terminal when the value of the second flag Flg_old after the delay is "0", and is electrically connected to the ninth input terminal when the value of the second flag Flg_old after the delay is "1". The output value ωest of the fifth output terminal is supplied to the current control unit 102 and the flag generation unit 108 as the output value of the rotational angle/speed calculation unit 106.

As described above, in the inverter control device and power conversion device of this embodiment, a pulsating current is used when calculating the rotation angle initial value θest_init and the rotation angular velocity initial value ωest_init in the angle/speed initial value calculation unit 107. This pulsating current approaches zero when the current control gain (PI control gain) used in the PI control in the current control unit 102 is increased, so that it becomes impossible to calculate the rotation angle initial value θest_init and the rotation angular velocity initial value ωest_init unless the current control gain in the initial value calculation period is appropriately set.
Therefore, in this embodiment, the current control unit 102 uses different current control gain values in the initial value calculation period and the normal control period.

FIG. 9 is a diagram illustrating an example of the configuration of a current control unit of the inverter control device according to the first embodiment.
In addition, since the configuration for calculating the d-axis voltage command V _{d_ACR} and the configuration for calculating the q-axis voltage command V _{q_ACR} are similar, both will be described here using the same diagram.

The current control unit 102 includes a subtraction unit 21, a gain change unit 22, a PI control unit 23, and an addition unit 24. The PI control unit 23 includes a proportional gain multiplication unit 231, an integral gain multiplication unit 232, an integration unit 233, and an addition unit 234.
A subtraction unit 21 outputs the difference obtained by subtracting the dc-axis current value _Idc from the d-axis current command _Idref , and the difference obtained by subtracting the qc-axis current value _Iqc from the q-axis current command _Iqref .

The gain change unit 22 changes the output value according to the value of the first flag flg1. For example, the gain change unit 22 outputs the current control cutoff frequency _ωACR1 when the value of the first flag flg1 is "1" (initial value calculation period), and outputs the current control cutoff frequency _ωACR2 (> _ωACR1 ) when the value of the first flag flg2 is "0" (normal control period).

The proportional gain multiplication unit 231 outputs the product of multiplying the value output from the subtraction unit 21 by the proportional gain Kp. The proportional gain Kp is set in accordance with the value output from the gain change unit 22.
The integral gain multiplication unit 232 outputs the product of multiplying the value output from the subtraction unit 21 by the integral gain Ki. The integral gain Ki is set in accordance with the value output from the gain change unit 22.
The integrator 233 integrates the value output from the integral gain multiplier 232 and outputs the result.
The adder 234 calculates and outputs the sum of the output value of the proportional gain multiplier 231 and the output value of the integrator 233 .

The proportional gain Kp and integral gain Ki used in the current control unit 102 will be described below.
When the estimation error Δθ is small, the speed estimate ωest approximately matches the motor angular velocity ω, the d-axis current command value _Idref and the q-axis current command value _Iqref match the actual values id and iq, and the parameter set values Ld_set and Lq_set match their true values, then by subtracting equation (14) from equation (1), the voltage equation of the controlled motor M can be replaced with a simple model.

Here, by replacing the differential operator p with the Laplace operator, we obtain the following:
According to equation (19), it can be seen that the current is a product of the voltage input multiplied by a first-order lag system of the motor parameter (motor resistance) R and the inductance L.

FIG. 10 is a diagram for explaining an example of a transfer function when the inverter control device of the first embodiment performs PI control on a current.
For example, when V _d ' and V _q ' are considered as PI controlled output voltages VdPI and VqPI, and the current is PI controlled as shown in FIG. 11, the closed loop transfer function is as follows:
where i: output current i _ref : current command R: motor resistance L: inductance τ: motor time constant (L/R) K _p , K _i : PI control gain

In this case, if a gain is set so that Kp/Ki=τ, it becomes possible to adjust the closed loop transfer function to a first-order lag system with an arbitrary time constant, as shown in the following equation (21).
τ _ACR : Current control time constant ω _ACR : Current control cutoff frequency

At this time, K _p and K _i are determined by comparing coefficients as follows:
In the calculation of _Kp and _Ki using the above formulas (22B) and (23C), the d-axis inductance Ld or the q-axis inductance Lq may be used as the inductance L in accordance with the current to be controlled. Also, the current control cutoff frequency _ωACR is the current control cutoff frequency _ωACR1 , _ωACR2 input from the gain change unit 22.

FIG. 11 is a diagram illustrating an example of timing for changing the current control gain in the current control unit of the inverter control device according to the first embodiment.
In the inverter control device of this embodiment, as described above, the current control gain is set to be lowered during the initial estimation period (the period during which the value of the first flag flg1 is "1"), but in that case, there is a possibility that the current cannot be controlled with high accuracy after the transition to the normal control period. In this case, for example, as shown in Fig. 12, after the initial estimation (calculation of the initial value) is completed (for example, one sampling period after the value of the first flag flg1 changes from "1" to "0"), the output value of the gain change unit 22 may be increased from the current control cutoff frequency _ωACR1 to the current control cutoff frequency _ωACR2 to increase the current control gain.

The cutoff frequency of the current control may be set based on a threshold value ωsh of the switching frequency between low-speed sensorless control and high-speed sensorless control. Since the current pulsation used in the initial value calculation occurs at twice the fundamental wave, for example, if the cutoff frequency _ωACR1 is set to ωsh/2, it can be determined that the speed is low when it is equal to or lower than the threshold value ωsh, and the low-speed sensorless control mode can be selected.

When the value of the threshold value ωsh is variable by the user, the current control cutoff frequency _ωACR1 may be determined based on the rated rotation speed. When the controlled object is a general motor, for example, the threshold value ωsh is often set to about 20% of the rated frequency, and the current control cutoff frequency _ωACR1 may be determined based on the relationship between the rated frequency and the threshold value ωsh.

Next, the operation of the inverter control device 100 of the first embodiment will be described.
FIG. 12 is a flowchart for explaining an example of the operation of the inverter control device of the first embodiment.
When the inverter control device 100 receives a restart command from a higher-level control device, the inverter control device 100 starts current control (step S1).

That is, the flag generation unit 108 sets the value of the first flag Flg1 to "1", the value of the second flag Flg2 to "0", the value of the third flag Flg3 to "0", and the value of the initialization flag Flg_init to "0".

When the value of the first flag Flg1 becomes "1" (step S2), the angle/speed initial value calculation unit 107 calculates the initial value θest of the rotational phase angle and the initial value ωest of the rotational angular velocity (step S3).

After a predetermined period of time has elapsed since the value of the first flag Flg1 was set to "1," the flag generation unit 108 sets the value of the first flag Flg1 to "0," sets the value of the second flag Flg2 to "1," and sets the value of the initialization flag Flg_init to "1." After one further sample period has elapsed, the flag generation unit 108 sets the value of the initialization flag Flg_init to "0."

In the inverter control device 100 of this embodiment, for example, when the value of the first flag Flg1 changes from "1" to "0", the value of the current control gain in the current control unit 102 is switched to be larger.

When the value of the second flag Flg2 changes from "0" to "1" (step S4), the value of the initialization flag Flg_init changes from "0" to "1", and the angle/speed initial value calculation unit 107 saves the initial value θest_init of the rotational phase angle and the initial value ωest_init of the rotational angular velocity, and outputs the initial value θest_init of the rotational phase angle and the initial value ωest_init of the rotational angular velocity to the rotational angle/speed calculation unit 106 (step S5).

When the value of the initialization flag Flg_init changes from "0" to "1", the rotation angle/speed calculation unit 106 initializes the initial value θest_init of the rotation phase angle and the initial value ωest_init of the rotation angular velocity supplied from the angle/speed initial value calculation unit 107 as the initial values of the PLL units 64, 66 and the integration unit 67 (step S6).

Then, the inverter control device 100 transitions from the initial value calculation period to the normal control period. The flag generation unit 108 sets the value of the initialization flag Flg_init to "0" and switches the third flag Flg3 according to the estimated value ωest of the rotational angular speed of the synchronous machine M. The flag generation unit 108 sets the value of the third flag Flg3 to "1" when the estimated value ωest of the rotational angular speed of the synchronous machine M is higher than a predetermined threshold, and sets the value of the third flag Flg3 to "0" when the estimated value ωest of the rotational angular speed of the synchronous machine M is lower than the predetermined threshold.

When the value of the third flag Flg3 is "1" (step S7), the rotational angle/speed calculation unit 106 outputs the rotational phase angle estimate θest and the rotational angular speed estimate ωest calculated using the phase angle error Δθ output from the second position error calculation unit 65 (step S8).

When the value of the third flag Flg3 is "0" (step S7), the rotational angle/speed calculation unit 106 outputs the rotational phase angle estimate θest and the rotational angular speed estimate ωest calculated using the phase angle error Δθ output from the first position error calculation unit 63 (step S9).

During the normal control period, the rotation angle/speed calculation unit 106 switches between steps S8 and S9 and controls the inverter main circuit INV depending on the judgment made in step S7.

FIG. 13 is a diagram for explaining an example of an effect of the inverter control device according to the first embodiment.
Here, the graph shows the time changes of the dq-axis voltage command values _Vdc , _Vqc , the phase currents (output currents of the inverter main circuit INV) _iu , iv, iw, the corrected phase currents iu_2, iv_2, iw_2, the actual angular velocity value ωact, the estimated angular velocity value ωest, the actual angle value θact, and the estimated angle value θest when the inverter INV is restarted with the q-axis current command set to zero and the d-axis current command set to Idref (DC current). The corrected phase currents iu_2, iv_2, iw_2 are waveforms of only the pulsating components excluding the DC components of the phase currents iu, iv, iw.

In this example, the control gain in the current control is set to a value smaller during the initial value calculation period than during the normal control period, and is set so as not to suppress the pulsation of the phase currents iu, iv, and iw while controlling the DC component of the current. Note that during the normal control period, low-speed sensorless control and high-speed sensorless control are switched, but the current control gain is set so as not to completely cut off current pulsation higher than the motor rotation speed corresponding to the switching frequency.

When flag Flg1 becomes "1", the calculation of the angular velocity estimate ωest and the phase angle estimate θest begins using the corrected phase currents iu_2, iv_2, and iw_2 (pulsating currents), and the angular velocity estimate ωest and the phase angle estimate θest are calculated so as to follow the actual values ωact and θact, respectively.

When a predetermined time has elapsed since the calculation of the angular velocity estimate ωest and the phase angle estimate θest started, the angular velocity estimate ωest and the phase angle estimate θest converge to the actual values ωact and θact. After that, the angular velocity estimate ωest and the phase angle estimate θest at the timing when the initialization flag Flg_init becomes "1" are initialized as the rotational angular velocity initial value ωest_init and the rotational phase angle initial value θest_init.

As described above, according to the inverter control device and power conversion device of this embodiment, the pulsating components of the phase current caused by magnetic salient poles can be extracted, thereby making it possible to accurately estimate the initial values of the rotational phase angle and the rotational angular velocity of the synchronous machine M.
That is, according to the present embodiment, it is possible to provide an inverter control device and a power conversion device that are capable of estimating the rotational phase angle of a rotating synchronous machine with high accuracy.

FIG. 14 is a diagram illustrating an example of another configuration of the angle/speed initial value calculation unit of the inverter control device according to the first embodiment. In FIG.
In the inverter control device 100 of this embodiment, the angle/speed initial value calculation unit 107 is not limited to the configuration shown in Fig. 7. For example, as shown in Fig. 14, it may be configured to include a multiplication unit 7M in the subsequent stage of an integration unit 79. With this configuration, it is possible to omit the integration unit 78 in Fig. 7.

Although several embodiments of the present invention have been described, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be embodied in various other forms, and various omissions, substitutions, and modifications can be made without departing from the gist of the invention. These embodiments and their modifications are included in the scope and gist of the invention, and are included in the scope of the invention and its equivalents described in the claims.

The program according to this embodiment may be transferred in a state where it is stored in an electronic device, or in a state where it is not stored in an electronic device. In the latter case, the program may be transferred via a network, or in a state where it is stored in a storage medium. The storage medium is a non-transitory tangible medium. The storage medium is a computer-readable medium. The storage medium may be in any form, such as a CD-ROM or a memory card, as long as it is capable of storing a program and is computer-readable.

M... synchronous machine, 10... stator, 20... rotor, INV... inverter main circuit, 100... inverter control device, 101... current command generation unit, 102... current control unit, 103, 105... coordinate conversion unit, 104... modulation unit, 106... rotation angle/speed calculation unit, 107... angle/speed initial value calculation unit, 108... flag generation unit, 109... high frequency voltage superposition unit, 110U, 110V, 110W... current detector, 70... hold unit, 71... delay unit, 72... coordinate conversion unit, 73... normalization unit, 74, 77... division unit, 75... subtraction unit, 76... PI control unit, 78, 79... integration unit, 111d, 111q... resistors.

Claims (6)

a current command generating unit that generates a current command;
a current detection unit that detects a value of a current flowing between the inverter main circuit and the synchronous machine;
a current control unit that calculates a voltage command corresponding to the current command;
a modulation unit that generates a drive signal for the inverter main circuit based on the value of the voltage command;
a coordinate conversion unit that converts the voltage command value and the detected current value into coordinates by using an estimated value of a rotational phase angle of the synchronous machine;
an initial value calculation unit that calculates initial values of a rotational phase angle and a rotational angular velocity of the synchronous machine based on a pulsation component of an output current of the synchronous machine generated due to magnetic salient pole of the synchronous machine when the estimated value of the rotational phase angle of the synchronous machine used in the coordinate transformation is set to a value that does not change;
an angle/speed calculation unit that calculates the estimated values of a rotational phase angle and a rotational angular speed of the synchronous machine using the initial values calculated by the initial value calculation unit. The inverter control device according to claim 1, wherein the current control unit includes a PI control unit that calculates the voltage command, and a gain change unit that sets a gain used by the PI control unit during the period in which the initial values of the rotation phase angle and the rotation angular velocity are calculated to a value that causes the DC component of the output current to match the current command value and does not suppress the pulsating component. The inverter control device according to claim 2, wherein the gain change unit increases the value of the gain after the calculation of the initial value is completed. The inverter control device according to claim 1, characterized in that the current command during the period in which the initial value is calculated in the initial value calculation unit is set so as to pass a current in the axial direction in which the magnetic salient pole of the synchronous machine is minimized. The inverter control device according to claim 1, characterized in that the current command value is a value that saturates the rotor bridge of the synchronous machine. An inverter main circuit;
a current command generating unit that generates a current command;
a current detection unit that detects a value of a current flowing between the inverter main circuit and a synchronous machine;
a current control unit that calculates a voltage command corresponding to the current command;
a modulation unit that generates a drive signal for the inverter main circuit based on the value of the voltage command;
a coordinate conversion unit that converts the voltage command value and the detected current value into coordinates by using an estimated value of a rotational phase angle of the synchronous machine;
an initial value calculation unit that calculates initial values of a rotational phase angle and a rotational angular velocity of the synchronous machine based on a pulsation component of an output current of the synchronous machine generated due to magnetic salient pole of the synchronous machine when the estimated value of the rotational phase angle of the synchronous machine used in the coordinate transformation is set to a value that does not change;
an angle/speed calculation unit that calculates the estimated values of a rotational phase angle and a rotational angular velocity of the synchronous machine using the initial values calculated by the initial value calculation unit.