WO2007059484A2 - Linear rf transmitter and method of operation - Google Patents

Linear rf transmitter and method of operation Download PDF

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Publication number
WO2007059484A2
WO2007059484A2 PCT/US2006/060866 US2006060866W WO2007059484A2 WO 2007059484 A2 WO2007059484 A2 WO 2007059484A2 US 2006060866 W US2006060866 W US 2006060866W WO 2007059484 A2 WO2007059484 A2 WO 2007059484A2
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WO
WIPO (PCT)
Prior art keywords
signal
channel
supply voltage
operable
linearity
Prior art date
Application number
PCT/US2006/060866
Other languages
French (fr)
Other versions
WO2007059484A3 (en
Inventor
Moshe Ben-Ayun
Ovadia Grossman
Shay Nir
Mark Rozental
Original Assignee
Motorola, Inc.
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Publication date
Application filed by Motorola, Inc. filed Critical Motorola, Inc.
Publication of WO2007059484A2 publication Critical patent/WO2007059484A2/en
Publication of WO2007059484A3 publication Critical patent/WO2007059484A3/en

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Classifications

    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
    • H03F1/0216—Continuous control
    • H03F1/0222—Continuous control by using a signal derived from the input signal
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H03F1/0211—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the supply voltage or current
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03C—MODULATION
    • H03C5/00—Amplitude modulation and angle modulation produced simultaneously or at will by the same modulating signal
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
    • H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
    • H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
    • H03F1/3247—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits using feedback acting on predistortion circuits
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
    • H03F1/32—Modifications of amplifiers to reduce non-linear distortion
    • H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
    • H03F1/3294—Acting on the real and imaginary components of the input signal
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
    • H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
    • H03F3/24—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
    • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
    • H04L27/36—Modulator circuits; Transmitter circuits
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/207—A hybrid coupler being used as power measuring circuit at the output of an amplifier circuit
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/321—Use of a microprocessor in an amplifier circuit or its control circuit
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/336—A I/Q, i.e. phase quadrature, modulator or demodulator being used in an amplifying circuit
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03F—AMPLIFIERS
    • H03F2200/00—Indexing scheme relating to amplifiers
    • H03F2200/504—Indexing scheme relating to amplifiers the supply voltage or current being continuously controlled by a controlling signal, e.g. the controlling signal of a transistor implemented as variable resistor in a supply path for, an IC-block showed amplifier

Definitions

  • the present invention relates to a linear RF (radio frequency) transmitter and a method of operation of the transmitter.
  • the invention relates to a linear RF transmitter including a non-linearity detector and a control loop to change an operating condition of the transmitter when the detector detects non-linearity.
  • RF communication terminals normally employ an RF transmitter to generate RF signals and a receiver to receive RF signals. Such terminals also normally employ an antenna to send signals produced by the transmitter over the air to another terminal and to receive signals sent over the air from another terminal for delivery to the receiver.
  • the transmitter normally includes an RF power amplifier to amplify the RF signals before they are coupled to the antenna for transmission.
  • the transmitter is desirable for the transmitter to be linear, i.e. for the RF power amplifier to produce a linear power amplification of the input signal provided to it, in order to prevent distortion of the input signal and to minimize inter-channel interference.
  • RF power amplifiers when RF power amplifiers are operated at high drive levels they may become non-linear. Similar non-linearity may be caused by other operating conditions.
  • an RF power amplifier may be susceptible to an increased antenna impedance that will cause RF energy to be reflected back from the antenna into the transmitter. Changes in the antenna impedance are indicated by a parameter known as VSWR (voltage standing wave ratio) .
  • an isolator In order to protect an RF power amplifier against changes in load impedance resulting from changes of antenna impedance or VSWR, an isolator is often inserted between the antenna and the power amplifier.
  • the isolator protects the power amplifier by absorbing the reflected energy and preventing it from reaching the power amplifier.
  • the isolator normally directs the reflected energy to an absorptive load.
  • an isolator generally works well, it adds significant cost, size, and weight to the design and construction of the communication terminal.
  • Linear RF transmitters which do not include an isolator are known in the art.
  • An example is described in Applicant's GB2403086.
  • the transmitter includes a Cartesian loop to provide linearization of the RF power amplifier of the transmitter.
  • the transmitter also includes an isolator eliminator which operates to detect non-linearity as indicated when excessive adjacent channel power is present. When non-linearity is detected the isolator eliminator applies an increase in an attenuation to an input baseband signal to reduce the signal level and thereby reduce the output signal power to restore linearity of the power amplifier.
  • the present invention seeks in one aspect to further improve the transmitter described in GB2403086, particularly by increasing the power of the output RF signal provided by the RF power amplifier of the transmitter.
  • a linear RF transmitter including a forward path for receiving and processing a baseband input signal, the forward path including an upconverting mixer operable to convert a baseband signal into a modulated RF signal and a RF (radio frequency) power amplifier operable to amplify a modulated RF signal produced by the upconverting mixer, a non-linearity detector connected to the forward path to sample a baseband signal for delivery to the upconverting mixer and operable to detect non-linearity of the transmitter and to provide in response a control signal to change operation of the transmitter to reduce the non-linearity, and characterised in that the transmitter further includes a regulator connectible to a supply voltage source and to the RF power amplifier to deliver a variable supply voltage to the RF power amplifier and a supply voltage controller operable to produce a control signal which controls adjustment by the regulator of the supply voltage, wherein the non-linearity detector is connected to the supply voltage controller and is operable upon detecting non-linearity to deliver a control
  • the forward path may include an I channel for receiving and processing an I (in phase) component of an input baseband signal and a Q channel for receiving and processing a Q (quadrature phase) component of an input baseband signal, and each of the I channel and the Q channel may include an upconverting mixer.
  • the non-linearity detector is connected to the I channel and to the Q channel to sample I and Q components of the baseband signal for delivery to the respective mixers.
  • the transmitter according to the first aspect of the invention may include an antenna to radiate output signals from the RF power amplifier with no isolator between the RF power amplifier and an antenna.
  • the non-linearity detector may thus comprise an isolator eliminator employed to detect non-linearity arising from a changed load impedance at the RF power amplifier caused by a change in antenna VSWR.
  • the non-linearity detector may be operable to detect, when non-linearity is detected by the non- linearity detector, whether a pre-determined maximum supply voltage is currently being delivered to the RF power amplifier.
  • the non-linearity detector may be connected to the supply voltage controller and may be operable to provide to the supply voltage controller, in response to detection of non-linearity and that a predetermined maximum supply voltage is not currently being delivered to the RF power amplifier, a control signal to the supply voltage controller which causes the supply voltage controller to adjust the regulator to increase the supply voltage to the RF power amplifier.
  • the non- linearity detector may be operable to produce a control signal indicating a value of a parameter to be employed by the supply voltage controller which may be increased in steps until a maximum value is reached, the maximum value corresponding to a predetermined maximum supply voltage .
  • the forward path may include an attenuator.
  • the forward path may include an I channel and a Q channel the forward path may include an attenuator in each of the I channel and the Q channel.
  • the attenuator or attenuators are operable to attenuate a baseband signal or a component thereof in the forward path.
  • the non-linearity detector may be connected to the attenuator or attenuators to deliver a control signal to control a level of attenuation applied by the attenuator or attenuators.
  • the non-linearity detector may be operable to apply a control signal to the attenuator or attenuators in response to detecting that a predetermined maximum supply voltage is currently being delivered to the RF power amplifier.
  • the transmitter according to the first aspect of the invention may include a supply voltage control loop connected to the forward channel to sample a baseband signal or components thereof delivered to the forward channel.
  • the supply voltage control loop may include a calculator operable to calculate a varying amplitude of the sampled baseband signal.
  • the supply voltage control loop may also include the supply voltage controller referred to earlier.
  • the supply voltage controller may be operable to control adjustment by the regulator of the supply voltage in a modulation which follows the varying amplitude of the sampled baseband signal.
  • the supply voltage controller may operate to multiply by a multiplication constant indicated by an output signal produced by the non-linearity detector an input provided by the calculator.
  • the supply voltage control loop including the supply voltage controller thereby forms an envelope follower in which the modulation applied by the voltage regulator follows the variation of the baseband signal amplitude, unless the supply voltage controller is instructed by the non-linearity detector to produce an output signal to provide an increase in supply voltage by the voltage regulator until a maximum supply voltage is reached.
  • a non-linearity detector e.g. to serve as an isolator eliminator
  • a supply voltage control loop e.g. to serve as an isolator eliminator
  • the calculator of the supply voltage control loop may include (i) an I 2 + Q 2 calculator which squares and adds samples of I and Q components of the baseband signal delivered to the I and Q channels and (ii) a square root calculator which calculates a square root of the calculation result x produced by the I 2 + Q 2 calculator.
  • the square root calculator may beneficially comprise an approximator which calculates an approximate value of the square root of the calculation result x produced by the I 2 + Q 2 calculator.
  • Such an approximator beneficially produces a signal which is narrow band compared with an accurate square root calculator which produces a signal with much larger bandwidth.
  • the calculator and the supply voltage controller of the supply voltage control loop may conveniently be incorporated in a single processor.
  • the single processor may be operable to employ a control signal delivered from the non-linearity detector to multiply, by a multiplication constant, the constants A 2 , Ai and Ao applied by the approximator so that an output control signal produced by the supply voltage controller produces a desired incremental increase in supply voltage by the voltage regulator.
  • the transmitter according to the first aspect of the invention may comprise a Cartesian loop linear transmitter.
  • the transmitter may include a coupler, e.g. a directional coupler, to sample an output of the RF power amplifier and a feedback path extending from the coupler to the forward path at a feedback path junction prior to the upconverting mixer, the feedback path including a downconverting mixer.
  • the forward path includes an I channel and a Q channel
  • the feedback path may include an I channel feedback loop including a first downconverting mixer and a Q channel feedback loop including a second downconverting mixer.
  • the I channel feedback loop may be connected to the I channel of the forward path at a first feedback path junction and the Q channel feedback loop may be connected to the Q channel of the forward path at a second feedback path junction.
  • the feedback path junction or each of the feedback path junctions may comprise a differential amplifier or a summing junction.
  • the non-linearity detector may be connected to the respective I and Q channels of the forward path at detector junctions which are located between the feedback path junctions and the upconverting mixers in the respective I and Q channels.
  • Each of the I channel and the Q channel may include a low pass filter and optionally an amplifier between the feedback path junction of the channel and the detector junction of the channel.
  • the non-linearity detector of the transmitter may include a first RMS estimator operable to estimate a root mean square signal level of an on-channel signal and a second RMS estimator operable to estimate a root mean square value of a noise signal at a predetermined frequency offset from a frequency of the on-channel signal, a divider operable to produce an output signal which represents a root mean square value estimated by the first RMS estimator divided by a root mean square value estimated by the second RMS estimator and a comparator operable to compare an output signal produced by the divider with a threshold signal.
  • the comparator may produce an output signal having one of two possible state values representing respectively non-linearity detected or not detected.
  • the transmitter according to the first aspect of the invention may be operable in accordance with TETRA or other standard operating protocols.
  • a method of operation in a linear RF transmitter which includes receiving a baseband input signal in a forward path, converting the baseband signal into a modulated RF signal in an upconverting mixer, and amplifying the modulated RF signal by a RF power amplifier, sampling the baseband signal for delivery to the upconverting mixer by a non-linearity detector connected to the forward path, detecting by the detector whether non-linearity is present in operation of the transmitter and providing in response to detected non-linearity a control signal to change operation of the transmitter to reduce the non-linearity, and characterised by delivering a supply voltage from a voltage source to the RF power amplifier, producing a modulation of the supply voltage by a voltage regulator and producing by a supply voltage controller a control signal which controls modulation by the regulator of the supply voltage, wherein the non-linearity detector is connected to the supply voltage controller and upon detecting non-linearity delivers a control signal to the supply voltage controller which causes the supply voltage controller to
  • the non-linearity detector may issue a signal to the voltage supply controller causing the supply voltage, as regulated by the regulator, to the RF power amplifier to be increased. This will make the RF power amplifier more linear and will allow the RF power amplifier to increase its nominal output power. If, but preferably only if, the supply voltage is increased to a maximum allowed value and the RF power amplifier is still not linear, e.g.
  • the non-linearity detector may issue a control signal to an attenuator in the forward path, or attenuators in each of an I channel and a Q channel of the forward path, to increase an attenuation applied by the attenuator or attenuators to a baseband signal or a component thereof in the forward path to reduce an output signal power.
  • FIG. 1 is a block schematic diagram of a Cartesian loop transmitter embodying the invention.
  • FIG. 2 is a block schematic diagram showing more detail of part of the transmitter of FIG. 1.
  • FIG. 3 is a flow chart of a method of operation embodying the invention in the transmitter of FIG. 1.
  • FIG. 1 is a block schematic diagram of a Cartesian loop linear transmitter 100 embodying the present invention.
  • the transmitter 100 includes (inter alia) a forward path 102 and a feedback path 104.
  • a DSP (digital signal processor) 101 generates a baseband input digital signal containing information to provide a modulation signal to be transmitted by the transmitter 100 by RF communication.
  • the input baseband digital signal comprises an I (in phase) signal component which is delivered via an I channel 105 and a corresponding Q signal component which is delivered via a Q channel 107.
  • the I signal component and the Q signal component are converted to analog form by a D/A (digital to analog) converter 103.
  • the D/A converter 103 delivers an I signal in analog form along the I channel 105 of the forward path 102 and a Q signal in analog form along the Q channel 107 of the forward path 102 in parallel with the I channel 105.
  • the I channel 105 includes, connected to the D/A converter 103 and connected together in turn, an I channel attenuator 108, a differential amplifier 112, an amplifier/ filter 116 and an upconverting mixer 120.
  • the upconverting mixer 120 is also connected to a local oscillator (carrier frequency synthesizer) 136.
  • the Q channel 107 includes, connected to the D/A converter 103 and connected together in turn, a Q channel attenuator 110, a differential amplifier 114, an amplifier/ filter 118 and an upconverting mixer 122.
  • the upconverting mixer 122 is also connected to the local oscillator 136 via a 90 degrees phase shifter 158. Output connections from the upconverting mixers 120 and 122 provide inputs to a summing junction 124 having an output connected in turn to a RFPA (radio frequency power amplifier) 126 and an antenna 128.
  • a summing junction 124 having an output connected in turn to a RFPA (radio frequency power amplifier) 126 and an antenna 128.
  • RFPA radio frequency power amplifier
  • an I signal produced digitally by the DSP 101 and converted to analog form by the D/A converter 103 is delivered to the I channel attenuator 108 and is attenuated by the I channel attenuator 108.
  • the level of attenuation by the I channel attenuator 108 is controlled in a manner to be described later.
  • the attenuated I signal is then delivered to the differential amplifier 112.
  • An error control signal produced in a manner to be described from a downconverting mixer 132 is subtracted from the I signal in the differential amplifier 112.
  • An output corrected I signal produced by the differential amplifier 112 is then amplified and filtered by the amplifier/ filter 116.
  • the amplifier/filter 116 comprises a low pass filter which serves as a slew rate limiter.
  • the amplified and filtered I signal produced by the amplifier/filter 116 is then mixed with a carrier frequency signal from the local oscillator 136 to upconvert the I signal from baseband to RF (radio frequency) .
  • a Q signal produced digitally by the DSP 101 and converted to analog form by the D/A converter 103 is delivered to the Q channel attenuator 110 and is attenuated by the Q attenuator 110 which is controlled in a manner to be described later.
  • the attenuated Q signal is then delivered to the differential amplifier 114.
  • An error control signal produced in a manner to be described later from a downconverting mixer 134 is subtracted from the Q signal in the differential amplifier 112.
  • An output corrected Q signal produced by the differential amplifier 114 is then amplified and filtered by the amplifier/ filter 118.
  • the amplifier/filter 118 comprises a low pass filter which serves as a slew rate limiter.
  • the amplified and filtered Q signal produced by the amplifier/filter 118 is then mixed by the upconverting mixer 122 with a carrier frequency signal (whose phase has been shifted by ninety degrees compared with the carrier frequency signal delivered to the upconverting mixer 120) delivered from the local oscillator 136 via the phase shifter 158 to upconvert the Q signal from baseband to RF (radio frequency) .
  • a carrier frequency signal (whose phase has been shifted by ninety degrees compared with the carrier frequency signal delivered to the upconverting mixer 120) delivered from the local oscillator 136 via the phase shifter 158 to upconvert the Q signal from baseband to RF (radio frequency) .
  • the RF signals produced as outputs by the upconverting mixer 120 of the I channel 105 and the upconverting mixer 122 of the Q channel 107 are combined at the summing junction 124, and the combined RF signal is amplified by the RFPA 126 to produce an amplified RF output signal.
  • the amplified RF output signal produced by the RFPA 126 is delivered to the antenna 128 and is sent over the air to a distant terminal (not shown) at which it is received.
  • the antenna 128 may (at times when the transmitter 100 is not in operation) also receive an incoming RF signal sent over the air from a distant terminal (not shown) and may deliver the received signal for processing to a RF receiver (not shown) .
  • the transmitter 100 includes no isolator between the RFPA 126 and the antenna 128.
  • a directional coupler 130 is connected between the RFPA 126 and the antenna 128 to sample the amplified RF output signal produced by the RFPA 126.
  • the directional coupler 130 is connected to the feedback path 104 at an attenuator 109.
  • the feedback path 104 leading from the attenuator 109 is branched to include an I channel feedback loop 115 and a Q channel feedback loop 117.
  • the I channel feedback loop 115 includes a downconverting mixer 132 connected to the attenuator 109 and also connected to the local oscillator 136.
  • the downconverting mixer 132 is connected to provide an output signal to the differential amplifier 112.
  • the Q channel feedback loop 117 includes a downconverting mixer 134 which is connected to the attenuator 109 and also is connected to the local oscillator 136 via a 90 degrees phase shifter 160.
  • the downconverting mixer 134 is connected to provide an output signal to the differential amplifier 114.
  • the directional coupler 130 supplies the I channel feedback loop 115 and the Q channel feedback loop 117 with a feedback signal representing the amplified output signal.
  • the feedback signal is attenuated by the attenuator 109.
  • the attenuated feedback signal is downconverted into I and Q feedback baseband components by the mixers 132 and 134.
  • the resultant baseband components are error control signals which are subtracted respectively in the differential amplifier 112 and the differential amplifier 114 from the I signal and the Q signal delivered to the differential amplifiers 112 and 114 respectively by the attenuators 108 and 110.
  • the feedback loops 115 and 117 provide a known mechanism for maintaining linear operation of the transmitter 100 by forcing the transmitter 100 to produce an RF output which follows the undistorted I signal and Q signal supplied respectively to the differential amplifiers 112 and 114.
  • the differential amplifiers 112 and 114 may be replaced by summing junctions, and an amplifier and low pass filter may be included in each of the loops 115 and 117 after the downconverting mixers 132 and 134. Such amplifiers and filters may be included instead of or in addition to the amplifier/ filters 116 and 118.
  • the transmitter 100 includes a supply modulation control loop 133 to improve efficiency of operation of the RFPA 126.
  • the loop 133 includes an I 2 + Q 2 calculator 119, a square root approximator 121, a supply voltage controller 127, a D/A converter 129 and a voltage regulator 111.
  • the calculator 119 is connected to the I channel 105 and to the Q channel 107 between the DSP 101 and the D/A converter 103.
  • the calculator 119 thereby samples the output baseband I and Q signals produced in digital form by the DSP 101.
  • the square root approximator 121 comprises a digital processor which calculates an approximation of the square root of x for each value of x provided by the calculator 119.
  • the square root approximator 121 uses a relationship as follows to calculate an approximation of the square root of each value of x:
  • F(x) A 2 X 2 + A 1 X + A 0 (Equation 2) where A 2 , Ai and Ao are pre-determined constants.
  • the approximator 121 is a preferred band limited form of calculator to calculate the square root of x. A more exact square root calculator could be used, but would result a greater signal bandwidth.
  • a narrower band calculator is preferred since for spectral efficiency it is desirable for the spectrum of the transmitted signal to be narrow band and thereby meet adjacent channel emission specifications defined in industry standard operating protocols.
  • the result of the calculation by the square root approximator 121 is provided as an output digital signal to the supply voltage controller 127.
  • the supply voltage controller 127 also receives an input digital control signal from a non-linearity detector 125. Such a signal is the output of an algorithm as described in more detail later with reference to FIGS . 2 and 3. When such an input signal is received by the supply voltage controller 127 it causes the controller 127 to multiply an output of the approximator 119 by a multiplication constant.
  • the supply voltage controller 127 produces in accordance with Equation 3 an output digital control signal, e.g. in the form of a digital control word, which is delivered to a D/A (digital to analog) converter 129 which converts the digital control signal into an analog control signal.
  • the analog control signal produced by the D/A converter 129 is delivered as an analog control signal to the voltage regulator 111.
  • the regulator 111 receives an input DC voltage from a voltage source 113 such as a battery.
  • the analog control signal received by the regulator 111 from the D/A converter 129 causes the regulator 111 to modulate the voltage from the voltage source 113 in a known manner to produce a modulated supply voltage V 8 which is applied to the RFPA 126.
  • the modulation follows the variation of the digital control signal produced by the supply voltage indicator 127.
  • An instantaneous increase in the supply voltage V 3 produced by the modulation causes an increase in the output power of the RFPA 126 and a reduction in the supply voltage causes a fall in the output power of the RFPA 126.
  • the control loop 133 improves efficiency of the RFPA by varying the supply voltage to suit the detected power level of the input signal.
  • the transmitter 100 includes the non-linearity detector 125 to serve as an isolator eliminator, which is included since the transmitter 100 includes no isolator between the RFPA 126 and the antenna 128.
  • the non- linearity detector 125 is connected to the I channel 105 between the amplifier/ filter 116 and the upconverter mixer 120 to sample the baseband filtered output I signal produced by the amplifier /filter 116, and is connected to the Q channel 107 between the amplifier/ filter 118 and the upconverter mixer 122 to sample the baseband filtered output Q signal produced by the amplifier /filter 118.
  • the detector 125 is also connected to the I channel attenuator 108 and the Q channel attenuator 110 to control operation of the attenuators 108 and 110 in a manner to be further described with reference to FIG. 2.
  • the non-linearity detector 125 is also connected to the supply voltage controller 127 to provide a control signal to the supply voltage controller 127 indicating a value of a multiplication constant ⁇ a' as in Equation 3.
  • the value of the multiplication constant is selected in a manner to be described later with reference to FIG. 3.
  • FIG. 2 shows in more detail an example of the non- linearity detector 125.
  • This example is based upon the isolator eliminator of the form described in GB 2403086 but is also adapted in accordance with an embodiment of the invention.
  • the non-linearity detector 125 is connected to an output of the amplifier/filter 116 in the I channel 105 and an output of the amplifier/filter 118 in the Q channel 107.
  • the non-linearity detector 125 thereby samples in baseband form the I signal and the Q signal in the channels 105 and 107 before these signals reach the upconverting mixers 120 and 122.
  • the non-linearity detector 125 detects whether there is non-linearity in operation of the RFPA 126.
  • the non- linearity detector 125 continuously samples from the information provided by the baseband I signal and the baseband Q signal corrected by the error control signals from the feedback loops 115 and 117 an on-channel baseband signal level as well as a noise (off channel signal) level at a predefined frequency offset in relation to a known desired transmission channel frequency.
  • An example of the frequency offset at which the noise level is measured is (+ or -) 13.5 kHz.
  • the sampled baseband I signal and the sampled baseband Q signal are passed through a low pass filter 138 and a low pass filter 142 respectively to obtain the on-channel signal components.
  • the sampled baseband I signal and the sampled baseband Q signal are passed through a band pass filter 140 and a band pass filter 144, in parallel respectively with the low pass filter 138 and the low pass filter 142, to obtain the noise (or off-channel signal) components.
  • Filtered output signals from the low pass filter 138 and the low pass filter 142 are delivered to a RMS (root mean square) estimator 146.
  • Filtered output signals from the band pass filter 140 and the band pass filter 144 are delivered to a RMS (root mean square) estimator 148.
  • the RMS estimator 146 comprises a digital processor which calculates the squares respectively of each consecutive incremental sample of the input I signal and the Q signal it receives.
  • the RMS estimator 146 also sums the values of the squares obtained for each sample in a given block of N samples, where N is a predetermined block size containing a given number of samples, e.g. from one to one hundred samples, and then divides the result by N. This gives for the block of N samples an estimate of the mean square value. Finally, the RMS estimator 146 calculates the square root of the mean square value obtained for the block of N samples giving a root mean square ( ⁇ RMS') value. This is an estimate of the current RMS signal level of the baseband signal having as components the I signal and the Q signal. An output signal representing a digital value of the calculated RMS signal level for each consecutive block of samples is passed from the RMS estimator 146 to a divider 150.
  • the RMS estimator 148 operates in a manner similar to the RMS estimator 146 to estimate an RMS value of the noise level (off-channel signal level) .
  • An output signal representing a digital value of the calculated RMS noise level for each consecutive block of samples is passed from the RMS estimator 148 to the divider 150.
  • the divider 150 calculates for each block value provided by the RMS estimator 146 and each corresponding block value provided by the RMS estimator 148 a ratio of the value of the RMS on-channel signal to the value of the RMS noise (off channel signal) level.
  • An output signal providing a signal to noise ratio value for each block considered by the divider 150 is provided by the divider 150 to a comparator 152.
  • the comparator 152 provides an output signal having a LOW state if the determined signal to noise ratio indicated by the output signal from the divider 150 is above a predefined threshold THR.
  • the comparator 152 provides an output signal having a HIGH state if the determined signal to noise ratio indicated by the output signal from the divider 150 is equal to or below a predefined threshold THR. If the comparator 152 produces an output signal having a LOW state it indicates that the transmitter 100 is operating linearly and no change in operation of the transmitter 100 is required. If the comparator 152 produces an output signal having a HIGH state it indicates non-linearity of the RFPA 126 of the transmitter 100 that needs to be corrected for.
  • the comparator 152 is connected to a microprocessor 154 which receives the output signal produced by the comparator 152 and initiates corrective action when the output signal from the comparator 152 has a HIGH state.
  • the microprocessor 154 is connected in turn to the I channel attenuator 108 and the Q channel attenuator 110 and also to the supply voltage controller 127. In some circumstances to be described in detail with reference to FIG. 3, if the output signal from the comparator 152 has a HIGH state the microprocessor 154 provides a control signal to the supply voltage controller 127 to produce a change in output signal from the controller 127 leading to an incremental change by the regulator 111 to a level of supply voltage Vs provided to the RFPA 126.
  • the microprocessor 154 provides to the I channel attenuator 108 and the Q channel attenuator 110 a control signal to make an incremental change to a level of attenuation applied by the I channel attenuator 108 and the Q channel attenuator 110.
  • the microprocessor 154 applies a control signal to the attenuators 108 and 110 it also applies a delay to issue of any further control signal based on a further HIGH state output signal from the comparator 152. This delay is applied to allow the transmitter 100 to settle after a step increase in attenuation by the attenuators 108 and 110 has been applied.
  • the delay is implemented by the microprocessor 154 not reading results of the comparator 152 for pre-defined period of time.
  • the microprocessor 154 stores a value indicating an attenuation setting of the attenuators 108 and 110 in a memory 156 to record how much attenuation is currently being applied.
  • FIG. 3 is a flow chart showing a method 200 embodying the invention used in operation by the transmitter 100.
  • the method 200 is an algorithm which runs in every transmission time slot, e.g. as indicated by a controller (not shown) which controls functional operations of the transmitter 100.
  • the method 200 thus begins at a step 201 which is a start of a transmission time slot.
  • a non-linearity detector algorithm run by the microprocessor 154 begins in a step 203.
  • the microprocessor 154 decides whether or not non-linearity is detected based on the current state of the output signal from the comparator 152.
  • the decision is ⁇ NO' when the signal received by the microprocessor 154 from the comparator 152 has a LOW state indicating linearity of operation of the transmitter 100, and the decision is ⁇ YES' when the signal received by the microprocessor 154 from the comparator 152 has a HIGH state indicating non-linearity of operation of the transmitter 100. If the decision taken by the microprocessor 154 in step 205 is ⁇ NO' the method 200 returns to step 203. If the decision taken by the microprocessor 154 in step 205 is ⁇ YES', i.e. that non-linearity is detected, a decision step 207 follows.
  • decision step 207 the microprocessor 154 decides whether the multiplication constant ⁇ a' (referred to earlier with reference to Equation 3) that is indicated by the microprocessor 154 to the supply voltage controller 127 is at its pre-determined maximum value.
  • the pre-determined maximum value of the multiplication constant ⁇ a' corresponds to a pre-determined maximum value of the supply voltage V s . If the decision taken by the microprocessor 154 in step 207 is ⁇ YES' , i.e.
  • a step 209 follows in which the microprocessor 154 issues a control signal to the attenuators 108 and 110 to increase the attenuation applied by the attenuators 108 and 110 in the manner described earlier. If the decision taken by the microprocessor 154 in step 207 is ⁇ NO' , i.e. the microprocessor 154 determines that a maximum value of the multiplication constant ⁇ a' has not already been reached, a step 211 follows. In step 211 the microprocessor 154 delivers a control signal to the supply voltage controller 127 to indicate to the supply voltage controller 127 that an incremental increase is required to the multiplication constant 'a' .
  • a decision step 213 is applied.
  • the microprocessor 154 decides whether or not non-linearity is detected. This is the same as in step 205 but is based on a later result from the comparator 152. If the decision taken by the microprocessor 154 in step 213 is ⁇ NO' , i.e. that linearity is detected, the method 200 returns to step 203. If the decision taken by the microprocessor 154 in step 213 is ⁇ YES' , i.e. that non-linearity is detected, the method 200 returns to decision step 207.
  • step 207 each time step 207 produces a ⁇ NO' result, the value of multiplication constant ⁇ a' indicated by the microprocessor 154 to the supply voltage controller 127 will be gradually increased in an iteration of step 211 until a maximum value of x a' is reached, when step 209 is alternatively applied instead of step 211.
  • the calculator comprising the square root approximator 121 (and optionally the calculator 119) and the supply voltage controller 127 may be combined in a single processor.
  • the combined single processor may apply an incremental increase to the constants Kz, Ai and Ao applied in accordance with
  • Equation 2 This may be done by multiplying the constants Kz, Ai and Ao by a multiplication constant indicated by an output signal from the microprocessor 154.
  • the multiplication constant may gradually increase for each iteration of step 211. This produces for each incremental increase to the constants A 2 , Ai and A 0 an incremental increase in the supply voltage V s until, in step 207 of the method 200, the microprocessor 154 detects that a maximum value of the multiplication constant has been reached indicating also that a maximum value of V s has been reached. When reaching of the maximum value is detected, step 209 is applied.
  • the microprocessor 154 issues a control signal to the supply voltage controller 127 which, by causing the regulator 111 to provide an increase in the supply voltage Vs, causes the output power capability of the RFPA 126 to be increased.
  • This corrective action is taken if it is still possible, i.e. if a maximum value of V s has not been reached, rather than the microprocessor 154 issuing a signal to increase the attenuation applied by the attenuators 108 and 110 to reduce the non-linearity.
  • Increasing the attenuation in order to reduce non-linearity is only applied when the maximum value of V s has been reached.
  • Increasing the supply voltage V 5 when still possible is more beneficial than simply increasing attenuation by the attenuators 108 and 110 as in the prior art, because increasing the supply voltage V 3 allows greater output power from the RFPA 126.
  • Increasing attenuation by the attenuators 108 and 110 reduces the output power of the transmitter 100.
  • reducing output power undesirably reduces the range of a RF signal produced by the transmitter 100.
  • Changes in the supply voltage V s in the manner described may beneficially be achieved more rapidly than changes which occur in the VSWR as experienced by the RFPA 126. This allows changes in the power level of the RFPA 126 due to changes in the VSWR to be minimized.
  • the tuning of the RFPA 126 may be set initially, e.g. in a factory in which the transmitter 100 is produced and tested, at a level assuming a nominal load termination employed instead of the antenna 128, e.g. a 50 ohm termination. This sets the initial nominal operating conditions under which the algorithm comprising the method 200 is applied, including an initial level of the multiplication constant ⁇ a' indicated by the signal provided by the microprocessor 154 to the supply voltage controller 127.
  • the transmitter 100 is suitable for use in an RF communication terminals for use in a number of communication applications, especially those that operate in narrow bandwidths and demand a high level of linearity. Of particular interest is use in a terminal in which the transmitter operates in accordance with pre-defined industry standard operating protocols or procedures such as the TETRA (Terrestrial Trunked Radio) standard protocols as defined by ETSI (European Telecommunication Standards Institute) .

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Abstract

A linear RF transmitter (100) includes a forward path (102, 105, 107) for receiving and processing a baseband input signal, including an upconverting mixer (120, 122) to convert a baseband signal into a modulated RF signal, and a RF (radio frequency) power amplifier (126) to amplify a modulated RF signal produced by the mixer, a non-linearity detector (125) connected to the forward path to sample a baseband signal for delivery to the mixer and to detect nonhneanty of the transmitter and provide in response a control signal to change operation of the transmitter to reduce the non-linearity. The transmitter further includes a regulator (111) connective to a supply voltage source (113) and to the RF power amplifier to deliver a vaπable supply voltage to the RF power amplifier and a controller (127) to produce a control signal which controls adjustment by the regulator (111) of the supply.

Description

LINEAR RF TRANSMITTER AND METHOD OF OPERATION
FIELD OF THE INVENTION
The present invention relates to a linear RF (radio frequency) transmitter and a method of operation of the transmitter. In particular, the invention relates to a linear RF transmitter including a non-linearity detector and a control loop to change an operating condition of the transmitter when the detector detects non-linearity.
BACKGROUND OF THE INVENTION
RF communication terminals normally employ an RF transmitter to generate RF signals and a receiver to receive RF signals. Such terminals also normally employ an antenna to send signals produced by the transmitter over the air to another terminal and to receive signals sent over the air from another terminal for delivery to the receiver. The transmitter normally includes an RF power amplifier to amplify the RF signals before they are coupled to the antenna for transmission. As modern RF communication systems operate in narrow frequency bands it is desirable for the transmitter to be linear, i.e. for the RF power amplifier to produce a linear power amplification of the input signal provided to it, in order to prevent distortion of the input signal and to minimize inter-channel interference.
However, it is well known that when RF power amplifiers are operated at high drive levels they may become non-linear. Similar non-linearity may be caused by other operating conditions. For example, an RF power amplifier may be susceptible to an increased antenna impedance that will cause RF energy to be reflected back from the antenna into the transmitter. Changes in the antenna impedance are indicated by a parameter known as VSWR (voltage standing wave ratio) .
In order to protect an RF power amplifier against changes in load impedance resulting from changes of antenna impedance or VSWR, an isolator is often inserted between the antenna and the power amplifier. The isolator protects the power amplifier by absorbing the reflected energy and preventing it from reaching the power amplifier. The isolator normally directs the reflected energy to an absorptive load. Although an isolator generally works well, it adds significant cost, size, and weight to the design and construction of the communication terminal.
Linear RF transmitters which do not include an isolator are known in the art. An example is described in Applicant's GB2403086. In that prior specification the transmitter includes a Cartesian loop to provide linearization of the RF power amplifier of the transmitter. The transmitter also includes an isolator eliminator which operates to detect non-linearity as indicated when excessive adjacent channel power is present. When non-linearity is detected the isolator eliminator applies an increase in an attenuation to an input baseband signal to reduce the signal level and thereby reduce the output signal power to restore linearity of the power amplifier. The present invention seeks in one aspect to further improve the transmitter described in GB2403086, particularly by increasing the power of the output RF signal provided by the RF power amplifier of the transmitter.
SUMMARY OF THE INVENTION
According to the present invention in a first aspect there is provided a linear RF transmitter including a forward path for receiving and processing a baseband input signal, the forward path including an upconverting mixer operable to convert a baseband signal into a modulated RF signal and a RF (radio frequency) power amplifier operable to amplify a modulated RF signal produced by the upconverting mixer, a non-linearity detector connected to the forward path to sample a baseband signal for delivery to the upconverting mixer and operable to detect non-linearity of the transmitter and to provide in response a control signal to change operation of the transmitter to reduce the non-linearity, and characterised in that the transmitter further includes a regulator connectible to a supply voltage source and to the RF power amplifier to deliver a variable supply voltage to the RF power amplifier and a supply voltage controller operable to produce a control signal which controls adjustment by the regulator of the supply voltage, wherein the non-linearity detector is connected to the supply voltage controller and is operable upon detecting non-linearity to deliver a control signal to the supply voltage controller which causes the supply voltage controller to adjust the regulator to increase the supply voltage.
The forward path may include an I channel for receiving and processing an I (in phase) component of an input baseband signal and a Q channel for receiving and processing a Q (quadrature phase) component of an input baseband signal, and each of the I channel and the Q channel may include an upconverting mixer. In this case, the non-linearity detector is connected to the I channel and to the Q channel to sample I and Q components of the baseband signal for delivery to the respective mixers.
The transmitter according to the first aspect of the invention may include an antenna to radiate output signals from the RF power amplifier with no isolator between the RF power amplifier and an antenna. In the transmitter according to the first aspect of the invention the non-linearity detector may thus comprise an isolator eliminator employed to detect non-linearity arising from a changed load impedance at the RF power amplifier caused by a change in antenna VSWR.
The non-linearity detector may be operable to detect, when non-linearity is detected by the non- linearity detector, whether a pre-determined maximum supply voltage is currently being delivered to the RF power amplifier. The non-linearity detector may be connected to the supply voltage controller and may be operable to provide to the supply voltage controller, in response to detection of non-linearity and that a predetermined maximum supply voltage is not currently being delivered to the RF power amplifier, a control signal to the supply voltage controller which causes the supply voltage controller to adjust the regulator to increase the supply voltage to the RF power amplifier. The non- linearity detector may be operable to produce a control signal indicating a value of a parameter to be employed by the supply voltage controller which may be increased in steps until a maximum value is reached, the maximum value corresponding to a predetermined maximum supply voltage .
In the transmitter according to the first aspect of the invention the forward path may include an attenuator. Where the forward path includes an I channel and a Q channel the forward path may include an attenuator in each of the I channel and the Q channel. The attenuator or attenuators are operable to attenuate a baseband signal or a component thereof in the forward path. The non-linearity detector may be connected to the attenuator or attenuators to deliver a control signal to control a level of attenuation applied by the attenuator or attenuators. The non-linearity detector may be operable to apply a control signal to the attenuator or attenuators in response to detecting that a predetermined maximum supply voltage is currently being delivered to the RF power amplifier.
The transmitter according to the first aspect of the invention may include a supply voltage control loop connected to the forward channel to sample a baseband signal or components thereof delivered to the forward channel. The supply voltage control loop may include a calculator operable to calculate a varying amplitude of the sampled baseband signal. The supply voltage control loop may also include the supply voltage controller referred to earlier. The supply voltage controller may be operable to control adjustment by the regulator of the supply voltage in a modulation which follows the varying amplitude of the sampled baseband signal. The supply voltage controller may operate to multiply by a multiplication constant indicated by an output signal produced by the non-linearity detector an input provided by the calculator. Thus, the controller may be operable to produce an output signal of the form z = a*y, where: y is an output value indicated by the calculator; z is an output of the supply voltage controller; a is a multiplication constant that is indicated by the non-linearity detector; and * represents multiplication.
The supply voltage control loop including the supply voltage controller thereby forms an envelope follower in which the modulation applied by the voltage regulator follows the variation of the baseband signal amplitude, unless the supply voltage controller is instructed by the non-linearity detector to produce an output signal to provide an increase in supply voltage by the voltage regulator until a maximum supply voltage is reached. Thus, such a transmitter provides the benefits of (i) a non-linearity detector, e.g. to serve as an isolator eliminator; and (ii) a supply voltage control loop; and (iii) synergistic interaction between the two.
The calculator of the supply voltage control loop may include (i) an I2 + Q2 calculator which squares and adds samples of I and Q components of the baseband signal delivered to the I and Q channels and (ii) a square root calculator which calculates a square root of the calculation result x produced by the I2 + Q2 calculator. The square root calculator may beneficially comprise an approximator which calculates an approximate value of the square root of the calculation result x produced by the I2 + Q2 calculator. The approximator may calculate a value of a parameter F(x) = A2X2 + Aix + Ao, where A2, Ai and Ao are pre-determined constants . Such an approximator beneficially produces a signal which is narrow band compared with an accurate square root calculator which produces a signal with much larger bandwidth.
The calculator and the supply voltage controller of the supply voltage control loop may conveniently be incorporated in a single processor. In this case the single processor may be operable to employ a control signal delivered from the non-linearity detector to multiply, by a multiplication constant, the constants A2, Ai and Ao applied by the approximator so that an output control signal produced by the supply voltage controller produces a desired incremental increase in supply voltage by the voltage regulator.
The transmitter according to the first aspect of the invention may comprise a Cartesian loop linear transmitter. The transmitter may include a coupler, e.g. a directional coupler, to sample an output of the RF power amplifier and a feedback path extending from the coupler to the forward path at a feedback path junction prior to the upconverting mixer, the feedback path including a downconverting mixer. Where the forward path includes an I channel and a Q channel the feedback path may include an I channel feedback loop including a first downconverting mixer and a Q channel feedback loop including a second downconverting mixer. The I channel feedback loop may be connected to the I channel of the forward path at a first feedback path junction and the Q channel feedback loop may be connected to the Q channel of the forward path at a second feedback path junction. The feedback path junction or each of the feedback path junctions may comprise a differential amplifier or a summing junction. The non-linearity detector may be connected to the respective I and Q channels of the forward path at detector junctions which are located between the feedback path junctions and the upconverting mixers in the respective I and Q channels.
Each of the I channel and the Q channel may include a low pass filter and optionally an amplifier between the feedback path junction of the channel and the detector junction of the channel.
The non-linearity detector of the transmitter according to the first aspect of the invention may include a first RMS estimator operable to estimate a root mean square signal level of an on-channel signal and a second RMS estimator operable to estimate a root mean square value of a noise signal at a predetermined frequency offset from a frequency of the on-channel signal, a divider operable to produce an output signal which represents a root mean square value estimated by the first RMS estimator divided by a root mean square value estimated by the second RMS estimator and a comparator operable to compare an output signal produced by the divider with a threshold signal. The comparator may produce an output signal having one of two possible state values representing respectively non-linearity detected or not detected.
The transmitter according to the first aspect of the invention may be operable in accordance with TETRA or other standard operating protocols.
According to the present invention in a second aspect there is provided a method of operation in a linear RF transmitter which includes receiving a baseband input signal in a forward path, converting the baseband signal into a modulated RF signal in an upconverting mixer, and amplifying the modulated RF signal by a RF power amplifier, sampling the baseband signal for delivery to the upconverting mixer by a non-linearity detector connected to the forward path, detecting by the detector whether non-linearity is present in operation of the transmitter and providing in response to detected non-linearity a control signal to change operation of the transmitter to reduce the non-linearity, and characterised by delivering a supply voltage from a voltage source to the RF power amplifier, producing a modulation of the supply voltage by a voltage regulator and producing by a supply voltage controller a control signal which controls modulation by the regulator of the supply voltage, wherein the non-linearity detector is connected to the supply voltage controller and upon detecting non-linearity delivers a control signal to the supply voltage controller which causes the supply voltage controller to adjust the regulator to increase the supply voltage. When the non-linearity detector detects that operation of the RF power amplifier is not linear, the non-linearity detector may issue a signal to the voltage supply controller causing the supply voltage, as regulated by the regulator, to the RF power amplifier to be increased. This will make the RF power amplifier more linear and will allow the RF power amplifier to increase its nominal output power. If, but preferably only if, the supply voltage is increased to a maximum allowed value and the RF power amplifier is still not linear, e.g. due to an increase in VSWR, the non-linearity detector may issue a control signal to an attenuator in the forward path, or attenuators in each of an I channel and a Q channel of the forward path, to increase an attenuation applied by the attenuator or attenuators to a baseband signal or a component thereof in the forward path to reduce an output signal power.
Further features of the invention are disclosed in the embodiments of the invention to be described.
Embodiments of the present invention will now be described by way of example with reference to the accompanying drawings, in which:
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 is a block schematic diagram of a Cartesian loop transmitter embodying the invention.
FIG. 2 is a block schematic diagram showing more detail of part of the transmitter of FIG. 1.
FIG. 3 is a flow chart of a method of operation embodying the invention in the transmitter of FIG. 1.
DESCRIPTION OF EMBODIMENTS OF THE INVENTION FIG. 1 is a block schematic diagram of a Cartesian loop linear transmitter 100 embodying the present invention. The transmitter 100 includes (inter alia) a forward path 102 and a feedback path 104. A DSP (digital signal processor) 101 generates a baseband input digital signal containing information to provide a modulation signal to be transmitted by the transmitter 100 by RF communication. The input baseband digital signal comprises an I (in phase) signal component which is delivered via an I channel 105 and a corresponding Q signal component which is delivered via a Q channel 107. The I signal component and the Q signal component are converted to analog form by a D/A (digital to analog) converter 103. The D/A converter 103 delivers an I signal in analog form along the I channel 105 of the forward path 102 and a Q signal in analog form along the Q channel 107 of the forward path 102 in parallel with the I channel 105. The I channel 105 includes, connected to the D/A converter 103 and connected together in turn, an I channel attenuator 108, a differential amplifier 112, an amplifier/ filter 116 and an upconverting mixer 120. The upconverting mixer 120 is also connected to a local oscillator (carrier frequency synthesizer) 136. The Q channel 107 includes, connected to the D/A converter 103 and connected together in turn, a Q channel attenuator 110, a differential amplifier 114, an amplifier/ filter 118 and an upconverting mixer 122. The upconverting mixer 122 is also connected to the local oscillator 136 via a 90 degrees phase shifter 158. Output connections from the upconverting mixers 120 and 122 provide inputs to a summing junction 124 having an output connected in turn to a RFPA (radio frequency power amplifier) 126 and an antenna 128. In operation, an I signal produced digitally by the DSP 101 and converted to analog form by the D/A converter 103 is delivered to the I channel attenuator 108 and is attenuated by the I channel attenuator 108. The level of attenuation by the I channel attenuator 108 is controlled in a manner to be described later. The attenuated I signal is then delivered to the differential amplifier 112. An error control signal produced in a manner to be described from a downconverting mixer 132 is subtracted from the I signal in the differential amplifier 112. An output corrected I signal produced by the differential amplifier 112 is then amplified and filtered by the amplifier/ filter 116. The amplifier/filter 116 comprises a low pass filter which serves as a slew rate limiter. The amplified and filtered I signal produced by the amplifier/filter 116 is then mixed with a carrier frequency signal from the local oscillator 136 to upconvert the I signal from baseband to RF (radio frequency) .
A Q signal produced digitally by the DSP 101 and converted to analog form by the D/A converter 103 is delivered to the Q channel attenuator 110 and is attenuated by the Q attenuator 110 which is controlled in a manner to be described later. The attenuated Q signal is then delivered to the differential amplifier 114. An error control signal produced in a manner to be described later from a downconverting mixer 134 is subtracted from the Q signal in the differential amplifier 112. An output corrected Q signal produced by the differential amplifier 114 is then amplified and filtered by the amplifier/ filter 118. The amplifier/filter 118 comprises a low pass filter which serves as a slew rate limiter. The amplified and filtered Q signal produced by the amplifier/filter 118 is then mixed by the upconverting mixer 122 with a carrier frequency signal (whose phase has been shifted by ninety degrees compared with the carrier frequency signal delivered to the upconverting mixer 120) delivered from the local oscillator 136 via the phase shifter 158 to upconvert the Q signal from baseband to RF (radio frequency) .
The RF signals produced as outputs by the upconverting mixer 120 of the I channel 105 and the upconverting mixer 122 of the Q channel 107 are combined at the summing junction 124, and the combined RF signal is amplified by the RFPA 126 to produce an amplified RF output signal. The amplified RF output signal produced by the RFPA 126 is delivered to the antenna 128 and is sent over the air to a distant terminal (not shown) at which it is received. The antenna 128 may (at times when the transmitter 100 is not in operation) also receive an incoming RF signal sent over the air from a distant terminal (not shown) and may deliver the received signal for processing to a RF receiver (not shown) . The transmitter 100 includes no isolator between the RFPA 126 and the antenna 128.
A directional coupler 130 is connected between the RFPA 126 and the antenna 128 to sample the amplified RF output signal produced by the RFPA 126. The directional coupler 130 is connected to the feedback path 104 at an attenuator 109. The feedback path 104 leading from the attenuator 109 is branched to include an I channel feedback loop 115 and a Q channel feedback loop 117. The I channel feedback loop 115 includes a downconverting mixer 132 connected to the attenuator 109 and also connected to the local oscillator 136. The downconverting mixer 132 is connected to provide an output signal to the differential amplifier 112. The Q channel feedback loop 117 includes a downconverting mixer 134 which is connected to the attenuator 109 and also is connected to the local oscillator 136 via a 90 degrees phase shifter 160. The downconverting mixer 134 is connected to provide an output signal to the differential amplifier 114. In operation, the directional coupler 130 supplies the I channel feedback loop 115 and the Q channel feedback loop 117 with a feedback signal representing the amplified output signal. The feedback signal is attenuated by the attenuator 109. The attenuated feedback signal is downconverted into I and Q feedback baseband components by the mixers 132 and 134. The resultant baseband components are error control signals which are subtracted respectively in the differential amplifier 112 and the differential amplifier 114 from the I signal and the Q signal delivered to the differential amplifiers 112 and 114 respectively by the attenuators 108 and 110. The feedback loops 115 and 117 provide a known mechanism for maintaining linear operation of the transmitter 100 by forcing the transmitter 100 to produce an RF output which follows the undistorted I signal and Q signal supplied respectively to the differential amplifiers 112 and 114. In alternative known forms of the forward path 102 and the feedback path 104 which will be readily apparent to those skilled in the art, the differential amplifiers 112 and 114 may be replaced by summing junctions, and an amplifier and low pass filter may be included in each of the loops 115 and 117 after the downconverting mixers 132 and 134. Such amplifiers and filters may be included instead of or in addition to the amplifier/ filters 116 and 118. The transmitter 100 includes a supply modulation control loop 133 to improve efficiency of operation of the RFPA 126. The loop 133 includes an I2 + Q2 calculator 119, a square root approximator 121, a supply voltage controller 127, a D/A converter 129 and a voltage regulator 111. The calculator 119 is connected to the I channel 105 and to the Q channel 107 between the DSP 101 and the D/A converter 103. The calculator 119 thereby samples the output baseband I and Q signals produced in digital form by the DSP 101. The calculator 119 calculates the value of a parameter x given by: x = I2 + Q2 (Equation 1)
, where I and Q are the sampled baseband I and Q signals over a given sampling period, e.g. to give for example a sampling rate of 96 ksamples/sec. The square root approximator 121 comprises a digital processor which calculates an approximation of the square root of x for each value of x provided by the calculator 119. The square root approximator 121 uses a relationship as follows to calculate an approximation of the square root of each value of x:
F(x) = A2X2 + A1X + A0 (Equation 2) where A2, Ai and Ao are pre-determined constants. Thus, the expression for F(x) given in Equation 2 gives a polynomial (quadratic) approximation of the square root of x. The approximator 121 is a preferred band limited form of calculator to calculate the square root of x. A more exact square root calculator could be used, but would result a greater signal bandwidth. A narrower band calculator is preferred since for spectral efficiency it is desirable for the spectrum of the transmitted signal to be narrow band and thereby meet adjacent channel emission specifications defined in industry standard operating protocols.
The result of the calculation by the square root approximator 121 is provided as an output digital signal to the supply voltage controller 127. The supply voltage controller 127 also receives an input digital control signal from a non-linearity detector 125. Such a signal is the output of an algorithm as described in more detail later with reference to FIGS . 2 and 3. When such an input signal is received by the supply voltage controller 127 it causes the controller 127 to multiply an output of the approximator 119 by a multiplication constant. Thus, the controller 127 is operable to produce an output digital signal of the form: z = a*y (Equation 3) , where : y is a value indicated by an output signal produced by the square root approximator 121; z is a value indicated by an output signal produced by the supply voltage controller 127; a is multiplication constant having a value indicated by a signal from the non-linearity detector 125; and * represents multiplication.
The supply voltage controller 127 produces in accordance with Equation 3 an output digital control signal, e.g. in the form of a digital control word, which is delivered to a D/A (digital to analog) converter 129 which converts the digital control signal into an analog control signal. The analog control signal produced by the D/A converter 129 is delivered as an analog control signal to the voltage regulator 111. The regulator 111 receives an input DC voltage from a voltage source 113 such as a battery. The analog control signal received by the regulator 111 from the D/A converter 129 causes the regulator 111 to modulate the voltage from the voltage source 113 in a known manner to produce a modulated supply voltage V8 which is applied to the RFPA 126. The modulation follows the variation of the digital control signal produced by the supply voltage indicator 127. An instantaneous increase in the supply voltage V3 produced by the modulation causes an increase in the output power of the RFPA 126 and a reduction in the supply voltage causes a fall in the output power of the RFPA 126. The control loop 133 improves efficiency of the RFPA by varying the supply voltage to suit the detected power level of the input signal.
The transmitter 100 includes the non-linearity detector 125 to serve as an isolator eliminator, which is included since the transmitter 100 includes no isolator between the RFPA 126 and the antenna 128. The non- linearity detector 125 is connected to the I channel 105 between the amplifier/ filter 116 and the upconverter mixer 120 to sample the baseband filtered output I signal produced by the amplifier /filter 116, and is connected to the Q channel 107 between the amplifier/ filter 118 and the upconverter mixer 122 to sample the baseband filtered output Q signal produced by the amplifier /filter 118. The detector 125 is also connected to the I channel attenuator 108 and the Q channel attenuator 110 to control operation of the attenuators 108 and 110 in a manner to be further described with reference to FIG. 2. As noted earlier, the non-linearity detector 125 is also connected to the supply voltage controller 127 to provide a control signal to the supply voltage controller 127 indicating a value of a multiplication constant Λa' as in Equation 3. The value of the multiplication constant is selected in a manner to be described later with reference to FIG. 3.
FIG. 2 shows in more detail an example of the non- linearity detector 125. This example is based upon the isolator eliminator of the form described in GB 2403086 but is also adapted in accordance with an embodiment of the invention. The non-linearity detector 125 is connected to an output of the amplifier/filter 116 in the I channel 105 and an output of the amplifier/filter 118 in the Q channel 107. The non-linearity detector 125 thereby samples in baseband form the I signal and the Q signal in the channels 105 and 107 before these signals reach the upconverting mixers 120 and 122.
The non-linearity detector 125 detects whether there is non-linearity in operation of the RFPA 126. The non- linearity detector 125 continuously samples from the information provided by the baseband I signal and the baseband Q signal corrected by the error control signals from the feedback loops 115 and 117 an on-channel baseband signal level as well as a noise (off channel signal) level at a predefined frequency offset in relation to a known desired transmission channel frequency. An example of the frequency offset at which the noise level is measured is (+ or -) 13.5 kHz. The sampled baseband I signal and the sampled baseband Q signal are passed through a low pass filter 138 and a low pass filter 142 respectively to obtain the on-channel signal components. The sampled baseband I signal and the sampled baseband Q signal are passed through a band pass filter 140 and a band pass filter 144, in parallel respectively with the low pass filter 138 and the low pass filter 142, to obtain the noise (or off-channel signal) components. Filtered output signals from the low pass filter 138 and the low pass filter 142 are delivered to a RMS (root mean square) estimator 146. Filtered output signals from the band pass filter 140 and the band pass filter 144 are delivered to a RMS (root mean square) estimator 148. The RMS estimator 146 comprises a digital processor which calculates the squares respectively of each consecutive incremental sample of the input I signal and the Q signal it receives. The RMS estimator 146 also sums the values of the squares obtained for each sample in a given block of N samples, where N is a predetermined block size containing a given number of samples, e.g. from one to one hundred samples, and then divides the result by N. This gives for the block of N samples an estimate of the mean square value. Finally, the RMS estimator 146 calculates the square root of the mean square value obtained for the block of N samples giving a root mean square (^RMS') value. This is an estimate of the current RMS signal level of the baseband signal having as components the I signal and the Q signal. An output signal representing a digital value of the calculated RMS signal level for each consecutive block of samples is passed from the RMS estimator 146 to a divider 150. The RMS estimator 148 operates in a manner similar to the RMS estimator 146 to estimate an RMS value of the noise level (off-channel signal level) . An output signal representing a digital value of the calculated RMS noise level for each consecutive block of samples is passed from the RMS estimator 148 to the divider 150. The divider 150 calculates for each block value provided by the RMS estimator 146 and each corresponding block value provided by the RMS estimator 148 a ratio of the value of the RMS on-channel signal to the value of the RMS noise (off channel signal) level. An output signal providing a signal to noise ratio value for each block considered by the divider 150 is provided by the divider 150 to a comparator 152. The comparator 152 provides an output signal having a LOW state if the determined signal to noise ratio indicated by the output signal from the divider 150 is above a predefined threshold THR. The comparator 152 provides an output signal having a HIGH state if the determined signal to noise ratio indicated by the output signal from the divider 150 is equal to or below a predefined threshold THR. If the comparator 152 produces an output signal having a LOW state it indicates that the transmitter 100 is operating linearly and no change in operation of the transmitter 100 is required. If the comparator 152 produces an output signal having a HIGH state it indicates non-linearity of the RFPA 126 of the transmitter 100 that needs to be corrected for. The comparator 152 is connected to a microprocessor 154 which receives the output signal produced by the comparator 152 and initiates corrective action when the output signal from the comparator 152 has a HIGH state. The microprocessor 154 is connected in turn to the I channel attenuator 108 and the Q channel attenuator 110 and also to the supply voltage controller 127. In some circumstances to be described in detail with reference to FIG. 3, if the output signal from the comparator 152 has a HIGH state the microprocessor 154 provides a control signal to the supply voltage controller 127 to produce a change in output signal from the controller 127 leading to an incremental change by the regulator 111 to a level of supply voltage Vs provided to the RFPA 126.
In other circumstances, to be described in detail with reference to FIG. 3, if the output signal from the comparator 152 has a HIGH state the microprocessor 154 provides to the I channel attenuator 108 and the Q channel attenuator 110 a control signal to make an incremental change to a level of attenuation applied by the I channel attenuator 108 and the Q channel attenuator 110. When the microprocessor 154 applies a control signal to the attenuators 108 and 110 it also applies a delay to issue of any further control signal based on a further HIGH state output signal from the comparator 152. This delay is applied to allow the transmitter 100 to settle after a step increase in attenuation by the attenuators 108 and 110 has been applied. The delay is implemented by the microprocessor 154 not reading results of the comparator 152 for pre-defined period of time. The microprocessor 154 stores a value indicating an attenuation setting of the attenuators 108 and 110 in a memory 156 to record how much attenuation is currently being applied.
FIG. 3 is a flow chart showing a method 200 embodying the invention used in operation by the transmitter 100. The method 200 is an algorithm which runs in every transmission time slot, e.g. as indicated by a controller (not shown) which controls functional operations of the transmitter 100. The method 200 thus begins at a step 201 which is a start of a transmission time slot. Next, a non-linearity detector algorithm run by the microprocessor 154 begins in a step 203. In a decision step 205, the microprocessor 154 decides whether or not non-linearity is detected based on the current state of the output signal from the comparator 152. The decision is λNO' when the signal received by the microprocessor 154 from the comparator 152 has a LOW state indicating linearity of operation of the transmitter 100, and the decision is ΛYES' when the signal received by the microprocessor 154 from the comparator 152 has a HIGH state indicating non-linearity of operation of the transmitter 100. If the decision taken by the microprocessor 154 in step 205 is λNO' the method 200 returns to step 203. If the decision taken by the microprocessor 154 in step 205 is ΛYES', i.e. that non-linearity is detected, a decision step 207 follows.
In decision step 207, the microprocessor 154 decides whether the multiplication constant Λa' (referred to earlier with reference to Equation 3) that is indicated by the microprocessor 154 to the supply voltage controller 127 is at its pre-determined maximum value. The pre-determined maximum value of the multiplication constant Λa' corresponds to a pre-determined maximum value of the supply voltage Vs. If the decision taken by the microprocessor 154 in step 207 is λYES' , i.e. the microprocessor 154 determines that a maximum value of the multiplication constant λa' has already been reached, a step 209 follows in which the microprocessor 154 issues a control signal to the attenuators 108 and 110 to increase the attenuation applied by the attenuators 108 and 110 in the manner described earlier. If the decision taken by the microprocessor 154 in step 207 is ΛNO' , i.e. the microprocessor 154 determines that a maximum value of the multiplication constant λa' has not already been reached, a step 211 follows. In step 211 the microprocessor 154 delivers a control signal to the supply voltage controller 127 to indicate to the supply voltage controller 127 that an incremental increase is required to the multiplication constant 'a' . This causes the output signal z provided by the supply voltage controller 127 to the voltage regulator 111 via the D/A converter 129 to indicate that the regulator 111 is to provide an incremental increase in the supply voltage V3. Following step 211, a decision step 213 is applied. In the decision step 213, the microprocessor 154 decides whether or not non-linearity is detected. This is the same as in step 205 but is based on a later result from the comparator 152. If the decision taken by the microprocessor 154 in step 213 is λNO' , i.e. that linearity is detected, the method 200 returns to step 203. If the decision taken by the microprocessor 154 in step 213 is λYES' , i.e. that non-linearity is detected, the method 200 returns to decision step 207.
As an example of applying steps 207, 209 and 211, the multiplication constant Λa' applied by the supply voltage controller 127 may initially be set at a = 1 by factory tuning. This value of Λa' is initially applied each time the method 200 runs. Then there may be different gradually increasing values of the constant λa' applied in turn by the controller 127, the value each time being as indicated by an output signal from the microprocessor 154 to the supply voltage controller 127. These different values may for example be a = 1.05; 1.1; 1.2; 1.3; 1.4; 1.5; and 1.6. The last value, a = 1.6, may be a maximum allowed value of the multiplication constant λa' corresponding to a maximum allowed value of supply voltage V3. Thus, each time step 207 produces a λNO' result, the value of multiplication constant Λa' indicated by the microprocessor 154 to the supply voltage controller 127 will be gradually increased in an iteration of step 211 until a maximum value of xa' is reached, when step 209 is alternatively applied instead of step 211. In a modified form of the transmitter 100 embodying the invention, the calculator comprising the square root approximator 121 (and optionally the calculator 119) and the supply voltage controller 127 may be combined in a single processor. In this case, when a signal is provided in step 211 to indicate to the supply voltage controller 127 to increase its output to increase the supply voltage V8, the combined single processor (approximator 121/controller 127) may apply an incremental increase to the constants Kz, Ai and Ao applied in accordance with
Equation 2. This may be done by multiplying the constants Kz, Ai and Ao by a multiplication constant indicated by an output signal from the microprocessor 154. The multiplication constant may gradually increase for each iteration of step 211. This produces for each incremental increase to the constants A2, Ai and A0 an incremental increase in the supply voltage Vs until, in step 207 of the method 200, the microprocessor 154 detects that a maximum value of the multiplication constant has been reached indicating also that a maximum value of Vs has been reached. When reaching of the maximum value is detected, step 209 is applied.
Thus, by the method 200, when non-linearity is detected in step 205 the microprocessor 154 issues a control signal to the supply voltage controller 127 which, by causing the regulator 111 to provide an increase in the supply voltage Vs, causes the output power capability of the RFPA 126 to be increased. This corrective action is taken if it is still possible, i.e. if a maximum value of Vs has not been reached, rather than the microprocessor 154 issuing a signal to increase the attenuation applied by the attenuators 108 and 110 to reduce the non-linearity. Increasing the attenuation in order to reduce non-linearity is only applied when the maximum value of Vs has been reached. Increasing the supply voltage V5 when still possible is more beneficial than simply increasing attenuation by the attenuators 108 and 110 as in the prior art, because increasing the supply voltage V3 allows greater output power from the RFPA 126. Increasing attenuation by the attenuators 108 and 110 reduces the output power of the transmitter 100. Where the transmitter 100 is used for example in a radio communication system, reducing output power undesirably reduces the range of a RF signal produced by the transmitter 100. Changes in the supply voltage Vs in the manner described may beneficially be achieved more rapidly than changes which occur in the VSWR as experienced by the RFPA 126. This allows changes in the power level of the RFPA 126 due to changes in the VSWR to be minimized. The tuning of the RFPA 126 may be set initially, e.g. in a factory in which the transmitter 100 is produced and tested, at a level assuming a nominal load termination employed instead of the antenna 128, e.g. a 50 ohm termination. This sets the initial nominal operating conditions under which the algorithm comprising the method 200 is applied, including an initial level of the multiplication constant λa' indicated by the signal provided by the microprocessor 154 to the supply voltage controller 127. The transmitter 100 is suitable for use in an RF communication terminals for use in a number of communication applications, especially those that operate in narrow bandwidths and demand a high level of linearity. Of particular interest is use in a terminal in which the transmitter operates in accordance with pre-defined industry standard operating protocols or procedures such as the TETRA (Terrestrial Trunked Radio) standard protocols as defined by ETSI (European Telecommunication Standards Institute) .

Claims

1. A linear RF transmitter including a forward path for receiving and processing a baseband input signal, the forward path including an upconverting mixer operable to convert a baseband signal into a modulated RF signal, and a RF (radio frequency) power amplifier operable to amplify a modulated RF signal produced by the upconverting mixer, a non-linearity detector connected to the forward path to sample a baseband signal for delivery to the upconverting mixer and operable to detect non- linearity of the transmitter and to provide in response a signal to change operation of the transmitter to reduce the non-linearity, and characterised in that the transmitter further includes a regulator connectible to a supply voltage source and to the RF power amplifier to deliver a variable supply voltage to the RF power amplifier and a supply voltage controller operable to produce a control signal which controls adjustment by the regulator of the supply voltage, wherein the non- linearity detector is connected to the supply voltage controller and is operable upon detecting non-linearity to deliver a control signal to the supply voltage controller which causes the supply voltage controller to adjust the regulator to increase the supply voltage.
2. A transmitter according to claim 1 wherein the forward path includes an I channel for receiving and processing an I (in phase) component of an input baseband signal and a Q channel for receiving and processing a Q (quadrature phase) component of an input baseband signal, wherein each of the I channel and the Q channel includes an upconverting mixer and wherein the non-linearity detector is connected to the I channel and to the Q channel to sample I and Q components of the baseband signal to be delivered to the respective mixers.
3. A transmitter according to claim 1 wherein the non- linearity detector comprises an isolator eliminator.
4. A transmitter according to claim 1 wherein the non- linearity detector is operable to detect, when non- linearity is detected by the non-linearity detector, whether a pre-determined maximum supply voltage is currently being delivered to the RF power amplifier.
5. A transmitter according to claim 4 wherein the non- linearity detector is connected to the supply voltage controller and is operable to provide, in response to detection of non-linearity and that a pre-determined maximum supply voltage is not currently being delivered to the RF power amplifier, a control signal to the supply voltage controller which causes the supply voltage controller to adjust the regulator to increase the supply voltage to the RF power amplifier.
6. A transmitter according to claim 5 wherein the non- linearity detector is operable to produce a control signal indicating a value of a parameter to be employed by the supply voltage controller which increases until a maximum value is reached, the maximum value corresponding to a predetermined maximum supply voltage .
7. A transmitter according to claim 1 wherein the forward path includes an attenuator, or, where the forward path includes an I channel and a Q channel, attenuators in each of the I channel and the Q channel to attenuate a baseband signal or a component thereof in the forward path.
8. A transmitter according to claim 7 wherein the non- linearity detector is connected to the attenuator or attenuators and is operable to produce a signal to control a level of attenuation applied by the attenuator or attenuators .
9. A transmitter according to claim 8 wherein the non- linearity detector is operable to apply a control signal to the attenuator or attenuators in response to detecting that a pre-determined maximum supply voltage is currently being delivered to the RF power amplifier.
10. A transmitter according to claim 1 including a supply voltage control loop connected to the forward channel to sample a baseband signal or components thereof delivered to the forward channel, the supply voltage control loop including a calculator operable to calculate a varying amplitude of the sampled baseband signal, and wherein the controller is operable to control adjustment by the regulator of the supply voltage in a modulation which follows the varying amplitude of the sampled baseband signal.
11. A transmitter according to claim 10 wherein the supply voltage controller is operable to multiply by a multiplication constant indicated by an output signal produced by the non-linearity detector an input provided by the calculator.
12. A transmitter according to claim 10 wherein the calculator includes an I2 + Q2 calculator which squares .. and adds samples of I and Q components of the baseband signal and a square root calculator which calculates a square root of a calculation result x produced by the I2 + Q2 calculator.
13. A transmitter according to claim 12 wherein the square root calculator is operable to calculate an approximate value of the square root of the calculation result x produced by the I2 + Q2 calculator by calculating a value of a parameter F(x) = A2X2 + Aix + A0, where A2, Ai and Ao are pre-determined constants.
14. A transmitter according to claim 13 wherein the calculator and the supply voltage controller are incorporated in a single processor and wherein the single processor is operable to employ a control signal delivered from the non-linearity detector to multiply the constants A2, Ai and Ao by a multiplication constant having a value indicated by the control signal from the non- linearity detector so that an output control signal produced by the controller produces an incremental increase in supply voltage by the voltage regulator.
15. A transmitter according to claim 1 which is a Cartesian loop linear transmitter and includes a coupler to sample an output of the RF power amplifier and a feedback path extending from the coupler to the forward path at a feedback path junction prior to the upconverting mixer, the feedback path including a downconverting mixer.
16. A transmitter according to claim 15 wherein the forward path includes an I channel and a Q channel and the feedback path includes an I channel feedback loop including a first downconverting mixer and a Q channel feedback loop including a second downconverting mixer, and the I channel feedback loop is connected to the I channel of the forward path at a first feedback path junction and the Q channel feedback loop is connected to the Q channel of the forward path at a second feedback path junction. miniiiraiira aiiiffiiuiuuTiiWiiii Iw)PlI mmύmmmmwm
33 CM07987EI
17. A transmitter according to claim 16 wherein the non- linearity detector is connected to the respective I and Q channels of the forward path at detector junctions located between the feedback path junctions and the upconverting mixers .
18. A transmitter according to claim 17 wherein each of the I channel and the Q channel includes a low pass filter between the feedback path junction of the channel 0 and the detector junction of the channel.
19. A transmitter according to claim 1 wherein the non- linearity detector includes a first RMS estimator operable to estimate a root mean square signal level of 5 an on-channel signal and a second RMS estimator operable to estimate a root mean square value of a noise signal at a predetermined frequency offset from a frequency of the on-channel signal, a divider operable to produce an output signal which represents a root mean square value 0 estimated by the first RMS estimator divided by a root mean square value estimated by the second RMS estimator and a comparator operable to compare an output signal produced by the divider with a threshold signal.
5 20. A transmitter according to claim 1 wherein the transmitter is operable in accordance with TETRA standard operating protocols.
PCT/US2006/060866 2005-11-15 2006-11-14 Linear rf transmitter and method of operation WO2007059484A2 (en)

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* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
GB2466218B (en) * 2008-12-12 2012-02-01 Motorola Solutions Inc Simultaneous supply modulation and circulator elimination with maximum efficiency
EP2663046A1 (en) * 2012-05-07 2013-11-13 Alcatel Lucent Apparatus and method for slew-rate limitation
WO2018223256A1 (en) 2017-06-05 2018-12-13 Telefonaktiebolaget Lm Ericsson (Publ) Method and controller for controlling power amplifier in radio transmitter as well as radio unit and radio device

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5699383A (en) * 1995-03-06 1997-12-16 Nec Corporation High-power linear amplification using periodically updated amplitude and phase correction values
US6594595B2 (en) * 2001-04-03 2003-07-15 Advantest Corporation Apparatus for and method of measuring cross-correlation coefficient between signals
US6731694B2 (en) * 2001-08-07 2004-05-04 Motorola, Inc. Isolator eliminator for a linear transmitter
US6819090B1 (en) * 2002-08-16 2004-11-16 Athena Controls, Inc. Line voltage regulator with RMS voltage approximation feedback control

Family Cites Families (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
GB2403086B (en) * 2003-06-17 2005-07-20 Motorola Inc Cartesian loop transmitter and method of adjusting an output level of such transmitter
US7259630B2 (en) * 2003-07-23 2007-08-21 Andrew Corporation Elimination of peak clipping and improved efficiency for RF power amplifiers with a predistorter
GB2406985A (en) * 2003-10-07 2005-04-13 Software Radio Technology Ltd An RF transmitter with optimal control of amplifier bias with respect to efficiency and adjacent channel power
JP2005197870A (en) * 2004-01-05 2005-07-21 Fujitsu Ltd Distortion compensation power amplifier

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5699383A (en) * 1995-03-06 1997-12-16 Nec Corporation High-power linear amplification using periodically updated amplitude and phase correction values
US6594595B2 (en) * 2001-04-03 2003-07-15 Advantest Corporation Apparatus for and method of measuring cross-correlation coefficient between signals
US6731694B2 (en) * 2001-08-07 2004-05-04 Motorola, Inc. Isolator eliminator for a linear transmitter
US6819090B1 (en) * 2002-08-16 2004-11-16 Athena Controls, Inc. Line voltage regulator with RMS voltage approximation feedback control

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GB2432271B (en) 2007-10-10

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