WO2007045071A1 - Recuperation d'horloge dans un signal optique reçu par un reseau de communication optique - Google Patents

Recuperation d'horloge dans un signal optique reçu par un reseau de communication optique Download PDF

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Publication number
WO2007045071A1
WO2007045071A1 PCT/CA2006/001459 CA2006001459W WO2007045071A1 WO 2007045071 A1 WO2007045071 A1 WO 2007045071A1 CA 2006001459 W CA2006001459 W CA 2006001459W WO 2007045071 A1 WO2007045071 A1 WO 2007045071A1
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WO
WIPO (PCT)
Prior art keywords
signal
phase
computing
polarization
clock
Prior art date
Application number
PCT/CA2006/001459
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English (en)
Inventor
Han Sun
Slobodan Jovanovic
Kuang Tsan Wu
Chandra Bontu
Kim B. Roberts
Jianzhong Xu
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Nortel Networks Limited
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Priority claimed from US11/315,345 external-priority patent/US7532822B2/en
Application filed by Nortel Networks Limited filed Critical Nortel Networks Limited
Publication of WO2007045071A1 publication Critical patent/WO2007045071A1/fr

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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L7/00—Arrangements for synchronising receiver with transmitter
    • H04L7/0004—Initialisation of the receiver
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
    • H03L7/00—Automatic control of frequency or phase; Synchronisation
    • H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
    • H03L7/08—Details of the phase-locked loop
    • H03L7/085—Details of the phase-locked loop concerning mainly the frequency- or phase-detection arrangement including the filtering or amplification of its output signal
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03L—AUTOMATIC CONTROL, STARTING, SYNCHRONISATION OR STABILISATION OF GENERATORS OF ELECTRONIC OSCILLATIONS OR PULSES
    • H03L7/00—Automatic control of frequency or phase; Synchronisation
    • H03L7/06—Automatic control of frequency or phase; Synchronisation using a reference signal applied to a frequency- or phase-locked loop
    • H03L7/08—Details of the phase-locked loop
    • H03L7/085—Details of the phase-locked loop concerning mainly the frequency- or phase-detection arrangement including the filtering or amplification of its output signal
    • H03L7/089—Details of the phase-locked loop concerning mainly the frequency- or phase-detection arrangement including the filtering or amplification of its output signal the phase or frequency detector generating up-down pulses
    • H03L7/0891—Details of the phase-locked loop concerning mainly the frequency- or phase-detection arrangement including the filtering or amplification of its output signal the phase or frequency detector generating up-down pulses the up-down pulses controlling source and sink current generators, e.g. a charge pump
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L7/00—Arrangements for synchronising receiver with transmitter
    • H04L7/02—Speed or phase control by the received code signals, the signals containing no special synchronisation information
    • H04L7/027—Speed or phase control by the received code signals, the signals containing no special synchronisation information extracting the synchronising or clock signal from the received signal spectrum, e.g. by using a resonant or bandpass circuit
    • H04L7/0278—Band edge detection

Definitions

  • the present invention relates to optical communications networks, and in particular to a method and apparatus for recovering a clock signal from optical signals received through an optical communications network.
  • the optical signal is mixed with a strong, narrow-line-width, local oscillator signal by an optical hybrid, and the combined signal made incident on one or more photodetectors.
  • the inbound optical signal is first split into orthogonal polarizations, and each polarization processed by a respective optical hybrid, fix-phase and Quadrature components of each polarization can be detected using respective photodetectors positioned to receive corresponding signals output by the optical hybrid.
  • the frequency spectrum of the electrical current appearing at the photodetector output(s) is substantially proportional to the convolution of the spectrum of the received optical signal and the local oscillator, and contains a signal component lying at an intermediate frequency that contains the data.
  • Coherent detection receivers offer numerous advantages over direct detection receivers, many of which follow from the fact that coherent detection techniques provide both phase and amplitude information of the optical signal. As such, more robust modulation schemes, such as phase shift keying (PSK), differential phase shift keying (DPSK) and quadrature phase shift keying (QPSK)can be used.
  • PSK phase shift keying
  • DPSK differential phase shift keying
  • QPSK quadrature phase shift keying
  • FIG. Ia schematically illustrates the system of No ⁇ [Supra, April 2005).
  • an optical signal received through an optical link 2 is divided by a polarization beam splitter 4 into orthogonal polarizations (nominally referred to as X and Y polarizations in FIG. 1), which are then mixed with a local oscillator (LO) 6 through a quadrature 90° optical hybrid 8.
  • the composite optical signals appearing at the output of the optical hybrid 8 are made incident on a set of photodetectors 10 to generate analog electrical signals respectively corresponding to real (Re) and imaginary (Im) parts of each polarization.
  • Analog-to-Digital (A/D) converters 12 are then sampled at the symbol rate by respective Analog-to-Digital (A/D) converters 12 to generate digital sample streams of each of the real (Re) and imaginary (Im) parts of each polarization.
  • the digital samples are then supplied to a 1:M DEMUXer 14, which splits the data path into M parallel sample streams having a lower sample rate (by a factor of M), each of which is supplied to a respective processing module 16.
  • an inverse Jones matrix that models the polarization performance of the optical link is used to compensate polarization distortions.
  • the polarization compensated samples can then be decoded for data recovery.
  • clock recovery is performed using either: a clock recovery block 18 inserted into the data path between the photodetectors and the A/D converters (FIG. Ia), or alternatively using an intensity modulation direct detection receiver 20 as shown in FIG. Ib.
  • Recovering the clock signals from the electrical I and Q signals generated by the photodetectors, as shown in FIG. Ia, is beneficial in that it keeps all of the received optical power within the main data path, and at the same time makes both amplitude and phase information of the received optical signal available to the clock recovery circuit.
  • this solution renders the system extremely sensitive to polarization impairments.
  • the use of a direct detection receiver 20 for clock recovery as shown in FIG.
  • Ib avoids i problems associated with the polarization sensitivity of the coherent receiver.
  • this solution diverts at least a portion of the energy of the received optical signal out of the data path, and is vulnerable to severe chromatic dispersion and polarization impairments at least in part due to inter-symbol interference (ISI).
  • ISI inter-symbol interference
  • the received optical signal contains a mixture of two versions of the transmitted data signals, separated by a half symbol differential delay. Interference between the two versions can prevent the clock recovery circuit from successfully achieving a phase/frequency locked state. Indeed, in this example, when the signal power is equally split between the two modes of the PMD, the recovered
  • clock tone goes to zero.
  • the amount of power in each mode varies with time. Each time the amount of power in one mode becomes greater than that of the other mode, the phase of the recovered clock jumps by 180 degrees.
  • the polarization impairments are generally time varying, with speeds as high as tens of kilohertz. This means that the phase of the recovered clock can be moved about by
  • an object of the present invention is to provide methods enabling clock recovery from a highly distorted optical signal.
  • an aspect of the present invention provides a method of recovering a clock signal from a high speed optical signal received through an optical communications network.
  • a stream of multi-bit digital samples of the optical signal is processed to generate a dispersion compensated signal.
  • the dispersion compensated signal is then processed to generate an optimized signal; and a clock phase of the optimized signal is then detected.
  • a clock recovery circuit which includes an optimization block implementing an inverse Jones matrix for processing a multi-bit digital signal corresponding to the optical signal; and an adaptation loop for computing updated coefficients of the inverse Jones matrix.
  • the adaptation loop In a locked state of the clock recovery circuit, the adaptation loop is controlled to compute updated coefficients of the inverse Jones matrix based on parameters of the optical signal.
  • the adaptation loop In an unlocked state of the clock recovery circuit, the adaptation loop is made to sweep the coefficients of the inverse Jones matrix at a predetermined rate.
  • FIGs. IA and IB are block diagrams schematically illustrating clock recovery methods for use in conjunction with coherent optical receivers known in the prior art
  • FIG. 2 is a block diagram schematically illustrating representative operations of a clock recovery circuit in accordance with an embodiment of the present invention
  • FIG. 3 is a block diagram schematically illustrating representative operations of the adaptation loop of the embodiment of FIG. 2;
  • FIGs. 4A-4B are block diagrams schematically illustrating representative operations of the phase detector, loop filter, VCO and lock detector of the embodiment of FIG. 2;
  • FIGs. 5A and 5B are block diagrams schematically illustrating representative operations a clock recovery circuit according to a second embodiment of the present invention.
  • FIG. 6 is a block diagram schematically illustrating representative operations of a third embodiment of the present invention.
  • FIG. 7 is a block diagram schematically illustrating representative operations of the sweeper block and coefficient calculator of FIG. 6 in greater detail.
  • FIG. 8 is a block diagram schematically illustrating a representative coherent optical receiver which incorporates a clock recovery circuit in accordance with an embodiment of the present invention.
  • the present invention provides methods and systems enabling clock recovery from a highly distorted optical signal. Embodiments of the invention are described below, by way of example only, with reference to FIGs. 2-8.
  • the present invention provides a clock recovery circuit in which a multi-bit sample stream of a received optical signal is digitally processed to compensate dispersion and/or polarization, and a clock signal recovered from the resulting compensated signal.
  • a clock recovery circuit in which a multi-bit sample stream of a received optical signal is digitally processed to compensate dispersion and/or polarization, and a clock signal recovered from the resulting compensated signal.
  • Various methods may be used to recover a clock from a compensated signal.
  • a nominal clock having a frequency that is sufficiently close to the optical signal symbol rate the could be used to enable the receiver to acquire the optical signal.
  • Over-sampling the optical signal e.g. at 3x or 4x the symbol rate
  • the distortion compensated signals generated by the receiver's data path can be used for clock recovery in "steady-state" operation of the receiver.
  • Another approach is to synthesize the clock signal by processing (e.g. by filtering, mixing etc) the digital signal without the use of a phase locked loop, and indeed without the use of an oscillator.
  • the derived clock need not be used to control the sampling, and need not be a periodic electrical signal. In fact, it may only exist as the phase of a mathematical interpolation function applied to a sequence of sets of digital values.
  • Still another approach is to select one sampling phase for clock recovery, and then monitor the mean power at the selected phase.
  • the initial selection of the sampling phase may be based on the mean power level, for example by selecting the sampling phase having the highest mean power level. Following the initial selection. If the mean power at the current sampling phase is less than the mean power at 180 degrees away, then the sign of the PLL feedback can be toggled so that the PLL will now track the sampling phase having a maximum mean power.
  • the mean power levels at each phase can be estimated directly from a portion of the samples. This arrangement reduces the sensitivity of the clock-recovery circuit to changes in the optical signal due to polarization effects.
  • Thresholds can be applied to the sample stream to augment or generate a clock component. Different polarization conditions cause different functions to have the best utility. One can select between a set of these functions in response to changing polarization conditions, or changing clock quality, to maintain adequate utility and therefore adequate clock quality. While the above approaches can be expected to work under at least some conditions, it is difficult to prove that they will work over a sufficiently wide range of conditions that are expected to be encountered in a real-world network, and this challenge increases with increasing dispersion and polarization distortion of the optical signal.
  • FIG. 2 illustrates a clock recovery circuit in accordance with an embodiment of the present invention which overcomes these difficulties.
  • an inbound optical signal is received through an optical link 2, split into orthogonal polarizations by a Polarization Beam Splitter 4, and then mixed with a Local Oscillator (LO) 6 signal by a conventional 90° optical hybrid 8.
  • the composite optical signals emerging from the optical hybrid are supplied to respective photodetectors 10, which generate corresponding analog signals.
  • the photodetector signals are sampled by respective Analog- to-Digital (A/D) converters 22 to yield sample streams corresponding to In-phase (T) and Quadrature (Q) components of each of the received polarizations.
  • A/D Analog- to-Digital
  • the multi-bit I and Q sample streams generated by the A/D converters 34 are supplied to Fast Fourier Transform (FFT) filters 24, which analyse the spectral content of each polarization and apply a first order dispersive function which compensates the expected chromatic dispersion of the link 2.
  • FFT Fast Fourier Transform
  • At the output of the FFT filters 24, respective Upper Side Band (USB) and Lower side Band (LSB) signals of each polarization are tapped and supplied to an optimization block 26, which implements a simplified polarization compensation function.
  • the compensated USB and LSB signals appearing at the output of the optimization block 48 are then combined (at 28), and the resulting signal supplied to a clock phase detector 30.
  • the phase detection result is then passed to a loop filter 32, which supplies respective control signals to coarse (C) and fine (F) tuning ports of a voltage controlled Oscillator (VCO) 34.
  • VCO voltage controlled Oscillator
  • the VCO output 36 is used as the A/D sample clock for driving the A/D converters 22.
  • the signal path from the A/D converters 22, through the FFT filters 24, the optimization block 26, phase detector 30, loop filter 32, VCO 34 and back to the A/D converters 22 defines a Phase Locked Loop (PLL) which tunes the VCO output 36 to phase and frequency match symbols modulated onto the received optical signal.
  • PLL Phase Locked Loop
  • the VCO output 36 can also be supplied to a Digital Signal processor (DSP) (not shown in FIG. 2) for controlling operation of the clock recovery circuit.
  • DSP Digital Signal processor
  • the FFT filters 24, optimization block 26, phase detector 30 and loop filter 32 are cascaded together to form a high speed data path.
  • Pipelining the signal processing and clock recovery functions in this manner minimises the response time of the PLL, and thereby facilitates effective "real-time” clock recovery.
  • the resolution of the A/D converters 22 is a balance between performance and cost. Increasing the resolution improves sampling accuracy, and thereby improves the extent to which signal distortions can be corrected by downstream dispersion and polarization compensators. However, this increased accuracy is obtained at a cost of increased complexity, silicon area and heat generation. It has been found that a resolution of 5 or 6 bits provides satisfactory performance, at an acceptable cost.
  • the sample rate of the A/D converters 22 is selected to satisfy the Nyquist criterion, for the highest anticipated symbol rate of the received optical signal. As will be appreciated, Nyquist sampling ensures that the sample streams generated at the A/D converter output contains all of the information content of each signal, even if the sample timing (with reference to each received symbol) is ambiguous and/or unknown.
  • the FFT filters 24 operate in a known manner to compute the spectrum of each received polarization, and apply a first order dispersive function which at least partially compensates chromatic dispersion.
  • the amount of dispersion that can be compensated will be a function of the width of the FFT filters 24, which will be a balance between performance and cost.
  • each FFT filter 24 has a width of 256 samples, which enables compensation of well over 10000 ps/nm of dispersion.
  • tapping the USB and LSB signals from the FFT output is a simple matter of tapping the appropriate points.
  • the upper and lower side band signals are obtained by tapping 16 points for each signal, respectively centered at ⁇ half the symbol rate.
  • Tapping the Upper and Lower side band signals at the output of the FFT filters 24, after substantial chromatic dispersion compensation has been applied has the effect of rendering the clock detectable, even in the presence of severe dispersion impairments in the received optical signal.
  • a clock can be accurately detected from a 10 Gigasymbol per second signal distorted by dispersion in excess of 50000 ps/nm. This is ten or twenty times the dispersion that can be tolerated by conventional clock recovery circuits operating on that signal.
  • the Upper and Lower side band signals appearing at the output of the FFT filters 24 are supplied to an adaptation loop 38 having a coefficient calculator 40 which determines filter coefficients that will compensate polarization impairments of the received optical signal.
  • the optimization block 26 preferably implements a simplified polarization compensation function, which partially compensates polarization impairments of the received optical signal.
  • a simplified polarization compensation function which partially compensates polarization impairments of the received optical signal.
  • the optimization block 26 implements an inverse Jones matrix using angle ⁇ (n + ⁇ ) and phase ⁇ (n + ⁇ ) filter coefficients generated by the coefficient calculator 40. These filter coefficients are iteratively recalculated to optimize a lock detection function f ⁇ p , ⁇ p ) value generated by the phase detector 30 (described in greater detail below).
  • FIG. 3 illustrates one method by which the filter coefficients ⁇ (n + ⁇ ) and ⁇ n + 1) may be generated.
  • the coefficient calculator 40 implements parallel feed-back optimization loops which respectively update the phase ⁇ p and angle ⁇ p coefficients in accordance with a pair of update equations:
  • the gradient functions Tj [ ⁇ p ) and Tf ( ⁇ p ) may be found from the lock detector function f( ⁇ p , ⁇ p ) using a steepest decent algorithm, in a manner that will be apparent to those of ordinary skill in the art.
  • the update step sizes u ⁇ and u ⁇ are programmable values that can be set based on a desired balance between response time and precision of the adaptation loop 38. This operation of the adaptation loop 38 automatically controls the values for the phase and angle coefficients ⁇ (n + l) and ⁇ (n+ ⁇ ) to optimise (in this case, to maximize) the lock detector function f ⁇ p , ⁇ p ) value.
  • each of the LSB and USB signals are added to yield corresponding TOTAL-LSB and TOTAL-USB signals that are substantially independent of polarization effects.
  • Each element of the TOTAL-LSB signal is then multiplied with the complex conjugate of the corresponding element of the TOTAL-USB signal.
  • the products are added together to form a composite signal that is a single multi-bit complex value, of which the mean of the imaginary term is proportional to the A/D sample clock (i.e. the VCO output 36) phase error relative to symbols of the received optical signal, and the mean of the real term is a "lock value" indicative of a degree to which the VCO output 36 is frequency and phase-locked to the received optical signal.
  • phase detector 30 may conveniently be implemented as a pair of summation circuits, as may be seen in FIG. 4A.
  • a first summation circuit 42 sums the imaginary term of the composite signal over a predetermined number of occurrences in time (e.g. 1 or 8 samples) to obtain a clock phase estimate 44, which is then passed to the loop filter 32.
  • the second summation circuit 46 sums the real term over a predetermined number of occurrences (e.g. 1 or 8 samples) to obtain a "lock value"
  • real and imaginary parts of the products are summed over the frequency range (e.g. 16 points of the FFT) and then summed over a time interval (e.g. 1 or 8 samples) to produce estimates proportional to the mean values.
  • the frequency range e.g. 16 points of the FFT
  • a time interval e.g. 1 or 8 samples
  • a representative lock detector 48 is illustrated in FIG. 4A.
  • the lock detector 48 is configured as an Infinite Impulse Response (IIR) filter which averages successive f ⁇ p , ⁇ p ) values to avoid spurious changes in the lock state of the IIR filter
  • the HR output is the lock indicator 50, and can be used by a control unit (not shown) for processing and control/monitoring functions.
  • the value of the lock indicator 50 signal can also be used to determine the PLL loop bandwidth, which enables the loop filter step size Ud k (described below) to be set based on a desired loop bandwidth.
  • the loop filter32 receives the phase estimate 44 from the phase detector 30, and supplies a pair of control signals to the coarse (C) and fine (F) tuning ports of the voltage controlled Oscillator (VCO) 34 which may, for example, be a conventional 1 IGHz VCO.
  • VCO voltage controlled Oscillator
  • the loop filter 32 adjusts damping and resonance frequency for the 2 nd order phase locked loop.
  • the loop filter 32 includes a coarse tuning path 52 and a fine tuning path 54.
  • the pulse sequencer 56 is shown in greater detail in FIG.
  • T c i k is a programmable parameter based on a desired sensitivity of the pulse sequencer.
  • the pulse sequencer 56 implements a digital integrator 60 which combines with the analog charge pump integrator 58 to form a single integrator with fine resolution and broad range, not limited by the precision of a Digital-to- Analog (D/A) circuit.
  • the feedback path 62 inside the pulse sequencer 56 subtracts off from the digital integrator 60 the digital value (TcIk) that corresponds to the analog charge that has been sent to the analog integrator 58, and thereby couples the two integrators to act as one.
  • the fine tuning path 54 comprises a multiplier 64 for scaling the phase estimate 44 from the phase detector 30, for example using a programmable step size U c i k , and an adder 66 for offsetting the scaled phase estimate to a desired value range (e.g. by adding a constant).
  • the step size ⁇ c i k may be selected based on a desired bandwidth of the phase locked loop, which will be a balance between pull in range, response time and sensitivity.
  • the output of the fine tuning path 54 is supplied to the fine tuning port of the VCO 34 via a digital-to-analog (D/A) converter 68.
  • D/A digital-to-analog
  • the 2-4A-B is capable of successfully recovering a clock signal from a received optical signal, even in the presence of severe impairments.
  • the FFT filters 24 substantially compensate dispersion of the inbound optical signal.
  • Selection and processing of the USB and LSB signals through the optimization block 26 enables the phase detector 30 to detect the A/D sample clock phase, in the presence of significant amounts of residual dispersion. For example, in some embodiments, successful phase detection in the presence of about 3000 ps/nm residual dispersion can be obtained.
  • the ability of the optimization block 26 to track, and thus compensate polarization transients, is a function of the adaptation time of the coefficient calculator.
  • a polarization update rate on the order of 100 MHz may be selected.
  • the coefficient calculator 40 can operate fast enough that the filter coefficients ⁇ p (n + l) and ⁇ p (n + ⁇ ) can be recomputed and uploaded to the optimization block 26 during each cycle. This translates into an adaptation time on the order of 100ns (equivalent to 1 OMHz), which is fast enough to accurately track, and thus compensate polarization transients of 5OkHz or higher.
  • ASE Amplified Spontaneous Emission
  • FIG. 5 illustrates a development of the system of FIGs. 2-4, in which a reference clock is used to generate tuning signals which can be used to tune the VCO 34 when there is no recoverable clock in the received optical signal.
  • FIG. 5 is similar to that of FIGs. 2-4, the primary difference being that the coarse tuning path 52 of the loop filter 32 receives a frequency error signal 72 from a timing reference 70, and a switch block 74 is inserted into the signal path at the output of the loop filter 32.
  • the switch block 74 is controlled by a select signal and operates to latch the PLL between a "test" state in which the VCO 34 is tuned to a reference clock 76, and an "operational" state in which the VCO 32 is tuned to the received optical signal.
  • FIG. 5A illustrates the "operational" state of the PLL
  • FIG. 5B illustrates the "test" state.
  • the timing reference 70 includes a reference clock 76, digital phase/frequency detector (DPFD) block 78 and a frequency detector 80.
  • the reference clock 76 is a comparatively stable oscillator which generates a clock signal having a frequency that is in known proportion to a desired operating frequency of the VCO 34.
  • the frequency detector 80 may conveniently be implemented using a digital counter 84 and a latch 86, and measures the frequency of the VCO output signal 36. Subtracting the measured VCO frequency from a "target" frequency value ⁇ yields a multi- bit estimate of the VCO frequency error 72. This error estimate is supplied to the coarse tuning path 52 of the loop filter 32, where it is added to the output 44 of the phase detector 30 immediately upstream of the pulse sequencer 56. This is generally useful to keep the VCO 34 centered on the desired frequency when the clock is not locked to a valid optical signal.
  • the tristate output 82 of the DPFD block 78 is discarded in favour of the course and fine tuning paths 52, 54 of the loop filter 32.
  • the clock recovery circuit operates substantially as described above with reference to FIGs. 2-4.
  • the pulse sequencer 56 also receives the frequency error estimate 72 from the frequency detector 80. In the absence of a useful received optical signal, the pulse sequencer 56 will operate entirely on the basis of this error estimate 72, which ensures that the coarse tuning path 52 can tune the VCO frequency to within the pull- in range of the fine tuning path 54 of the PLL.
  • the VCO output 36 provides a nominal clock which can be used for initial acquisition of an optical signal.
  • the frequency detector and coarse tuning path 52 of the PLL will hold the VCO frequency within about ⁇ 120ppm of the received optical signal. With Nyquist sampling, this error is sufficiently small that the sample streams generated by the A/D converters 22 will contain sufficient information to enable the receiver to acquire the optical signal
  • the VCO frequency may be as much as ⁇ 10% off the target frequency. This places a lower bound on the required pull-in range of the coarse tuning path 52 of the PLL, so that the VCO frequency can be acquired at start-up and then pulled into the pull-in range of the fine tuning path 54.
  • the phase detector output44 will contain useful phase information, and the pulse sequencer can successfully operate on the basis of that information, locking the VCO 34 to twice the symbol rate of the received optical signal.
  • the response of the frequency detector 80 it is possible to arrange that the error estimate 72 under these frequency conditions is zero, so that the frequency detector output does not interfere with valid phase estimates output by the phase detector 30.
  • clock circuits used in communications systems typically have a frequency tolerance of about ⁇ 40ppm. This means that, under normal operating conditions, tuning the VCO 34 to the received optical signal (and thus the Tx local clock at the transmitter end of the optical link) may result in the frequency detector 80 measuring a frequency "error" of up to 80ppm, due to accumulated tolerances of both the Tx local clock and the reference clock 76. However, if the frequency detector response is chosen to be insensitive to frequency errors of less than ⁇ 80ppm, then the error estimate produced by the frequency detector 80 will be suppressed. One method of accomplishing this is to apply a thresholding function to the frequency detector output, which forces detector output values of between -80p ⁇ m and +80ppm to zero.
  • An alternative method is to clip a number of least significant bits representing at least ⁇ 80ppm from the multibit frequency detector output, so that only the most significant bits are supplied to the loop filter.
  • the insensitivity of the frequency detector 80 means that, when a useful optical signal is received, there could be as much as 120ppm (combining the ⁇ 80ppm insensitivity of the frequency detector 80 with the ⁇ 40ppm frequency tolerance of the Tx local clock) difference between the received optical signal and the VCO frequency.
  • this value will be different in embodiments in with clock frequency tolerances are other than ⁇ 40p ⁇ m. However, in all cases, this frequency difference sets a lower limit on the required pull-in range of the fine tuning path 54 of the PLL.
  • the tristate output of the DPFD block 78 is supplied to the coarse tuning port of the VCO 34 via the charge pump 58, and thereby tunes the VCO 34 to the reference clock 76.
  • the D/A converter 68 connected to the fine tuning port can also be supplied with a constant value (e.g. zero) so that the D/A output does not float undesirably.
  • the loop filter 32 receives phase information from the phase detector 30 as well as frequency error information from the frequency detector 80. However, the loop filter output is not used to tune the VCO 34.
  • the VCO frequency may differ from the Tx local clock by as much as ⁇ 80ppm, depending on the frequency tolerances of the Tx local clock and the reference clock 76, which is within the pull-in range of the fine tuning path 54 of the PLL. Accordingly, the clock recovery circuit can be latched between the "Test” and "Operational" states without requiring a reset.
  • the clock recovery circuits of FIGs. 2-5 cannot reliably obtain a lock condition. For example, consider a condition in which two components of the received optical signal are out of phase (due to PMD) by about V 2 of a symbol period. In an out-of lock condition, clock phase estimates 44 generated by the phase detector 30 will exhibit random excursions and average near zero.
  • the lock indicator signal 50 obtained by averaging successive f ⁇ p , ⁇ p ) values in the lock detector 48 will also be near zero, as will. be the gradient functions Tf ( ⁇ p ) and Tf ⁇ p ).
  • This situation may be prevented by forcing the filter coefficients ⁇ p (n + ⁇ ) and ⁇ p ⁇ n +1) to slowly sweep the Poincare sphere, and examining the phase detector output for a locked condition using a high speed lock detector.
  • a representative implementation of this approach is illustrated in FIGs. 6 and 7.
  • the ER filter-based lock detector 48 of FIGs. 2-5 is retained to provide a coarse lock indication. However, this is supplemented with a fine filter 88, which has a lower memory effect and thus faster response than the DR. filter 48.
  • various types of filter circuits may be used, provided that they exhibit a sufficiently fast response and are relatively insensitive to noise.
  • a commonly known filter type that can be used for this purpose is a so-called “leaky bucket" filter, which is illustrated in the figures.
  • the output of the Leaky Bucket filter 88 provides a fine lock indicator signal 90, which is used to enable operation of a sweeper circuit 92 for driving the coefficient calculator 40. This operation is illustrated in greater detail in FIG. 7.
  • the Leaky Bucket filter 88 provides a low-memory averaging function 94 and a threshold comparison block 96 to provide a high-speed, "clean" lock indication.
  • the leak rate ( ⁇ i 0Ck ) is a programmable negative value that may be selected based on a desired response of the filter.
  • the governing thresholds T s and T 1 - are also programmable. Representative state transitions of the Thresholding block 96 are as follows:
  • the sweeper 92 provides a pair of parallel frequency dividers 98a,b which respectively divide a clock signal frequency by pi and p2.
  • the clock signal may conveniently be provided by the reference clock 76 of the timing reference 70, but a different clock signal source may be used, if desired.
  • the divisors pi and p2 are preferably prime numbers, which prevents aliasing effects, and have different values so that the filter coefficients ⁇ p (n+ ⁇ ) and ⁇ p (n+ ⁇ ) will be swept at correspondingly different rates.
  • the output of each frequency divider 98a,b is a trigger signal which causes a respective buffer 100a,b to output multibit increments ⁇ + and ⁇ + at its trigger signal frequency.
  • the value of each increment is preferably a programmable value, which is set based on a desired step size. The increment values ⁇ + and ⁇ + generated by the sweeper 92 are then supplied to the coefficient calculator 40.
  • the increment values ⁇ + and ⁇ + are supplied to respective latch circuits 102a,b, which also receive the summed gradient values
  • Each latch circuit 102 is controlled by an "enable signal"
  • the coefficient calculator 40 updates the filter coefficients ⁇ p (n+ ⁇ ) and ⁇ n+l) according to the equations:
  • the choice of prime numbers for the divisors pi and p2 results in the filter coefficients ⁇ p (n+l) and ⁇ p (n+ ⁇ ) evenly covering the Poincare sphere with a pattern that repeats every 20 seconds.
  • the forced sweeping of the filter coefficients moves them away from whatever regions of the Poincare sphere that would otherwise have currently prevented reacquisition of "lock".
  • Embodiments using the above parameters can attain a worst case average hunt time of less than 5ms.
  • FIG. 8 illustrates a representative coherent optical receiver in which the clock recovery circuit of FIGs. 2-7 is implemented.
  • an inbound optical signal is received through an optical link 2, split into orthogonal polarizations by a Polarization Beam Splitter 4, and then mixed with a Local Oscillator (LO) signal 6 by a conventional 90° optical hybrid 8.
  • the composite optical signals emerging from the optical hybrid 8 are supplied to respective photodetectors 10, which generate corresponding analog signals.
  • the photodetector signals are sampled by respective Analog-to-Digital (AfD) converters 22 to yield multi-bit digital sample streams corresponding to In-phase (I) and Quadrature (Q) components of each of the received polarizations.
  • AfD Analog-to-Digital
  • each dispersion compensator 104 is implemented using a Fast Fourier Transform (FFT) filter 24 cascaded with an Inverse Fast Fourier Transform (IFFT) filter 106.
  • FFT Fast Fourier Transform
  • IFFT Inverse Fast Fourier Transform
  • the dispersion-compensated sample streams appearing at the output of the dispersion compensators 104 are then supplied to a 1:M distribution unit 108, which operates to divide the signal path, by selectively routing blocks of samples from the dispersion compensators 104 into each one of the M paths.
  • a polarization compensator 110 operates to de-convolve the transmitted I and Q signal components of each polarization from the dispersion-compensated sample streams.
  • the distortion-compensated sample streams appearing at the output of each polarization compensator 110 are then supplied to a respective decoder 112 for detection of data symbols and recovery of data.
  • the clock recovery circuit utilizes the receiver "front end" to receive and detect the optical signal.
  • the Upper Side Band (USB) and Lower Side Band (LSB) signals are tapped at the output of the dispersion compensator FFTs 24, and supplied to a signal processor block 114 which includes the optimization block 26, phase detector 30, loop filter 32 and VCO 34 all of which operate as described above with reference to FIGs. 2-7 to generate a recovered clock signal, which is used to drive the A/D converters 22 and other operations of the receiver.
  • the signal processor block 114 can be incorporated within a controller unit 116 of the receiver, which can also provide the functionality of the timing reference described above with reference to FIGs. 5-7.
  • This controller unit 116 is advantageously implemented as part of an ASIC or FPGA, but some or all can be implemented in Digital Signal Processor (DSP) firmware.
  • DSP Digital Signal Processor
  • an alternative clock recovery method may be implemented in which a clock signal is recovered by a clock recovery circuit 118 using the fully compensated signals available at the output of the polarization compensators 110.
  • conventional clock recovery methods may be used to obtain a valid clock signal from the fully compensated signals, subject to the limitation that the delay between the sampling instant and the application of the feedback to the VCO limits the achievable bandwidth and transfer function of the PLL. More processing introduces more delay within this loop.
  • another method may be used, which operates on the sample blocks generated by the distribution unit 108.
  • phase and/or frequency of a symbol clock representative of the symbol rate within a block of samples can be determined using methods described by L.E. Franks in Carrier and Bit Synchronization in Data Communication - a Tutorial Review. IEEE Transactions on Communications, Vol.. Com-28, No. 8, August 1980. This information can be used to adjust the A/D converter sampling clock for future samples, thereby forming a phase-locked loop.
  • the symbol clock can be used in performing interpolation between the existing samples of the sample block to derive interpolated samples which closely approximate the timing (i.e. the phase) of symbols within the sample block. Interpolation methods such as Fourier interpolation are known in the art of digital signal processing.
  • the symbol rate for example, interpolation may not be required and a reasonably close sample can be chosen for each symbol. Once a set of samples that are adequately aligned with the symbols have been identified (or derived by interpolation) the set of samples can then be decoded to determine the values of those symbols.
  • the clock recovery circuit of FIGs. 2-7 can be used to provide a valid clock signal during a start-up phase of the receiver, following which it is possible to switch to a clock signal recovered from the polarization compensator output.
  • the clock signal recovered from the polarization compensator output can be used for monitoring a quality of operation of the clock recovery circuit of FIGs. 2-7
  • the multi-bit digital sample stream generated by the A/D converter 22 block is digitally processed by an FFT filter 24 cascaded with an optimization block 26, and the resulting optimized (that is, compensated) signal supplied to the phase detector 30 for clock phase detection and control of the VCO 34.
  • the FFT filter 24 performs a spectral analysis of the digital sample stream, and applies a dispersive function to compensate dispersion of the optical link 2.
  • the optimization block implements a simplified inverse Jones matrix to compensate polarization impairments. It will be appreciated, however, that there are means by which dispersion and polarization compensation may be provided. For example, filter types other than a FFT filter may be used for dispersion compensation. Similarly, filter types other than an inverse Jones matrix may be used for polarization compensation.
  • the optical link 2 may be provided with optical dispersion compensation in a manner well known in the art.
  • the FFT filter 24 (or any other electronic dispersion compensation) may be redundant, and thus omitted.
  • the digital sample streams emerging from the A/D converter 22 block can be supplied directly to the optimization block 26 for polarization compensation.
  • an FFT filter can be retained to perform a spectral analysis (i.e. without applying a dispersive function), but this is not essential.
  • the tapping of upper side band *USB and lower side band (LSB) signals is advantageous because it reduces the required size of the optimization block, and thus system cost.
  • the clock phase can equally be detected from the entire digital sample stream, if desired.
  • USB and LSB signals can be tapped at the output of the FFT filter 24 and supplied to the phase detector, if desired, but the entire dispersion compensated signal may equally be used.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Optical Communication System (AREA)
  • Synchronisation In Digital Transmission Systems (AREA)

Abstract

La présente invention concerne un procédé de récupération d'un signal d'horloge dans un signal optique reçu par l'intermédiaire d'un système de communication optique. Un flux d'échantillonnage numérique est traité pour générer un signal compensé en dispersion. Le signal compensé en dispersion est ensuite divisé pour obtenir des signaux de bande latérale supérieure et de bande latérale inférieure de chaque polarisation reçue du signal optique. Les signaux de bande latérale supérieure et de bande latérale inférieure sont ensuite traités pour compenser les dégradations dépendant de la polarisation et l'horloge est récupérée du signal résultant optimisé.
PCT/CA2006/001459 2005-10-21 2006-09-05 Recuperation d'horloge dans un signal optique reçu par un reseau de communication optique WO2007045071A1 (fr)

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US72875105P 2005-10-21 2005-10-21
US60/728,751 2005-10-21
US11/315,345 US7532822B2 (en) 2005-02-28 2005-12-23 Clock recovery from an optical signal with polarization impairments
US11/315,342 2005-12-23
US11/315,342 US7627252B2 (en) 2005-02-28 2005-12-23 Clock recovery from an optical signal with dispersion impairments
US11/315,345 2005-12-23

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EP1942590A1 (fr) * 2007-01-03 2008-07-09 Alcatel Lucent Récepteur optique cohérent et procédé pour compensation de la distorsion liée à la polarisation des signaux dans un système de transmission à fibre optique
JP2009253972A (ja) * 2008-04-01 2009-10-29 Fujitsu Ltd フィルタ係数変更装置および方法
WO2010131323A1 (fr) * 2009-05-11 2010-11-18 三菱電機株式会社 Appareil de demultiplexage de polarisation
WO2011022869A1 (fr) 2009-08-24 2011-03-03 Huawei Technologies Co., Ltd. Appareil de récupération d'horloge
US8712247B2 (en) 2010-02-20 2014-04-29 Huawei Technologies Co., Ltd. Clock phase recovery apparatus
EP2482486A3 (fr) * 2011-01-31 2017-05-31 Fujitsu Limited Appareil de synchronisation d'horloge d'échantillonnage, appareil de réception cohérent numérique et procédé de synchronisation d'horloge d'échantillonnage
WO2023020677A1 (fr) * 2021-08-16 2023-02-23 Huawei Technologies Co., Ltd. Détecteur de verrouillage de récupération de synchronisation, appareil récepteur de signal et procédé de détection de verrouillage de récupération de synchronisation

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WO2000060776A1 (fr) * 1999-04-01 2000-10-12 Optical Technologies U.S.A. Corp. Dispositif et procede de compensation de dispersion de polarisation de mode dans un systeme de communication optique
WO2002027994A1 (fr) * 2000-09-26 2002-04-04 Celight, Inc. Systeme et procede destines a la communication optique multiplexee par repartition de code
US20020186435A1 (en) * 2000-09-26 2002-12-12 Isaac Shpantzer System and method for orthogonal frequency division multiplexed optical communication
US20030123884A1 (en) * 2001-07-09 2003-07-03 Willner Alan E. Monitoring optical dispersion based on vestigial side band optical filtering
US20030175034A1 (en) * 2000-10-09 2003-09-18 Reinhold Noe Method and apparatus for optical information transmission

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WO2000060776A1 (fr) * 1999-04-01 2000-10-12 Optical Technologies U.S.A. Corp. Dispositif et procede de compensation de dispersion de polarisation de mode dans un systeme de communication optique
WO2002027994A1 (fr) * 2000-09-26 2002-04-04 Celight, Inc. Systeme et procede destines a la communication optique multiplexee par repartition de code
US20020186435A1 (en) * 2000-09-26 2002-12-12 Isaac Shpantzer System and method for orthogonal frequency division multiplexed optical communication
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US20030123884A1 (en) * 2001-07-09 2003-07-03 Willner Alan E. Monitoring optical dispersion based on vestigial side band optical filtering

Cited By (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1942590A1 (fr) * 2007-01-03 2008-07-09 Alcatel Lucent Récepteur optique cohérent et procédé pour compensation de la distorsion liée à la polarisation des signaux dans un système de transmission à fibre optique
JP2009253972A (ja) * 2008-04-01 2009-10-29 Fujitsu Ltd フィルタ係数変更装置および方法
WO2010131323A1 (fr) * 2009-05-11 2010-11-18 三菱電機株式会社 Appareil de demultiplexage de polarisation
WO2011022869A1 (fr) 2009-08-24 2011-03-03 Huawei Technologies Co., Ltd. Appareil de récupération d'horloge
CN102405614A (zh) * 2009-08-24 2012-04-04 华为技术有限公司 时钟恢复设备
US8781333B2 (en) 2009-08-24 2014-07-15 Huwei Technologies Co., Ltd. Clock recovery apparatus
US8712247B2 (en) 2010-02-20 2014-04-29 Huawei Technologies Co., Ltd. Clock phase recovery apparatus
EP2482486A3 (fr) * 2011-01-31 2017-05-31 Fujitsu Limited Appareil de synchronisation d'horloge d'échantillonnage, appareil de réception cohérent numérique et procédé de synchronisation d'horloge d'échantillonnage
WO2023020677A1 (fr) * 2021-08-16 2023-02-23 Huawei Technologies Co., Ltd. Détecteur de verrouillage de récupération de synchronisation, appareil récepteur de signal et procédé de détection de verrouillage de récupération de synchronisation

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