WO2006018035A1 - Appareil et procede de reduction de derive de phase - Google Patents

Appareil et procede de reduction de derive de phase Download PDF

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Publication number
WO2006018035A1
WO2006018035A1 PCT/EP2004/009373 EP2004009373W WO2006018035A1 WO 2006018035 A1 WO2006018035 A1 WO 2006018035A1 EP 2004009373 W EP2004009373 W EP 2004009373W WO 2006018035 A1 WO2006018035 A1 WO 2006018035A1
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WO
WIPO (PCT)
Prior art keywords
signal
phase
phase drift
time
domain
Prior art date
Application number
PCT/EP2004/009373
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English (en)
Inventor
Günther Auer
Original Assignee
Ntt Docomo, Inc.
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Publication date
Application filed by Ntt Docomo, Inc. filed Critical Ntt Docomo, Inc.
Priority to JP2007526210A priority Critical patent/JP2008511196A/ja
Priority to EP04764355A priority patent/EP1782594A1/fr
Priority to PCT/EP2004/009373 priority patent/WO2006018035A1/fr
Publication of WO2006018035A1 publication Critical patent/WO2006018035A1/fr

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2657—Carrier synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2662—Symbol synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2668—Details of algorithms
    • H04L27/2673—Details of algorithms characterised by synchronisation parameters
    • H04L27/2676—Blind, i.e. without using known symbols
    • H04L27/2679—Decision-aided
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L25/00—Baseband systems
    • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
    • H04L25/0202—Channel estimation
    • H04L25/0204—Channel estimation of multiple channels
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2656—Frame synchronisation, e.g. packet synchronisation, time division duplex [TDD] switching point detection or subframe synchronisation

Definitions

  • the present invention is in the field of telecommunications and, in particular, in the field of digital signal process- ing.
  • Multi-carrier modulation in particular, orthogonal fre ⁇ quency division multiplexing (OFDM) has been successfully applied for transmission in a wide variety of digital com- munication systems.
  • OFDM orthogonal fre ⁇ quency division multiplexing
  • QAM Quadrature Amplitude Modulation
  • cyclic prefix is inserted into a guard interval (GI) of the transmit signal, the guard interval being preferably longer than a maximum delay of the communication channel through which the transmit signal is to be transmitted.
  • Fig. 13 shows, by the way of example, a block diagram of a conventional OFDM receiver.
  • the guard interval is removed from a signal received by an antenna.
  • a Fourier Transform for example a Fast Fourier Transform (FFT) is performed in or- der to obtain a received version of the multi-carrier sig ⁇ nal in frequency domain.
  • FFT Fast Fourier Transform
  • the received signal When transmitting an OFDM modulated signal over a multi- path fading channel, the received signal will have unknown amplitude and phase variations.
  • CTF channel transfer func ⁇ tion
  • phase drift defines the average change of phase between two adjacent sub-carriers. For many applications this phase drift should be as small as possi ⁇ ble.
  • Phase drift in frequency domain results e.g. from frame synchronization errors, i.e. from timing errors introducing a phase shift in spectral domain, the phase shift increas ⁇ ing e.g. linearly over frequency.
  • phase drift .E[A ⁇ ] if the timing offset, T off , does not exceed the guard interval length, T GI , minus the maximum delay of the chan- nel ⁇ max > so T off ⁇ T GI ⁇ ⁇ ma* ' there will be no loss in or ⁇ thogonality of the OFDM signal.
  • T off ⁇ 0 will re ⁇ sult in a different phase drift E[Ap 1 ], as is described in M. Hsieh and C. Wei, "Channel Estimation for OFDM Systems Based on Comb-Type Pilot Arrangement in Frequency Selective Fading Channels," IEEE Transactions on Consumer Electron ⁇ ics, vol. 44, pp. 217-225, Feb. 1998.
  • phase drift may degrade the performance if space fre ⁇ quency codes or differential modulation is used. Further ⁇ more, the phase drift also degrades the performance of polynomial interpolation algorithms, e.g. linear or spline interpolation.
  • Fig. 14 shows a phase and a magnitude of a snapshot of a channel transfer function (CTF) .
  • Fig. 15 shows a corre- spending time domain snapshot of a channel impulse response (CIR) .
  • the non-zero phase drift E[A ⁇ d ] is typical for any OFDM system. It is due to the structure of the channel impulse response (CIR) , which is the inverse Fourier Transform of the CTF.
  • CIR channel impulse response
  • the CIR which generates the CTF in Fig. 14 is shown in Fig. 15.
  • the CIR is only non-zero within the range f ⁇ ,r, 1, where ⁇ ma denotes the maximum delay of the channel, or T O ff being a frame offset.
  • the CIR is related to the CTF by a Fourier Transform it may be viewed as the spectrum of the CTF.
  • a non-zero phase drift can degrade a performance of an OFDM system, for example in a case of a differential phase cod ⁇ ing. This is due to the fact that a phase drift introduces an additional phase term leading to phase errors when per- forming a differential phase demodulation, so that informa ⁇ tion detection errors occur.
  • phase drift can introduce channel estima ⁇ tion errors when estimating the communication channel, for example when estimating the channel transfer function, in order to e.g. equalize the received multi-carrier signal in frequency domain.
  • an OFDM transmitter may introduce so-called pilot symbols be ⁇ ing known in the receiver for channel estimation.
  • the pilot symbols are used for modulating sub-carriers of a OFDM signal to be transmitted.
  • the pilot symbols are removed from the modulated sub-carriers by the means of demodulation, e.g. by dividing the modulated sub- carriers by corresponding pilot symbols in order to obtain sub-carrier values comprising information with respect to the channel transfer function.
  • Fig. 16 a block diagram of pilot symbol based channel estimation for OFDM is depicted.
  • a transformed signal After transforming the re ⁇ ceived signal to frequency domain by the means of the Fast Fourier Transform (FFT) a transformed signal is obtained having values associated with sub-carriers, wherein only certain sub-carriers are modulated by pilot symbols for channel estimation.
  • FFT Fast Fourier Transform
  • a de-multiplexer can be used in order to de- multiplex (DMUX) the modulated sub-carriers, which are, subsequently, provided to a channel estimator being config ⁇ ured for channel estimation.
  • the channel estimator may per ⁇ form the pilot demodulation mentioned above in order to ex ⁇ tract the sub-carrier values, wherein, after having per- formed the de-modulation, the sub-carrier values are esti ⁇ mates of the channel transfer function at frequency points associated with sub-carriers modulated by the pilot sym ⁇ bols.
  • the channel estimator may, for example, be con- figured for performing an interpolation in order to provide channel estimates for all sub-carriers by means of interpo ⁇ lating between two subsequent channel estimates obtained from the modulated sub-carriers being spaced apart by a number of sub-carriers.
  • a detection unit In order to detect information con- tained by the transformed signal bypassed by the de ⁇ multiplexer, a detection unit (DT) is used.
  • the detection unit is configured for receiving the channel estimates in order to equalize the transformed signal. How ⁇ ever, if a phase drift occurs, then the channel estimates are erroneous due to an additional phase term. This leads to a performance degradation while detecting the informa ⁇ tion comprised by the transform signal after having equal ⁇ ized the transformed signal using the erroneous channel es ⁇ timates.
  • a phase compensation can be performed, as is described e.g. in M. Hsieh and C. Wei, "Channel Estimation for OFDM Systems Based on Comb- Type Pilot Arrangement in Frequency Selective Fading Chan ⁇ nels," IEEE Transactions on Consumer Electronics, vol. 44, pp. 217-225, Feb. 1998, for the case of channel estimation.
  • a least squares estimate of pilot sig ⁇ nals is performed in order to provide an estimated channel transfer function.
  • a change in phase caused by frame error is determined from subsequent values of the channel transfer function.
  • the estimated change in phase is removed from the estimated channel transfer function and the resulting channel transfer func ⁇ tion is provided to a minimum mean squared error estimator for further channel estimation.
  • the channel transfer functions of data carriers are interpolated using linear or higher order interpola ⁇ tions.
  • a phase post-compensation is per ⁇ formed where the previously removed phase change is re ⁇ stored in order to provide an interpolated channel transfer function comprising the change in phase.
  • the present invention is based on the finding that the phase drift can efficiently be reduced when a feed-back loop phase drift reduction structure is used. More specifi ⁇ cally, the phase drift can efficiently be reduced when a detector for detecting the phase drift in a signal is ar ⁇ ranged after a phase drift compensating entity in order to detect the phase drift or a remaining phase drift in the signal provided by the phase drift compensating entity and in order to control the phase drift compensating entity in dependence on a phase drift detected in the signal via a 'feed-back loop so that e.g. in each compensation step a further phase drift reduction can be achieved.
  • the functionality of the phase drift compensating entity is comprised by a transformer for time-frequency transforming a time domain signal into a transformed signal in frequency domain.
  • the time domain signal may be a received version of a multi-carrier signal, the received version suffering from e.g. frame synchronization error. Therefore, the transformed signal representing the spectrum of the time domain signal suffers from the phase drift influencing a phase of each sub-carrier.
  • the detector may be configured for directly detecting the phase drift in frequency domain in the transformed signal provided by the transformer.
  • the detector may be configured for detecting the phase drift indirectly in time domain from a pre- processed time-domain signal.
  • the detector may be configured for detecting a delay resulting from a frame synchronisation error, and for calculating the phase drift resulting from the delay sing e.g. a Fourier transform.
  • the transformed signal is provided to a detector connected to an output of the transformer for detecting the phase drift in the transformed signal, wherein the detection of the phase drift is preferably performed in frequency do- main.
  • the detec ⁇ tor is configured for generating a control signal indicat ⁇ ing the phase drift upon detecting the phase drift.
  • the control signal is then fed-back to the transformer which is configured, in response to the control signal, for pre- processing the time-domain signal in time-domain before time-frequency transforming in order to introduce a correc ⁇ tion phase drift to the transformed signal in frequency do ⁇ main, or for post-processing the transformed signal after time-frequency transforming the time domain signal in order to directly introduce the correction phase drift to the transformed signal, the correction phase drift, in both cases, at least partly compensating the phase drift so that the phase drift is reduced.
  • the transformed signal obtained from time-frequency transforming the time domain signal may be post-processed in frequency domain by e.g. manipulating the phase of the transformed signal so that a correction phase drift is in ⁇ troduced in frequency domain to the transformed signal, i.e. so that a correction phase is introduced to the phase of the transformed signal in order to reduce the phase drift.
  • the inventive transformer has the ability of pre-processing the time domain signal in the time domain in order to indirectly introduce the correction phase drift to the transformed signal resulting from transforming the time domain signal into frequency domain after pre-processing.
  • the time delay-phase shift correspondence of a time-frequency transform for example of a Fourier Trans ⁇ form, is exploited since e.g. a time delay always intro- prises an additional phase shift in a spectrum of a time de ⁇ layed signal.
  • the detector is configured for detect- ing the phase drift in the transformed signal, the trans ⁇ formed signal resulting from transforming the time domain signal into frequency domain signal after pre-processing the time domain signal or resulting from transforming the time domain signal into frequency domain. Therefore, the phase drift can also adaptively be reduced, wherein, in a plurality of reduction steps, a phase drift remaining after a previous compensation step may be detected and reduced.
  • the detector receives the transformed signal comprising the uncompensated phase drift. Then, the detector determines, on the basis of the unprocessed transformed signal, the phase drift comprised by the transformed signal and generates a control signal, which is provided to the transformer for controlling the pre-processing or post-processing or both. In response to the control signal comprising e.g. information on the de ⁇ tected phase drift, the transformer is configured for pre ⁇ processing or post-processing or pre- and/or post ⁇ processing the transformed signal in order to reduce the phase drift. In a next step, the detector receives the processed transformed signal comprising reduced phase drift.
  • the detector detects the remaining phase drift comprised by the (processed) transformed signal provided by the transformer, generates a further control signal indicating the remaining phase drift and feeds the control signal back to the transformer for further reduc ⁇ tion of the remaining phase drift and so forth.
  • the phase drift can effectively be reduced when the transformer applies all inventive processing func ⁇ tionalities for phase drift reduction.
  • the transformer may be configured for coarsely reducing the phase drift by the means of pre-processing the time domain signal before time frequency transforming, for transforming the time domain signal after pre-processing and for post ⁇ processing the resulting transformed signal in frequency- domain for obtaining a fine phase drift reduction, or vice versa.
  • Fig. 1 shows a block diagram of an apparatus for reduc ⁇ ing a phase drift in accordance with an embodi ⁇ ment of the present invention.
  • Fig. 2a shows the inventive post-processing approach
  • Fig. 2b shows the inventive pre-processing approach
  • Fig. 3a shows a phase of a channel transfer function
  • Fig. 3b shows a magnitude of a channel transfer function
  • Fig. 4 shows a corresponding spectrum of the channel transfer function of Figs. 3a and 3b;
  • Fig. 5 shows a block diagram of an apparatus for reduc ⁇ ing a phase drift in accordance with a further embodiment of the present invention
  • Fig. 6 shows a block diagram of an apparatus for reduc- ing a phase drift in accordance with a further embodiment of the present invention
  • Fig. 7 shows OFDM system parameters
  • Fig. 8 shows a power delay profile of a channel
  • Fig. 9 demonstrates a performance of the inventive ap ⁇ proach
  • Fig. 10 shows an effective channel impulse response re ⁇ sulting when cyclically shifting a received sig ⁇ nal in time domain
  • Fig. 11 demonstrates the performance of the inventive ap ⁇ proach
  • Fig. 12 shows a block diagram of the inventive apparatus for reducing a phase shift in accordance with a further embodiment of the present invention
  • Fig. 13 shows a block diagram of an OFDM receiver
  • Fig. 14 shows a phase and a magnitude of a channel trans ⁇ form functions
  • Fig. 15 shows a corresponding time domain channel impulse response
  • Fig. 16 shows a channel estimation approach.
  • Fig. 1 shows a block diagram of the inventive apparatus for reducing a phase drift in accordance with an embodiment of the present invention.
  • the apparatus shown in Fig. 1 comprises a transformer 101, the transformer 101 having an input 103, a control input 105 and a plurality of outputs 107 connected to a plurality of inputs of a detector 109.
  • the detector 109 comprises a plurality of outputs 111 and a control output 113, the con ⁇ trol output 113 being coupled to the control input 105 of the transformer 101.
  • the transformer 101 is configured for receiving a time- domain signal via the input 103. If, for example, during frame synchronization being performed in order to obtain the time-domain signal an error occurs, then a spectrum of the time-domain signal comprises a phase drift superimpos ⁇ ing the phase of the spectrum of the time-domain signal.
  • the phase drift may be described as a linearly increasing phase ramp superimposing the phase of the spec- trum of the time domain signal. In this case, the phase change between two subsequent spectral values is constant.
  • the transformer is configured for transforming the time-domain signal into a transformed sig ⁇ nal in frequency domain, the transformed signal represent ⁇ ing a spectrum of the time-domain signal.
  • the transformed signal in frequency domain comprises a num ⁇ ber of spectral coefficients which are provided via the plurality of outputs 107 to the detector 109 for detecting the phase drift in the transformed signal.
  • the detector 109 is configured for detecting the phase drift in the trans- formed signal, for generating a control signal indicating the phase drift and to providing the control signal via the control output 113 to the transformer 101.
  • the transformer 101 is configured for pre-processing the time domain signal before time-frequency transforming or for post-processing the transformed signal after time-frequency transforming the time domain signal in order to introduce a correction phase drift to the trans ⁇ formed signal, the correction phase drift at least partly compensating the phase drift for phase drift reduction.
  • the transformer may comprise a delay element for delaying the time-domain sig ⁇ nal so that, upon exploiting the time delay/frequency shift correspondence the correction phase shift is introduced.
  • the transformer comprises, as a delay element, a cyclic shift element for pre-processing the time-domain signal, wherein the cyclic shift element is configured for receiving the control signal from the detector 109 and, in response to the control signal, for cyclically shifting the time domain signal in order to introduce the correction phase drift to the transformed signal.
  • the cyclic shift element is configured for cy ⁇ tunally shifting the time-domain signal by a number of values, wherein the number of values depends on the phase drift detected by the detector.
  • the detector 109 is configured for determining the number of values the time-domain signal is to be shifted by using the detected phase drift and exploiting the cyclic shift/frequency shift property of a time-frequency transform, for example of a Fourier Transform.
  • the control signal provided to the transformer may in this case comprise the information on the number of values the time domain signal is to be shifted by.
  • the detector is configured for detecting the phase shift from the transformed signal and for providing infor ⁇ mation on the detected phase drift to the transformer.
  • the transformer in response to the received information on the phase drift, may be configured for determining the number of values the time-domain signal is to be shifted by in ex- actly the same way as described above in connection with the detector 109.
  • the transformer comprises a Fourier transformer coupled to an output of the delay elements, e.g. to an out- put of the cyclic shift element, for transforming the time- domain signal provided by the cyclic shift element into the transformed signal.
  • the plurality of outputs 107 comprised by the transformer 101 corresponds to a plurality of outputs of the Fourier transformer, the Fourier transformer outputting the transformed signal comprising a number of spectral val- ues, wherein a number of the outputs 107 corresponds to the number of spectral values.
  • the plurality of inputs of the detector is coupled to the plurality of out ⁇ puts of the Fourier transformer.
  • the plurality of outputs of the Fourier transformer is directly connected with the plurality of inputs of the detector 109.
  • the control output 113 of the detector 109 may be connected to a control input comprised by the cyclic shift element, so that the detector 109 directly controls the pre- processing of the time-domain signal.
  • either the detector 109 or the transformer 101 may be configured for determining the num ⁇ ber of values the time-domain signal is to be shifted by.
  • the number of values is determined based on an expectation value for a phase drift, which expectation value is calculated in frequency domain by, for example, averaging over a phase change between two subsequent sub- carriers.
  • the expectation value for the phase drift may be multiplied by a number of values comprised by the transformed signal and divided by a factor of 2 ⁇ , wherein the number of values of the transformed signal corresponds for example to a number of sub-carriers associated with a multi-carrier transmission scheme.
  • the cyclic shift element is configured for performing a left shift or a right shift in dependence on a sign of the phase drift.
  • the cyclic shift ele- ment is configured for performing a left shift in order to introduce a positive correction phase drift to the trans ⁇ formed signal for compensating a negative phase drift, or for performing a right shift in order to introduce a nega- tive correction phase drift to the transformed signal for compensating a positive phase drift.
  • the cyclic shift element may comprise a shift register, the shift register having an input for receiving the time-domain signal and an output coupled to the input, so that a cyclic shift can be performed.
  • the transformer may be config ⁇ ured for post-processing the transformed signal in order to introduce the correction phase drift directly in the fre ⁇ quency domain for compensating the phase drift.
  • the transformer comprises a phase compensator for post-processing the transformed signal in frequency domain, the phase compensator being configured for changing a phase of the transformed signal in frequency domain in order to introduce the correction phase drift.
  • the phase compensator is configured for multiplying each trans- formed signal value by a complex value for phase shifting the transformed signal values in frequency domain.
  • the transformer 101 comprises either the cyclic shift ele- ment for pre-processing the time domain signal or the phase compensator for post-processing the transformed signal.
  • the transformer may comprise a Fourier transformer for time-frequency transforming the time domain signal, wherein the phase compensator is connected to an output or to a plurality of outputs of the Fourier trans ⁇ former, wherein the detector 109 is connected to an output or to a plurality of outputs of the phase compensator, so that the detector 109 only receives the transformed signal provided by the phase compensator, wherein .
  • the phase com- pensator is configured for receiving the control signal generated by the detector 109 upon detecting a phase drift in the transformed signal.
  • the phase compensator may comprise a control input for receiving the control signal from the detector 109.
  • the control input of the phase compensator is di ⁇ rectly connected to the control input 105 of the trans ⁇ former 101.
  • the compensator may be configured for adding the correction phase drift to a phase of the transformed signal for com ⁇ pensating a negative phase drift, or for subtracting the correction phase drift from the phase of the transformed signal for compensating a positive phase drift.
  • the correction phase drift to be introduced by the phase compensator may be determined by the detector 109 upon de- tecting the phase drift. Furthermore, the correction phase drift may be determined by the transformer 109 in response to information on the phase drift provided by the detector 109. For example, the correction phase drift to be intro ⁇ pokerd by the phase compensator corresponds to the phase drift detected by the detector 109, except for the sign.
  • the transformer may comprise both: the cyclic shift element for pre-processing the time domain signal and the phase compensator for post-processing the transformed signal for two-stage reduction of the phase drift.
  • the cyclic shift element cyclically shifts the time domain sig ⁇ nal in order to coarsely reduce the phase drift.
  • the time domain signal provided by the cyclic shift element is then transformed into frequency domain by e.g. a Fourier trans- former coupled to the cyclic shift element in order to ob ⁇ tain the transformed signal.
  • the phase compensator is cou ⁇ pled to an output or to a plurality of outputs of a Fourier transformer for post-processing the transformed signal, wherein an output or a plurality of outputs of the phase compensator is coupled to the detector 109 for detecting the phase drift.
  • the detector 109 simultane ⁇ ously controls the cyclic shift element and the phase com ⁇ pensator for phase drift reduction.
  • the cyclic shift element is configured for re ⁇ ceiving the control signal from the detector, wherein the control signal indicates a number of values the time domain signal is to be shifted by in order to introduce a correc ⁇ tion phase drift to the transformed signal for coarsely compensating the phase drift.
  • the cyclic shift element may be configured for cy ⁇ tunally shifting the time domain signal by the number of values in order to coarsely compensate the phase drift.
  • the phase compensator may be configured for receiving a further control signal from the detector 109, the further control signal indicating a further correction phase drift to be introduced to a phase of the transformed signal for a fine compensation of the phase drift.
  • the phase compensator may be configured for introducing the further correction phase drift to the phase of the transformed signal.
  • control signal and the further control signal may be comprised by a common control signal generated by the detector 109 upon detecting phase drifts affecting the transformed signal.
  • the number of values the time domain signal is to be shifted by or the correction phase drift to be added to a phase of the transformed signal may be determined by the detector 109 upon detecting the phase drift or by the transformer 101 upon receiving a control signal indicating a detected phase drift.
  • the detector 109 may be configured for determining an average change in phase be- tween two subsequent values of the transformed signal, wherein the average change in phase represents the phase drift.
  • the transformed signal may comprise all spec ⁇ tral values resulting from transforming the time domain signal into frequency domain.
  • the transformed signal may be composed of a set of spectral values associated with sub-carriers being, in a transmitter, modulated by pilot symbols for channel estimation purposes. For example, two subsequent sub-carriers being modulated by pilot symbols may be spaced apart by a number of sub-carriers being used e.g. for information transmission.
  • the detec ⁇ tor is operative for detecting the phase drift from the spectral values associated with sub-carriers being modu ⁇ lated by the pilot symbols.
  • the transformer may further comprise a selector, the selector being configured for selecting the sub-carriers modulated by the pilot symbols from spectral values obtained from time-frequency transforming the time- domain signal.
  • the set of spectral values representing the sub-carriers modulated by the pilot sym- bols constitutes the transformed signal.
  • the transformer may further be configured for demodulating the set of spectral values constituting the transformed signal using pilot sym ⁇ bols wherein, by way of example only, each spectral value may be divided by an associated pilot symbol. Alterna ⁇ tively, each spectral value may be multiplied by a conju ⁇ gate complex version of the associated pilot symbol.
  • the detector 109 may be configured for determining an average value of a product between two subsequent values of the transformed signal, wherein one of these values may complex conjugated. For ex ⁇ ample, the detector 109 may be configured for determining an average value of a product between a value associated with a certain sub-carrier and a complex conjugate version of a value associated with a sub-carrier preceding the sub- carrier. In order to determine the phase drift, the detector 109 may be configured for determining a phase of the average value, the phase of the average value representing the phase drift.
  • the phase of the average value is preferably defined by a spacing, the spacing indicating a number of sub- carriers separating two subsequent sub-carriers used for pilot symbol transmission.
  • the detector may be configured for determining an av ⁇ erage phase change between two values of the transformed signal, the two values being spaced apart by a number of values, and for dividing the average phase change by the number of values in order to detect the phase shift for each value of the transformed signal.
  • This case may, for example, correspond to the above-mentioned case when only certain sub-carriers are used for pilot symbol transmis- sion, and, unlike in the above embodiment, the transformer does not comprise any selector.
  • the phase drift may be also derived from spectral values of the transformed signal, which are not associated with sub-carriers being used for pilot symbol transmission. In any case, a complex- ity reduction can be introduced since only a certain subset of values of the transformed signal will be used for de ⁇ tecting the phase drift.
  • the present invention further provides a multi-carrier re- ceiver comprising the apparatus for reducing a phase drift as has been described above.
  • the apparatus for reducing the phase drift is configured for reducing the phase drift in a receivable multi-carrier signal in time domain and for providing a transformed signal representing a receivable multi-carrier signal in frequency domain, the transformed signal having reduced phase drift.
  • the inventive appara- tus for reducing the phase drift is utilized for transform ⁇ ing the receivable multi-carrier signal in time domain into frequency domain in accordance with a multi-carrier receiv ⁇ ing scheme, for example OFDM, and simultaneously, for re- ducing a phase drift.
  • the multi-carrier receiver may comprise means for extracting information comprised by the transformed signal.
  • the multi-carrier receiver may comprise a synchro ⁇ nizer for performing a frame synchronization in time do ⁇ main, wherein the phase drift results from a frame synchro- nization error due to an erroneous sample time instant be ⁇ ing considered the time instant determining a beginning of a frame in time domain.
  • the de- tector 109 comprised by the apparatus for reducing the phase drift may be configured for generating a frame syn ⁇ chronization control signal indicating the phase drift, and providing the frame synchronization control signal to the synchronizer.
  • the synchronizer may be configured, in re- sponse to the frame synchronization control signal, for ad ⁇ justing a sample time instant in order to reduce the frame synchronization error. Therefore, the detector may also control the operation of the synchronizer so that frame synchronization errors are reduced in dependence on a de- tected phase drift, wherein the synchronizer is arranged before the inventive transformer for performing a frame synchronization in time domain.
  • the means for extracting information may comprise a differential de-modulator for differentially de-modulating the transformed signal in order to extract information from a phase change between two subsequent values of the trans ⁇ formed signal.
  • the means for extracting information may comprise a low-pass filter for filtering the transformed signal in frequency domain.
  • the low-pass filter comprises real valued coefficients representing a real part of a fil ⁇ ter response, wherein filter coefficients representing the imaginary part of the filter response are set to zero.
  • the transformer is further configured for pre ⁇ processing the receivable multi-carrier signal in time do ⁇ main or for post-processing the transformed signal in order to introduce a phase shift to the transformed signal for shifting a spectrum of the transformed signal towards the passband of the low-pass filter.
  • frequency domain filtering using real valued filters having symmetrical two-sided filter transfer func ⁇ tion can be performed which further reduces a receiver's complexity.
  • the low-pass filter comprised by the means for ex ⁇ tracting information may be configured for channel estima- tion in order to extract channel information from, for ex ⁇ ample, the receivable multi-carrier signal in time domain is a receivable version of a transmit signal being trans ⁇ mitted through a communication channel.
  • the low-pass filter is a Wiener filter for channel estimation.
  • the low-pass filter is an interpolation filter for interpolating between estimated values of the channel transfer function, when the estimated values of the channel transfer function in frequency domain are associated with sub-carriers which are spaced apart by a number of sub- carriers. Tn this case, the interpolation process is per- formed in order to obtain estimates of intermediate values of the channel transfer function.
  • the means for extracting information comprises a space frequency block decoder for space-frequency block de ⁇ coding the transformed signal in frequency domain when, in a receiver, space-frequency block coding is employed.
  • Fig. 2a shows an OFDM receiver with phase compensation af ⁇ ter Fourier Transform
  • the receiver comprises an antenna 201 coupled to a means 203 for removing guard in- terval, the means 103 for removing the guard interval being coupled to a serial to parallel converter 205 (S/P) .
  • the serial to parallel converter 205 has a plurality of outputs coupled to a plurality of inputs of a Fourier transformer 207, the Fourier transformer 207 being configured for per- forming a Fast Fourier Transform (FFT) .
  • the Fourier trans ⁇ former 207 has a plurality of outputs coupled to a phase compensator 209, the phase compensator 209 having a plural ⁇ ity of inputs being coupled to a detector not shown in Fig. 2a. It is to be noted that the Fourier transformer 207 and the phase compensator 209 are comprised by the inventive transformer mentioned above.
  • Fig. 2b shows a block diagram of an OFDM receiver having cyclic shift before performing the Fourier Transform by the Fourier transformer 207.
  • the cyclic shift is performed by a cyclic shift element 301 coupled between the means 203 for removing the guard interval and the serial to parallel con ⁇ verter 205.
  • the cyclic shift element 301, the serial to parallel converter 205 and the Fourier transformer 207 constitute the inventive transformer, wherein the plurality of outputs of the transformer 207 is coupled to the detector, which is not shown in Fig. 2b.
  • the detector may be configured for controlling the phase com ⁇ pensator 209 and the cyclic shift element 301.
  • Figs. 2a and 2b the solutions shown therein are applicable to a wide range of OFDM receivers due to the OFDM standard conform structure. Moreover, if the phase compensation is applied according to the embodi ⁇ ment of Fig. 2a, there is no need to compensate the induced phase shift after channel estimation and interpolation, contrary to the prior art approaches mentioned above.
  • the negative effects of a non-zero phase drift from (1) can be compensated by a phase compensation unit after the FFT at the OFDM receiver, as is shown in Figs. 2a and 2b.
  • Cy ⁇ tract shifts in the receiver are known from M. I. Rahman, K. Witrisal, D. Prasad, O. Olsen, and R. Prasad, "Performance Comparison between MRC Receiver Diversity and Cyclic Delay Diversity in OFDM WLAN Systems, "in Proc. Int. Symposium on Wireless Personal Multimedia Communications (WPMC'03), Yo- kosuka, Japan, Oct. 2003 and from A. Damrtiann and S.
  • Fig. 3a shows a phase of a channel transfer function (CTF) after cyclically shifting a time domain signal before FFT.
  • Fig. 3b shows a corresponding magnitude of the channel transfer function.
  • a phase shift according to Fig. 2a is more compu ⁇ tationally complex. While a cyclic shift can be performed very efficiently using a shift register of — S cyc samples, a phase compensation of ⁇ cyc degree per sub-carrier requires one multiplication by e j ⁇ cyai per sub-carrier. On the other hand, given the overall complexity of an OFDM receiver, one additional multiplication per sub-carrier may not be that significant.
  • the cyclic shift only changes the phase of the CTF, the magnitude remains unaffected. This can be checked by comparing the CTF of an OFDM signal without and with cyclic shift shown in Figs. 14, 3a and 3b. Since the effects of the frequency selective channel are compensated by the channel estimator anyway, no other operations are neces- sary.
  • FIG. 3a A snapshot of the magnitude and phase of the CTF and the corresponding CIR after cyclically shifting the received signal is shown in Figs. 3a, 3b and Fig. 4, respectively. While there are still strong variations in amplitude and phase due to frequency selective fading, the phase drift E[A ⁇ 1 ] is compensated. The effective CIR of Fig. 4 is shifted towards negative delays. Instead of a one-sided spectrum, the received signal now has a two-sided spectrum.
  • the cyclic shift at the receiver side can also be applied to channel estimation based on the discrete cosine trans ⁇ form (DCT) . Since the DCT operates on a two-sided spectrum, the cyclic shift before OFDM demodulation may be very bene- ficial.
  • DCT discrete cosine trans ⁇ form
  • the coefficients of FIR interpolation and/or smoothing filters will be real valued. In general a real valued filter will only have half the computational cost of a complex valued filter.
  • the filter W ⁇ i / '" / W M j is designed such that it covers a great variety of power delay pro ⁇ files. Accordingly, a rectangular shaped power delay pro ⁇ file with maximum delay T w fulfils this requirement.
  • a complexity of a filtering operation or of a channel estimation operation when using filter having com ⁇ plex valued coefficients is insignificantly increased when compared with filtering or channel estimation using filters having real valued coefficients only.
  • T w T GJ .
  • the WIF matched to the uniform power delay pro ⁇ file [-T w /2,T w /2] is also real valued. This means that the computational cost is cut by half.
  • any FIR low pass interpo- lation filter benefits from the proposed cyclic shift.
  • the filter is matched to a passband of the filter will be real valued.
  • the per ⁇ formance will be optimum if the signal to be filtered passes through the filter unchanged.
  • Fig. 5 shows an OFDM receiver with phase compensation after the FFT in accordance with a further embodiment of the pre ⁇ sent invention.
  • the OFDM receiver shown in Fig. 5 comprises a phase compensator 501 having a plu ⁇ rality of outputs coupled to a detector 503, the detector 503 having a control output 505 coupled back to a control input of the phase compensator for providing information on the detected phase drift ⁇ cyc -
  • Fig. 6 shows an OFDM receiver with cyclic shift before the FFT.
  • the OFDM re ⁇ DCVER shown in Fig. 6 comprises a cyclic shift element 601 coupled between the means 203 for removing guard interval and the serial to parallel converter 205, and a detector 603 coupled to the plurality of outputs of the Fourier transformer 207, wherein the detector 603 comprises a con ⁇ trol output 605 coupled back to a control input of the cy ⁇ tun shift element 601 for providing information on the number of values the time domain signal provided by the means 203 for removing the guard interval is to be shifted by.
  • the cyclic shift may be chosen to compensate the phase drift of (1) , that is
  • OFDM systems with differential modulation or space- frequency coded OFDM systems will have an improved perform- ance if E[ ⁇ ] is estimated sufficiently well.
  • ap ⁇ pears attractive as shown in Fig. 5.
  • this solu ⁇ tion can be implemented as shown in Fig. 6.
  • the cyclic shift, ⁇ cyc may be set to a default value within the range [ ⁇ , T GI /2] .
  • ⁇ cyc can be estimated and fed back to the cyclic shifting unit. Note, the phase drift is expected to change only on a long term basis, so no frequent updates are necessary.
  • the system parameters of the OFDM system and of the channel model are shown in Fig. 7.
  • the total transmit power of the system is fixed, such that the total transmit power of a N ⁇ antenna system is equivalent to a single antenna sys ⁇ tem. No outer channel coding has been employed.
  • the BER floor can be somewhat reduced by optimizing the cyclic shift ⁇ cyc .
  • the optimum cyclic shift is about 44T spi , which result in the best performance. It can also be seen that the accuracy of 6 cyc does not need to be high.
  • a performance of a polynomial interpolator can significantly be optimized.
  • PACE interpolation in frequency and time direction is necessary. While in time direction the Doppler power spec ⁇ trum has in general a symmetric two-sided and real valued profile (at least approximately) , the power delay profile is real valued but one-sided.
  • S ⁇ 0 the benefit of inserting a cyclic shift, S ⁇ 0 will be described for a linear interpolator. The results are applicable for higher order polynomial interpolators as well.
  • a performance of a polynomial interpolator can significantly be optimised.
  • two successive pilot sub-carriers are used to determine the channel response for sub-carrier located in between these two pilots.
  • sub-carrier i the channel estimate is given by
  • Fig. 10 shows a time domain channel impulse response snap shot after cyclically shifting the signal before the FFT. In Fig. 10, also a frequency response of a linear interpo ⁇ lator is shown.
  • Possible applications of the proposed cyclic shift at the OFDM receiver are e.g.:
  • orthogonal frequency division multiplex ⁇ ing OFDM
  • the signal stream is divided into N c parallel sub-streams, typically for any multi-carrier modulation scheme.
  • An inverse DFT with N FFT points is performed on each block, and subsequently the guard interval having N GI sam ⁇ ples is inserted to obtain x lrn .
  • the signal x(t) is transmitted over a mobile radio channel with response h(t, ⁇ ) .
  • n(t) represents additive white Gaussian noise
  • the guard interval is removed and the information is recovered by performing a DFT on the re ⁇ ceived block of signal samples, to obtain the output of the OFDM demodulation Y t/i .
  • the received signal after OFDM de ⁇ modulation is given by
  • X l ⁇ and H 11 denotes the transmitted information sym ⁇ bol and the channel transfer function (CTF) at sub-carrier i of the £ th OFDM symbol, respectively.
  • CTF channel transfer function
  • N ti ac ⁇ counts for additive white Gaussian noise (AWGN) with zero mean and variance W 0 . It is assumed that the transmitted signal consists of L OFDM symbols, each having N c sub- carriers.
  • the guard interval is longer than the maximum delay of the channel, i.e. T GI > ⁇ max , the orthogonality at the re- ceiver after OFDM demodulation is maintained, and the re ⁇ ceived signal of (11) is obtained.
  • the FFT translates a cyclic delay into phase shifts.
  • the effective CTF of the cyclic receiver is described by
  • the received signal Y 1 of (11) is obtained.
  • the received signal at the pilot positions are de-multiplexed from the data stream, to obtain the received pilot sequence
  • G is the subset of the OFDM frame containing the pi ⁇ lots.
  • the first step in the channel estimation process is to re ⁇ move the modulation of the pilot symbols, which provides an initial estimate of the CTF at pilot positions
  • the channel estimator uses the demodulated pilots H 1 from (17) to yield the channel estimate where M f denotes the filter order, i.e. the number of co ⁇ efficients of the FIR filter W 1n .
  • the FIR filter W [W 0 , • •• r W M ⁇ 1 J may be implemented as e.g. low-pass interpolation filters, polynomial interpolators, or Wiener interpolation filters.
  • the Wiener interpolation filter minimizes the mean squared error (MSE) between the desired response H 1 and the observation, i.e. the received pilot symbols. This means that knowledge about the channel statistics is required. In contrast, low-pass interpolation filters and polynomial interpolators do not assume any knowledge of the channel statistics.
  • the observed channel is typically correlated in two dimensions, frequency and time.
  • the extension to PACE in two dimensions is possible.
  • the Wiener interpolation filter (WIF) is implemented by a FIR filter with M f taps, according to (18) .
  • the WIF, W'[Ai] is obtained by solving the Wiener-Hopf equation
  • the filter W is designed such that it covers a great variety of power delay profiles. For exam ⁇ ple, a rectangular shaped power delay profile with maximum delay T w fulfils this requirement.
  • This assumption provides the frequency correlation function of the mismatched esti- mator, E ⁇ yc) [ ⁇ i], from (6) .
  • the mismatched estimator is de ⁇ termined by substituting from (6) into (21) and (23) . Then the Wiener-Hopf equation (19) needs to be deter ⁇ mined only once.
  • the filter coefficients can be pre-computed and stored.
  • the parameters of the robust estimator should always be equal or larger than the worst case channel conditions, i.e. largest propagation delays and maximum expected velocity of the mobile user.
  • the average SNR at the filter input, ⁇ w which is used to generate the filter coefficients, should be equal or larger than actual average SNR, so ⁇ n ⁇ ⁇ c .
  • T n the maximum delay of the channel ⁇ max is not known it can be upper bounded by the guard interval dura ⁇ tion T GI . Since the filter should also satisfy the sampling theorem, the filter pass-bands can be chosen within the range
  • phase drift specifies the average change in phase between two adjacent sub-carriers, as defined in (1) .
  • One possibility to estimate the phase drift is f 1 w 1
  • phase drift may be estimated by
  • Fig. 12 shows a further embodiment of an OFDM receiver in accordance with the present invention.
  • the OFDM receiver shown in Fig. 12 comprises a de-multiplexer 1201 for de- multiplexing sub-carriers being modulated by pilot symbols for channel estimation.
  • the de-multiplexer 1201 (DMUX pilots) is configured for demodulating the sub- carriers being modulated by the pilot symbols.
  • the de ⁇ multiplexer has an output coupled to the inventive appara- tus 1203 for reducing the phase drift.
  • the apparatus 1203 for reducing the phase drift is coupled to a channel esti ⁇ mator 1205 being configured for estimating the channel transfer function in frequency domain.
  • the channel estima ⁇ tor is coupled to means 1207 being configured for introduc- ing back the compensated phase shift so that all channel influences may be taken into account.
  • the means 1207 for phase post-compensation is coupled to a means 1209 for ex ⁇ tracting information comprised by a transformed signal pro ⁇ vided by the FFT 207.
  • the means 1209 for extracting information is configured for determining an in ⁇ formation amount comprised by a signal provided by the de ⁇ multiplexer 1201.
  • phase drift compensator may be configured for providing a signal containing information on the phase drift to the means 1207 for phase post-compensation, so that in response to the signal provided by the apparatus 1203, phase post- compensation can be performed.
  • the present invention provides concepts for fil- tering and interpolation for OFDM.
  • the frequency response of the received signal after OFDM demodulation has a one-sided spectrum.
  • the one-sided spectrum can be trans ⁇ formed into a symmetric two-sided spectrum.
  • the performance of standard interpolation algorithms such as linear or spline interpolation can be improved if the cy ⁇ barn shift is appropriately chosen.
  • the per ⁇ formance of space frequency codes and differential modula ⁇ tion can also be improved.
  • the inventive methods can be im ⁇ plemented in hardware or in software.
  • the implementation can be performed using a digital storage medium, in par ⁇ ticular a disk or a CD having electronically readable con- trol signals stored thereon, which can cooperate with a programmable computer system such that the inventive meth ⁇ ods are performed.
  • the present invention is, therefore, a computer program product with a program code stored on a machine-readable carrier, the program code be- ing configured for performing at least one of the inventive methods, when the computer program products runs on a com ⁇ puter.
  • the inventive methods are, there ⁇ fore, a computer program having a program code for perform ⁇ ing the inventive methods, when the computer program runs on a computer.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Synchronisation In Digital Transmission Systems (AREA)

Abstract

La présente invention concerne un appareil permettant de réduire une dérive de phase dans un spectre d'un signal de domaine temporel, qui comprend un transformateur (101) permettant une transformation temps-fréquence du signal de domaine temporel en signal transformé dans le domaine de fréquence, ce signal transformé représentant un spectre du signal de domaine temporel, un détecteur (109) permettant de détecter la dérive de phase, ce détecteur (109) étant aussi configuré pour générer un signal de commande indiquant la dérive de phase, le transformateur (101) étant aussi configuré, en réponse au signal de commande, pour prétraiter le signal de domaine temporel avant la transformation temps-fréquence ou pour traiter a posteriori le signal transformé après la transformation temps-fréquence du signal de domaine temporel afin d'introduire une dérive de phase de correction au signal transformé pour une réduction de dérive de phase, la dérive de phase de correction compensant au moins partiellement la dérive de phase.
PCT/EP2004/009373 2004-08-20 2004-08-20 Appareil et procede de reduction de derive de phase WO2006018035A1 (fr)

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EP04764355A EP1782594A1 (fr) 2004-08-20 2004-08-20 Appareil et procede de reduction de derive de phase
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WO2010022007A2 (fr) * 2008-08-20 2010-02-25 Qualcomm Incorporated Estimation de pilote sdma de liaison montante
WO2011094599A1 (fr) * 2010-01-29 2011-08-04 Qualcomm Incorporated Réduction des erreurs de phase sur un dispositif de communication
WO2012134677A2 (fr) * 2011-03-29 2012-10-04 Intel Corporation Symétrisation d'une réponse impulsionnelle de canal
CN109120347A (zh) * 2018-09-19 2019-01-01 南京信息工程大学 一种时频动态变化的多频多概率光载毫米波产生方法

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US20100182899A1 (en) * 2009-01-17 2010-07-22 Qualcomm Incorporated OFDM Time Basis Matching With Pre-FFT Cyclic Shift
CN102158436B (zh) * 2010-02-11 2014-01-08 富士通株式会社 信道频域相关性计算方法及装置、信道估计方法及装置
CN102387110B (zh) * 2010-09-06 2014-11-05 日电（中国）有限公司 用于生成导频序列的设备和方法
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WO2010022007A2 (fr) * 2008-08-20 2010-02-25 Qualcomm Incorporated Estimation de pilote sdma de liaison montante
WO2010022007A3 (fr) * 2008-08-20 2010-09-23 Qualcomm Incorporated Estimation de pilote sdma de liaison montante
US8654911B2 (en) 2008-08-20 2014-02-18 Qualcomm Incorporated Uplink SDMA pilot estimation
WO2011094599A1 (fr) * 2010-01-29 2011-08-04 Qualcomm Incorporated Réduction des erreurs de phase sur un dispositif de communication
US9031122B2 (en) 2010-01-29 2015-05-12 Qualcomm Incorporated Reducing phase errors on a communication device
WO2012134677A2 (fr) * 2011-03-29 2012-10-04 Intel Corporation Symétrisation d'une réponse impulsionnelle de canal
WO2012134677A3 (fr) * 2011-03-29 2012-11-22 Intel Corporation Symétrisation d'une réponse impulsionnelle de canal
CN109120347A (zh) * 2018-09-19 2019-01-01 南京信息工程大学 一种时频动态变化的多频多概率光载毫米波产生方法
CN109120347B (zh) * 2018-09-19 2021-02-09 南京信息工程大学 一种时频动态变化的多频多概率光载毫米波产生方法

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