Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2614—Peak power aspects
  • H04L27/2615—Reduction thereof using coding
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2602—Signal structure
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2614—Peak power aspects
  • H04L27/2621—Reduction thereof using phase offsets between subcarriers

Definitions

• the present invention relates to a method and a device for encoding and decoding for multi-carrier transmission.
• a multicarrier transmission method is known as a modulation method excellent in multipath fusing resistance. This method divides the transmission band into multiple carriers (called subcarriers) to obtain a frequency diversity effect for frequency selective fusing and enables high-quality wireless transmission. is there.
• Orthogonal frequency division multiplexing (OFDM) technology is one form of this.
• Solutions to this problem can be broadly divided into (1) limitations on input signals and (2) limitations on output signals.
• the former is based on the encoding process It does not generate a signal pattern with a large peak power, and does not cause any deterioration in characteristics. Furthermore, if these codes can extend the minimum distance, it is possible to improve the reception characteristics (bit error rate, BER).
• BER bit error rate
• the latter takes advantage of the small probability of signal peaks that generate peak power.For example, if a peak power exceeding a certain threshold value occurs, it is expressed as a threshold value. Forcibly cutting things, such as clipping, are applicable.
• This technology naturally causes an increase in the sidelobe level due to nonlinear distortion, that is, degradation of characteristics by causing interference between carriers.
• There is also a method of normalizing the entire envelope level of the signal to a threshold value but this eventually degrades the SN, and similarly degrades the characteristics. For realizing wideband and high-quality wireless transmission, the former method is more desirable.
• a complementary sequence As a peak suppression code, a complementary sequence (Complementary code) is well known, and its application to a multicarrier modulation scheme is being studied. This code can be applied to polyphase modulation (Mary PSK, PSK).
• the present invention increases N while maintaining high quality by suppressing peak power and increasing the minimum distance between codes. Achieve high-efficiency transmission by providing codes with higher coding rates
• k is an integer equal to or greater than 1 based on the input signal
• 2k Determine a plurality of phases including at least one kernel consisting of the first to fourth phases satisfying a phase condition that I is a predetermined value, and determine a first phase for a plurality of carrier frequencies.
• Ri third phase is split the difference between the second 2 Bok phase is assigned 'number of carriers frequency has the same frequency interval and the first phase for 2 Bok one conveying wave frequencies assigned 2 a fourth phase for 2 Bok 1 carrier frequencies devoted was assigned k - 1 so as to satisfy the frequency condition that equal to the difference of the carrier frequency, said by assigning said plurality of phases
• a multi-carrier encoding method is provided that includes generating a code corresponding to an input signal.
• the plurality of input signals based on the input signal, k and integer of 1 or more, 2 k - 2 1 of the second phase of the phase difference for one of the first phase ( and 2 k), 2 k - 1 pieces of the absolute value of the difference between the phase Sa ⁇ 0 * (2 k) of the third 2 against the phase of the k 1 pieces of the fourth phase
• I is determining a plurality of phases containing one or more first to kernel consisting of the fourth phase satisfy matter phase condition that is a predetermined value, a plurality against carrier frequency difference between the first phase assignment was 2 Bok one 2 k 1 carrier frequencies to second phase is assigned for carrier frequency first phase and the same frequency the difference between the third 2 Bok one carrier frequency fourth phase is assigned against two single carrier frequency phase is assigned with intervals
• a plurality of codes corresponding to the input signal are generated, and a code distance between each of the plurality of codes and the received code is calculated.
• a method for decoding a multi-carrier code comprising decoding an received code by determining an input signal that provides a code having a minimum code distance from the received code.
• the present invention based on an input signal, k and integer of 1 or more, 2 k - the phase difference between the 'number of the first of the 2 k 1 pieces to the phase the second phase delta ⁇ (2 k) If the absolute value I ⁇ e (2 k) of the difference between the 2 k 'number of the third phase difference of 2 Bok one fourth phase relative to the phase ⁇ ⁇ * (2 k) - (2 k) I Is determined to be a predetermined value, and a plurality of phases including one or more kernels of first to fourth phases satisfying a phase condition is determined, and a first phase is assigned to a plurality of carrier frequencies.
• a coder comprises a sub Buse' Bok mapping unit for pine Bing a phase subset selection unit assigned to the quadrature signal is also provided.
• the plurality of input signals based on the input signal, k and integer of 1 or more, the phase difference of 2 k-number of the second phase for 2 k 1 pieces of first phase (2 and k), 2 k-number of the fourth phase difference of the phase delta 0 * 'number of the third 2 k against the phase' (2 k) and the difference between the absolute value ⁇ a theta of (2 ") - a multiple comprising ⁇ * (2 k) I 1 or a kernel consisting of first to fourth phase satisfy the phase condition that is a predetermined value
• the first 2 k phases are assigned - the difference in 'number of 2 Bok second phase is assigned for the carrier frequency' number of carrier frequencies
• the frequency condition that the fourth phase is equal to the difference between the 2 k carrier frequencies assigned to the fourth phase and the 21 carrier frequencies assigned to the third phase having the same frequency interval as the first phase.
• An encoding unit that generates a plurality of codes corresponding to the input signal by allocating the plurality of phases so as to satisfy, and a code that calculates a code distance between each of the plurality of codes and the received code.
• a multi-carrier code comprising: a distance calculation unit; and a minimum distance code determination unit that decodes the received code by determining an input signal that gives a code having a minimum code distance to the received code. Decoder Is also provided.
• Figure 1 shows 16 signal point patterns that can be given to 2 carriers in the case of QPSK
• Fig. 2 is a diagram illustrating 4 carriers that can achieve a 3 dB PEP suppression effect
• Fig. 3 is a diagram illustrating allocation of 4 carrier kernels on the frequency axis
• Figure 4 illustrates the reordering of two 4-carrier kernels on the frequency axis
• Figure 5 illustrates the extension from a 4-carrier kernel to an 8-carrier kernel
• Fig. 6 is a graph showing the relationship between the phase difference ⁇ and the degree of PEP suppression
• Fig. 7 is a diagram for explaining the extension of the code length
• Fig. 9 is a diagram illustrating a code length extension using an extension subset
• Figure 14 is a graph showing the relationship between the number of carriers and the coding rate
• Figure 15 is a graph showing the relationship between the number of carriers and the coding rate
• FIG. 16 is a block diagram showing an example of the configuration of an encoder according to one embodiment of the present invention.
• Figure 17 is a block diagram showing a variation of the encoder of Figure 16;
• FIG. 18 is a block diagram showing a more generalized encoder of FIG. 16;
• FIG. 19 is a block diagram showing a more generalized encoder of FIG. 17;
• FIG. 20 is an embodiment of the present invention.
• FIG. 21 is a block diagram showing a configuration of the decoder which is a more generalized version of FIG. 20;
• FIG. 22 is a block diagram showing a configuration of the decoder according to the present invention when a thermal noise disturbance is applied; Graph showing C / N vs. BER characteristics;
• FIG. 23 is a graph showing CZN vs. BER characteristics of the code of the present invention in a fading environment
• FIG. 24 is a diagram showing four forms of ⁇ 0 (4) in QPSK. BEST MODE FOR CARRYING OUT THE INVENTION
• the 4-carrier peak envelope power (PEP) is four times (+ 6 dB) the 1-carrier PEP if the phases of the 4 carrier signal points are independent of each other.
• the relative phase (2) of the signal points of the two carriers and the other provided that the frequency difference between the two carriers is equal to each other, as shown in Fig. 2.
• the PEP is suppressed twice as much as one carrier. That is, a code composed of 4 carriers is converted to a signal pattern of 4 carriers.
• Such a code is a combination of two carriers whose relative phase is 0 and 7 ⁇ (combination with any of (a) to (d) and any of (e) to (h) in Fig. 1). And the combination whose relative phase is ⁇ / 2 and 1 7 ⁇ 2 (combination with any of (i) to () and any of (m) to (p)). Therefore, if the input signal is mapped and mapped to a set of codes corresponding to these, a PEP suppression effect of 3 dB can be obtained.
• two codes whose signal points are different only in one carrier and the signal points of the other three carriers are the same, one of them always satisfies the expression (1). It cannot exist because it cannot be obtained. In other words, to satisfy Equation (1), the signal points of two or more carriers must be different. Therefore, for the minimum distance d min
• the code that satisfies (1) is the relative value of two carriers. There are two groups: a combination with phases 0 and ⁇ ⁇ and a combination with 7 ⁇ 2 and —. Therefore, the number of groups G (2) when the modulation scheme is QPSK is
• G (2) 2 (3). Also, since each group is composed of a combination of four types of two-carrier signal patterns and four types of two-carrier signal patterns, the number of combinations 2 4 (2) is
• two codes are generated from each of the two two-carrier signal pattern combinations, so that the code pattern (in one kernel, QPSK) is generated.
• the number P 4 (1, 2) is
• each group consists of 2 m types of 2 carrier signal patterns and 2 m types of 2 carrier signal patterns, so the number of combinations K 4 (m) is
• a 4-carrier signal pattern that satisfies the expression (1) is set to 1 Carneore, and a 4n carrier code consisting of n power channels is used. PEP suppression effect and code distance expansion effect are obtained.
• the two carrier pairs of phase difference 0 (2) and 0 * (2) in equation (1) are called the two subsets that make up the 4-carrier kernel.
• the only condition concerning the frequency is that the frequency difference between each two carriers (each subset) in each kernel is equal to each other. May be placed anywhere on the frequency axis. In other words, as shown in Fig. 3, if these conditions are met, each kernel can be assigned to any of the four carriers to create a code. Therefore, the number of force Since the degree of freedom increases in the rearrangement on the frequency axis and the number of code patterns increases, the coding rate approaches 1.
• n— ⁇ ⁇ is the maximum value of the range of each n
• the total number of code patterns P 4 (n, m) is the sum of the number of obtained code patterns P 4 (n, i, m) for each number i of types.
• Table 1 shows the results of the ideal coding rate R calculated for various values of m and n, together with the amount of PEP suppression and d mi ⁇ .
• ⁇ any type of the following groups: ⁇ in any combination (e.g., configure the kernel using only two It is also possible to limit the type used under other conditions (eg, ⁇
• Fig. 5 shows eight carriers 10 to 17 on the frequency axis, and above them is a schematic diagram showing an example of the positions on the two-dimensional plane of the signal points assigned to each carrier. ing.
• the phase difference ⁇ 0 (2) between the carrier 10 and the carrier 12 is zero, the power for the carriers 11 and 13 is seven, and the power for the carrier 14 is 16 is given 0 and Carriers 15 and 17 are given 7 ⁇ . That is, Carriers 11-13 constitute a four-carrier kernel that satisfies equation (1), and Carriers 14-17 also constitute a four-carrier kernel that satisfies equation (1). And carriers having the same phase difference
• phase difference between the pair 10 and 12 and the pair 14 and 16 is 0, and between the pair 11 and 13 and the pair 15 and 17 with the same phase difference. Has a phase difference of 7 °.
• the frequency difference between carriers 10-13 is equal to the frequency difference between carriers 14-17.
• a 3 dB PEP suppression effect can be obtained in addition to the 3 dB PEP suppression effect obtained by the expression (1).
• two four-carrier kernels are each used as two four-carrier subsets.
• Eq. (18) the relationship in Eq. (18) is established between the two 4-carrier subsets, a PEP suppression effect of 3 dB can be obtained.
• the carrier frequency difference in each subset must be the same between the subsets. In general,
• phase difference ⁇ (2 k ) of the 2 k carrier is defined by two (2 k ⁇ 1 ) having the same phase difference (for 0 m (2 k ) The same applies).
• FIG. 6 shows the calculation results of the relationship between the phase difference ⁇ and the PEP suppression degree P pep (2 k , 2 k + 1 ).
• the minimum phase angle at 2 m -PSK is
• phase difference that can be expressed by 2 m -PSK is
• ⁇ 8
• the sub-set B k — is given a phase difference of 2 (2 k + 1 ).
• A. , B. Represents a code given to one carrier, respectively, and it is assumed that the phase difference ⁇ ((4) in ⁇ , ⁇ is limited to a predetermined value (for example, 7 ⁇ ).
• a k B ⁇ A k- , B k- , ⁇ ⁇ -, ⁇ - ⁇ (21)
• K 2- . (M) 2 m x K 2k (m) (22)
• the subset interval of the 2 k carrier is arbitrary. 3 k (dB) suppression of PEP It is possible to In other words, a subset Bok the (2 k calibration Li a) it is possible to perform the rearrangement as a single place.
• FIG. 8 shows examples of signal points applied to each carrier above each carrier.
• the two kernel subsets have the same phase difference from each other.
• each force pair is ⁇
• the number of code patterns of this code is calculated by using i pairs of groups instead of (m) pairs in Eq. (15).
• the number of the j (1 ⁇ i) th group is assigned to n W n
• Table 3 shows the calculation results of the ideal coding rate of this code for QPSK and 8PSK, together with the amount of PEP suppression and the value of d min .
• the pair of channels (Fig. 8) is defined as a pair of extended subsets A, B, and the code pattern of eight carriers composed of them is used as a basic unit.
• An example of such a code for k2 2 is shown in FIG. A. , B. Is not defined.
• the internal phase difference (4) between each of the extended subsets A and B is 7 ⁇ , and the phase difference ⁇ ⁇ (4 ) Is 7 ⁇ / 2.
• Carrier subset A. , B. Is generated starting from The subsets A k , B k generated starting from the extended subsets A,, ⁇ , which are distinguished from the other subsets, are also called extended subsets.
• the force of the 2 k + 2 carrier is not 2 k + l — the extended subset (A k , B) with the carrier, but the 2 k — carrier Reordering in the extended subsets (A k —,, B k — is possible, which improves the number of code patterns.
• the number of code patterns obtained is the value obtained by dividing P 4 ′ (2, 1, m) by the number of combinations of pair group selection, that is,
• extended subset 8 Calibration Li A force one panel of possible number of patterns in consideration of the overlap due to rearrangement Bok (K 8 '(m)), since the half of the path evening first number of emissions,
• the extension method of this code is the same as the above-mentioned extension of the 4-carrier code.
• the ideal coding rate (R 2 ' ⁇ 2 (n, m) (k ⁇ 2)) is log 2 P 2 '" 2 (n, m)
• the numerator of the ideal coding rate of the present invention log 2 P ⁇ (n, m) is not always an integer. This is useful for digital signal processing based on binary numbers.
• the coding rate is 20Z24.
• the 2 carrier mapping unit 24 performs mapping to signal points according to the input mapping control signal. Specifically, for example, if the mapping to (0, 0) shown in Fig. 1 (a) is specified, all the I and Q phases of the two carriers are at a high level (for example, +1). When the mapping to (1, 1) shown in Fig. 1 (b) is specified, the I phase of the two carriers is set to a high level (for example, +1), and the Q phase is set to a low level. (For example, 1 1).
• the frequencies f 1, to f 2 of the 12 quadrature modulators 25 are, for example, equally spaced, At least f! And f, f and f, f and f, f and f, f and, f and, f and,. , And the frequency difference between f and f, are equal.
• the two-carrier mapping unit 24 is a circuit that maps a 4-bit input to a 4-bit output on a one-to-one basis, so that it can be easily realized by a simple combinational logic circuit. 2
• the carrier selection unit 22 can be easily realized by using a ROM. Also, since there is the above-mentioned logical relationship between the input and the output, it is possible to realize the combination logic circuit.
• the two-carrier phase generating unit 26 determines the combination of phases to be applied to the carrier, and the two-carrier rearranging unit 30 performs rearranging in units of two carriers on the frequency axis.
• the phase difference between two carriers ⁇ (2) is 0 and 7 ⁇
• the phase difference between 4 carrier kernels and 2 carriers ⁇ ⁇ (2) is ⁇ 2 and 1 7 ⁇ ⁇ 2
• each combination of signal points given to them is identified by 12 bits.
• the number of cases in which the two carrier pairs are rearranged in the carrier rearranging section 30 is the number of cases in which three carrier pairs are assigned to any three of the six locations.
• the input 20 bits are divided into 12 bits for determining the combination of carrier phases and 8 bits for rearrangement.
• One of the eight bits (for example, the most significant bit) is also used to determine whether to use one or two types.
• the two-carrier phase generator One of the eight divided bits is input to 26, and the number of groups is determined by this.
• the combination of phases given to the six two carriers is determined.
• the two carrier mapping unit 28 is the same as the two carrier mapping unit 24 in FIG.
• the outputs are rearranged on a frequency axis according to an 8-bit input in a 2-carrier rearranging section 30.
• the two-carrier phase generator 26 and the two-carrier rearranger 30 can be easily realized by a combinational logic circuit.
• FIG. 18 is a more generalized version of the encoder of FIG. 16, represented as representing encoders for all the types of codes described so far.
• the subset selecting unit 32 selects and assigns a combination of signal points satisfying, for example, ⁇ for 2 n subsets or extended subsets according to the input of the C bit, and assigns the assigned signal points. Outputs a mapping control signal indicating. Since the signal point assigned to each subset (or extension subset) is identified by m / 2 bits, the (extension) subset selection unit 32 outputs mcZ for each subset or extension subset. 2-bit mapping control signal Is output.
• the subset mapping section 34 outputs the m-two bits indicating the values of the I-phase and Q-phase signal points of the two cZ carriers in accordance with the input mcZ2-bit mapping control signal. Outputs the mating signal.
• FIG. 19 shows a variation of the encoder of FIG. 18, which represents a more generalized representation of the encoder of FIG. 17 to represent encoders for all the types of codes described so far. Is expressed.
• the (extended) subset phase generator 36 determines the combination of phases to be given to the carrier, and the (extended) subset rearrangement section 40 arranges the subsets on the frequency axis in units of subsets or extended subsets. Make a replacement.
• C input information bits
• X bits are input to the (extended) subset rearranging section 40 as a subset rearranging control signal, and the remaining C—X bits Is input to the (extended) subset phase generator 36.
• Extension Part of the rearrangement control signal is input to the subset phase generation unit 36, and the type and number of power channels used are determined accordingly.
• the combination of phases of each subset or extended subset is determined according to the input C-n bits from those satisfying ⁇ 0 ⁇ 7 ⁇ .
• the (extended) subset mapping section 38 is the same as the (extended) subset mapping section 34 in FIG.
• the subset rearranging section 40 rearranges the subsets or extended subsets on the frequency axis in accordance with the X bit rearrangement control signal.
• the transmission signal encoded by the encoder at the transmission side is received at the reception side via a transmission path composed of N carriers, and is decoded by, for example, the maximum likelihood decoding method.
• a signal with C-bit width information which is a signal in one symbol section, is received and a D-bit width / cn carrier received signal (()
• the timing control unit 42 increments or decrements by one clock timing, and makes 20 rounds in one symbol period.
• a signal with a cut width is output.
• Encoding section 44 has the same configuration as the encoder of FIG. 16 or FIG. 17, and has the same correspondence as the one used on the transmitting side with the 20-bit width signal from timing control section 42. And outputs a 24-bit width code.
• the code distance calculator 46 calculates the code distance from the received signal and the output of the encoder 44 according to equation (41), and outputs the result as 16-bit width data.
• the code distance comparing section 48 compares the value of the minimum distance stored in the minimum distance memory section 50 and updates the minimum distance when updating the minimum distance. Then, the output of the timing controller 42 at that time is stored in the timing memory unit 52. When the output of the timing control unit 42 makes one round, the contents of the timing memory unit 52 become decoded data.
• FIG. 21 shows a configuration of a decoder for decoding the code obtained by the encoder shown in FIG. 18 or FIG.
• Encoding section 44 has the same configuration as the encoder of FIG. 18 or FIG.
• the operation of each unit is the same as that of the decoder in FIG. 20 except that the bit width is generalized, the description is omitted.

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• 1999
  • 1999-12-17 WO PCT/JP1999/007123 patent/WO2000038363A1/ja active IP Right Grant
  • 1999-12-17 DE DE69937161T patent/DE69937161T2/de not_active Expired - Lifetime
  • 1999-12-17 JP JP2000590335A patent/JP4338318B2/ja not_active Expired - Fee Related
  • 1999-12-17 EP EP99959897A patent/EP1152560B1/en not_active Expired - Lifetime
• 2001
  • 2001-06-15 US US09/882,907 patent/US6678335B2/en not_active Expired - Fee Related
• 2003
  • 2003-11-07 US US10/704,435 patent/US7254179B2/en not_active Expired - Fee Related

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