Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/69—Spread spectrum techniques
  • H04B1/707—Spread spectrum techniques using direct sequence modulation
  • H04B1/7097—Interference-related aspects
  • H04B1/71—Interference-related aspects the interference being narrowband interference
  • H04B1/7101—Interference-related aspects the interference being narrowband interference with estimation filters
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04M—TELEPHONIC COMMUNICATION
  • H04M3/00—Automatic or semi-automatic exchanges
  • H04M3/42—Systems providing special services or facilities to subscribers
  • H04M3/487—Arrangements for providing information services, e.g. recorded voice services or time announcements
  • H04M3/493—Interactive information services, e.g. directory enquiries ; Arrangements therefor, e.g. interactive voice response [IVR] systems or voice portals
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03C—MODULATION
  • H03C3/00—Angle modulation
  • H03C3/38—Angle modulation by converting amplitude modulation to angle modulation
  • H03C3/40—Angle modulation by converting amplitude modulation to angle modulation using two signal paths the outputs of which have a predetermined phase difference and at least one output being amplitude-modulated
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
  • H04L27/20—Modulator circuits; Transmitter circuits
  • H04L27/2032—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner
  • H04L27/2053—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases
  • H04L27/206—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers
  • H04L27/2067—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers with more than two phase states
  • H04L27/2071—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers with more than two phase states in which the data are represented by the carrier phase, e.g. systems with differential coding
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B2201/00—Indexing scheme relating to details of transmission systems not covered by a single group of H04B3/00 - H04B13/00
  • H04B2201/69—Orthogonal indexing scheme relating to spread spectrum techniques in general
  • H04B2201/707—Orthogonal indexing scheme relating to spread spectrum techniques in general relating to direct sequence modulation
  • H04B2201/7097—Direct sequence modulation interference
  • H04B2201/709709—Methods of preventing interference
• • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
  • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
  • Y02D—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN INFORMATION AND COMMUNICATION TECHNOLOGIES [ICT], I.E. INFORMATION AND COMMUNICATION TECHNOLOGIES AIMING AT THE REDUCTION OF THEIR OWN ENERGY USE
  • Y02D30/00—Reducing energy consumption in communication networks
  • Y02D30/70—Reducing energy consumption in communication networks in wireless communication networks

Definitions

• This invention relates to modulation systems and methods, and more particularly to modulation systems and methods that modulate digital input signals.
• Modulation systems and methods are widely used in transmitters to modulate an information input including voice and/or data onto a carrier.
• the carrier may be a final carrier or an intermediate carrier.
• the carrier frequency can be in UHF, VHF, RF, microwave or any other frequency band.
• Modulators are also referred to as “mixers” or “multipliers”. For example, in a mobile radiotelephone, a modulator is used for the radiotelephone transmitter.
• modulation systems and methods for digital input signals generally include a Digital-to-Analog Converter (DAC) that converts the digital input signal into an analog signal.
• a low pass filter also referred to as an "anti-aliasing filter” filters the analog signal to produce a filtered analog signal.
• a modulator modulates the filtered analog signal onto a carrier.
• the modulator includes a multiplier that is coupled to a local oscillator, such as a Voltage Controlled Oscillator (VCO), and to the filtered analog signal.
• VCO Voltage Controlled Oscillator
• the carrier including the filtered analog signal may then be transmitted by an antenna.
• VCO Voltage Controlled Oscillator
• CDMA Code Division Multiple Access
• DSSS Direct Sequence Spread Spectrum
• the narrowband AMPS FM signal has a bandwidth of about 12.5KHz
• the wideband CDMA signal has a bandwidth of about 615KHz, or about an order of magnitude wider.
• FIG. 1 illustrates a first conventional dual-mode modulator.
• an IQ modulator 10 also referred to as a "quadraphase modulator” or a “quadrature modulator” includes a quadrature splitter 20, also known as a 90° phase shifter, and a pair of multipliers 16a, 16b coupled to the quadrature splitter.
• a local oscillator 15, such as a Voltage Controlled Oscillator (VCO), is coupled to the quadrature splitter 20 to produce 90° phased shifted local oscillator signals.
• I data 11a and Q data lib are coupled to a respective multiplier or mixer 16a, 16b respectively.
• VCO Voltage Controlled Oscillator
• Digital input data is converted to analog data by I Digital-to-Analog Converter (DAC) 14a and Q DAC 14b, respectively.
• the outputs of the DACs 14a and 14b respectively are applied to low pass filters 12a and 12b respectively to provide the I and Q data inputs 11a and lib respectively.
• the modulator modulates the input data on a carrier 13, by summing the outputs of the multipliers 16a, 16b at summing node 218, and transmits the modulated carrier 13 via an antenna.
• the DACs 14a and 14b, low pass filters 12a and 12b and IQ modulator 10 may be used to modulate a high bandwidth CDMA signal such as a Direct Sequence Spread Spectrum (DSSS) signal onto a carrier.
• DSSS Direct Sequence Spread Spectrum
• the signal Since the signal is generated digitally, it is low pass filtered by filters 12a and 12b to let the information through while removing digitally generated spurs and noise.
• a separate FM DAC 19 and a separate FM low pass filter 17 may be provided.
• Baseband circuitry generates an FM voltage signal that is applied to the tune line of the VCO, to modulate the FM information onto the carrier for transmission according to the AMPS standard. Since the FM voltage signal is generated digitally, it is low pass filtered by FM low pass filter 17 to let the information through while removing digitally generated spurs and noise.
• the low pass filter 17 generally has a different bandpass characteristic than the low pass filters 12a and 12b that are part of the CDMA modulator, due to the widely differing bandwidths of the FM and CDMA signals. Accordingly, in this dual-mode embodiment, a separate FM DAC 19 and a separate FM low pass filter 17 is provided. Modulation systems according to Figure 1 have been designed into many integrated circuit chip sets developed for CDMA standards that also include AMPS functionality. Unfortunately, this technique uses separate DACs and low pass filters, which may increase the size, cost and/or power consumption of the modulator.
• FIG. 2 A second dual-mode modulation system is illustrated in Figure 2.
• an IQ modulator 210 including a quadrature splitter 220, a pair of multipliers 216a and 216b, a summing node 218 and a VCO 215 are provided to produce a modulated carrier 213.
• the DACs and low pass filters are shared for the dual-mode operation.
• the I DAC and Q DAC 214a and 214b respectively are used for both wideband CDMA and narrowband FM operation.
• Low pass filters 212a and 212b are also used for wideband CDMA and narrowband FM operation.
• the low pass filters 212a and 212b should have different band pass characteristics when in the different modes.
• the band pass frequency is switched depending upon mode. Accordingly, while these switched filters 212a, 212b are used in both modes, they may be expensive to implement and may consume excessive power and/or area in a radiotelephone.
• the required carrier suppression for FM modulation in an IS-19 AMPS analog system may be approximately -35dBc.
• the nominal carrier suppression may be -40dBc, which can translate into a 14mV differential DC offset signal when a 2V peak-to-peak differential information signal is generated in a balanced system.
• Low DC offset in the digital input signal may be provided using conventional techniques.
• the modulation system may generate its own DC offset. More specifically, the digital-to-analog converter and/or the low pass filter may generate DC offsets.
• the DC offset that is generated in the digital-to-analog converter can be reduced using high performance digital-to-analog converters.
• these digital-to-analog converters may be costly and complex.
• DC offset can be reduced in the low pass filter by providing a passive, off-chip filter with tight tolerance components.
• a passive off-chip filter may be costly and complex, and may consume excessive space in a portable radiotelephone.
• a narrow bandwidth signal such as a narrowband FM signal
• a modulator that modulates a wide bandwidth signal, such as a CDMA signal
• oversampling the narrow bandwidth signal the same fixed low pass filter can be used for both the wide bandwidth signal and the oversampled narrow bandwidth signal. Accordingly, different low pass filters or switched low pass filters are not needed.
• a CDMA modulator including a sampler is used for dual-mode modulation by applying a narrow bandwidth FM signal to the CDMA modulator, such that the CDMA modulator oversamples the FM signal and modulates the oversampled FM signal.
• the CDMA modulator includes a fixed low pass filter having a passband that encompasses a CDMA signal and the oversampled FM signal, so that the same fixed low pass filter is used to filter both the CDMA signal and an FM signal.
• the CDMA modulator may be particularly useful in a radiotelephone where the CDMA signal may be a direct sequence spread spectrum signal and the FM signal may be an analog cellular telephone signal.
• Dual-mode modulation systems include means for modulating an applied signal onto a carrier and means for applying a first signal to the modulating means, to thereby modulate the first signal onto a carrier.
• Oversampling means is included for oversampling a second signal of narrower bandwidth than the first signal.
• the systems also include means for applying the oversampled second narrower bandwidth signal to the modulating means, to thereby modulate the second narrower bandwidth signal onto a carrier.
• the modulating means preferably comprises a digital-to-analog converter and a low pass filter that filters the analog output of the digital-to-analog converter, wherein the low pass filter has a passband that encompasses the first signal, and the oversampled second narrower bandwidth signal, such that the same fixed low pass filter is used to filter both the first signal and the oversampled second narrower bandwidth signal.
• the oversampling means preferably comprises first and second samplers.
• Dual-mode modulation systems also include means for sampling an applied signal, means for converting the sampled signal to an analog signal, means for low pass filtering the analog signal and means for modulating the low pass filtered analog signal onto a carrier.
• Dual-mode modulation systems also include means for applying a first signal to the sampling means, to thereby modulate the first signal on a carrier using the sampling means, the converting means and the low pass filtering means, and for applying a second signal of narrower bandwidth than the first signal to the sampling means, to thereby oversample the second signal and modulate the second signal on a carrier using the sampling means, the converting means and the low pass filtering means. Accordingly, the same unswitched filters may be used for both the wide and narrow bandwidth signals, to thereby allow reduction in cost, space and/or power consumption.
• Dual-mode modulation systems and methods according to the invention also compensate for the DC offset that is introduced by the digital-to-analog converter and/or the low pass filter thereof. Compensation is preferably provided in the digital domain, to thereby reduce DC offset to within acceptable limits for the modulation that is being used. More preferably, compensation is provided by subtracting from the sampled signal, a digital value representing the DC offset in the filtered analog signal that is introduced by the digital-to-analog converter and/or the low pass filter.
• Modulation systems according to the invention include a digital-to-analog converter that converts the sampled signal into an analog signal. The analog signal is filtered by a low pass filter to produce a filtered analog signal.
• the digital-to-analog converter and/or the low pass filter introduce DC offset into the filtered analog signal.
• a modulator modulates the filtered analog signal onto a carrier.
• a DC offset compensator compensates for the DC offset in the filtered analog signal that is introduced by the digital-to-analog converter and/or the low pass filter.
• DC offset compensators preferably include a sensor that senses the DC offset in the filtered analog signal.
• An analog-to-digital converter is responsive to the sensor, to convert the sensed DC offset into a digital offset signal.
• a subtractor is responsive to the analog-to-digital converter, to subtract the digital DC offset signal from the sampled signal, and to apply the sampled signal less (minus) the digital DC offset signal, to the digital-to-analog converter. Accordingly, the sensed offset is subtracted in the digital domain.
• a sealer may also be included that is responsive to the analog-to-digital converter, to scale the digital DC offset signal into a scaled digital DC offset signal. The subtractor is then responsive to the sealer, to subtract the scaled digital DC offset signal from the sampled signal.
• the subtractor need not continuously sense the DC offset in the filtered analog signal, but rather may do so on an intermittent and preferably periodic basis.
• the DC offset compensator may include a latch that is responsive to the analog-to-digital converter to intermittently latch the digital DC offset signal and to apply the latched digital DC offset signal to the subtractor, such that the latched digital DC offset signal is subtracted from the sampled signal.
• the latch can be clocked at a second clock rate that is lower than the first clock rate.
• the sensor may comprise a low pass filter that senses the DC offset in the filtered analog signal.
• the analog-to-digital converter is a one bit delta-sigma analog-to-digital converter.
• a polarity inverter is responsive to the sensor, to periodically invert the polarity of the sensed DC offset signal.
• the analog-to-digital converter converts the periodically polarity inverted sensed DC offset signal into the digital offset signal, thus reducing the effect of the internal DC offset of the analog-to-digital converter.
• DC Offset compensation may be advantageously used with dual bandwidth modulators wherein the sampled signal comprises a selected one of a first digital input signal and a second digital input signal of narrower bandwidth than the first digital input signal.
• the invention may be used with a first digital input signal that is a CDMA signal, and with a second digital input signal that is an FM signal.
• the CDMA signal may be a direct sequence spread spectrum signal
• the FM signal may be an analog cellular telephone signal.
• the present invention may also be used in IQ modulators, also referred to as "quadraphase modulators” or “quadrature modulators” that modulate in-phase and quadrature filtered analog signals onto a carrier. Analogous modulation methods may also be provided. Accordingly, modulation systems and methods for a digital input signal can provide low DC offset notwithstanding the introduction of DC offset by the digital-to- analog converter and/or the low pass filters thereof. High performance and costly digital-to-analog converters need not be used. High performance off-chip low pass filters also need not be used.
• Figure 1 is a block diagram of a first conventional dual-mode modulation system and method.
• Figure 2 is a block diagram of a second conventional dual-mode modulation system and method.
• Figure 3 is a block diagram of dual-mode modulation systems and methods including DC offset compensation according to the present invention.
• Figures 4 A and 4B which when placed together as shown form Figure 4, are a block diagram of a second embodiment of dual-mode modulation systems and methods including DC offset compensation according to the present invention.
• Figure 5 is a block diagram of a one bit- delta-sigma analog-to-digital converter that may be used to compensate for DC offset according to the present invention.
• Figure 6 illustrates another embodiment of an analog-to-digital converter than can be used to compensate for DC offset according to the present invention.
• Figure 7 illustrates operation of a polarity inverter than may be used in Figure 6.
• FIG. 8 is a block diagram of modulation systems and methods according to the present invention.
• FIGS 9 A and 9B which when placed together as indicated form Figure 9, illustrate a first embodiment of dual mode IQ modulation systems and methods according to the present invention.
• Figure 10 illustrates a second embodiment of dual mode IQ modulation systems and methods according to the present invention.
• FIGS 11 A and 1 IB which when placed together as indicated form Figure 11 , illustrate an embodiment of single mode IQ modulation systems and methods according to the present invention.
• dual-mode modulation systems and methods include an IQ modulator 310 that includes a VCO 315, a pair of multipliers 316a and 316b, a quadrature splitter 320 and a summing node 318.
• the interconnection of these elements to form a quadrature modulator are well known to those having skill in the art, and need not be described further herein.
• IQ modulator 310 accepts I inputs 311a and Q inputs 311b and produces an output 313 that modulates these inputs on a carrier.
• dual-mode modulation systems and methods according to the invention include a pair of fixed low pass filters 312a and 312b.
• the fixed low pass filters include a passband which can pass the wide bandwidth signals, such as the CDMA DSSS signals.
• An I DAC 314a and a Q DAC 314b are also included.
• a pair of samplers 330a and 330b also referred to as an I sampler and a Q sampler, respectively.
• a source of wide bandwidth signals such as DSSS signal source 350 and a source of narrow bandwidth signals, such as FM signal source 340 are both applied to the samplers 330a and 330b.
• DSSS signal source 350 may produce a DSSS-I signal and DSSS-Q signal that are applied to sampler 330a and sampler 330b respectively.
• FM signal source 330 may produce an FM I signal and an FM Q signal that are applied to the I sampler 330a and the Q sampler 330b respectively. It will be understood that DSSS signal source 350 and FM signal source 340 may be generated as baseband signals in dual-mode radiotelephones. The generation of DSSS signals and FM signals in dual-mode radiotelephones are well known to those having skill in the art and need not be described further herein.
• the samplers 330a and 330b and the DACs 314a and 314b operate at a sample rate T sam pie-
• the sample rate T sa mpi e may be controlled by a signal that is applied to control line 335.
• the sample rate is generally set by the sample rate for the wide bandwidth signals 350. Accordingly, when the narrow bandwidth signals 340 are applied to samplers 330a and 330b, the sampler acts to oversample the narrow bandwidth signals. By oversampling the narrow bandwidth signals, the same DACs and low pass filters 314 and 312 respectively, may be used for the wide and narrow bandwidth signals.
• the wide bandwidth signals also may be oversampled by samplers 330a and 330b.
• the narrow bandwidth signals are highly oversampled by samplers 330a and 330b.
• the sample rate need not be identical for the wide and narrow bandwidth signals.
• the sample rate T samp i e is generally maintained at a rate such that the same fixed low pass filters 312a and 312b can be used for the wide and narrow bandwidth signals. Accordingly, a dual-mode modulator may use the same unsw ⁇ tched low pass filters for modulating the wide bandwidth signals and the oversampled narrow bandwidth signals, to thereby save cost, space and/or power.
• the present invention can use the same fixed low pass filter (without switching) for both the FM signal and spread spectrum signal fed into the IQ modulator.
• Variable low pass filters such as switched low pass filters, are not needed.
• the FM signal is highly oversampled in the DAC, preferably at the same sample rate, T samP i e> as the spread spectrum signal. To be sampled at this rate in the DAC, it is upsampled/interpolated to this highly oversampled rate.
• the sample rate can be 8X or 4.9152MHz.
• the sample rate for the FM signal can be a divider of a reference clock rate that is near the IS-95 sample rate (19.2MHz/4 or 4.8MHz).
• the sampling spur ( ⁇ 5MHz) can be reduced by the low pass filter to meet the spur performance specifications of the transmitter for both AMPS and CDMA.
• the sampling noise floor from the DAC can be decreased to acceptable levels for the AMPS transmission ( ⁇ -60dBc).
• the spread spectrum signal is not highly oversampled, rather it is normally oversampled (8X).
• the sampling noise floor for the DAC for this mode can be decreased to acceptable levels for the IS-95 transmission ( ⁇ -45dBc).
• the spurious free dynamic range for either mode is preferably greater than the difference between the largest signal and the noise floor (>60dB for AMPS and >45dB for CDMA).
• the quality of the IQ modulator is preferably sufficient to meet the modulation specifications in the AMPS mode, which are generally more severe than the CDMA mode.
• the present invention may be applied to modulation systems and methods that share an IQ modulator with an FM signal (narrow bandwidth) and a direct sequence spread spectrum signal (wide bandwidth).
• the analog FM signal is converted to a digital signal at some rate. If the converted rate is low, then it can be upsampled/interpolated to the final desired rate, T samp ⁇ e . During A/D conversion, it can immediately be highly oversampled at the final desired rate.
• the present invention may also be applied to modulation systems and methods that share an IQ modulator with an unspread digital signal (narrow bandwidth) and a direct sequence spread spectrum signal (wide bandwidth).
• the present invention may be used in mixed AMPS/CDMA radiotelephones (IS-95), mixed GSM/WCDMA radiotelephones (third generation wideband cellular standards), and other radiotelephones that combine wideband and narrowband signals.
• DC offset compensation according to the invention will now be described.
• the sampled I and Q signals 333a and 333b may have low DC offset
• the DC offset that is introduced by at least one of the digital-to- analog converters 314a and 314b and the low pass filters 312a and 312b may produce unacceptably high DC offset.
• the digital-to-analog converters 314a and 314b and low pass filter 312a and 312b are collectively labeled as sources of DC offset by dashed blocks 324a and 324b.
• DC offset compensators 322a and 322b are provided, that compensate for the DC offset in the filtered analog signal 311 and 311b that is introduced by the at least one of the digital-to-analog converters 314a and 314b and the low pass filter 312a and 312b.
• the DC offset compensators 322a and 322b act upon the sampled signals 333a and 333b in the digital domain to compensate for the DC offset in the filtered analog signals 311a and 311b that is introduced by at least one of the digital-to-analog converters 314a and 314b and the low pass filters 312a and 312b, to thereby reduce the DC offset to within acceptable limits for the modulation scheme that is used. More specifically, as shown in Figure 3, each of the DC offset compensators
• 322a and 322b preferably include a DC sensor 321a, 321b that senses the DC offset in the filtered analog signal 311a, 311b.
• An Analog-to-Digital Converter (ADC) 323a, 323b converts the sensed DC offset 329a, 329b into a digital DC offset signal 331a and 331b.
• a subtractor 326a, 326b subtracts the digital DC offset signal 331a, 331b from the sampled signal 333a, 333b and applies the sampled signal less (minus) the DC offset signal 327a, 327b, to the digital-to-analog converter 314a, 314b. Accordingly, the DC offset that is generated in the forward path is sensed and converted to a digital value. The digital value representing the DC offset is then subtracted from the incoming sampled signal 333a, 333b.
• the digital DC offset signal need not be calculated at the same frequency as the digital-to-analog conversion by the digital-to-analog converters 314a and 314b, in order to compensate for the DC offset. Rather, the DC offset may be determined intermittently, preferably periodically. Accordingly, as shown in Figure 1, latches 325a and 325b may be used to intermittently and periodically latch the digital DC offset signals 311a and 311b, such that the latched digital DC offset signals 332a and 332b are subtracted from the sampled signals 333a and 333b.
• the latches 325a and 325b and/or the analog-to-digital converters 323a, 323b may be clocked at lower frequencies than the digital-to-analog converters 314a and 314b, because the DC offset will generally not vary as rapidly as the sampled signals 333a and 333b.
• the DC offset may be sensed once per second, or at other intervals. It will be understood that in Figure 3, two separate DC offset compensators
• 322a and 322b are provided for the two input signal paths. However, it will be understood that a single DC offset compensator may be used for both the I input signal path and the Q input signal path.
• an optional sealer 460a, 460b is used between a respective analog-to-digital converter 323a, 323b and a respective latch 325a, 325b.
• the sealer scales the digital DC offset signals 329a, 329b into scaled digital DC offset signals 329a', 329b'.
• Scaling may be used in order to apply a scaling factor to the digital signal produced by the analog-to-digital converters 323a, 323b. For example, when a differential DC voltage is sensed by the DC sensors 321a, 321b of Figure 4, a scaling factor may need to be used.
• an optional amplifier 461a, 461b is used between a respective low pass filter 312a, 312b and a respective modulator 316a, 316b, to provide amplification if necessary.
• the DC offset compensators 322a, 322b can also compensate for DC offset in the amplifiers 461a, 461b.
• timing for the analog-to-digital converters 323a, 323b, for the digital-to-analog converters 314a, 314b and for the latches 325a, 325b may also be provided from a common clock 462. It will be understood that the voltage controlled oscillator 315 is also preferably locked to the same reference as the common clock 462. As shown in Figure 4, the digital-to-analog converters 314a, 314b are preferably clocked by the clock 462. The analog-to-digital converters 323a, 323b are preferably clocked at a first clock rate that is lower than that of clock 462 using divide by M ( ⁇ M) circuits 463a, 463b.
• ⁇ M divide by M
• the latches 425a, 425b are latched at a second clock rate that is even lower than the first clock rate using divide by N ( ⁇ N) circuits 464a, 464b.
• First and second clock rates that are lower than the speed of clock 462 may be used, because the DC offset need not be sensed as often as the sampled signals 333a, 333b are converted to digital, since the DC offset generally changes less rapidly than the sampled signal itself.
• the DC sensors 321a, 321b of Figure 4 may be provided by passive low pass filters with a low corner frequency. A low corner frequency may be used because only the DC component needs to be sensed. Subtraction in the digital domain may also occur at a low rate, that is preferably gated by dividing the clock 462.
• DC offset compensators of Figure 4 may be able to cancel a DC offset that is as large as the Least Significant Bit (LSB) of the digital-to-analog converter 314a, 314b.
• LSB Least Significant Bit
• the DC offset cancellation that can be accomplished may be limited by the DC offset that is introduced by the analog-to-digital converters 323a, 323 b in the DC offset compensators 322a, 322b. Accordingly, low DC offset analog-to-digital converters 323a, 323b are preferably used.
• a preferred analog-to-digital converter 323a, 323b is a one bit delta-sigma converter. A one bit converter may be used so that the feedback path inside the delta-sigma converter may be obtained using a straight feedback connection without the need for intervening components.
• Figure 5 illustrates a block diagram of a one bit delta-sigma analog-to-digital converter 323'.
• the analog-to-digital converter 323' includes an integrator 70, a comparator 71 and a decimation/low pass filter 72.
• the output of the comparator 71 is fed back to the input via a summing node 73.
• the design of a one bit delta-sigma digital-to-analog converter is well known to those having skill in the art, and is described for example in the textbook by Candy and Temes entitled “Oversampling Delta-Sigma Data Converters", IEEE Press, 1992, in the chapter entitled “Oversampling Methods for A/D and D/A Conversion", -pp.
• DC offsets introduced by elements in the forward path of the delta-sigma converter can be nulled.
• the only DC offset contributor may be the analog difference amplifier represented by the summing node 73.
• DC offset can be reduced in this difference amplifier 73, for example by selecting the decimation low pass filter 72 to have a corner at 10 Hz.
• the difference amplifier 73 may operate at a very low frequency, for example 10 kHz.
• the difference amplifier 73 thus can be made with large geometry, low frequency transistors that may have excellent component matching when used in a process that also includes the high frequency digital-to-analog converters 314a, 314b. This good matching can produce low DC offset in the delta-sigma converter.
• the corner frequency of the sensor 321 may be chosen to be low enough to roll off modulation, and thereby keep modulation out of the feedback loop.
• the frequency of the one bit delta-sigma converter is preferably selected so as to remove this modulation without aliasing errors.
• the rate of latching by the latch 325 is preferably low enough so that the open loop system has settled.
• the resolution of the feedback is preferably within one least significant bit of the digital-to-analog converter 314. This resolution may be determined by the oversampling ratio between the input and output of the decimation/low pass filter 72.
• the largest error correction may be available for the first subtraction. This correction may be accomplished in one step.
• this correction can be made smoother through a digital filter that may be placed between the latch 325 and the subtractor 326.
• the transition can be made smoother by constraining the latch to move only one least significant bit per latch instant. The selection may be made based on the desired speed to initially set the compensation. Thereafter, the DC offset compensator may operate in tracking mode, and the same techniques may be used.
• Figure 6 illustrates another embodiment of an analog-to-digital converter system 80 that can use any type of analog-to-digital converter 323.
• a polarity inverter 81 is included that periodically inverts the polarity of the sensed DC offset signal. Operation of the polarity inverter 81 is described schematically in Figure 7.
• the analog-to-digital converter 323 converts the periodically polarity-inverted sensed DC offset signal 82 into a digital offset signal.
• first and second latches 82a, 82b are then latched in first and second latches 82a, 82b.
• a subtractor 83 Prior to subtraction by a subtractor 83, one of the samples, such as the odd samples in latch 82b, are delayed by a delay 84.
• Subtraction of the polarity inversion samples in block 85 may be clocked using the clock 462, wherein the clock 462 is divided by R and applied to the polarity inverter 81 and analog-to-digital converter 323 by first and second divide by R ( ⁇ R) circuits 86, 87.
• a slower clock that is produced by divide by R times P ( ⁇ R*P) circuit 88 may be used to clock the subtractor 83.
• the analog-to-digital converter system 80 of Figure 6 can produce low DC offset due to the following relationships:
• DC offS et (Mo - M,)/2, where M 0 is the error measured in one state of the inverter 81, Mj is the error measured in the other state of the inverter 81, DC 0 f se t is the DC offset signal 311 and Emea s ur e m e n t is the DC error of the measurement system of Figure 6, that is assumed to be constant between measurements.
• modulation systems and methods according to the present invention include a Digital-to- Analog Converter (DAC) 814 that converts a digital input signal 833 into an analog signal 828.
• DAC Digital-to- Analog Converter
• a low pass filter 812 also referred to as - lo an anti-aliasing filter, filters the analog signal 828 to produce a filtered analog signal
• a modulator 816 modulates the filtered analog signal 811 onto a carrier that is generated by a controlled source, such as a Voltage Controlled Oscillator (VCO) 815.
• VCO Voltage Controlled Oscillator
• the modulated input signal 813 is then transmitted by a transmit antenna 834.
• Other transmitting circuitry may also be included, as is well known to those having skill in the art.
• a DC offset compensator 822 that compensates for the DC offset in the filtered analog signal 811 that is introduced by the at least one of the digital-to-analog converter 814 and the low pass filter 812. As shown, the DC offset compensator 822 acts upon the digital input signal 833 in the digital domain to compensate for the DC offset in the filtered analog signal 811 that is introduced by at least one of the digital-to-analog converter 814 and the low pass filter
• the DC offset compensator 822 preferably includes a DC sensor 821 that senses the DC offset in the filtered analog signal 811.
• An Analog-to-Digital Converter (ADC) 823 converts the sensed DC offset 829 into a digital DC offset signal 831.
• a subtractor 826 subtracts the digital DC offset signal 831 from the digital input signal 833 and applies the digital input signal less (minus) the DC offset signal 827, to the digital-to-analog converter 814. Accordingly, the DC offset that is generated in the forward path is sensed and converted to a digital value. The digital value representing the DC offset is then subtracted from the incoming digital input signal 833.
• the digital DC offset signal need not be calculated at the same frequency as the digital-to-analog conversion by the digital-to-analog converter 814, in order to compensate for the DC offset. Rather, the DC offset may be determined intermittently, preferably periodically. Accordingly, as shown in Figure 8, a latch 825 may be used to intermittently and periodically latch the digital DC offset signal 831, such that the latched digital DC offset signal 832 is subtracted from the digital input signal 833. Thus, as will be described below, the latch 825 and/or the analog-to-digital converter 823 may be clocked at lower frequencies than the digital-to-analog converter 814, because the DC offset will generally not vary as rapidly as the digital input signal 833. In a particular example, the DC offset may be sensed once per second, or at other intervals.
• FIG. 9 illustrates a dual mode IQ modulation system according to the present invention.
• an IQ modulator 910 includes a quadrature splitter 920, also known as a 90° phase shifter, and a pair of multipliers 916a, 916b coupled to the quadrature splitter.
• the VCO 915 is coupled to the quadrature splitter 920, to produce 90° phase shifted local oscillator signals.
• a digital I input signal 933a and a digital Q input signal 933b are provided to respective I and Q paths of the IQ modulation system.
• the elements in the I path are designated by a reference character a and the elements in the Q path are designated by the reference character b.
• the IQ modulator 910 modulates the I and Q filtered analog signals 911a and 911b respectively, on a carrier, by summing the outputs of the multipliers 916a and 916b at a summing node 918.
• the modulated input signal is transmitted via an antenna 934.
• the digital-to-analog converters 914a and 914b, low pass filters 912a and 912b and IQ modulator 910 may be used to modulate a high bandwidth CDMA signal, such as a Direct Sequence Spread Spectrum (DSSS) signal, onto a carrier. Since the signal is generated digitally, it is low pass filtered by filters 912a and 912b, to let the information through while removing digitally generated spurs and noise.
• DSSS Direct Sequence Spread Spectrum
• a separate FM digital-to-analog converter 914c and a separate FM low pass filter 912c may be provided.
• the components in the modulation path of the digital FM input signal 933c are labeled with a reference character c.
• Baseband circuitry generates the digital FM input signal 933c that is applied to the tune line of the VCO 915, to modulate the FM information onto the carrier for transmission according to the AMPS standard.
• the low pass filter 912c generally has a different bandpass characteristic than the low pass filters 912a and 912b that are part of the CDMA modulator, due to the widely differing bandwidths of the FM and CDMA signals.
• FIG. 10 A second dual mode modulation system is illustrated in Figure 10.
• the digital-to-analog converters and low pass filters are shared for the dual- mode operation.
• the I DAC 1014a and the Q DAC 1014b are used for both wideband CDMA and narrowband FM operation.
• Low pass filters 1012a' and 1012b' are also used for wideband CDMA and narrowband FM operation. Due to the widely disparate bandwidths of the CDMA signal and the FM signal, the low pass filters 1012a' and 1012b' should have different bandpass characteristics when in the different modes. In order to share the low pass filters, the bandpass frequency is switched depending upon mode. Referring now to Figure 11 , an embodiment of a single mode IQ modulator according to the present invention will now be described.
• an optional sealer 1160a, 1160b is used between a respective analog-to-digital converter 1123a, 1123b and a respective latch 1125a, 1125b.
• the sealer scales the digital DC offset signal 1129a, 1129b into a scaled digital DC offset signal 1129a', 1129b'.
• Scaling may be used in order to apply a scaling factor to the digital signal produced by the analog-to-digital converter 1123a, 1123b. For example, when a differential DC voltage is sensed by the DC sensors 1121a, 1121b of Figure 11, a scaling factor may need to be used.
• an optional amplifier 1161a, 1161b is used between the respective low pass filter 1112a, 1112b and a respective modulator 1116a, 1116b, to provide amplification if necessary.
• the DC offset compensators 1122a, 1122b can also compensate for DC offset in the amplifiers 1161a, 1161b.
• timing for the analog-to-digital converters 1123a, 1123b, for the digital-to-analog converters 1114a, 1114b and for the latches 1125a, 1125b may also be provided from a common clock 1162. It will be understood that the voltage controlled source 1115 is also preferably locked to the same reference as the common clock 1162. As shown in Figure 11, the digital-to-analog converters 1114a and 1114b are preferably clocked by the clock 1162. The analog-to-digital converters 1123a, 1123b are preferably clocked at a first clock rate that is lower than that of clock 1162 using divide by M ( ⁇ M) circuits 1163a, 1163b.
• ⁇ M divide by M
• the latches 1125a and 1125b are latched at a second clock rate that is even lower than the first clock rate using divide by N ( ⁇ N) circuits 1164a, 1164b.
• First and second clock rates that are lower than the speed of clock 1162 may be used, because the DC offset need not be sensed as often as the input signals 1133a, 1133b are converted to digital, since the DC offset generally changes less rapidly than the input signal itself.
• the DC sensors 1121a, 1121b of Figure 11 may be provided by passive low pass filters with a low corner frequency. A low corner frequency may be used because only the DC component needs to be sensed. Subtraction in the digital domain may also occur at a low rate that is preferably gated by dividing the clock 1162.
• DC offset compensators of Figure 11 may be able to cancel a DC offset that is as small as the Least Significant Bit (LSB) of the digital-to-analog converter 1114a, 1114b.
• LSB Least Significant Bit
• the DC offset cancellation that can be accomplished may be limited by the DC offset that is introduced by the analog-to-digital converters 1123a, 1123b in the DC offset compensators 1122a, 1122b. Accordingly, low DC offset analog-to-digital converters 1123a, 1123b are preferably used.
• a preferred analog-to-digital converter 1123a, 1123b is a one bit delta-sigma converter. A one bit converter may be used so that the feedback path inside the delta-sigma converter may be obtained using a straight feedback connection without the need for intervening components.
• a one bit delta-sigma analog-to-digital converter 23' of Figure 5 may be used.
• An analog-to- digital converter system 80 that can use any type of analog-to-digital converter 23 as shown in Figure 6 also may be used.

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• 1998
  • 1998-11-05 CA CA002310220A patent/CA2310220A1/en not_active Abandoned
  • 1998-11-05 AU AU13821/99A patent/AU1382199A/en not_active Abandoned
  • 1998-11-05 WO PCT/US1998/023566 patent/WO1999026361A1/en active IP Right Grant
  • 1998-11-05 JP JP2000521604A patent/JP2002542633A/ja active Pending
  • 1998-11-05 CN CNB988131668A patent/CN1150694C/zh not_active Expired - Fee Related
  • 1998-11-05 BR BR9814200-3A patent/BR9814200A/pt not_active IP Right Cessation
• 2000
  • 2000-05-16 FI FI20001178A patent/FI20001178A/fi not_active Application Discontinuation
• 2001
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