WO1996010866A1 - Circuit tampon d'entree cmos a verrouillage dynamique - Google Patents

Circuit tampon d'entree cmos a verrouillage dynamique Download PDF

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Publication number
WO1996010866A1
WO1996010866A1 PCT/US1995/012094 US9512094W WO9610866A1 WO 1996010866 A1 WO1996010866 A1 WO 1996010866A1 US 9512094 W US9512094 W US 9512094W WO 9610866 A1 WO9610866 A1 WO 9610866A1
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WO
WIPO (PCT)
Prior art keywords
transistor
terminal
coupled
input buffer
reference voltage
Prior art date
Application number
PCT/US1995/012094
Other languages
English (en)
Inventor
Mark G. Johnson
Original Assignee
Rambus, Inc.
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Rambus, Inc. filed Critical Rambus, Inc.
Priority to AU37218/95A priority Critical patent/AU3721895A/en
Publication of WO1996010866A1 publication Critical patent/WO1996010866A1/fr

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Classifications

    • HELECTRICITY
    • H03ELECTRONIC CIRCUITRY
    • H03KPULSE TECHNIQUE
    • H03K3/00Circuits for generating electric pulses; Monostable, bistable or multistable circuits
    • H03K3/02Generators characterised by the type of circuit or by the means used for producing pulses
    • H03K3/353Generators characterised by the type of circuit or by the means used for producing pulses by the use, as active elements, of field-effect transistors with internal or external positive feedback
    • H03K3/356Bistable circuits
    • H03K3/356104Bistable circuits using complementary field-effect transistors
    • H03K3/356113Bistable circuits using complementary field-effect transistors using additional transistors in the input circuit
    • H03K3/356147Bistable circuits using complementary field-effect transistors using additional transistors in the input circuit using pass gates

Definitions

  • This invention pertains to the field of input buffer circuitry for interfacing integrated circuits with external circuitry. More particularly, this invention relates to improving the input buffer circuitry on an integrated circuit that receives signals from an external source. BACKGROUND OF THE INVENTION
  • High performance digital integrated circuits including dynamic RAMs, have traditionally employed specialized circuits for receiving data from an external source.
  • the digital signals produced by the external source are continuously changing and are considered to be valid only during certain intervals.
  • Data receiving circuits also known as input buffers
  • input buffers are used to capture the data from the external source during these periods of validity so that the integrated circuit may continue to utilize the captured data even though the external source is no longer providing a valid digital data signal.
  • timing signals produced by clocks. These timing signals (or latch signals) have a high level, a low level, and a latch edge. The latch edge is created when the latch signal transitions from a high level to a low level or from a low level to a high level.
  • the latch signals and their edges are known as "strobes.” For example, in dynamic RAM circuitry, these signals are called Row Address Strobe ("RAS”) and Column Address Strobe (“CAS").
  • RAS Row Address Strobe
  • CAS Column Address Strobe
  • Data input to the integrated circuit is considered to be valid and stable only during a brief interval of time near a latch edge. At other times (known as transitioning times), the data is transitioning between valid states. During these transitioning times, the data input to the integrated circuit is not valid and should not be utilized.
  • Latching input buffers can be constructed in metal oxide semiconductor (MOS) technologies using dynamic circuit design techniques, such as precharging and latching cross-coupled pairs of MOS transistors with positive feedback.
  • MOS metal oxide semiconductor
  • Figure 1 illustrates one prior art dynamic latching input buffer.
  • An external signal Input is compared to a reference voltage VREF- If the signal Input has a voltage level which exceeds VREF- then the signal
  • Latchl is a latching signal. When the latching signal transitions from a high level to a low level, the latch signal is referred to as a falling edge signal. When the latching signal transitions from a low level to a high level, the latch signal is referred to as a rising edge signal.
  • the prior art shown in Figure 1 requires multiple latch signals (Latchl and Latch2).
  • Latchl and Latch2 are often collectively referred to as multi-phase latch signals because of their timing relationship to each other. Latchl and Latch2 should operate in a precisely timed relationship for maximum performance.
  • the circuit shown in Figure 1 may suffer from charge "kick-back.” After Latch2 has risen, the voltages on nodes 130 and 140 have been amplified to approximately the full power supply levels. For example, if Input > VREF- then the voltage at node 130, V130, is approximately VDD.
  • transistor 1 10 couples node 130 to Input
  • transistor 120 couples node 140 to VREF-
  • the charge on node 130 flows back to the input terminal 1 12.
  • the charge on node 140 similarly flows back to the reference voltage terminal 122.
  • This charge "kick-back" can inject high frequency noise onto the Input signal and the VREF signal. Since a number of data receivers are typically coupled to receive VREF. this may result in substantial high frequency noise injected back onto the VREF signal. This can create a data- dependent disturbance on VREF which can be a factor in causing the data receivers to malfunction on the next latch cycle.
  • FIG. 2 illustrates another dynamic latching input buffer according to the prior art. As in Figure 1 , the external signal Input is compared against a reference voltage VREF-
  • a second disadvantage of this circuit is the capacitive loading on the critical cross-coupled nodes Output 250 and Output 260.
  • these nodes are connected to two MOSFET gates and four MOSFET drains (in addition to the next stage of logic circuitry which is not shown). This capacitance may slow down circuit operation.
  • One of the objectives of the present invention is to provide latching input buffer circuitry that requires only a single latch signal.
  • Another object of the invention is to provide latching input buffer circuitry that dissipates substantially zero DC power.
  • Another object of the invention is to provide latching input buffer circuitry that minimizes capacitance loading on critical (positive feedback) nodes of the input buffer circuitry.
  • Another object of the invention is to provide latching input buffer circuitry that minimizes charge "kick-back" into the input buffer circuitry inputs.
  • Another object of the invention is to provide latching input buffer circuitry that minimizes capacitive "kick-back" into the input buffer circuitry inputs.
  • a metal-oxide semiconductor dynamic latching input buffer latches a data signal in response to a single latch signal.
  • a precharging circuit precharges an output terminal.
  • An input sampler provides a sampled data voltage from the data signal in response to an edge of the latch signal. The sampled data voltage is substantially independent of the data signal after the latch signal is received.
  • a reference voltage sampler provides a sampled reference voltage from a reference voltage in response to the latch edge. The sampled reference voltage is substantially independent of the data signal after the latch signal is received.
  • a comparator provides an output signal indicating the greater of the sampled reference voltage and the sampled data voltage. The comparator and the input and reference voltage samplers cooperate such that the input buffer d.c. power consumption is substantially zero.
  • Figure 1 is a schematic of a prior art dynamic latching input buffer.
  • Figure 2 is a schematic of a prior art improvement of the circuitry of Figure 1.
  • Figure 3 illustrates one embodiment of the dynamic latching input buffer.
  • Figure 4 illustrates an alternative embodiment of the dynamic latching input buffer.
  • Figure 5 illustrates an alternative embodiment of the dynamic latching input buffer.
  • Figure 6 illustrates an alternative embodiment of the dynamic latching input buffer.
  • transistors 520 and 530 turn on. Current begins to flow through transistors 540, 520, and 500 as well as through 550, 530, and 510. These currents cause the voltage levels of the Output 560 and Output 565 nodes to be substantially less than VDD.
  • the Latch signal is applied to gates 522, 532 of transistors 520 and 530, respectively.
  • transistors 520 and 530 are turned on.
  • transistor 520 is turned on, current can flow from the drain 524 to the source 526.
  • transistor 530 is applied to gates 522, 532 of transistors 520 and 530, respectively.
  • transistor 500 will be turned on "harder” than transistor 510 (i.e., the drain-to-source resistance of 500 will be less than the drain-to-source resistance of 510). Thus more current will flow through 540, 520, and 500 than through 550, 530, and 510. This causes the voltage at the Output node 565 to decrease faster than the voltage at the Output node 560. Thus a voltage difference is established between the Output 560 and Output 565 nodes.
  • Cross-coupled transistors 575 and 580 amplify this voltage difference.
  • Cross-coupled transistors 540 and 550 also amplify this voltage difference.
  • Transistors 575 and 580 are connected in a positive feedback configuration. As voltage at the Output node 565 decreases, the gate- to-source voltage on transistor 580, VQS-580- increases. This in turn causes the voltage at the Output node 560 to rise. When the Output voltage rises, the gate-to-source voltage on transistor 575, VQS-575. decreases. The decreasing voltage VQS-575 causes the voltage on the
  • Transistors 540 and 550 are also cross-coupled in a positive feedback configuration. This increases the gain and the gain- bandwidth product of the circuit, thus helping to increase the speed and the sensitivity of the latching input buffer circuitry.
  • Transistors 500 and 510 should operate in the "triode region" (nonsaturation region).
  • the voltage drop across 500 should be smaller than the voltage drop across 540.
  • the drain-to-source voltage of 500, VDS-500. should be smaller than the drain-to-source voltage of 540, VDS-540-
  • the voltage drop across 510 should be smaller than the drop across 550. (In other words, VQS-510 ⁇ VDS-550)- This allows cross-coupled amplifier transistors 540 and 550 to have increased gain and bandwidth. This in turn helps to increase the speed and the sensitivity of the latching input buffer circuitry.
  • the lower voltage drop across 500 and 510 also minimizes charge kick-back.
  • the voltage difference between the Output node 560 and Output node 565 is amplified until the Output and Output signal voltages approximate power supply limits. Assume the voltage level at the Output node 560 is approximately VDD and that the voltage level at the
  • Output node 565 is approximately VSS.
  • transistors 550 and 575 will be turned off and transistors 540 and 580 will be on.
  • transistor 550 is turned off while 540 is on.
  • the Output node cannot be forced from a high level to a low level because transistor 550 is cut off.
  • the voltage level at the Output node 565 is approximately VDD and the voltage level at the Output node 560 is approximately VSS, there is still no DC current path. In this case 540 will be turned off and the Output node 565 cannot be forced to a low level.
  • Output 560 and Output 565 is also important for speed concerns.
  • the Output node is connected to two MOSFET gates (552 and 582), three MOSFET drains (544, 578, 573) and the next logic stage (not shown). The count is identical for the Output node 560.
  • the Output node 560 is connected to two MOSFET gates (542 and 577), three MOSFET drains (554, 583, 588) and the next logic stage (not shown).
  • the present embodiment uses only a single latch signal, Latch. As discussed previously, this is advantageous when the highest performance is desired because it eliminates problems with phase-to-phase skew, especially in the presence of long latch signal lines having non-negligible metallization RC delay and waveshape distortion.
  • the present embodiment has minimal charge kick-back into the input nodes 502 and 512.
  • Input and VREF were connected to nodes that were previously charged to VDD or VSS thus forcing charge to be kicked back into Input and VREF- No such connection is illustrated in Figure 3.
  • the present embodiment also has minimal capacitive kick-back into the inputs.
  • the input signals, Input and VREF are coupled to the gates of g rounded-source transistors, so the primary mechanism for capacitive kick-back is gate-to-drain capacitive coupling.
  • the transistor sizes are selected so that the voltage at the drains 526, 514 of the input transistors 500 and 510, respectively, does not fluctuate significantly and therefore the drain-to-gate charge injection is small.
  • the latching circuit is activated by a rising edge of the Latch signal.
  • the latching circuit is activated by a falling edge of the Latch signal.
  • the NMOS transistors 520, 530, 540, and 550 of Figure 3 tend to be faster than the PMOS transistors 620, 630, 640, and 650 of Figure 4 due to carrier mobility in NMOS transistors being higher than carrier mobility in PMOS transistors. Use of NMOS transistors helps to ensure faster latching operation.
  • the common-mode voltage of the input signals, Input and VREF. should be near VDD (or VSS when using the circuitry of Figure 4). This helps to ensure that transistors 500 and 510 receive a gate-to-source voltage which will place each of them in the triode region of operation.
  • One way of achieving such a common-mode input voltage is to use the circuitry in an application where the interface levels are defined to be at or near the power supply voltages. Some examples include current mode logic (CML), Gunning transceiver logic (GTL), Rambus signaling levels (RSL), and so forth.
  • Another way of helping to ensure that the circuitry of Figure 3 receives a large common-mode input voltage is to precede the circuitry by a preamplifier which has a large (near power supply levels) common-mode output voltage.
  • CMOS fabrication processes contain an optional step which can be advantageously exploited in the present embodiment.
  • these processes offer a second NMOS transistor (variously called “natural,” “unimplanted,” or “zero threshold”) with a low threshold voltage (-0.15 ⁇ VJH ⁇ +0.15 volts).
  • a low threshold voltage transistor -0.15 ⁇ VJH ⁇ +0.15 volts.
  • transistors 500 and 510 have a threshold voltage less than zero volts, there will still not be any substantial DC power dissipation. This is because transistors 540 and 550 cut off the DC current path. As long as 540 and 550 are enhancement mode transistors (i.e. having a threshold voltage above zero), the DC power dissipation will be approximately zero.
  • transistors 500 and 520 have switched positions from their respective locations in Figure 3.
  • transistors 510 and 530 have also exchanged positions from their respective locations in Figure 3.
  • FIG. 6 Another such embodiment is shown in Figure 6 where the series order of transistors 500, 520, and 540 in Figure 3 has been altered so that transistor 540 is serially coupled between transistors 520 and 500. Similarly, transistor 550 has been repositioned between transistors 530 and 510.

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  • Logic Circuits (AREA)

Abstract

Circuit tampon d'entrée CMOS à verrouillage dynamique servant d'interface entre des circuits intégrés et des circuits extérieurs. Les circuits dudit tampon n'utilisent qu'un seul signal de verrouillage. Ce tampon réduit la capacité en excès sur les noeuds critiques (560, 565) ainsi que les retours de charge et de capacité sur les entrées (502, 512) du circuit de verrouillage. De plus, le circuit tampon d'entrée ne dissipe pratiquement pas de courant continu.
PCT/US1995/012094 1994-09-30 1995-09-18 Circuit tampon d'entree cmos a verrouillage dynamique WO1996010866A1 (fr)

Priority Applications (1)

Application Number Priority Date Filing Date Title
AU37218/95A AU3721895A (en) 1994-09-30 1995-09-18 Cmos dynamic latching input buffer circuit

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US31591394A 1994-09-30 1994-09-30
US08/315,913 1994-09-30

Publications (1)

Publication Number Publication Date
WO1996010866A1 true WO1996010866A1 (fr) 1996-04-11

Family

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Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/US1995/012094 WO1996010866A1 (fr) 1994-09-30 1995-09-18 Circuit tampon d'entree cmos a verrouillage dynamique

Country Status (3)

Country Link
AU (1) AU3721895A (fr)
TW (1) TW280027B (fr)
WO (1) WO1996010866A1 (fr)

Cited By (10)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5838177A (en) * 1997-01-06 1998-11-17 Micron Technology, Inc. Adjustable output driver circuit having parallel pull-up and pull-down elements
EP0896430A2 (fr) * 1997-08-06 1999-02-10 Lucent Technologies Inc. Circuit de verrouillage haute vitesse activé par horloge
US5872736A (en) * 1996-10-28 1999-02-16 Micron Technology, Inc. High speed input buffer
US5917758A (en) * 1996-11-04 1999-06-29 Micron Technology, Inc. Adjustable output driver circuit
US5949254A (en) * 1996-11-26 1999-09-07 Micron Technology, Inc. Adjustable output driver circuit
FR2948809A1 (fr) * 2009-07-31 2011-02-04 St Microelectronics Rousset Amplificateur de lecture faible puissance auto-minute
US8433023B2 (en) 1999-03-01 2013-04-30 Round Rock Research, Llc Method and apparatus for generating a phase dependent control signal
US8565008B2 (en) 1997-06-20 2013-10-22 Round Rock Research, Llc Method and apparatus for generating a sequence of clock signals
US8892974B2 (en) 2003-06-12 2014-11-18 Round Rock Research, Llc Dynamic synchronization of data capture on an optical or other high speed communications link
KR20200084907A (ko) * 2017-12-05 2020-07-13 마이크론 테크놀로지, 인크. 입력 버퍼 회로

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4973864A (en) * 1988-07-13 1990-11-27 Kabushiki Kaisha Toshiba Sense circuit for use in semiconductor memory
EP0407591A1 (fr) * 1988-10-11 1991-01-16 Oki Electric Industry Co., Ltd. Circuit amplificateur differentiel
US5132567A (en) * 1991-04-18 1992-07-21 International Business Machines Corporation Low threshold BiCMOS circuit

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4973864A (en) * 1988-07-13 1990-11-27 Kabushiki Kaisha Toshiba Sense circuit for use in semiconductor memory
EP0407591A1 (fr) * 1988-10-11 1991-01-16 Oki Electric Industry Co., Ltd. Circuit amplificateur differentiel
US5132567A (en) * 1991-04-18 1992-07-21 International Business Machines Corporation Low threshold BiCMOS circuit

Cited By (21)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US5872736A (en) * 1996-10-28 1999-02-16 Micron Technology, Inc. High speed input buffer
US5910920A (en) * 1996-10-28 1999-06-08 Micron Technology, Inc. High speed input buffer
US6326810B1 (en) 1996-11-04 2001-12-04 Micron Technology, Inc. Adjustable output driver circuit
US5917758A (en) * 1996-11-04 1999-06-29 Micron Technology, Inc. Adjustable output driver circuit
US6437600B1 (en) 1996-11-04 2002-08-20 Micron Technology, Inc. Adjustable output driver circuit
US5949254A (en) * 1996-11-26 1999-09-07 Micron Technology, Inc. Adjustable output driver circuit
US6084434A (en) * 1996-11-26 2000-07-04 Micron Technology, Inc. Adjustable output driver circuit
US5838177A (en) * 1997-01-06 1998-11-17 Micron Technology, Inc. Adjustable output driver circuit having parallel pull-up and pull-down elements
US6069504A (en) * 1997-01-06 2000-05-30 Micron Technnology, Inc. Adjustable output driver circuit having parallel pull-up and pull-down elements
US8565008B2 (en) 1997-06-20 2013-10-22 Round Rock Research, Llc Method and apparatus for generating a sequence of clock signals
US6018260A (en) * 1997-08-06 2000-01-25 Lucent Technologies Inc. High-speed clock-enabled latch circuit
EP0896430A3 (fr) * 1997-08-06 1999-03-10 Lucent Technologies Inc. Circuit de verrouillage haute vitesse activé par horloge
EP0896430A2 (fr) * 1997-08-06 1999-02-10 Lucent Technologies Inc. Circuit de verrouillage haute vitesse activé par horloge
US8433023B2 (en) 1999-03-01 2013-04-30 Round Rock Research, Llc Method and apparatus for generating a phase dependent control signal
US8892974B2 (en) 2003-06-12 2014-11-18 Round Rock Research, Llc Dynamic synchronization of data capture on an optical or other high speed communications link
FR2948809A1 (fr) * 2009-07-31 2011-02-04 St Microelectronics Rousset Amplificateur de lecture faible puissance auto-minute
EP2287850A1 (fr) * 2009-07-31 2011-02-23 STMicroelectronics (Rousset) SAS Amplificateur de lecture faible puissance auto-minuté
US8363499B2 (en) 2009-07-31 2013-01-29 STMicroelectrics (Rousset) SAS Self-timed low power sense amplifier
KR20200084907A (ko) * 2017-12-05 2020-07-13 마이크론 테크놀로지, 인크. 입력 버퍼 회로
KR102373860B1 (ko) 2017-12-05 2022-03-14 마이크론 테크놀로지, 인크. 입력 버퍼 회로
US11349479B2 (en) 2017-12-05 2022-05-31 Micron Technology, Inc. Input buffer circuit

Also Published As

Publication number Publication date
TW280027B (fr) 1996-07-01
AU3721895A (en) 1996-04-26

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