WO1992006542A1 - A carrier recovery method and apparatus having an adjustable response time determined by carrier signal parameters - Google Patents
A carrier recovery method and apparatus having an adjustable response time determined by carrier signal parameters Download PDFInfo
- Publication number
- WO1992006542A1 WO1992006542A1 PCT/US1991/006924 US9106924W WO9206542A1 WO 1992006542 A1 WO1992006542 A1 WO 1992006542A1 US 9106924 W US9106924 W US 9106924W WO 9206542 A1 WO9206542 A1 WO 9206542A1
- Authority
- WO
- WIPO (PCT)
- Prior art keywords
- signal
- phase
- carrier
- tdma
- carrier signal
- Prior art date
Links
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
- H04L27/22—Demodulator circuits; Receiver circuits
- H04L27/227—Demodulator circuits; Receiver circuits using coherent demodulation
- H04L27/2271—Demodulator circuits; Receiver circuits using coherent demodulation wherein the carrier recovery circuit uses only the demodulated signals
- H04L27/2273—Demodulator circuits; Receiver circuits using coherent demodulation wherein the carrier recovery circuit uses only the demodulated signals associated with quadrature demodulation, e.g. Costas loop
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/06—Receivers
- H04B1/10—Means associated with receiver for limiting or suppressing noise or interference
Definitions
- the present invention relates generally to communication receivers, and, more particularly, to a communication receiver with a carrier recovery method and apparatus having an adjustable response time determined by carrier signal parameters.
- a communication receiver with a carrier recovery method and apparatus having an adjustable response time determined by carrier signal parameters.
- a radio communication system is comprised, at minimum, of a transmitter and a receiver.
- the transmitter and the receiver are interconnected by a radio-frequency channel to permit transmission of an information signal therebetween.
- the information signal is impressed upon a radio-frequency electromagnetic wave by a process referred to as modulation to permit transmission of the information signal between the transmitter and the receiver.
- the radio- frequency electromagnetic wave is referred to as a carrier wave which is of a particular frequency
- the carrier wave, once modulated by the information signal is referred to as a modulated information signal.
- the modulated information signal may be transmitted through free space to transmit thereby the information between the transmitter and the receiver.
- AM Amplitude modulation
- FM frequency modulation
- PM phase modulation
- CM composite modulation
- an amplitude modulated signal is formed by impressing (i.e., modulating) an information signal upon a carrier wave such that the information signal modifies the amplitude of the carrier wave corresponding to the value of the information signal.
- An angle modulated signal formed is formed by impressing (i.e., modulating) an information signal upon a carrier wave such that the information signal modifies the phase (or the time differential of phase, frequency) of the carrier wave corresponding to the value of the information signal.
- Angle modulation does not cause the amplitude of the carrier wave to vary, and the information content of the modulated information signal is contained in the variation of the phase (or frequency) of the signal. Because the amplitude of an angle modulated signal does not vary, an angle modulated signal is referred to as a constant envelope signal.
- a composite modulated signal is formed by impressing
- the carrier wave (or a carrier intermediate frequency, i.e., IF, source) is first separated into sine wave and cosine wave carrier portions. Separate portions, referred to as the in-phase (or I) and the quadrature (or Q) components, of the information signal are impressed upon the cosine wave and sine wave carrier portions of the carrier wave, respectively. The sine wave and cosine wave components are then recombined, and the resultant signal, the composite modulated signal, varies in both amplitude, and, additionally, phase.
- I in-phase
- Q quadrature
- Composite modulation is advantageous in that a composite modulated signal permits a greater amount of information to be transmitted within a frequency bandwidth than a signal generated by any of the previously mentioned modulation techniques. See, for instance, a discussion in the text Introduction to Communication Systems, 2nd Ed, by Ferrel G. Stremmler, ISBN 0-201-07251-3, pages 590-596.
- a family type of composite modulation is quadrature amplitude modulation (QAM).
- QAM quadrature amplitude modulation
- the binary data stream is separated into bit pairs.
- the individual bits of the bit pairs are converted from unipolar to bipolar format, passed through a pair of electric wave filters, and applied to the multiplier pair whose other inputs are the sine and cosine components of the carrier or carrier IF signal.
- a particular type of QAM is ⁇ /4-shift DQPSK (for differential quadrature phase shift keying), in which the input data stream is encoded so that the composite modulated carrier shifts in increments of ⁇ /4 or ⁇ 3 ⁇ /4 according to the input bit pairs.
- a receiver which receives a modulated information signal includes circuitry to detect, or otherwise to recreate, the information signal modulated upon the carrier wave. This process is referred to as demodulation.
- demodulation As many different modulated information signals may be simultaneously transmitted by a plurality of transmitters at a plurality of different frequencies, a receiver contains tuning circuitry to demodulate only those signals received by the receiver which are of certain desired frequencies.
- the broad range of frequencies at which modulated information signals may be transmitted is referred to as the electromagnetic frequency spectrum. Regulation of radio-frequency communications in certain frequency bands of the electromagnetic frequency spectrum minimizes interference between simultaneously transmitted signals.
- portions of a 100 MHz band of the electromagnetic frequency spectrum are allocated for radiotelephone communication, such as, for example, communication effectuated by radiotelephones utilized in a cellular, communication system.
- Existing radiotelephones contain circuitry both to generate and to receive radio-frequency modulated information signals.
- a cellular communications system is created by positioning numerous base stations at specific locations throughout a geographical area.
- Each of the base stations is constructed to receive and to transmit modulated information signals simultaneously to and from radiotelephones to permit two-way communication there between.
- Each of the base stations is provided with means to communicate with one or more switching offices which permit connection to the conventional telephone network.
- the base stations are positioned at locations such that a radiotelephone at any location throughout the geographical area is within the reception range of at least one of the base station receivers.
- the geographical area is divided into portions, and one base station is positioned in each portion. Each portion of the geographical area defined thereby is referred to as a "cell".
- each modulated information signal when transmitted, occupies a finite portion of the frequency band.
- Substantial overlapping of simultaneously transmitted modulated information signals at the same frequency in the same geographic area is impermissible as interference between overlapping signals at the same frequency could prevent detection of either of the transmitted modulated information signals by a receiver. Frequency re-use is permitted if sufficient geographic separation exists between base sites using the same frequency, because of the attenuation of signals with distance.
- the frequency band allocated for radiotelephone communication in the U.S. is divided into channels, each of which is of a 30 KHz bandwidth.
- a first portion, extending between 824 MHz and 849 MHz of the frequency band, is allocated for the transmission of modulated information signals from a radiotelephone to a base station.
- a second portion, extending between 869 MHz and 894 MHz of the frequency band is allocated for the transmission of modulation information signals from a base station to a radiotelephone.
- the modulation technique utilized by radiotelephone communication systems to form the modulated information signal thereby is angle modulation.
- an angle modulated signal impresses an information signal upon a carrier wave to modify the frequency (FM) or phase (PM) of the carrier wave according to the value of the information signal.
- FM frequency
- PM phase
- conventional angle modulation techniques use spectral resources inefficiently.
- the voice signal to be transmitted which contains substantial redundant information, is modulated onto the carrier without substantial removal of the redundancy.
- the total bandwidth required for transmission of information for a given modulation method is directly proportional to the amount of information to be transmitted.
- an information signal (such as a voice signal) which is to be transmitted is first encoded according to a redundancy-reduction scheme. Once encoded, the information signal, in encoded form, is modulated upon a carrier wave and is transmitted in sequential intermittent time segments. Other information signals may similarly be encoded, modulated, and transmitted in intermittent bursts at the same frequency by other transmitters. Thus, a greater number information signals may be transmitted within a particular frequency bandwidth. When the information signals are generated by users of radiotelephones forming a portion of a cellular communications system, a greater number of radiotelephones may be operated within a particular frequency bandwidth when such a TDMA technique is utilized.
- a receiver constructed to receive a TDMA signal, such as a TDMA composite modulated signal, reconstructs the original information signal by decoding the TDMA signal transmitted to the receiver in one of the sequential time segments.
- a TDMA signal such as a TDMA composite modulated signal
- a receiver constructed to receive TDMA composite- modulated signals may also require circuitry to perform channel equalization in the receiver. Equalizer circuitry is required to correct for delay problems associated with reflections of signals transmitted to the receiver which arrive at the receiver at different times. Because the signal received by a receiver is actually a vector sum of all signals received at a particular frequency, the signal received by a receiver may actually be comprised of the same signal received at different times as the signal may be reflected off objects prior to reception thereof by the receiver. The signal actually received by the receiver is, therefore, the sum of all signals which are transmitted to the receiver along many different paths. The path lengths may vary, and hence the signal actually received by the receiver may vary, responsive to repositioning of the receiver.
- Equalizer circuitry is oftentimes formed by a processor having an appropriate software process embodied therein.
- the receiver In order to permit optimal operation of the equalizer circuitry, the receiver should be constructed to be linear (i.e., the demodulated signals should represent accurately the original I and Q portions modulated onto the carrier).
- the number of, phase of, and intensity of, signals actually received by a receiver in a multipath environment may vary over time as a result of repositioning of the receiver, or of the objects from which a transmitted signal is reflected. As a result, the phase and signal level of a received signal varies over time. This variance is referred to as "fading" of the signal.
- the resultant signal strength and rate of change of signal strength at the receiver is predominantly determined by how rapidly the receiver is moving through its environment, and the frequency of the channel being used. For instance, in the cellular frequency band, and when a cellular radiotelephone is positioned in a vehicle travelling at sixty miles per hour, the signal strength of the received signal can vary by approximately twenty decibels during a five millisecond period.
- the MAHO test also requires gain control circuitry which permits rapid and continuous tracking of a signal.
- Optimal receiver performance is realized for composite modulation if the receiver incorporates a means for generating an estimate of the carrier phase of the received signal.
- Receivers which generate such an estimate are known as coherent receivers.
- the process of generating the phase estimate is known as carrier recovery.
- Several methods of carrier recovery are known.
- One such method applicable to carrier recovery for receiving a signal under fading conditions in a TDMA system with ⁇ 4-shift DQPSK modulation is a decision feedback phase lock loop (DFPLL).
- a DFPLL determines what the phase-error of the received signal is relative to an ideal received signal.
- the phase-error signal is coupled through a loop filter to remove noise.
- the phase-error, with a reduced noise level is coupled to a voltage controlled oscillator (VCO).
- VCO voltage controlled oscillator
- the phase of the VCO is adjusted based on the phase-error input.
- the corrected phase out of the VCO is multiplied with the received signal's quadrature components to correct the phase of the
- Another method applicable to carrier recovery for the aforementioned system is to raise the received signal components to the 4th power, which removes a substantial portion of the modulation, low-pass filter the resultant, and apply the low-pass filter output to a phase-correction input of the reference phase source.
- This can be generalized to an M- th power carrier recovery apparatus for M-ary signaling.
- Another method for carrier recovery is called the generalized-Costas loop. This method requires multiplying the received signal by M-phase shifted reference signals. Where M equals 8 for ⁇ /4 DQPSK signaling.
- the reference phase signal is separated into eight components phase shifted by 0, ⁇ 8, ⁇ /4, 3 ⁇ /8, ⁇ _2, 5 ⁇ /8, 3 ⁇ /4 and 7 ⁇ /8 radians. These components multiply the received signal; the products generated are low-pass filtered, the filter outputs are then multiplied to generate a phase-correction signal which is applied to a phase correction input of the reference phase source.
- the response time of the carrier recovery process is determined by the filtering (or averaging) applied in the generation of the phase correction signal.
- filtering or averaging
- Such methods were suitable for and commonly applied to systems where substantial multipath effects did not exist, such as satellite communications links, or fixed terrestrial point-to-point links. Because of the aforementioned fading effects, it is undesirable to perform carrier recovery using methods which do not adapt to variations of the received signal due to fading effects.
- the signal, during a fade event is corrupted to some degree by noise energy. Thus, it corrupts the phase estimation of the received signal.
- a carrier recovery apparatus for receiving a carrier signal which has a carrier signal phase and carrier signal parameters has a phase-error estimate signal generated by a comparator which compares the carrier signal phase to a reference signal phase generated by a reference signal source.
- At least one of the carrier signal parameters is determined from the carrier, and the phase-error estimate signal is adjustably coupled to the reference signal source in response to the determined carrier signal parameter.
- FIG. 1 is a block diagram of a transmitter and receiver which may employ the present invention.
- FIG. 2 is a block diagram of a signal processor which may employ the present invention.
- FIG. 3 is a block diagram of an energy estimator which may employ the present invention.
- Figure 4 is a channel state diagram for a TDMA event sequence which may employ the present invention.
- a system which may employ the present invention is shown in the block diagram of Figure 1.
- the figure shows a transmitter 101 and a receiver 103.
- the receiver 103 comprises a carrier recovery apparatus for ⁇ 4-shift DQPSK modulation.
- Receiver 103 elements comprise a signal processor 141, an information decoder 143 and an adjustable gain preamplifier 105. All other elements in the receiver collectively comprise a decision feedback phase lock loop.
- the decision feedback phase lock loop (DFPLL) generates a phase- error term at the output of summer block 121 which corrects the phase of the local estimate of received phase generated by voltage controlled oscillator 109.
- the signal processor 141 uses quadrature component signals, I'(t) and Q'(t), and synchronous signal 144 to generate a control signal 142 for adjustable gain preamplifier 105 and synchronous TDMA timing reference signal 140 for the frequency response of loop filter 123.
- the control signal 142 for adjustable gain preamplifier 105 is adjusted based on the energy level of the received signal 102.
- the control signal 140 for the frequency response of loop filter 123 is based on parameters of the received signal. These parameters comprise: the energy level, the rate of change of the energy level and the clocked TDMA event sequence of the received signal 102.
- the control signal 140 for loop filter 123 adjusts the response time for the DFPLL. An adjustable response time may improve the quality of the received information signal 102.
- the response time in a conventional DFPLL is not adjustable during fading conditions.
- An adjustable response time is desirable during fading conditions.
- the received signal is entering a fade, the energy level of the signal is gets weaker thereby approaching the noise floor. Under this condition, it is desirable to slow down the loop response time by narrowing the loop bandwidth filter.
- the narrow loop bandwidth filter increases the signal to noise ratio for the received signal.
- the slower loop response time enables the last good received signal's phase-error estimate to remain as long as possible before the next signal is received.
- the received signal is exiting a fade the energy level of the signal gets stronger thereby rising higher above the noise floor. Under this condition, it is desirable to speed up the loop response time by widening the loop bandwidth filter. The faster loop response time enables the next good received signal's phase-error estimate to be acquired as rapidly as possible.
- the response time in the DFPLL phase-error correction system is not dependent on the timing of the clocked TDMA event sequence. There are times during the TDMA event sequence when it is desirable to have a fast loop response time such as when we first lock on to a received signal. There are other times during the TDMA event sequence when it is desirable to have a slow loop response time such as when we are receiving a signal which is entering a fade.
- a TDMA transmitter 101 transmits a digital information signal which is modulated on the I(t) and Q(t) quadrature component signals described previously. This information signal is broadcast to a receiver 103 which receives and demodulates the information signal. As previously noted, the information signal may be subject to random phase delays and multipath fading which degrade the quality of the information signal.
- the phase error in the received information signal 102 is denoted by I e (t) and Q e (t).
- the information signal 102 is coupled to adjustable gain preamplifier 105 which reduces input signal energy level variations as discussed in U.S. Patent Application No. (CE00347R) "Input-Estimating Fast AGC System With Adaptive Gain Adjustment".
- the output signal of adjustable gain preamplifier 105 is subsequently separated into quadrature signal components I(t) and Q(t).
- Quadrature signal component, Q(t) is generated by applying the amplified signal to a conventional mixer 107 where a local oscillator signal generated by VCO 109 is shifted by a fixed 90° phase shifter 111 to create the Q(t) signal.
- the I(t) signal is created from mixer 113 and the output local oscillator signal from VCO 109.
- Quadrature signal components, I(t) and Q(t) are now phase coherent with the transmitted signal.
- Quadrature signal component, Q(t), is filtered through matched filter 131 and subsequently coupled to analog to digital converter 132. Digitized signal, Q(t), is split into two paths. One path is mixed in mixer 117 with an estimated phase signal processed by sine determinator 119. The other path is sampled by conventional sampler 133 before being applied to a first port of phase estimator 135, a first port of signal processor 141 and a first port of information decoder 143. likewise, quadrature signal component, I(t), is filtered through matched filter 137 and subsequently coupled to analog to digital converter 138. Digitized signal, I(t), is split into two paths. One path is mixed in mixer 127 with an estimated phase signal processed by cosine determinator 129. The other path is sampled by conventional sampler 139 before being applied to a second port of phase estimator 135, a second port of signal processor 141 and a second port of information decoder 143.
- the resultant mixed signals from mixers 117 and 127 are coupled to a conventional summer 121.
- the output phase- error signal of summer 121 is coupled to a digital to analog converter 124 through loop filter 123.
- the analog phase-error signal is applied to VCO 109.
- VCO 109 generates a local oscillator signal to be applied to mixer 113 and serially applied to 90° phase shifter 111 first then to mixer 107.
- the phase-error estimate signal of summer 121 is generated by a comparator which compares the received carrier signal phase (first input to mixer 107) to a reference signal phase (second input to mixer 107) generated by a reference signal source.
- the comparator comprises: mixer 107, matched filter 131, A/D converter 132, sampler 133, mixer 117, summer 121, sine and cosine determinator 119 and 129, mixer 127 and phase estimator 135.
- the reference signal source comprises: voltage controlled oscillator 109, D/A converter 124 and 90° phase shifter 111. Analogous terms may also be described for the quadrature signal I(t) side of the DFPLL.
- Signal processor 141 receives inputs I'(t), Q'(t) and information decoder output 144. Signal processor 141 generates two outputs. The first output is a control signal 142 for adjustable gain preamplifier 105. The second output 140 is a control signal for the frequency response of loop filter 123. Information decoder 143 receives inputs I'(t) and Q'(t) and generates outputs to the signal processor 141 and speaker 145. The information decoder represents elements of a receiver not shown in FIG. 1. The first output 144 to signal processor 141 provides information about the position of a received signal 102 in the TDMA event sequence. The second output 102 is an analog output which may be coupled to a conventional speaker 145.
- FIG. 2 a block diagram of signal processor 141 is shown.
- Digitally sampled signals, I'(t) and Q'(t) are coupled to energy estimator 205.
- a cellular radiotelephone transceiver in compliance with IS-54 must determine an estimate of the energy level of the received signal and process it in accordance with Section 2.4.5.4.1.2.1. Further details on the function of the energy estimator 205 will be discussed with FIG. 3.
- the output of the energy estimator 205 is coupled to adjustable gain preamplifier 105 using control signal 142, energy level detector 207, and rate of change of the energy level detector 209.
- the output 142 from energy estimator 205 to the adjustable gain preamplifier 105 is a feedback path which reduces received input signal variations.
- a software process in the signal processor determines the energy level of the received information signal.
- the energy level determinator 207 can be described by the following equation:
- Vctl is the feedback signal 142 also used to adjust the input gain to reduce input signal variations.
- Limiter 211 creates a threshold level for high levels of received information signals. Limiter 211 only allows measured energy levels up to a maximum predetermined value. The output of 211 is coupled to product combiner 213.
- the output of energy estimator 205 is also coupled into a rate of change of energy level determinator 209.
- the rate of change of energy level determinator describes how the energy level of the received information signal changes with time.
- the rate of change of energy level can be described by the following equation:
- Limiter 215 creates a low threshold for the measure of the rate of change of energy level.
- the low threshold is set to a predetermined value to prevent the calculation of rate of change of the energy level from approaching zero.
- the output of limiter 215 is coupled to product combiner 213.
- TDMA time clock 219 provides information about the position of the received signal in the TDMA event sequence.
- the output of the TDMA time clock 219 is coupled to look-up table 221.
- Look-up table 221 contains a number of predetermined values based on the receiver design and location within the TDMA event sequence.
- the output of look-up table 221 is coupled to product combiner 213.
- the product combiner 213 has as its inputs the energy level determinator 207 processed through a high threshold limiter 211, the rate of change of energy level determinator 209 processed through a low threshold limiter 215 and a predetermined value from look-up table 221 based on the clocked TDMA event sequence 221.
- the product combiner 213 is a software process which multiplies its three inputs to produce an output. The output of the product combiner is described by the following equation.
- the output of product combiner 213 is coupled to a digital to analog converter 223.
- the digital to analog converter 223 converts the digital signal from the signal processor 141 into an analog signal to adjust the frequency response time of the loop filter 225.
- the response time of the carrier recovery apparatus is adjusted with a variable bandwidth filter. Adjusting the response time may also be accomplished by varying the gain of the phase-error signal coupled to a fixed bandwidth filter or by averaging the phase- error signal over an adjustable time period.
- signal processor 141 is implemented as a digital signal processor (DSP) employing a DSP 56001 available from Motorola, Inc.
- DSP digital signal processor
- the DSP combines these parameters to achieve an output which controls the decision feedback phase lock loop frequency response time for loop filter 225.
- the DSP also produces an output to control the adjustable gain preamplifier 105.
- FIG. 3 there is shown a block diagram of the contents of the energy estimator 205.
- Received quadrature component signals, I'(t) and Q'(t) are coupled to averager 301 which calculates an average signal energy from the addition of sample pairs of the squared amplitude of I'(t) and Q'(t).
- the averaged output is coupled to feedback gain adjust circuit 303 which adjusts the value to which I' 2 (t) and Q' 2 (t) are normalized.
- the logarithm (base 10) is conventionally taken (in 305) to produce the signal having logarithmic characteristics to match the exponential control function characteristics of the adjustable gain preamplifier 105.
- the output of logio (x) calculator 305 is coupled to voltage control sensitivity estimator 307 and mixer 309.
- the output of mixer 309 which represents a control signal corrected for rate of change of gain versus control voltage of the adjustable gain preamplifier 105, is coupled to delay and compare function 311.
- the output of delay and compare function 311 is coupled to control voltage sensitivity estimator 307 to correct for errors in the estimate of the rate of change of energy level of the adjustable gain preamplifier 105, and is also output to the other functions of the receiver.
- This output of the energy estimator 205 is provided as an automatic gain control (AGO signal to adjustable gain preamplifier 105.
- AGO signal to adjustable gain preamplifier 105 The voltage control sensitivity estimator 307 and the energy estimator 205 are further described in U.S. Patent Application No. (CE00347R) "Automatic Gain Control Apparatus and Method", filed on the same date herewith in behalf of Cahill.
- FIG 4 there is shown a timed sequence of events for mobile transmit and receive channels in a TDMA system.
- the mobile channel state 401 for a TDMA event sequence is generally divided into three time slots.
- the three time slots comprise transmitting (T) time slot 403, receiving (R) time slot 405, and idle (I) time slot 407.
- Each time slot is approximately 6.66 milliseconds long thereby generating a clocked sequence of events of 20 milliseconds for all three time slots.
- the transmitting 403, receiving 405, and idle 407 time slots are sequentially repeated in a TDMA event sequence.
- the advantage of a TDMA is to multiplex the number of users operating on the same frequency channel.
- the idle time slot 407 defines a time in the TDMA event sequence in which a particular mobile unit is neither transmitting nor receiving. During this idle time slot 407, the mobile unit intermittently measures the energy levels of transmitters located in one or more cells. This process of testing signal energy level is referred to as mobile-assisted hand-off or MAHO.
- the mobile receive 409 describes the receive time slots for three different mobile receivers described as x 411, Ry 413 and Rz 415. Each mobile receiver can only receive information during its allocated time slot.
- the mobile transmit channel 417 is likewise divided into three time slots. The three time slots comprise T x 419, T y 421, and T z 423. Each mobile unit may only transmit during its own allocated time slot.
- the transmit 417 and receive 419 channel pattern of the three adjacent time slots is sequentially repeated over time.
- mobile transmit time slot T x 419 is offset from mobile receive time slot Rx 411. This offset is primarily needed to account for real world system considerations such as propagation delay and the physical distance between transmitting and receiving units.
- the values in the look-up table 221 and the position within a TDMA event sequence help determine response time.
- a position in a TDMA event sequence defines a specific location in a time slot of a mobile channel.
- a fast loop response time is desirable immediately prior to the receive time slot 405 in order to rapidly acquire the phase-error and lock on to the received signal.
- the loop response time is determined by the energy level of the received signal and the rate of change of the energy level of the received signal.
- the control signal to adjust the loop response time is minimized or eliminated.
- the quality of the received information signal in a TDMA system can be improved in the presence of fading by adjusting the loop response time of the decision feedback phase lock loop.
- the loop response time is adjusted based on the energy level of the received signal, the rate of change of energy level of the received signal and the timed sequence of events in a TDMA system.
Landscapes
- Engineering & Computer Science (AREA)
- Computer Networks & Wireless Communication (AREA)
- Signal Processing (AREA)
- Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
- Mobile Radio Communication Systems (AREA)
- Cable Transmission Systems, Equalization Of Radio And Reduction Of Echo (AREA)
Priority Applications (3)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
DE4192400A DE4192400C2 (de) | 1990-09-28 | 1991-09-24 | Demodulationsverfahren- und Vorrichtung |
BR919105906A BR9105906A (pt) | 1990-09-28 | 1991-09-24 | Aparelho e processo de recuperacao de portadora,e receptor de radio |
GB9211281A GB2255867B (en) | 1990-09-28 | 1992-05-28 | Carrier recovery method and apparatus having an adjustable response time determined by carrier signal parameters |
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US07/590,401 US5150384A (en) | 1990-09-28 | 1990-09-28 | Carrier recovery method and apparatus having an adjustable response time determined by carrier signal parameters |
US590,401 | 1990-09-28 |
Publications (1)
Publication Number | Publication Date |
---|---|
WO1992006542A1 true WO1992006542A1 (en) | 1992-04-16 |
Family
ID=24362111
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
PCT/US1991/006924 WO1992006542A1 (en) | 1990-09-28 | 1991-09-24 | A carrier recovery method and apparatus having an adjustable response time determined by carrier signal parameters |
Country Status (7)
Cited By (1)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5828710A (en) * | 1995-12-11 | 1998-10-27 | Delco Electronics Corporation | AFC frequency synchronization network |
Families Citing this family (48)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5287351A (en) * | 1990-11-27 | 1994-02-15 | Scientific-Atlanta, Inc. | Method and apparatus for minimizing error propagation in correlative digital and communication system |
DK0587620T3 (da) * | 1991-06-03 | 1998-09-07 | British Telecomm | Radiosystem |
US5311545A (en) * | 1991-06-17 | 1994-05-10 | Hughes Aircraft Company | Modem for fading digital channels affected by multipath |
FR2681994B1 (fr) * | 1991-09-26 | 1994-09-30 | Alcatel Telspace | Dispositif de transmission numerique comportant un recepteur a demodulation coherente realisee directement en hyperfrequence. |
US5353311A (en) * | 1992-01-09 | 1994-10-04 | Nec Corporation | Radio transmitter |
US5271041A (en) * | 1992-03-16 | 1993-12-14 | Scientific-Atlanta, Inc. | Method and apparatus for QPR carrier recovery |
US5255290A (en) * | 1992-08-21 | 1993-10-19 | Teknekron Communications System, Inc. | Method and apparatus for combined frequency offset and timing offset estimation |
JP2797916B2 (ja) * | 1993-08-05 | 1998-09-17 | 日本電気株式会社 | 搬送波再生回路 |
KR970000661B1 (ko) * | 1993-12-29 | 1997-01-16 | 현대전자산업 주식회사 | 위상오차 검출기를 사용한 판정의거 반송파 복원회로 |
US5606581A (en) * | 1994-03-17 | 1997-02-25 | Myers; Glen A. | Method and apparatus for the cancellation of interference in electrical systems |
SE502813C2 (sv) * | 1994-05-04 | 1996-01-22 | Ericsson Telefon Ab L M | Metod och anordning vid analog-digitalomvandlare |
US5666429A (en) * | 1994-07-18 | 1997-09-09 | Motorola, Inc. | Energy estimator and method therefor |
US5692014A (en) * | 1995-02-03 | 1997-11-25 | Trw Inc. | Subsampled carrier recovery for high data rate demodulators |
US5703597A (en) * | 1995-12-22 | 1997-12-30 | Alliedsignal, Inc. | Adaptive carrier phase lock loop in a GPS receiver |
JP3310160B2 (ja) * | 1996-03-29 | 2002-07-29 | 松下電器産業株式会社 | スペクトラム拡散方式受信装置 |
WO1997050187A1 (en) * | 1996-06-27 | 1997-12-31 | Philips Electronics N.V. | Satellite receiver |
US6097768A (en) * | 1996-11-21 | 2000-08-01 | Dps Group, Inc. | Phase detector for carrier recovery in a DQPSK receiver |
US5901173A (en) * | 1996-12-09 | 1999-05-04 | Raytheon Company | Noise Estimator |
US6067328A (en) * | 1996-12-12 | 2000-05-23 | Alliedsignal | High precision hardware carrier frequency and phase aiding in a GPS receiver |
US6694154B1 (en) * | 1997-11-17 | 2004-02-17 | Ericsson Inc. | Method and apparatus for performing beam searching in a radio communication system |
JP3646010B2 (ja) * | 1998-09-18 | 2005-05-11 | 株式会社ケンウッド | デジタル衛星放送受信機 |
US6229841B1 (en) * | 1998-12-11 | 2001-05-08 | Qualcomm Incorporated | Method and apparatus for energy estimation in a wireless receiver capable of receiving multiple instances of a common signal |
DE10000008A1 (de) * | 2000-01-03 | 2001-07-12 | Alcatel Sa | Verfahren zur aufwandsarmen Signal-, Ton- und Phasenwechseldetektion |
DE60137657D1 (en) * | 2000-08-03 | 2009-03-26 | Infineon Technologies Ag | Flexible tdma systemarchitektur |
US6868129B2 (en) * | 2001-03-12 | 2005-03-15 | Freescale Semiconductor, Inc. | Demodulator for a radio receiver and method of operation |
EP1868382A3 (en) * | 2002-09-13 | 2008-02-27 | Sharp Kabushiki Kaisha | Broadcast program recording method, communication control device, and mobile communication device |
CN101233678B (zh) * | 2005-07-25 | 2011-05-04 | Nxp股份有限公司 | 用于调幅信号的接收器和检测接收器中的相位误差的方法 |
US8345801B2 (en) * | 2005-11-10 | 2013-01-01 | Weon-Ki Yoon | Apparatus and method for signal mismatch compensation in a wireless receiver |
US20070160168A1 (en) * | 2006-01-11 | 2007-07-12 | Beukema Troy J | Apparatus and method for signal phase control in an integrated radio circuit |
US7701371B2 (en) * | 2006-04-04 | 2010-04-20 | Qualcomm Incorporated | Digital gain computation for automatic gain control |
US8212941B2 (en) * | 2008-04-30 | 2012-07-03 | Mediatek Inc. | Digitized analog TV signal processing system |
US9288089B2 (en) | 2010-04-30 | 2016-03-15 | Ecole Polytechnique Federale De Lausanne (Epfl) | Orthogonal differential vector signaling |
US9251873B1 (en) | 2010-05-20 | 2016-02-02 | Kandou Labs, S.A. | Methods and systems for pin-efficient memory controller interface using vector signaling codes for chip-to-chip communications |
US8488697B2 (en) | 2011-05-06 | 2013-07-16 | Northrop Grumman Systems Corporation | Universal timing recovery circuit |
US8712361B1 (en) * | 2012-11-21 | 2014-04-29 | Broadcom Corporation | Method and system for reciprocal mixing cancellation of wideband modulated blockers |
US9806761B1 (en) | 2014-01-31 | 2017-10-31 | Kandou Labs, S.A. | Methods and systems for reduction of nearest-neighbor crosstalk |
CN110266615B (zh) | 2014-02-02 | 2022-04-29 | 康杜实验室公司 | 低isi比低功率芯片间通信方法和装置 |
US9509437B2 (en) | 2014-05-13 | 2016-11-29 | Kandou Labs, S.A. | Vector signaling code with improved noise margin |
US9900186B2 (en) * | 2014-07-10 | 2018-02-20 | Kandou Labs, S.A. | Vector signaling codes with increased signal to noise characteristics |
US9432082B2 (en) | 2014-07-17 | 2016-08-30 | Kandou Labs, S.A. | Bus reversable orthogonal differential vector signaling codes |
EP3152879B1 (en) | 2014-07-21 | 2019-09-04 | Kandou Labs S.A. | Multidrop data transfer |
KR101949964B1 (ko) | 2014-08-01 | 2019-02-20 | 칸도우 랩스 에스에이 | 임베딩된 클록을 갖는 직교 차동 벡터 시그널링 코드 |
US9674014B2 (en) | 2014-10-22 | 2017-06-06 | Kandou Labs, S.A. | Method and apparatus for high speed chip-to-chip communications |
EP3700154A1 (en) | 2015-06-26 | 2020-08-26 | Kandou Labs, S.A. | High speed communications system |
US10055372B2 (en) | 2015-11-25 | 2018-08-21 | Kandou Labs, S.A. | Orthogonal differential vector signaling codes with embedded clock |
US10116468B1 (en) | 2017-06-28 | 2018-10-30 | Kandou Labs, S.A. | Low power chip-to-chip bidirectional communications |
US10347283B2 (en) * | 2017-11-02 | 2019-07-09 | Kandou Labs, S.A. | Clock data recovery in multilane data receiver |
WO2019133897A1 (en) | 2017-12-28 | 2019-07-04 | Kandou Labs, S.A. | Synchronously-switched multi-input demodulating comparator |
Citations (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4336616A (en) * | 1978-12-15 | 1982-06-22 | Nasa | Discriminator aided phase lock acquisition for suppressed carrier signals |
US4466108A (en) * | 1981-10-06 | 1984-08-14 | Communications Satellite Corporation | TDMA/PSK Carrier synchronization without preamble |
US4473801A (en) * | 1979-12-17 | 1984-09-25 | Robert Maurer | Demodulator circuit with phase control loop |
US4485487A (en) * | 1981-05-26 | 1984-11-27 | U.S. Philips Corporation | Method of, and a receiver for, demodulating a double sideband amplitude modulated signal in a quasi-synchronous area coverage scheme utilizing sideband diversity |
Family Cites Families (8)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3447084A (en) * | 1966-01-03 | 1969-05-27 | Bell Telephone Labor Inc | Correction of frequency shift in carrier systems |
US3768030A (en) * | 1972-05-08 | 1973-10-23 | Motorola Inc | Automatic signal acquisition means for phase-lock loop with anti- sideband lock protection |
US4091410A (en) * | 1976-11-08 | 1978-05-23 | Zenith Radio Corporation | Frequency and phase lock loop synchronous detecting system having a pair of phase lock conditions |
US4419759A (en) * | 1980-08-05 | 1983-12-06 | Communications Satellite Corporation | Concurrent carrier and clock synchronization for data transmission system |
DE3114063A1 (de) * | 1981-04-07 | 1982-10-21 | Licentia Patent-Verwaltungs-Gmbh, 6000 Frankfurt | Empfangssystem |
US4595927A (en) * | 1984-07-05 | 1986-06-17 | Motorola, Inc. | Loran C cycle slip reduction technique |
US4847869A (en) * | 1987-12-04 | 1989-07-11 | Motorla, Inc. | Rapid reference acquisition and phase error compensation for radio transmission of data |
US4829543A (en) * | 1987-12-04 | 1989-05-09 | Motorola, Inc. | Phase-coherent TDMA quadrature receiver for multipath fading channels |
-
1990
- 1990-09-28 US US07/590,401 patent/US5150384A/en not_active Expired - Lifetime
-
1991
- 1991-09-24 BR BR919105906A patent/BR9105906A/pt not_active IP Right Cessation
- 1991-09-24 CA CA002071869A patent/CA2071869C/en not_active Expired - Fee Related
- 1991-09-24 WO PCT/US1991/006924 patent/WO1992006542A1/en active Application Filing
- 1991-09-24 DE DE4192400A patent/DE4192400C2/de not_active Expired - Fee Related
- 1991-09-24 DE DE19914192400 patent/DE4192400T/de active Pending
- 1991-09-27 MX MX9101319A patent/MX9101319A/es not_active IP Right Cessation
-
1992
- 1992-05-28 GB GB9211281A patent/GB2255867B/en not_active Expired - Fee Related
Patent Citations (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4336616A (en) * | 1978-12-15 | 1982-06-22 | Nasa | Discriminator aided phase lock acquisition for suppressed carrier signals |
US4473801A (en) * | 1979-12-17 | 1984-09-25 | Robert Maurer | Demodulator circuit with phase control loop |
US4485487A (en) * | 1981-05-26 | 1984-11-27 | U.S. Philips Corporation | Method of, and a receiver for, demodulating a double sideband amplitude modulated signal in a quasi-synchronous area coverage scheme utilizing sideband diversity |
US4466108A (en) * | 1981-10-06 | 1984-08-14 | Communications Satellite Corporation | TDMA/PSK Carrier synchronization without preamble |
Cited By (1)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5828710A (en) * | 1995-12-11 | 1998-10-27 | Delco Electronics Corporation | AFC frequency synchronization network |
Also Published As
Publication number | Publication date |
---|---|
DE4192400T (US07413550-20080819-C00001.png) | 1992-10-08 |
US5150384A (en) | 1992-09-22 |
CA2071869C (en) | 1998-09-15 |
CA2071869A1 (en) | 1992-03-29 |
MX9101319A (es) | 1992-05-04 |
GB9211281D0 (en) | 1992-07-22 |
BR9105906A (pt) | 1992-11-03 |
GB2255867B (en) | 1995-03-08 |
DE4192400C2 (de) | 1995-06-14 |
GB2255867A (en) | 1992-11-18 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US5150384A (en) | Carrier recovery method and apparatus having an adjustable response time determined by carrier signal parameters | |
KR960012425B1 (ko) | 통신채널의 채널 이득 및 노이즈 분산을 계산하기 위한 시스템 및 방법 | |
EP0601855B1 (en) | Adaptive equalizer capable of compensating for carrier frequency offset | |
CA1184249A (en) | High capacity digital mobile radio system | |
EP0715440B1 (en) | Synchronous detector and synchronizing method for digital communication receiver | |
US5187809A (en) | Dual mode automatic gain control | |
KR960000607B1 (ko) | 코우히어런트 무선수신기의 주파수를 급속하게 제어하는 방법 및 그 방법을 실시하기 위한 장치 | |
US4899367A (en) | Multi-level quadrature amplitude modulator system with fading compensation means | |
JP3910443B2 (ja) | 自動周波数制御装置 | |
CA2211291C (en) | Clock timing recovery methods and circuits | |
US6590945B1 (en) | Method and apparatus for frequency offset compensation | |
CA2065647A1 (en) | Tangental type differential detector for pulse shaped pi/4 shifted differentially encoded quadrature phase shift keying | |
US6047023A (en) | Swept frequency modulation and demodulation technique | |
US5259005A (en) | Apparatus for and method of synchronizing a clock signal | |
US5200977A (en) | Terminal unit apparatus for time division multiplexing access communications system | |
JP4338310B2 (ja) | 通信システムのための高速同期 | |
US5970102A (en) | Circuit and method for detecting frequency correction burst in TDMA digital mobile communication system | |
WO1988005981A1 (en) | Tdma communications system with adaptive equalization | |
US6861900B2 (en) | Fast timing acquisition for multiple radio terminals | |
US7474718B2 (en) | Frequency control for a mobile communications device | |
CN101010871B (zh) | 用于无线通信终端的接收机和方法 | |
US5261120A (en) | Method and apparatus for transmitting a signal with an offset which follows a received signal | |
JPH09214406A (ja) | 非線形信号相関器およびその方法 | |
JP3178138B2 (ja) | フレーム同期回路及びフレーム同期方法 | |
AU8307691A (en) | Dual mode automatic gain control |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
AK | Designated states |
Kind code of ref document: A1 Designated state(s): BR CA DE GB |
|
WWE | Wipo information: entry into national phase |
Ref document number: 2071869 Country of ref document: CA |
|
RET | De translation (de og part 6b) |
Ref document number: 4192400 Country of ref document: DE Date of ref document: 19921008 |
|
WWE | Wipo information: entry into national phase |
Ref document number: 4192400 Country of ref document: DE |