USRE40255E1 - Communication system - Google Patents

Communication system Download PDF

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USRE40255E1
USRE40255E1 US09/686,467 US68646700A USRE40255E US RE40255 E1 USRE40255 E1 US RE40255E1 US 68646700 A US68646700 A US 68646700A US RE40255 E USRE40255 E US RE40255E
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signal
data
level
transmission
points
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US09/686,467
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Mitsuaki Oshima
Seiji Sakashita
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Panasonic Corp
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Panasonic Corp
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Priority to JP9581391 priority
Priority to JP15565091 priority
Priority to JP18223691 priority
Priority to JP6073992 priority
Priority to US85762792A priority
Priority to JP13298493 priority
Priority to JP26161293 priority
Priority to JP34997293 priority
Priority to JP7966894A priority patent/JPH07264148A/en
Priority to US08/240,521 priority patent/US5600672A/en
Priority to US09/244,037 priority patent/USRE40241E1/en
Application filed by Panasonic Corp filed Critical Panasonic Corp
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Abstract

At the transmitter side, carrier waves are modulated according to an input signal for producing relevant signal points in a signal space diagram. The input signal is divided into, two, first and second, data streams. The signal points are divided into signal point groups to which data of the first data stream are assigned. Also, data of the second data stream are assigned to the signal points of each signal point group. A difference in the transmission error rate between first and second data streams is developed by shifting the signal points to other positions in the space diagram expressed at least in the polar coordinate system. At the receiver side, the first and/or second data streams can be reconstructed from a received signal. In TV broadcast service, a TV signal is divided by a transmitter into low and high frequency band components which are designated as first and second data streams respectively. Upon receiving the TV signal, a receiver can reproduce only the low frequency band component or both the low and high frequency band components, depending on its capability. Furthermore, a communication system based on an OFDM system is utilized for data transmission of a plurality of subchannels, wherein the subchannels are differentiated by changing the length of a guard time slot or a carrier wave interval of a symbol transmission time slot, or changing the transmission electric power of the carrier.

Description

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part of application Ser. No. 07/857,627, filed Mar. 25, 1992, pending.

This is a reissue application of U.S. Pat. No. 5,600,672, issued Feb. 4, 1997, and a divisional application of reissue application Ser. No. 09/244,037, filed Feb. 4, 1999, which is also a reissue application of U.S. Pat. No. 5,600,672, issued Feb. 4, 1997 which is a Continuation-In-Part of application Ser. No. 07/857,627, filed Mar. 25, 1992 now abandoned. Further reissue divisional applications have been filed, all of which are reissues of U.S. Pat. No. 5,600,672. These further applications Ser. Nos. are: 09/677,421, filed Oct. 5, 2000; 09/678,014, filed Oct. 5, 2000; 09/677,420, filed Oct. 5, 2000; 09/680,177, filed Oct. 5, 2000; 09/680,176, filed Oct. 5, 2000; 09/662,695, filed Sep. 15, 2000; 09/686,463, filed Oct. 12, 2000; 09/686,466, filed Oct. 12, 2000; 09/688,028, filed Oct. 12, 2000; 09/686,464, filed Oct. 12, 2000; 09/686,465, filed Oct. 12, 2000; 09/666,012, filed Sep. 19, 2000; 09/667,525, filed Sep. 21, 2000; 09/667,438, filed Sep. 21, 2000; 09/668,068, filed Sep. 25, 2000; 09/669,916, filed Sep 25, 2000; 09/672,948, filed Sep. 29, 2000; 09/672,946, filed Sep. 29, 2000; 09/672,947, filed Sep. 29, 2000; 10/133,347, filed Apr. 29, 2002; 10/133,364, filed Apr. 29, 2002; 10/692,469, filed Oct. 24, 2003; 10/693,526, filed Oct. 27, 2003; 10/635,468, filed Aug. 7, 2003; 10/690,297, filed Oct. 27, 2003; 10/860,666, filed Jun. 4, 2004; 10/782,411, filed Feb. 20, 2004; 10/783,588, filed Feb. 23, 2004; 10/773,811, filed Feb. 9, 2004; 10/882,126, filed Jun. 30, 2004; 10/885,572, filed Jul. 7, 2004; 10/911,680, filed Aug. 5, 2004; and 11/038,006, filed Jan. 19, 2006.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a communication system for transmission and reception of a digital signal through modulation of its carrier wave and demodulation of the modulated signal.

2. Description of the Prior Art

Digital signal communication systems have been used in various fields. Particularly, digital video signal transmission techniques have been improved remarkably.

Among them is a digital TV signal transmission method. So far, such digital TV signal transmission systems are in particular use for transmission between TV stations. They will soon be utilized for terrestrial and/or satellite broadcast service in every country of the world.

The TV broadcast systems including HDTV, PCM music, FAX, and other information services are now demanded to increase desired data in quantity and quality for satisfying millions of sophisticated viewers. In particular, the data has to be increased in a given bandwidth of frequency allocated for TV broadcast service. The data to be transmitted is always abundant and provided as much as handled with up-to-date techniques of the time. It is ideal to modify or change the existing signal transmission system corresponding to an increase in the data amount with time.

However, the TV broadcast service is a public business and cannot go further without considering the interests and benefits of viewers. It is essential to have any new service compatible with existing TV receivers and displays. More particularly, the compatibility of a system is much desired for providing both old and new services simultaneously or one new service which can be intercepted by both the existing and advanced receivers.

It is understood that any new digital TV broadcast system to be introduced has to be arranged for data extension in order to respond to future demands and technological advantages and also, for compatibility to allow the existing receivers to receive transmissions.

The expansion capability and compatible performance of the prior art digital TV system will be explained.

A digital satellite TV system is known in which NTSC TV signals compressed to an about 6 Mbps are muitiplexed multiplexed by time division modulation of 4 PSK and transmitted on 4 to 20 channels while HDTV signals are carried on a signal channel. Another digital HDTV system is provided in which HDTV video data compressed to as small as 15 Mbps are transmitted on a 16 or 32 QAM signal through ground stations.

Such a known satellite system permits HDTV signals to be carried on the channel by a conventional manner, thus occupying a band of frequencies equivalent to the same channels of NTSC signals. This causes the corresponding NTSC channels to be unavailable during the transmission of the HDTV signal. Also, the compatibility between NTSC and HDTV receivers or displays is hardly concerned and data expansion capability needed for matching a future advanced mode is utterly disregarded.

Such a common terrestrial HDTV system offers an HDTV service on conventional 16 or 32 QAM signals without any modification. In any analogue TV broadcast service, there are developed a lot of signal attenuating or shadow regions within its service area due to structural obstacles, geographical inconveniences, or signal interference from a neighbor station. When the TV signal is an analogue from form, it can be intercepted more or less at such signal attenuating regions although its reproduced picture is low in quality. If the TV signal is a digital form, it can rarely be reproduced at an acceptable level within the regions. This disadvantage is critically hostile to the development of any digital TV system.

SUMMARY OF THE INVENTION

It is an object of the present invention, for solving the foregoing disadvantages, to provide a communication system arranged for compatible use for both the existing NTSC and newly introduced HDTV broadcast services, particularly via satellite and also, for minimizing signal attenuating or shadow region of its service area on the grounds ground.

A communication system according to the present invention intentionally varies signal points, which used to be disposed at uniform intervals, to perform the signal transmission and reception. For example, if applied to a QAM signal, the communication system comprises two major sections: a transmitter having a signal input circuit, a modulator circuit for producing m numbers of signal points, in a signal vector field through modulation of a plurality of out-of-phase carrier waves using an input signal supplied from the input circuit, and a transmitter circuit for transmitting a resultant modulated signal; and a receiver having an input circuit for receiving the modulated signal, a demodulator circuit for demodulating one-bit signal points of a QAM carrier wave, and an output circuit.

In operation, the input signal containing a first data stream of n values and a second data stream is fed to the modulator circuit of the transmitter where a modified m-bit QAM carrier wave is produced representing m signal points in a vector field. The m signal points are divided into n signal point groups to which the n values of the first data stream are assigned respectively. Also, data of the second data stream are assigned to m/n signal points or sub groups of each signal point group. Then, a resultant transmission signal is transmitted from the transmitter circuit. Similarly, a third data stream can be propagated.

At the p-bit demodulator circuit, p>m, of the receiver, the first data stream of the transmission signal if is first demodulated through dividing p signal points in a signal space diagram into n signal point groups. Then, the second data stream is demodulated through assigning p/n values to p/n signal points of each corresponding signal point group for reconstruction of both the first and second data streams. If the receiver is at P=n, the n signal point groups are reclaimed and assigned the n values for demodulation and reconstruction of the first data stream.

Upon receiving the same transmission signal from the transmitter, a receiver equipped with a large sized antenna and capable of large-data modulation can reproduce both the first and second data streams. A receiver equipped with a small sized antenna and capable of small-data modulation can reproduce the first data stream only. Accordingly, the compatibility of the signal transmission system will be ensured. When the first data stream is an NTSC TV signal or low frequency band component of an HDTV signal and the second data stream is a high frequency band component of the HDTV signal, the small-data modulation receiver can reconstruct the NTSC TV signal and the large-data modulation receiver can reconstruct the HDTV signal. As understood, a digital NTSC/HDTV simultaneous broadcast service will be feasible using the compatibility of the signal transmission system of the present invention.

More specifically, the communication system of the present invention comprises: a transmitter having a signal input circuit, a modulator circuit for producing m signal point, points in a signal vector field through modulation of a plurality of out-of-phase carrier waves using an input signal supplied from the input, and a transmitter circuit for transmitting a resultant modulated signal, in which the main procedure includes receiving an input signal containing a first data stream of n values and a second data stream, dividing the m signal points of the signal into n signal point groups, assigning the n values of the first data stream to the n signal point groups respectively, assigning data of the second data stream to signal points of each signal point group respectively, and transmitting the resultant modulated signal; and a receiver having an input circuit for receiving the modulated signal, a demodulator circuit for demodulating p signal points of a QAM carrier wave, and an output circuit, in which the main procedure includes dividing the p signal points into n signal point groups, demodulating the first data stream of which n values are assigned to the n signal point groups respectively, and demodulating the second data stream of which p/n values are assigned to p/n signal points of each signal point group respectively. For example, a transmitter produces a modified m-bit QAM signal of which first, second, and third data streams, each carrying n values, are assigned to relevant signal point groups with a modulator. The signal can be intercepted and the first data stream only reproduced by a first receiver, both the first and second data streams can be reproduced by a second receiver, and all the first, second, and third streams can be reproduced by a third receiver.

More particularly, a receiver capable of demodulation of n-bit data can reproduce n bits from a multiple-bit modulated carrier wave carrying m-bit data where m>n, thus allowing the communication system to have compatibility and capability of future extension. Also, a multi-level signal transmission will be possible by shifting the signal points of QAM so that a nearest signal point to the origin point of I-axis and Q-axis coordinates is spaced nf from the origin where f is the distance of the nearest point from each axis and n is more than 1.

Accordingly, a compatible digital satellite broadcast service for both the NTSC and HDTV systems will be feasible when the first data stream carries an NTSC signal and the second data stream carries a difference signal between NTSC and HDTV. Hence, the capability of corresponding to an increase in the data amount to be transmitted will be ensured. Also, on the ground, the service area will be increased while signal attenuating areas are decreased.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic view of the entire arrangement of a signal transmission system showing a first embodiment of the present invention;

FIG. 2 is a block diagram of a transmitter of the first embodiment;

FIG. 3 is a vector diagram showing a transmission signal of the first embodiment;

FIG. 4 is a vector diagram showing a transmission signal of the first embodiment;

FIG. 5 is a view showing an assignment of binary codes to signal points according to the first embodiment;

FIG. 6 is a view showing an assignment of binary codes to signal point groups according to the first embodiment;

FIG. 7 is a view showing an assignment of binary codes to signal points in each signal point group according to the first embodiment;

FIG. 8 is a view showing another assignment of binary codes to signal point groups and their signal points according to the first embodiment;

FIG. 9 is a view showing threshold values of the signal point groups according to the first embodiment;

FIG. 10 is a vector diagram of a modified 16 QAM signal of the first embodiment;

FIG. 11 is a graphic diagram showing the relationship between antenna radius r2 and transmission energy ratio n according to the first embodiment;

FIG. 12 is view showing the signal points of a modified 64 QAM signal of the first embodiment;

FIG. 13 is a graphic diagram showing the relationship between antenna radius r3 and transmission energy ratio n according to the first embodiment;

FIG. 14 is a vector diagram showing signal point groups and their signal points of the modified 64 QAM signal of the first embodiment;

FIG. 15 is an explanatory view showing the relationship between A1 and A2 of the modified 64 QAM signal of the first embodiment;

FIG. 16 is a graph diagram showing the relationship between antenna radius r2 and r3 and transmission energy ratio n16 and n64 respectively according to the first embodiment;

FIG. 17 is a block diagram of a digital transmitter of the first embodiment;

FIG. 18 is a signal space diagram of a 4 PSK modulated signal of the first embodiment;

FIG. 19 is a block diagram of a first receiver of the first embodiment;

FIG. 20 is a signal space diagram of a 4 PSK modulated signal of the first embodiment;

FIG. 21 is a block diagram of a second receiver of the first embodiment;

FIG. 22 is a vector diagram of a modified 16 QAM signal of the first embodiment;

FIG. 23 is a vector diagram of a modified 64 QAM signal of the first embodiment;

FIG. 24 is a flowchart showing the operation of the first embodiment;

FIG. 25(a) and 25(b) are vector diagrams respectively showing an 8 and a 16 QAM signal of the first embodiment;

FIG. 26 is a block diagram of a third receiver of the first embodiment;

FIG. 27 is a view showing signal points of the modified 64 QAM signal of the first embodiment;

FIG. 28 is a flowchart showing another the operation of the first embodiment;

FIG. 29 is a schematic view of the entire arrangement of a signal transmission system showing a third embodiment of the present invention;

FIG. 30 is a block diagram of a first video encoder of the third embodiment;

FIG. 31 is a block diagram of a first video decoder of the third embodiment;

FIG. 32 is a block diagram of a second video decoder of the third embodiment;

FIG. 33 is a block diagram of a third video decoder of the third embodiment;

FIG. 34 is an explanatory view showing a time multiplexing of D1, D2, and D3 signals according to the third embodiment;

FIG. 35 is an explanatory view showing another time multiplexing of D1, D2, and D3 signals according to the third embodiment;

FIG. 36 is an explanatory view showing a further time multiplexing of D1, D2, and D3 signals according to the third embodiment;

FIG. 37 is a schematic view of the entire arrangement of a signal transmission system showing a fourth embodiment of the present invention;

FIG. 38 is a vector diagram of a modified 16 QAM signal of the third embodiment;

FIG. 39 is a vector diagram of the modified 16 QAM signal of the third embodiment;

FIG. 40 is a vector diagram of a modified 64 QAM signal of the third embodiment;

FIG. 41 is a diagram of assignment of data components on a time base according to the third embodiment;

FIG. 42 is a diagram of assignment of data components on a time base in TDMA action according to the third embodiment;

FIG. 43 is a block diagram of a carrier reproducing circuit of the third embodiment;

FIG. 44 is a diagram showing the principle of carrier wave reproduction according to the third embodiment;

FIG. 45 is a block diagram of a carrier reproducing circuit for reverse modulation of the third embodiment;

FIG. 46 is a diagram showing an assignment of signal points of the 16 QAM signal of the third embodiment;

FIG. 47 is a diagram showing an assignment of signal points of the 64 QAM signal of the third embodiment;

FIG. 48 is a block diagram of a carrier reproducing circuit for 16× multiplication of the third embodiment;

FIG. 49 is an explanatory view showing a time multiplexing of DV1, DH1, DV2, DH2, DV3, and DH3 signals according to the third embodiment;

FIG. 50 is an explanatory view showing a TDMA time multiplexing of DV1, DH1, DV2, DH2, DV3, and DH3 signals according to the third embodiment;

FIG. 51 is an explanatory view showing another TDMA time multiplexing of the DV1, DH1, DV2, DH2, DV3, and DH3 signals according to the third embodiment;

FIG. 52 is a diagram showing a signal interference region in a known transmission method according to the fourth embodiment;

FIG. 53 is a diagram showing signal interference regions in a multi-level signal transmission method according to the fourth embodiment;

FIG. 54 is a diagram showing signal attenuating regions in the known transmission method according to the fourth embodiment;

FIG. 55 is a diagram showing signal attenuating regions in the multi-level signal transmission method according to the fourth embodiment;

FIG. 56 is a diagram showing a signal interference region between two digital TV stations according to the fourth embodiment;

FIG. 57 is a diagram showing an assignment of signal points of modified 4 ASK signal of the fifth embodiment;

FIG. 58 is a diagram showing another assignment of signal points of the modified 4 ASK signal of the fifth embodiment;

FIGS. 59(a) and 59(b) are diagrams showing assignment of signal points of the modified 4 ASK signal of the fifth embodiment of FIGS. 59(c) and 59(d) are diagrams respectively showing the slice levels of the modulated 4 ASK signal in subchannels 1 and 2;

FIG. 60 is a diagram showing another assignment of signal points of the modified 4 ASK signal of the fifth embodiment when the C/N rate is low;

FIG. 61 shows a 4- and 8-level VSB transmitter according to the fifth embodiment of the invention;

FIG. 62(a) is a wave spectrum diagram of the ASK signal, i.e., a multi-value VSB signal before filtering, in the fifth embodiment of the invention and FIG. 62(b) is a wave spectrum diagram showing the characteristics of the filtered VSB signal;

FIG. 63 is a block diagram of a 4-, 8-, and 16-level VSB receiver in the fifth embodiment of the invention;

FIG. 64 is a block diagram of a video signal transmitter of the fifth embodiment;

FIG. 65 is a block diagram of a TV receiver of the fifth embodiment;

FIG. 66 is a block diagram of another TV receiver of the fifth embodiment;

FIG. 67 is a block diagram of a satellite-to-ground TV receiver of the fifth embodiment;

FIG. 68(a) is an 8-level VSB constellation map in the fifth and sixth embodiments of the invention;

FIG. 68(b) is an 8-level VSB constellation map in the fifth and sixth embodiments of the invention;

FIG. 68(c) is an 8-level VSB signal-time waveform diagram in the fifth and sixth embodiments of the invention;

FIG. 69 is a block diagram of a video encoder of the fifth embodiment;

FIG. 70 is a block diagram of a video encoder of the fifth embodiment containing one divider circuit;

FIG. 71 is a block diagram of a video decoder of the fifth embodiment;

FIG. 72 is a block diagram of a video decoder of the fifth embodiment containing one mixer circuit;

FIG. 73 is a diagram showing a time assignment of data components of a transmission signal according to the fifth embodiment;

FIG. 74(a) is a block diagram of a video decoder of the fifth embodiment;

FIG. 74(b) is a diagram showing another time assignment of data components of the transmission signal according to the fifth embodiment;

FIG. 75 is a diagram showing a time assignment of data components of a transmission signal according to the fifth embodiment;

FIG. 76 is a diagram showing a time assignment of data components of a transmission signal according to the fifth embodiment;

FIG. 77 is a diagram showing a time assignment of data components of a transmission signal according to the fifth embodiment;

FIG. 78 is a block diagram of a video decoder of the fifth embodiment;

FIG. 79 is a diagram showing a time assignment of data components of a three-level transmission signal according to the fifth embodiment;

FIG. 80 is a block diagram of another video decoder of the fifth embodiment;

FIG. 81 is a diagram showing a time assignment of data components of a transmission signal according to the fifth embodiment;

FIG. 82 is a block diagram of a video decoder for D1 signal of the fifth embodiment;

FIG. 83 is a graphic diagram showing the relationship between frequency and time of a frequency modulated signal according to the fifth embodiment;

FIG. 84 is a block diagram of a magnetic record/playback apparatus of the fifth embodiment;

FIG. 85 is a graphic diagram showing the relationship between C/N and level according to the second embodiment;

FIG. 86 is a graphic diagram showing the relationship between C/N and transmission distance according to the second embodiment;

FIG. 87 is a block diagram of a transmissiontransmitter of the second embodiment;

FIG. 88 is a block diagram of a receiver of the second embodiment;

FIG. 89 is a graphic diagram showing the relationship between C/N and error rate according to the second embodiment;

FIG. 90 is a diagram showing signal attenuating regions in the three-level transmission of the fifth embodiment;

FIG. 91 is a diagram showing signal attenuating regions in the four-level transmission of athe sixth embodiment;

FIG. 92 is a diagram showing the four-level transmission of the sixth embodiment;

FIG. 93 is a block diagram of a divider of the sixth embodiment;

FIG. 94 is a block diagram of a mixer of the sixth embodiment;

FIG. 95 is a diagram showing another four-level transmission of the sixth embodiment;

FIG. 96 is a view of signal propagation of a known digital TV broadcast system;

FIG. 97 is a view of signal propagation of a digital TV broadcast system according to the sixth embodiment;

FIG. 98 is a diagram showing a four-level transmission of the sixth embodiment;

FIG. 99 is a vector diagram of a 16 SRQAM signal of the third embodiment;

FIG. 100 is a vector diagram of a 32 SRQAM signal of the third embodiment;

FIG. 101 is a graphic diagram showing the relationship between C/N and error rakerate according to the third embodiment;

FIG. 102 is a graphic diagram showing the relationship between C/N and error rate according to the third embodiment;

FIG. 103 is a graphic diagram showing the relationship between shift distance n and C/N needed for transmission according to the third embodiment;

FIG. 104 is a graphic diagram showing the relationship between shift distance n and C/N needed for transmission according to the third embodiment;

FIG. 105 is a graphic diagram showing the relationship between signal level and distance from a transmitter antenna in terrestrial broadcast service according to the third embodiment;

FIG. 106 is a diagram showing a service area of the 32 SRQAM signal of the third embodiment;

FIG. 107 is a diagram showing a service area of the 32 SRQAM signal of the third embodiment;

FIG. 108(a) is a diagram showing a frequency distribution profile of a conventional TV signal;

FIG. 108(b) is a diagram showing a frequency distribution profile of a conventional two-layer TV signal;

FIG. 108(c) is a diagram showing threshold values of the third embodiment;

FIG. 108(d) is a diagram showing a frequency distribution profile of two-layer OFDM carriers of the ninth embodiment, and FIG. 108(e) is a diagram showing threshold values for three-layer OFDM of the ninth embodiment;

FIG. 109 is a diagram showing a time assignment of the TV signal of the third embodiment;

FIG. 110 is a diagram showing a principle of C-CDM of the third embodiment;

FIG. 111 is a view showing an assignment of codes according to the third embodiment;

FIG. 112 is a view showing an assignment of an extended 36 QAM according to the third embodiment;

FIG. 113 is a view showing a frequency assignment of a modulation signal according to the fifth embodiment;

FIG. 114 is a block diagram showing a magnetic recording/playback apparatus according to the fifth embodiment;

FIG. 115 is a block diagram showing a transmitter/receiver of a portable telephone according to the eighth embodiment;

FIG. 116 is a block diagram showing base stations according to the eighth embodiment;

FIG. 117 is a view illustrating communication capacities and traffic distribution of a conventional system;

FIG. 118 is a view illustrating communication capacities and traffic distribution according to the eighth embodiment;

FIG. 119(a) is a diagram showing a time slot assignment of a conventional system;

FIG. 119(b) is a diagram showing a time slot assignment according to the eighth embodiment;

FIG. 120(a) is a diagram showing a time slot assignment of a conventional TDMA system;

FIG. 120(b) is a diagram showing a time slot assignment according to a TDMA system of the eighth embodiment;

FIG. 121 is a block diagram showing a one-level transmitter/receiver according to the eighth embodiment;

FIG. 122 is a block diagram showing a two-level transmitter/receiver according to the eighth embodiment;

FIG. 123 is a block diagram showing an OFDM type transmitter/receiver according to the ninth embodiment;

FIG. 124 is a view illustrating a principle of the OFDM system according to the ninth embodiment;

FIG. 125(a) is a view showing a frequency assignment of a modulation signal of a conventional system;

FIG. 125(b) is a view showing a frequency assignment of a modulation signal according to the ninth embodiment;

FIG. 126(a) is a view showing a frequency assignment of an OFDM signal of the ninth embodiment, wherein no weighting is applied;

FIG. 126(b) is a view showing a frequency assignment of an OFDM signal of the ninth embodiment, wherein two channels of two-layer OFDM are weighted by transmission electric power;

FIG. 126(c) is a view showing a frequency assignment of an OFDM signal of the ninth embodiment, wherein carrier intervals are doubled by weighting;

FIG. 126(d) is a view showing a frequency assignment of an OFDM signal of the ninth embodiment, wherein carrier intervals are not weighted;

FIG. 127 is a block diagram showing a transmitter/receiver according to the ninth embodiment;

FIG. 128(a) is a block diagram of a trellis encoder (ratio ½) in embodiments 2, 4, and 5,

FIG. 128(b) is a block diagram of a trellis encoder (ratio ⅔) in embodiments 2, 4, and 5,

FIG. 128(c) is a block diagram of a trellis encoder (ratio ¾) in embodiments 2, 4, and 5,

FIG. 128(d) is a block diagram of a trellis decoder (ratio ½) in embodiments 2, 4, and 5,

FIG. 128(e) is a block diagram of a trellis decoder (ratio ⅔) in embodiments 2, 4, and 5,

FIG. 128(f) is a block diagram of a trellis decoder (ratio ¾) in embodiments 2, 4, and 5;

FIG. 129 is a view showing a time assignment of effective symbol periods and guard intervals according to the ninth embodiment;

FIG. 130 is a graphic diagram showing a relationship between C/N rate and error rate according to the ninth embodiment;

FIG. 131 is a block diagram showing a magnetic recording/playback apparatus according to the fifth embodiment;

FIG. 132 is a view showing a recording format of track on the magnetic tape and a travellingtraveling of a head;

FIG. 133 is a block diagram showing a transmitter/receiver according to the third embodiment;

FIG. 134 is a diagram showing a frequency assignment of a conventional broadcasting;

FIG. 135 is a diagram showing a relationship between service area and picture quality in a three-level signal transmission system according to the third embodiment;

FIG. 136 is a diagram showing a frequency assignment in case the multi-level signal transmission system according to the third embodiment is combined with FDM;

FIG. 137 is a block diagram showing a transmitter/receiver according to the third embodiment, in which Trellis encoding is adopted;

FIG. 138 is a block diagram showing a transmitter/receiver according to the ninth embodiment, in which a part of low frequency band signal is transmitted by OFDM;

FIG. 139 is a diagram showing an assignment of signal points of the 8-PS-APSK signal of the first embodiment;

FIG. 140 is a diagram showing an assignment of signal points of the 16-PS-APSK signal of the first embodiment;

FIG. 141 is a diagram showing an assignment of signal points of the 8-PS-PSK signal of the first embodiment;

FIG. 142 is a diagram showing an assignment of signal points of the 16-PS-PSK (PS type) signal of the first embodiment;

FIG. 143 is a graphic diagram showing the relationship between antenna radius of satellite and transmission capacity according to the first embodiment;

FIG. 144 is a block diagram showing a weighted OFDM transmitter/receiver according to the ninth embodiment;

FIG. 145(a) is a diagram showing the waveform of the guard time and the symbol time in the multi-level OFDM according to the ninth embodiment, wherein multipath is short;

FIG. 145(b) is a diagram showing the waveform of the guard time and the symbol time in the multi-level OFDM according to the ninth embodiment, wherein multipath is long;

FIG. 146 is a diagram showing a principle of the multi-level OFDM according to the ninth embodiment;

FIG. 147 is a diagram showing subchannel assignment of a two-layer signal transmission system, weighted electric power according to the ninth embodiment;

FIG. 148 is a diagram showing relationship among the D/V ratio, the multipath delay time, and the guard time according to the ninth embodiment;

FIG. 149(a) is a diagram showing time slots of respective layers according to the ninth embodiment;

FIG. 149(b) is a diagram showing time distribution of guard times of respective layers according to the ninth embodiment;

FIG. 149(c) is a diagram showing time distribution of guard times of respective layers according to the ninth embodiment;

FIG. 150 is a diagram showing the relationship between multipath delay time and transfer rate according to the ninth embodiment, wherein a three-layer signal transmission effective to multipath is realized; and

FIG. 151 is a diagram showing the relationship between multipath delay time and C/N ratio according to the ninth embodiment, wherein two-dimensional, matrix type, multi-layer broadcast service can be realized by combining the GTW-OFDM and the C-CDM (or the CSW-OFDM).

FIG. 152 is a timing chart of a 3-level hierarchical television signal at each time slot when GTW-OFDM of the ninth embodiment is combined with C-CDM (or CSW-OFDM);

FIG. 153 shows the relationship between the multipath signal delay time, C/N ratio, and transmission rate when GTW-OFDM of the ninth embodiment is combined with C-CDM (or CSW-OFDM), and is used to describe the hierarchical broadcasting method using three-dimensional matrix structure;

FIGS. 154A-C together form a frequency distribution graph of power weight OFDM in the ninth embodiment;

FIG. 155 shows the position on the time axis of a 3-level hierarchical television signal at each time slot when guard time-OFDM of the ninth embodiment is combined with C-CDM;

FIG. 156 is a block diagram of the transmitter and the receiver in the fourth and fifth embodiments of the invention;

FIG. 157 is a block diagram of the transmitter and the receiver in the fourth and fifth embodiments of the invention;

FIG. 158 is a block diagram of the transmitter and the receiver in the fourth and fifth embodiments of the invention;

FIG. 159(a) is a signal point positioning diagram in 16-level VSB in the fifth embodiment of the invention;

FIG. 159(b) is a signal point positioning (8-level VSB) diagram in 16-level VSB in the fifth embodiment of the invention;

FIG. 159(c) is a signal point positioning (4-level VSB) diagram in 16-level VSB in the fifth embodiment of the invention;

FIG. 159(d) is a signal point positioning (16-level VSB) diagram in 16-level VSB in the fifth embodiment of the invention;

FIG. 160(a) is a block diagram of an ECC encoder in the fifth and sixth embodiments of the invention;

FIG. 160(b) is a block diagram of an ECC decoder in the fifth and sixth embodiments of the invention;

FIG. 161 is an overall block diagram of a VSB receiver in the fifth embodiment of the invention;

FIG. 162 is a block diagram of a the receiver in the fifth embodiment of the invention;

FIG. 163 is a graph of the error rate and C/N ratio curve in 4-level VSB and TC-8-level VSB in the fourth embodiment of the invention;

FIG. 164 is an error rate curve of subchannel 1 and subchannel 2 in 4-level VSB and TC-8-level VSB in the fourth embodiment of the invention;

FIG. 165(a) is a block diagram of the Reed-Solomon encoder in the second, fourth, and fifth embodiments of the invention;

FIG. 165(b) is a block diagram of the Reed-Solomon decoder in the second, fourth, and fifth embodiments of the invention;

FIG. 166 is a flowchart of Reed-Solomon error correction and operation in the second, fourth and fifth embodiments of the invention;

FIG. 167 is a block diagram of the deinterleaver in the second, third, fourth, fifth and sixth embodiments of the invention;

FIG. 168(a) is an interleave/deinterleave table for the second, third, fourth, and fifth embodiments of the invention;

FIG. 168(b) shows the interleave distance in the second, third, fourth, and fifth embodiments of the invention;

FIG. 169 is a comparison of redundancy in 4-level VSB, 8-level VSB, and 16-level VSB in the fifth embodiment of the invention;

FIG. 170 is a block diagram of a television receiver for receiving the high priority signal of the second, third, fourth, and fifth embodiments of the invention;

FIG. 171 is a block diagram of the receiver and transmitter in the second, third, fourth, and fifth embodiments of the invention;

FIG. 172 is a block diagram of the receiver and transmitter in the second, third, fourth, and fifth embodiments of the invention; and

FIG. 173 is a block diagram of an ASK magnetic recording and reproducing apparatus according to the sixth embodiment of the invention;

FIG. 174 is a block diagram showing a circuitry arrangement of QAM/VSB compatible modulator for multi-level transmission according to Embodiment 5.

FIG. 175 is a block diagram showing another circuitry arrangement of the QAM/VSB modulator for multi-level transmission according to Embodiment 5.

FIG. 176 illustrates a third modification of the QAM/VSB modulator of Embodiment 5.

FIG. 177 is a block diagram showing a Trellis decoder in the demodulator of Embodiment 5.

FIG. 178 is a block diagram of a receiver of Embodiment 5 for interception of VSB multi-level transmitted signals emitted in the air.

FIG. 179is illustrates another arrangement of the QAM/VSB compatible receiver of Embodiment 5.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS EMBODIMENT 1

One embodiment of the present invention will be described referring to the relevant drawings.

In the preferred embodiment of the invention both the transmission apparatus, which comprises a transmitter for transmitting a digital HDTV signal or other digital signal and a receiver for receiving the transmitted signal, and the recording and reproducing apparatus, which records the digital HDTV signal or other digital signal on a magnetic tape or other recording medium and reproduces the recorded signal from said medium, are described.

It should be noted, however, that the configuration, operation, and principle of the digital modulator and demodulator, error correction encoder and decoder, and the encoder and decoder for image coding the HDTV signal are common to the transmission apparatus and the recording and reproducing apparatus, and apply essentially the same technologies. Therefore, to more concisely describe each embodiment, the block diagrams for either the transmission apparatus or the recording and reproducing apparatus are referenced in the description of each embodiment. In addition, the configuration of each embodiment of the invention can be achieved by means of any multi-value digital modulation method, e.g., QAM, ASK and PSK, positioning signal points in a constellation, and for brevity the embodiments of the present invention are described using only one modulation method. FIG. 1 shows the entire arrangement of a signal transmission system according to the first embodiment of the present invention. A transmitter 1 comprises an input unit 2, divider circuit 3, a modulator 4, and a transmitter unit 5. In operation, each input multiplex signal is divided by the divider circuit 3 into three groups, a first data stream D1, a second data stream and, a third data stream D3, which are then modulated by the modulator 4 before being transmitted from the transmitter unit 5. The modulated signal is sent up from an antenna 6 through an uplink 7 to a satellite 10 where it is intercepted by an uplink antenna 11 and amplified by a transponder 12 before being transmitted from a downlink antenna 13 towards the ground.

The transmission signal is then sent down through three downlinks 21, 32 31, and 41 to a first 23, a second 33, and a third receiver 43 respectively. In the first receiver 23, the signal intercepted by an antenna 22 is fed through an input unit 24 to a demodulator 25 where its first data stream only is demodulated, while the second and third data streams are not recovered, before being transmitted further from an output unit 26.

Similarly, the second receiver 33 allows the first and second data streams of the signal intercepted by an antenna 32 and fed from an input unit 34 to be demodulated by a demodulator 35 and then, combined by a mixer 37 into a single data stream which is then transmitted further from an output unit 36.

The third receiver 43 allows all of the first, second, and third data streams of the signal intercepted by an antenna 42 and fed from an input unit 44 to be demodulated by a demodulator 45 and then, combined by a mixer 47 into a single data stream which is then transmitted further from an output unit 46.

As understood, the three discrete receivers 23, 33, and 43 have their respective demodulators of different characteristics such that their outputs demodulated from the same frequency band signal of the transmitter 1 contain data of different sizes. More particularly, three different but compatible data can simultaneously be carried on a given frequency band signal to their respective receivers. For example, each of three, existing NTSC, HDTV, and super HDTV, digital signals is divided into low, high, and super high frequency band components which represent the first, the second, and the third data stream respectively. Accordingly, the three different TV signals can be transmitted on a one-channel frequency band carrier for simultaneous reproduction of medium, high, and super high resolution TV images respectively.

The NTSC TV signal is intercepted by a receiver accompanied by a small antenna for demodulation of small-sized data; the HDTV signal is intercepted by a receiver accompanied by a medium antenna for demodulation of medium-sized data, and the super HDTV signal is intercepted by a receiver accompanied by a large antenna for demodulation of large-sized data. Also, as illustrated in FIG. 1, a digital NTSC TV signal containing only the first data stream for digital NTSC TV broadcasting service is fed to a digital transmitter 51 where it is received by an input unit 52 and modulated by a demodulator modulator 54 before being transmitted further from a transmitter unit 55. The demodulated modulated signal is then sent up from an antenna 56 through an uplink 57 to the satellite 10 which in turn transmits the same through a downlink 58 to the first receiver 23 on the ground.

The first receiver 23 demodulates with its demodulator 25 the modulated digital signal supplied from the digital transmitter 51 into the original first data stream signal. Similarly, the same modulated digital signal can be intercepted and demodulated by the second receiver 33 or third receiver 43 into the first data stream or NTSC TV signal. In summary, the three discrete receivers 23, 33, and 43 all can intercept and process a digital signal of the existing TV system for reproduction.

The arrangement of the signal transmission system will be described in more detail.

FIG. 2 is a block diagram of the transmitter 1, in which an input signal is fed across the input unit 2 and divided by the divider circuit 3 into three digital signals containing a first, a second, and a third data stream respectively.

Assuming that the input signal is a video signal, its low frequency band component is assigned to the first data stream, its high frequency band component to the second data stream, its super-high frequency band component to the third data stream. The three different frequency band signals are fed to a modulator input 61 of the modulator 4. Here, a signal point shifting circuit 67 shifts the positions of the signal points according to an externally given signal. The modulator 4 is arranged for amplitude modulation on two 90°-out-of phase carriers respectively which are then combined into a multiple QAM signal. More specifically, the signal from the modulator input 61 is fed to both a first AM modulator 64 62 and a second AM modulator 63. Also, a carrier wave of cos(2πfct) produced by a carrier generator 64 is directly fed to the first AM modulator 64 62 and also, to a π/2 phase shifter 66 where it is 90° shifted in phase to a sin(2πfct) form prior to being transmitted to the second AM modulator 63. The two amplitude modulated signals from the first and second AM modulators 64 62, 63 are combined by a summer 65 into a transmission signal which is then transferred to the transmitter unit 5 for output. The procedure is well known and will not be further explained.

The QAM signal will now be described in a common 4×4 or 16 state constellation referring to the first quadrant of a space diagram in FIG. 3. The output signal of the modulator 4 is expressed by a sum vector of two, Acos2πfct and Bcos Bsin 2πfct, vectors 81 and 82 which respectively represent the two 90°-out-of-phase carriers. When the distal point of a sum vector from the zero point represents a signal point, the 16 QAM signal has 16 signal points determined by a combination of four horizontal amplitude values a1, a2, a3, and a4 and four vertical amplitude values b1, b2, b3, and b4. The first quadrant in FIG. 3 contains four signal points 83 at c11, 84 at c12, 85 at c22, and 86 at c21.

c11 is a sum vector of a vector 0-a1 and a vector 0-b1 and thus, expressed as c11=a1cos2πfct−b1sin2πfct=Acos (2πfct+dπ/2).

It is now assumed that the distance between 0 and a1 in the orthogonal coordinates of FIG. 3 is A1, between a1 and a2 is A2, between 0 and b1 is B1, and between b1 and b2 is B2.

As shown in FIG. 4, the 16 signal points are allocated in a vector coordinate, in which each point represents a four-bit pattern thus to allow the transmission of four bit data per period or time slot.

FIG. 5 illustrates a common assignment of two-bit patterns to the 16 signal points.

When the distance between two adjacent signal points is great, it will be identified by the receiver with much ease. Hence, it is desirable to space the signal points at greater intervals. If two particular signal points are allocated near to each other, they are rarely distinguished and the error rate will be increased. Therefore, it is most preferable to have the signal points spaced at equal intervals as shown in FIG. 5, in which the 16 QAM signal is defined by A1=A2/2.

The transmitter 1 of the embodiment is arranged to divide an input digital signal into a first, a second, and a third data or bit stream. The 16 signal points or groups of signal points are divided into four groups. Then, 4 two-bit patterns of the first data stream are assigned to the four signal point groups respectively, as shown in FIG. 6. More particularly, when the two-bit pattern of the first data stream is 11, one of four signal points of the first signal point group 91 in the first quadrant is selected depending on the content of the second data stream for transmission. Similarly, when 01, one signal point of the second signal point group 92 in the second quadrant is selected and transmitted. When 00, one signal point of the third signal point group 93 in the third quadrant is transmitted and when 10, one signal point of the fourth signal point group 94 in the fourth quadrant is transmitted. Also, 4 two-bit patterns in the second data stream of the 16 QAM signal, or e.g. 16 four-bit patterns in the second data stream of a 64-state QAM signal, are assigned to four signal points or sub signal point groups of each of the four signal point groups 91, 92, 93, and 94 respectively, as shown in FIG. 7. It should be understood that the assignment is symmetrical between any two quadrants. The assignment of the signal points to the four groups 91, 92, 93, and 94 is determined by priority to the two-bit data of the first data stream. As the result, two-bit data of the first data stream and two-bit data of the second data stream can be transmitted independently. Also, the first data stream will be demodulated by using a common 4 PSK receiver having a given antenna sensitivity. If the antenna sensitivity is higher, a modified type of the 16 QAM receiver of the present invention will intercept and demodulate both the first and second data stream streams with equal success.

FIG. 8 shows an example of the assignment of the first and second data streams in two-bit patterns.

When the low frequency band component of an HDTV video signal is assigned to the first data stream and the high frequency component to the second data stream, the 4 PSK receiver can produce an NTSC-level picture from the first data stream and the 16- or 64-state QAM receiver can produce an HDTV picture from a composite reproduction signal of the first and second data streams.

Since the signal points are allocated at equal intervals, there is developed in the 4 PSK receiver a threshold distance between the coordinate axes and the shaded area of the first quadrant, as shown in FIG. 9. If the threshold distance is ATO, a PSK signal having an amplitude of ATO will successfully be intercepted. However, the amplitude has to be increased to a three times greater value or 3ATO for transmission of a 16 QAM signal while the threshold distance ATO is maintained. More particularly, the energy needed for transmitting the 16 QAM signal is nine times greater than that for sending the 4 PSK signal. Also, when the 4 PSK signal is transmitted in a 16 QAM mode, energy waste will be high and reproduction of a carrier signal will be troublesome. Above all, the energy available for satellite transmitting is not abundant but strictly limited to minimum use. Hence, no large-energy-consuming signal transmitting system will be put into practice until more energy for satellite transmission is available. It is expected that a great number of the 4 PSK receivers will be introduced into the market as digital TV broadcasting is placed in service. After introduction to the market, the 4 PSK receivers will hardly be shifted to higher sensitivity models because a signal intercepting characteristic gap between the two, old and new, models is high. Therefore, the transmission of the 4 PSK signals must not be abandoned. In this respect, a new system is desperately needed for transmitting the signal point data of a quasi 4 PSK signal in the 16 QAM mode using less energy. Otherwise, the limited energy at a satellite station will degrade the entire transmission system.

The present invention resides in a multiple signal level arrangement in which the four signal point groups 91, 92, 93, and 94 are allocated at a greater distance from each other, as shown in FIG. 10, for minimizing the energy consumption required for 16 QAM modulation of quasi 4 PSK signals.

For clearing the relationship between the signal receiving sensitivity and the transmitting energy, the arrangement of the digital transmitter 51 and the first receiver 23 will be described in more detail referring to FIG. 1.

Both the digital transmitter 51 and the first receiver 2 3 23 are formed of known types for data transmission or video signal transmission e.g. in TV broadcasting service. As shown in FIG. 17, the digital transmitter 51 is a 4 PSK transmitter equivalent to the multiple-bit QAM transmitter 1, shown in FIG. 2, without AM modulation capability. In operation, an input signal is fed through an input unit 52 to a modulator 54 where it is divided by a modulator input 121 into two components. The two components are then transferred to a first two-phase modulator circuit 122 for phase modulation of a base carrier and a second two-phase modulator circuit 123 for phase modulation of a carrier which is 90° out of phase with the base carrier respectively. The two outputs of the first and second two-phase modulator circuits 122 and 123 are then combined by a summer 65 into a composite modulated signal which is further transmitted from a transmitter unit 55.

The resultant modulated signal is shown in the space diagram of FIG. 18.

It is known that the four signal points are allcated allocated at equal distances for achieving optimum energy utilization. FIG. 18 illustrates an example where the four signal points 125, 126, 127, and 128 represent 4 two-bit patterns, 11, 01, 00, and 10 respectively. It is also desirable for successful data transfer from the digital transmitter 51 to the first receiver 23 that the 4 PSK signal from the digital transmitter 51 has an amplitude of not less than a given level. More specifically, when the minimum amplitude of the 4 PSK signal needed for transmission from the digital transmitter 51 to the first receiver 23 of 4 PSK mode, or the distance between 0 and a1 in FIG. 18 is ATO, the first receiver 23 must successfully intercept any 4 PSK signal having an amplitude of more than ATO.

The first receiver 23 is arranged to receive at its small-diameter antenna 22 a desired or 4 PSK signal which is transmitted from the transmitter 1 or digital transmitter 51 respectively through that transponder 12 of the satellite 10 and demodulate it with the demodulator 24 25. In more detail, the first receiver 23 is substantially designed for interception of a digital TV or data communications signal of 4 PSK or 2 PSK mode.

FIG. 19 is a block diagram of the first receiver 23 in which an input signal received by the antenna 22 from the satellite 12 10is fed through the input unit 24 to a carrier reproducing circuit 131 where a carrier wave is demodulated and to a π/2 phase shifter 132 where a 90° phase carrier wave is demodulated. Also, two 90°-out-of-phase components of the input signal are respectively detected by a first phase detector circuit 133 and a second phase detector circuit 134 and are respectively transferred to first 136 and second discrimination/demodulation circuits 136 and 137. Two demodulated components from their respective discrimination/demodulation circuits 136 and 137, which have separately been discriminated at units of time slot by means of timing signals from a timing wave extracting circuit 135, are fed to a first data stream reproducing unit 232 where they are combined into a first data stream signal which is then delivered as an output from the output unit 26.

The input signal to the first receiver 23 will now be explained in more detail referring to the vector diagram of FIG. 20. The 4 PSK signal received by the first receiver 23 from the digital transmitter 51 is expressed in an ideal form without transmission distortion and noise, using four signal points 151, 152, 153, and 154, as shown in FIG. 20.

In practice, the real four signal points appear in particular extended areas about the ideal signal positions 151, 152, 153, and 154 respectively due to noise, amplitude distortion, and phase error developed during transmission. If one signal point is unfavorably displaced from its original position, it will hardly be distinguished from its neighboring signal point and the error rate will thus be increased. As the error rate increases to a critical level, the reproduction of data becomes less accurate. For enabling the data reproduction at a maximum acceptable level of the error rate, the distance between any two signal points should be far enough to be distinguished from each other. If the distance is 1AR0, the signal point 151 of a 4 PSK signal close to a critical error level has to stay in a first discrimination area 155 denoted by the hatching of FIG. 20 and determined by |0-aR1|>AR0 and |0-bR1|>AR0. This allows the signal transmission system to reproduce carrier waves and thus, demodulate a wanted signal. When the minimum radius of the antenna 22 is set to r0, the transmission signal of more than a given level can be intercepted by any receiver of the system. The amplitude of a 4 PSK signal of the digital transmitter 51 shown in FIG. 18 is minimum at AT0 and thus, the minimum amplitude AR0 of a 4 PSK signal to be received by the first receiver 23 is determined to be equal to AT0. As a result, the first receiver 23 can intercept and demodulate the 4 PSK signal from the digital transmitter 51 at the maximum acceptable level of the error rate when the radius of the antenna 22 is more than r0. If the transmission signal is of a modified 16- or 64-state QAM mode, the first receiver 23 may find it difficult to reproduce its carrier wave. For compensation, the signal points are increased to eight which are allocated at angles of (π/4+nπ/2) as shown in FIG. 25(a) and its carrier wave will be reproduced by a 16× multiplication technique. Also, if the signal points are assigned to 16 locations at angles of nπ/8 as shown in FIG. 25(b), the carrier of a quasi 4 PSK mode 16 QAM modulated signal can be reproduced with the carrier reproducing circuit 131 which is modified for performing 16× frequency multiplication. At the time, the signal points in the transmitter 1 should be arranged to satisfy A1/(A1+A2)=tan(π/8).

Here, a case of receiving a QPSK signal will be considered. Similarly to the manner performed by the signal point setting circuit 67 in the transmitter shown in FIG. 2, it is also possible to modulate the positions of the signal points of the QPSK signal shown in FIG. 18 (amplitude-modulation, pulse-modulation, or the like). In this case, the signal point demodulating unit 138 in the first receiver 23 demodulates the position modulated or position changed signal. The demodulated signal is outputted together with the first data stream.

The 16 PSK signal of the transmitter 1 will now be explained referring to the vector diagram of FIG. 9. When the horizontal vector distance A1 of the signal point 83 is greater than ATO of the minimum amplitude of the 4 PSK signal of the digital transmitter 51, the four signal points 83, 84, 85, and 86 in the first quadrant of FIG. 9 stay in the shaded or first 4 PSK signal receivable area 87. When received by the first receiver 23, the four points of the signal appear in the first discriminating area of the vector field shown in FIG. 20. Hence, any of the signal points 83, 84, 85, and 86 of FIG. 9 can be translated into the signal level 151 of FIG. 20 by the first receiver 23 so that the two-bit pattern of 11 is assigned to a corresponding time slot. The two-bit pattern of 11 is identical to 11 of the first signal point group 91 or first data stream of a signal from the transmitter 1. Equally, the first data stream will be reproduced at the second, third, or fourth quadrant. As the result, the first receiver 23 reproduces two-bit data of the first data stream out of the plurality of data streams in a 16-, 32-, or 64-state QAM signal transmitted from the transmitter 1. The second and third data streams are contained in four segments of the signal point group 91 and thus, will not affect the demodulation of the first data stream. They may however affect the reproduction of a carrier wave and an adjustment, described later, will be needed.

If the transponder of a satellite supplies an abundance of energy, the forgoing technique of 16 to 64-state QAM mode transmission will be feasible. However, the transponder of the satellite in any existing satellite transmission system is strictly limited in the power supply due to its compact size and the capability of solar batteries. If the transponder or satellite is increased in size and thus weight, its launching cost will soar. This disadvantage will rarely be eliminated by traditional techniques unless the cost of launching a satellite rocket is reduced by to a considerable level. In the existing system, a common communications satellite provides as low as 20 W of power and a common broadcast satellite offers 100 W to 200 W at best. For transmission of such a 4 PSK signal in the symmetrical 16-state QAM mode as shown in FIG. 9, the minimum signal point distance is needed needed is 3AT0 as the 16 QAM amplitude is expressed by 2A1=A2. Thus, the energy needed for the purpose is nine times greater than that for transmission of a common 4 PSK signal, in order to maintain compatibility. Also, any conventional satellite transponder can hardly provide a power for enabling such a small antenna of the 4 PSK first receiver to intercept a transmitted signal therefrom. For example, in the existing 40 W system, 360 W is needed for appropriate signal transmission and will be unrealistic with respect to cost.

It would be understood that the symmetrical signal state QAM technique is most effective when the receivers equipped with the same sized antennas are employed corresponding to a given transmitting power. Another novel technique will however be preferred for use with receivers equipped with different sized antennas.

In more detail, while the 4 PSK signal can be intercepted by a common low cost receiver system having a small antenna, the 16 QAM signal is intended to be received by a high cost, high quality, multiple-bit modulating receiver system with a medium or large sized antenna which is designed for providing highly valuable services, e.g. HDTV entertainment, to a particular person who invests more money. This allows both 4 PSK and 16 QAM signals, if desired, with a 64 DMA QAM, to be transmitted simultaneously with the help of a small increase in the transmitting power.

For example, the transmitting power can be maintained low when the signal points are allocated at A1=A2 as shown in FIG. 10. The amplitude A(4) for transmission of 4 PSK data is expressed by a vector 96 equivalent to the square root of (A1+A2)2+(B1+B2)2. Then,
|A(4)|2=A1 2+B1 2=A2 TO+A2 TO=2A2 TO
|A(16)|2=(A1+A2)2+(B1+B2)2=4A2 TO+4A2 TO=8ATO
|A(16)|/|A(4)|=2

Accordingly, the 16 QAM signal can be transmitted at a two times greater amplitude and a four times greater transmitting energy than those needed for the 4 PSK signal. A modified 16 QAM signal according to the present invention will not be demodulated by a common receiver designed for symmetrical, equally distanced signal point QAM. However, it can be demodulated with the second receiver 33 when two threshold values A1 and A2 are preset to appropriate values. In FIG. 10, the minimum distance between two signal points in the first segment of the signal point group 91 is A1 and A2/2A1 is established as compared with the distance 2A1 of 4 PSK. Then, as A1=A2, the distance becomes ½. This explains that the signal receiving sensitivity has to be two times greater for the same error rate and four times greater for the same signal level. For having a four times greater value of sensitivity, the radius r2 of the antenna 32 of the second receiver 33 has to be two times greater than the radius r1 of the antenna 22 of the first receiver 23 thus satisfying r2=2r1. For example, the antenna 32 of the second receiver 33 is 60 cm diameter when the antenna 22 if the first receiver 23 is 30 cm. In this manner, the second data stream representing the high frequency component of an HDTV will be carried on a signal channel and demodulated successfully. As the second receiver 33 intercepts the second data stream or a higher data signal, its owner can enjoy a of high return of investment return . Hence, the second receiver 33 of a high price may be accepted. As the minimum energy for transmission of 4 PSK data is predetermined, the ratio n16 of modified 16 APSK transmitting energy to 4 PSK transmitting energy will be calculated according to the antenna radius r2 of the second receiver 33 using a ratio between A1 and A2 shown in FIG. 10.

In particular, n16 is expressed by ((A1+A2)/A1)2 which is the minimum energy for transmission of 4 PSK data. As the signal point distance suited for modified 16 QAM interception is A2, The the signal point distance for 4 PSK interception is 2A1, and the signal point distance ratio is A2/2A1, the antenna radius r2 is determined as shown in FIG. 11, in which the curve 101 represents the relationship between the transmitting energy ratio n16 and the radius r2 of the antenna 22 of the second receiver 23.

Also, the point 102 indicates transmission of common 16 QAM at the equal distance signal state mode where the transmitting energy is nine times greater and thus will no more be practical. As apparent from the graph of FIG. 11, the antenna radius r2 of the second receiver 23 cannot be reduced further even if n16 is increased more than 5 times.

The transmitting energy at the satellite is limited to a small value and thus, n16 preferably stays not more than 5 times the value, as denoted by the hatching of FIG. 11. The point 104 within the hatching area 103 indicates, for example, that the antenna radius r2 of a two times greater value is matched with a 4× value of the transmitting energy. Also the point 105 represents that the transmission energy should be doubled when r2 is about 5× greater. Those values are all within a feasible range.

The value of n16 not greater than 5× value is expressed using A1 and A2 as:
n16=((A1+A2)/A1)2<5
Hence, A2<1.23A1.

If the distance between any two signal point group segments shown in FIG. 10 is 2A(4) and the maximum amplitude is 2A(16), A(4) and A(16)−A(4) are proportional to A1 A1 and A2 A2 respectively. Hence, (A(16))2<5(A(14))2 is established.

The action of a modified 64 ASPK transmission will be described as the third receiver 43 can perform 64-state QAM demodulation.

FIG. 12 is a vector diagram in which each signal point group segment contains 16 signal points as compared with 4 signal points of FIG. 10. The first signal point group segment 91 in FIG. 12 has a 4×4 matrix of 16 signal points allocated at equal intervals including the point 170. For providing compatibility with 4 PSK, A1>AT0 has to be satisfied. If the radius of the antenna 42 of the third receiver 43 is r3 and the transmitting energy is n64, the equation is expressed as:
r3 2={62/(n−1)}r1 2

This relationship between r3 r3 and n of a 64 QAM signal is also shown in the graphic representation of FIG. 13.

It is understood that the signal point assignment shown in FIG. 12 allows the second receiver 33 to demodulate only two-bit patterns of 4 PSK data. Hence, it is desirable for to have compatibility between among the first, second, and third receivers that the second receiver 33 is capable of demodulating a modified 16 QAM form from the 64 QAM modulated signal.

The compatibility between among the three discrete receivers can be implemented by a three-level grouping of signal points, as illustrated in FIG. 14. A description follows referring to the first quadrant in which the first signal point group segment 91 represents the two-bit pattern 11 of the first data stream.

In particular, a first sub segment 181 in the first signal point group segment 91 is assigned the two-bit pattern 11 of the second data stream. Equally, a second 182, a third 183, and a fourth sub segment 184 are assigned 01, 00, and 10 of the same respectively. This assignment is identical to that shown in FIG. 7.

The signal point allocation of the third data stream will now be explained referring to the vector diagram of FIG. 15 which shows the first quadrant. As shown, the four signal points 201, 205, 209, and 213 represent the two-bit pattern of 11, the signal points 202, 206, 210, and 214 represent 01, the signal points 203, 207, 211, and 215 represent 00, and signal points 204, 208, 212, and 216 represent 10. Accordingly, the two-bit patterns of the third data stream can be transmitted separately of the first and second data streams. In other words, two-bit data of the three different signal levels can be transmitted respectively.

As understood, the present invention permits not only transmission of six-bit data but also interception of three, two-bit, four-bit, and six-bit, different bit length data with their respective receivers while the signal compatibility remains between these levels.

The signal point allocation for providing compatibility between among the three levels will be described.

As shown in FIG. 15, A1>AT0 is essential for allowing the first receiver 23 to receive the first data stream.

It is necessary to space any two signal points from each other by such a distance that the sub segment signal points, e.g. 182, 183, 184, of the second data stream shown in FIG. 15 can be distinguished from the signal point 91 shown in FIG. 10.

FIG. 15 shows that they are spaced by ⅔A2. In this case, the distance between the two signal points 201 and 202 in the first sub segment 181 is A2/6. The transmitting energy needed for signal interception with the third receiver 43 is now calculated. If the radius of the antenna 32 42is r3 and the needed transmitting energy is n64 times the 4 PSK transmitting energy, the equation is expressed as:
R3 2(12r1)2/(n−1)R3 2=( 12r 1)2/(n−1 )

This relationship is also denoted by the curve 211 in FIG. 16. For example, if the transmitting energy is 6 or 9 times greater than that for 4 PSK transmission at the point 223 or 222, the antenna 32 having a radius of 8× or 6× value respectively can intercept the first, second, and third data streams for demodulation. As the signal point distance of the second data stream is close to ⅔A2, the relationship between r1 and r2 is expressed by:
R22=(3r1)2/(n−1) R2 2=( 3r 1)2/(n−1 )
Therefore, the antenna 32 of the second receiver 33 has to be slightly increased in radius as denoted by the curve 223.

As understood, while the first and second data streams are transmitted through a traditional satellite which provides a small signal transmitting energy, the third data stream can also be transmitted through a future satellite which provides a greater signal transmitting energy without interrupting the action of the first and second receivers 23 or 33 or with no need of modification of the same and thus, both the compatibility and the advancement is ensured.

The signal receiving action of the second receiver 33 will first be described. As compared with the first receiver 23 arranged for interception with a small radius r1 antenna and demodulation of the 4 PSK modulated signal of the digital transmitter 51 or the first data stream of the signal of the transmitter 1, the second receiver 33 is adopted for perfectly demodulating the 16 signal state two-bit data, shown in FIG. 10, or second data stream of the 16 QAM signal from the transmitter 1. In total, four-bit data including also the first data stream can be demodulated. The ratio between A1 and A2 is however different in the two transmitters. The two different data are loaded to a demodulation controller 231 of the second receiver 33, shown in FIG. 21, which in turn supplies their respective threshold values to the demodulating circuit for AM demodulation.

The block diagram of the second receiver 33 in FIG. 21 is similar in basic construction to that of the first receiver 23 shown in FIG. 19. The difference is that the radius r2 of the antenna 32 is greater than r1 of the antenna 22. This allows the second receiver 33 to identify a signal component involving a smaller signal point distance. The demodulator 35 of the second receiver 33 also contains first and second data stream reproducing units 232 and 233 in addition to the demodulation controller 231. There is provided a first discrimination/demodulation circuit 136 for AM demodulation of modified 16 QAM signals. As understood, each carrier is a four-bit signal having two, positive and negative, threshold values about the zero level. AS As apparent from the vector diagram, of FIG. 22, the threshold values are varied depending on the transmitting energy of a transmitter since the transmitting signal of the embodiment is a modified 16 QAM signal. When the reference threshold is TH16, it is determined by, as shown in FIG. 22:
TH16=(A1+A2/2)/(A1+A2)

The various data for demodulation including A1 and A2 or TH16, and the value m for multiple-bit modulation are also transmitted from the transmitter 1 as carried in the first data stream. The demodulation controller 231 may be arranged for recovering such demodulation data through statistical process of the received signal.

A way of determining the shift factor A1/A2 will be described with reference to FIG. 26. A change of the shift factor A1/A2 causes a change of the threshold value. Increase of a difference of a value of A1/A2 set at the receiver side from a value of A1/A2 set at the transmitter side will increase the error rate. Referring to FIG. 26, the demodulated signal from the second data stream reproducing unit 233 may be fed back to the demodulation controller 231 to change the shift factor A1/A2 in a direction to increase the error rate. By this arrangement, the third receiver 43 may not demodulate the shift factor A1/A2, so that the circuit construction can be simplified. Further, the transmitter may not transmit the shift factor A1/A2, so that the transmission capacity can be increased. This technique can be applied also to the second receiver 33.

FIGS. 25(a) and 25(b) are views showing signal point allocations for the C-CDM signal points, wherein signal points are added by shifting in the polar coordinate direction (r, θ). The previously described C-CDM is characterized in that the signal points are shifted in the rectangular coordinate direction, i.e. XY direction; therefore it is referred to as rectangular coordinate system C-CDM. Meanwhile, this C-CDM characterized by the shifting of signal points in the polar coordinate direction, i.e. r, θ direction, is referred to as polar coordinate system C-CDM.

FIG. 25(a) shows the signal allocation of 8PS-APSK signals, wherein four signal points are added by shifting each of 4 QPSK signals in the radius r direction of the polar coordinate system. In this manner, the APSK of polar coordinate system C-CDM having 8 signal points is obtained from the QPSK as shown in FIG. 25(a). As the pole is shifted in the polar coordinate system to add signal points in this APSK, it is referred to as shifted pole-APSK, i.e. SP-APSK in the abbreviated form. In this case, coordinate values of the newly added four QPSK signals 85 are specified by using a shift factor S1 as shown in FIG. 139. Namely, 8PS-APSK signal points includes ordinary QPSK signal points 83 (r0, θ0) and a signal point ((S1+1)(r0, θ0) obtained by shifting the signal point 83 in the radius r direction by an amount of S1r0. Thus, a 1-bit subchannel 2 is obtained in addition to a 2-bit subchannel 1 identical with the QPSK.

Furthermore, as shown in the constellation diagram of FIG. 140, new eight signal points, represented by coordinates (r0+S2r0, θ0) and (r0+S1r0+S2r0, θ0), can be added by shifting the eight signal points (r0, θ0) and (r0+S1r0, θ0) in the radius r direction. As this allows two kinds of allocations, a 1-bit subchannel is obtained and is referred to as 16PS-APSK which provides the 2-bit subchannel 1, 1-bit subchannel 2, and 1-bit subchannel 3. As the 16-PS-APSK disposes the signal points on the lines of θ=¼(2n+1)π, it allows the ordinary QPSK receiver explained with reference to FIG. 19 to reproduce the carrier wave to demodulate the first 2-bit subchannel although the second subchannel cannot be demodulated. As described above, the C-CDM method of shifting the signal points in the polar coordinate direction is useful in expanding the capacity of information data transmission while assuring compatibility to the PSK, especially to the QPSK receiver, a main receiver for the present satellite broadcast service. Therefore, without losing the first generation viewers of the satellite broadcast service based on the PSK, the broadcast service will advance to a second generation stage wherein the APSK will be used to increase transmittable information amount by use of the multi-level modulation while maintaining compatibility.

In FIG. 25(b), the signal points are allocated on the lines of θ=π/8. With this arrangement, the 16 PSK signal points are reduced or limited to 12 signal points, i.e. 3 signal points in each quadrant. With this limitation, these three signal points in each quadrant are roughly regarded as one signal point for 4 QPSK signals. Therefore, this enables the QPSK receiver to reproduce the first subchannel in the same manner as in the previous embodiment.

More specifically, the signal points are disposed on the lines of θ=π/4, θ=π/4+π/8, and θ=π/4−π/8. In other words, the added signals are offset by an amount ±θ in the angular direction of the polar coordinate system from the QPSK signals disposed on the lines of θ=π/4. Since all the signals are in the range of θ=π/4±π/8, they can be regarded as one of the QPSK signal points on the line of θ=π/4. Although the error rate is lowered a little bit in this case, the QPSK receiver 23 shown in FIG. 19 can discriminate these points as four signal points angularly allocated. Thus, 2-bit data can be reproduced. In case of the angular shift C-CDM, if signal points are disposed on the lines of π/n, the carrier wave reproduction circuit can reproduce the carrier wave by the use of an n-multiplier circuit in the same manner as in other embodiments. If the signal points are not disposed on the lines of π/n, the carrier wave can be reproduced by transmitting several pieces of carrier information within a predetermined period in the same manner as in other embodiment embodiments. Assuming that an angle between two signal points of the QPSK or 8-SP-APSK is 2θ0 in the polar coordinate system and a first angular shift factor is P1, two signal points (r0, θ0+P1θ0) and (r0, % θ0−P1θ0) are obtained by shifting the QPSK signal point in the angular θ direction by an amount ±P1θ0. Thus, the number of signal points are doubled. Thus, the 1-bit subchannel 3 can be added and is referred to as 8-SP-PSK of P=P1. If eight signal points are further added by shifting the 8-SP-PSK signals in the radius r direction by an amount S1r0, it will become possible to obtain 16-SP-APSK (P, S1 type) as shown in FIG. 142. The subchannels 1 and 2 can be reproduced by two 8PS-PSKs having the same phase. Returning to FIG. 25(b), as the C-CDM based on the angular shift in the polar coordinate system can be applied to the PSK as shown in FIG. 141, this will be adopted to the first generation satellite broadcast service. However, if adopted to the second generation satellite broadcasting based on the APSK, this polar coordinate system C-CDM is inferior in that signal points in the same group cannot be uniformly spaced as shown in FIG. 142. Accordingly, utilization efficiency of electrical power is worsened. On the other hand, the rectangular coordinate system C-CDM has good compatibility to the PSK.

The system shown in FIG. 25(b) is compatible with both the rectangular and polar coordinate systems. As the signal points are disposed on the angular lines of the 16 PSK, they can be demodulated by the 16 PSK. Furthermore, as the signal points are divided into groups, the QPSK receiver can be used for demodulation. Still further, as the signal points are also allocated to suit the rectangular coordinate system, the demodulation will be performed by the 16-SRQAM. Consequently, the compatibility between the rectangular coordinate system C-CDM and the polar coordinate system C-CDM can be assured in any of the QPSK, 16PSK, and 16-SRQAM.

The demodulation controller 231 has a memory 231a for storing therein different threshold values (i.e., the shift factors, the number of signal points, the synchronization rules, etc.) which correspond to different TV broadcast channels. When receiving one of the channels again, the values corresponding to the receiving channel will be read out of the memory to thereby stabilize the reception quickly.

If the demodulation data is lost, the demodulation of the second data stream will hardly be executed. This will be explained referring to a flowchart shown in FIG. 24.

Even if the demodulation data is not available, demodulation of the 4 PSK at Step 313 and of the first data stream at Step 301 can be implemented. At Step 302, the demodulation data retrieved by the first data stream reproducing unit 232 is transferred to the demodulation controller 231. If m is 4 or 2 at Step 303, the demodulation controller 231 triggers demodulation of 4 PSK or 2 PSK at Step 313. If not, the procedure moves to Step 310. At Step 305, two threshold values TH8 and TH16 are calculated. The threshold value TH16 for AM demodulation is fed at Step 306 from the demodulation controller 231 to both the first 136 and the second discrimination/demodulation circuit 137. Hence, demodulation of the modified 16 QAM signal and reproduction of the second data stream can be carried out at Steps 307 and 315 respectively. At Step 308, the error rate is examined and if high, the procedure returns to Step 313 for repeating the 4 PSK demodulation.

As shown in FIG. 22 and the signal points 85, 83, are aligned on a line at an angle of cos(ωt+nπ/2) while 84 and 86 are off the line. Hence, the feedback of a second data stream transmitting carrier wave data from the second data stream reproducing unit 233 to a carrier reproducing circuit 131 is carried out so that no carrier needs to be extracted at the timing of the signal points 84 and 86.

The transmitter 1 is arranged to transmit carrier timing signals at intervals of a given time with the first data stream for the purpose of compensation for no demodulation of the second data stream. The carrier timing signal enables one to identify the signal points 83 and 85 of the first data stream regardless of demodulation of the second data stream. Hence, the reproduction of carrier wave can be triggered by the transmitting of carrier data to the carrier reproducing circuit 131.

A determination then made at Step 304 of the flowchart of FIG. 24 as to whether or not m is 16 upon receipt of such a modified 64 QAM signal as shown in FIG. 23. At Step 310, a determination is also made as to whether or not m is more than 64. If it is determined at Step 311 that the received signal has no equal distance signal point constellation, the procedure goes to Step 312. The signal point distance TH64 of the modified 64 QAM signal is calculated from:
TH64=(A1+A2/2)/(A1+A2)
This calculation is equivalent to that of TH16 but its resultant distance between signal points is smaller.

If the signal point distance in the first sub segment 181 is A3, the distance between the first 181 and the second sub segment 182 is expressed by (A22A3). Then, the average distance is (A22A3)/(A1+A2) which is designated as D64, when D64 is smaller than T2 which represents the signal point discrimination capability of the second receiver 33, any two signal points in the segment will hardly be distinguished from each other. This judgement is executed at Step 313. If D64 is out of a permissive range, the procedure moves back to Step 313 for 4 PSK mode demodulation. If D64 is within the range, the procedure advances to Step 305 for allowing the demodulation of 16 QAM at Step 307. If it is determined at Step 308 that the error rate is too high, the procedure goes back to Step 313 for 4 PSK mode demodulation.

When the transmitter 1 supplied a modified 8 QAM signal such as shown in FIG. 25(a) in which all the signal points are at angles of cos (2πf+π/4), the carrier waves of the signal are lengthened to the same phase and will thus be reproduced with much ease. At the time, two-bit data of the first data stream are demodulated by the 4-PSK receiver while one-bit data of the second data stream is demodulated by the second receiver 33 and the total of three-bit data can be reproduced.

The third receiver 43 will be described in more detail. FIG. 26 shows a block diagram of the third receiver 43 similar to that of the second receiver 33 in FIG. 21. The difference is that a third data stream reproducing unit 234 is added and also, the discrimination/demodulation circuit has a capability of identifying eight-bit data. The antenna 42 of the third receiver 43 has a radius r3 greater than r2 thus allowing smaller distance state signals, e.g. 32- or 64-state QAM signals, to be demodulated. For demodulation of the 64 QAM signal, the first discrimination/reproduction circuit 136 has to identify 8 digital levels of the detected signal in which seven different threshold levels are involved. As one of the threshold values is zero, three are contained in the first quadrant.

FIG. 27 shows a space diagram of the signal in which the first quadrant contains three different threshold values.

As shown in FIG. 27, when the three normalized threshold values are TH1 64, TH2 64, and TH3 64 they are expressed by:
TH164=(A1+A3/2)/(A1+A2)
TH264=(A1+A2/2)/(A1+A2) and
TH364=(A1+A2−A3/2)/(A1+A2)

Through AM demodulation of a phase detected signal using the three threshold values, the third data stream can be reproduced like the first and second data stream explained with FIG. 21. The third data stream contains e.g. four signal points 201, 202, 203, and 204 at the first sub segment 181 shown in FIG. 23 which represent 4 values of two-bit pattern. Hence, six digits or modified 64 QAM signals can be demodulated.

The demodulation controller 231 detects the value m, A1, A2, and A3 from the demodulation data contained in the first data stream demodulated by the first data stream reproducing unit 232 and calculates the three threshold values TH1 64, TH2 64, and TH3 64 which are then fed to the first 136 and the second discrimination/demodulation circuit 137 so that the modified 64 QAM signal is demodulated with certainty. Also, if the demodulation data have been scrambled, the modified 64 QAM signal can be demodulated only with a specific or subscriber receiver. FIG. 28 is a flowchart showing the action of the demodulation controller 231 for modified 64 QAM signals. The difference from the flowchart for demodulation of 16 QAM shown in FIG. 24 will be explained. The procedure moves from Step 304 to Step 320 where it is determined whether or not m=32 or not. If m=32, demodulation of 32 QAM signals is executed at Step 322. If not, the procedure moves to Step 321 where it is determined whether or not m=64. If yes, A3 is examined at Step 323. If A3 is smaller than a predetermined value, the procedure moves to Step 305 and the same sequence as of FIG. 24 is implemented. If it is judged at Step 323 that A3 is not smaller than the predetermined value, the procedure goes to Step 324 where the threshold values are calculated. At Step 325, the calculated threshold values are fed to the first and second discrimination/demodulation circuits and at Step 326, the demodulation of the modified 64 QAM signal is carried out. Then, the first, second, and third data streams are reproduced at Step 327. At Step 328, the error rate is examined. If the error rate is high, the procedure moves to Step 305 where the 16 QAM demodulation is repeated and if low, the demodulation of the 64 QAM is continued.

The action of carrier wave reproduction needed for execution of a satisfactory demodulating procedure will now be described. The scope of the present invention includes reproduction of the first data stream of a modified 16 or 64 QAM signal using a 4 PSK receiver. However, a common 4 PSK receiver rarely reconstructs carrier waves, thus failing to perform a correct demodulation. For compensation, some arrangements are necessary at both the transmitter and receiver sides.

Two techniques for compensation are provided according to the present invention. A first technique relates to transmission of signal points aligned at angles of (2n−1)π/4 at intervals of a given time. A second technique offers transmission of signal points arranged at intervals of an angle of nπ/8.

According to the first technique, the eight signal points including 83 and 85 are aligned at angles of π/4, 3π/4, 5π/4, and 7π/4, as shown in FIG. 38. In action, at least one of the eight signal points is transmitted during sync time slot periods 452, 453, 454, and 455 arranged at equal intervals of time in a time slot gap 451 shown in the time chart of FIG. 38. Any desired signal points are transmitted during the other time slots. The transmitter 1 is also arranged to assign a data for the time slot interval to the sync timing data region 499 of a sync data block, as shown in FIG. 41.

The content of a transmitting signal will be explained in more detail referring to FIG. 41. The time slot group 451 containing the sync time slots 452, 453, 454, and 455 represents a unit data stream or block 491 carrying a data of Dn.

The sync time slots in the signal are arranged at equal intervals of a given time determined by the time slot interval or sync timing data. Hence, when the arrangement of the sync time slots is detected, reproduction of carrier waves will be executed slot by slot through extracting the sync timing data from their respective time slots. Such a sync timing data S is contained in a sync block 493 at the front end of a data frame 492, which consists of a number of sync time slots denoted by the hatching in FIG. 41. Accordingly, the data to be extracted for carrier wave reproduction are increased, thus allowing the 4 PSK receiver to reproduce desired carrier waves at higher accuracy and efficiency.

The sync block 493 comprises sync data regions 496, 497, and 498, —containing sync data S1, S2, and S3, —respectively which include unique words and demodulation data. The phase sync signal assignment region 499 is at the end of the sync block 493, which holds a data of IT including information about interval arrangement and assignment of the sync time slots.

The signal point data in the phase sync time slot has a particular phase and can thus be reproduced by the 4 PSK receiver. Accordingly, IT in the phase sync signal assignment region 499 can be retrieved without error thus ensuring the reproduction of carrier waves at with accuracy.

As shown in FIG. 41, the sync block 493 is followed by a demodulation data block 501 which contains demodulation data about threshold voltages needed for demodulation of the modified multiple-bit QAM signal. This data is essential for demodulation of the multiple-bit QAM signal and may preferably be contained in a region 502 which is a part of the sync block 493 for ease of retrieval.

FIG. 42 shows the assignment of signal data for transmission of burst form signals through a TDMA method.

The assignment is distinguished from that of FIG. 41 by the fact that a guard period 521 is inserted between any two adjacent Dn data blocks 491 and 491 for interruption of the signal transmission. Also, each data block 491 is at the front end of a sync region 522, the signal points at a phase of (2n−1)π/4 are only transmitted. Accordingly, the carrier wave reproduction will be feasible with the 4 PSK receiver. More specifically, the sync signal and carrier waves can be reproduced through the TDMA method.

The carrier wave reproduction of the first receiver 23 shown in FIG. 19 will be explained in more detail referring to FIGS. 43 and 44. As shown in FIG. 43, an input signal is fed through the input unit 24 to a sync detector circuit 541 where it is sync detected. A demodulated signal from the sync detector 541 is transferred to an output circuit 542 for reproduction of the first data stream. A data of the phase sync signal assignment data region 499 (shown in FIG. 41) is retrieved by an extracting timing controller circuit 543 so that the timing of sync signals of (2n−1)π/4 data can be acknowledged and transferred as a phase sync control pulse 561 shown in FIG. 44 to a carrier reproduction controlling circuit 544. Also, the demodulated signal of the sync detector circuit 541 is fed to a frequency multiplier circuit 545 where it is 4× multiplied prior to being transmitted to the carrier reproduction controlling circuit 544. The resultant signal denoted by 562 in FIG. 44 contains true phase data 563 and other data. As illustrated by 564 in the time chart 564 of FIG. 44, the phase sync time slots 452 carrying the (2n−1)π/4 data are also contained at equal intervals. In the carrier reproducing controlling circuit 544, the signal 562 is sampled by the phase sync control pulse 561 to produce a phase sample signal 565 which is then converted through a sample and hold operation into a phase signal 566. The phase signal 566 of the carrier reproduction controlling circuit 544 is fed through a loop filter 546 to a VCO 547 where its relevant carrier wave is reproduced. The reproduced carrier is then sent to the sync detector circuit 541.

In this manner, the signal point data of the (2n−1)π/4 phase denoted by the shaded areas in FIG. 39 is recovered and utilized so that a correct carrier wave can be reproduced by 4× or 16× frequency multiplication. Although a plurality of phases are reproduced at the time, the absolute phases of the carrier can be successfully be identified using a unique word assigned to the sync region 496 shown in FIG. 41.

For transmission of a modified 64 QAM signal such as shown in FIG. 40, signal points in the phase sync areas 471 at the (2n−1)π/4 phase denoted by the hatching are assigned to the sync time slots 452, 452b, etc. Its carrier can hardly be reproduced with a common 4 PSK receiver but can be successfully reproduced with the first receiver 23 of 4 PSK mode provided with the carrier reproducing circuit of the embodiment.

The foregoing carrier reproducing is of COSTAS type. A carrier reproducing circuit of the reverse modulation type will now be explained according to the embodiment.

FIG. 45 shows a reverse modulation type carrier reproducing circuit according to the present invention, in which a received signal is fed from the input unit 24 to a sync detector circuit 541 for producing a demodulated signal. Also, the input signal is delayed by a first delay circuit 591 to a delay signal. The delay signal is then transferred to a quadrature phase modulator circuit 592 where it is reverse demodulated by the demodulated signal from the sync detector circuit 541 to a carrier signal. The carrier signal is fed through a carrier reproduction controller circuit 544 to a phase comparator 593. A carrier wave produced by a VCO 547 is delayed by a second delay circuit 594 into a delay signal which is also fed to the phase comparator 593. At the phase comparator 593, the reverse demodulated carrier signal is compared in phase with the delay signal thus producing a phase difference signal. The phase difference signal is fed through a loop filter 546 to the VCO 547 which in turn produces a carrier wave arranged in phase with the received carrier wave. In the same manner as of the COSTAS carrier reproducing circuit shown in FIG. 43, an extracting timing controller circuit 543 performs sampling of signal points contained in the hatching areas of FIG. 39. Accordingly, the carrier wave of a 16 or 64 QAM signal can be reproduced with the 4 PSK demodulator of the first receiver 23.

The reproduction of a carrier wave by 16× frequency multiplication will be explained. The transmitter 1 shown in FIG. 1 is arranged to modulate and transmit a modified 16 QAM signal with assignment of its signal points at nπ/8 phase as shown in FIG. 46. At the first receiver 23 shown in FIG. 19, the carrier wave can be reproduced with its COSTAS carrier reproduction controller circuit containing a 16× multiplier circuit 661 shown in FIG. 48. The signal points at each nπ/8 phase shown in FIG. 46 are processed at the first quadrant b the action of the 16× multiplier circuit 661, whereby the carrier will be reproduced by the combination of a loop filter 546 and a VCO 541 547. Also, the absolute phase may be determined from 16 different phases by assigning a unique word to the sync region.

The arrangement of the 16× multiplier circuit will be explained referring to FIG. 48. A sum signal and a difference signal are produced from the demodulated signal by an adder circuit 662 and a subtractor circuit 663 respectively and then, multiplied together by a multiplier 664 into a cos 2θ signal. Also, a multiplier 665 produces a sin 2θ signal. The two signals are then multiplied by a multiplier 666 into a sin 4θ signal.

Similarly, a sin 8θ signal is produced from the two, sin 2θ and cos 2θ, signals by the combination of an adder circuit 667, a subtracter circuit 668, and a multiplier 670. Furthermore, a sin 16θ signal is produced by the combination of an adder circuit 671, a subtractor circuit 672, and a multiplier 673. Then, the 16× multiplication is completed.

Through the foregoing 16× multiplication, the carrier wave of all the signal points of the modified 16 QAM signal shown in FIG. 46 will successfully be reproduced without extracting particular signal points.

However, reproduction of the carrier wave of the modified 64 QAM signal shown in FIG. 47 can involve an increase in the error rate due to dislocation of some signal points from the sync areas 471.

Two techniques are known for compensation for the consequences. One is inhibiting transmission of the signal points dislocated from the sync areas. This causes the total amount of transmitted data to be reduced but allows the arrangement to be facilitated. The other is providing the sync time slots as described in FIG. 38. In more particular, the signal points in the nπ/8 sync phase areas, e.g. 471 and 471a, are transmitted during the period of the corresponding sync time slots in the time slot group 451. This triggers an accurate synchronizing action during the period thus minimizing phase error.

As now understood, the 16× multiplication allows the simple 4 PSK receiver to reproduce the carrier wave of a modified 16 or 64 QAM signal. Also, the insertion of the sync time slots causes the phasic accuracy to be increased during the reproduction of carrier waves from a modified 64 QAM signal.

As set forth above, the signal transmission system of the present invention is capable of transmitting a plurality of data on a single carrier wave simultaneously in the multiple signal level arrangement.

More specifically, three different level receivers which have discrete characteristics of signal intercepting sensitivity and demodulating capability are provided in relation to one single transmitter so that any one of them can be selected depending on a wanted data size to be demodulated which is proportional to the price. When the first receiver of low resolution quality and low price is acquired together with a small antenna, its owner can intercept and reproduce the first data stream of a transmission signal. When the second receiver of medium resolution quality and medium price is acquired together with a medium antenna, its owner can intercept and reproduce both the first and second data streams of the signal. When the third receiver of high resolution quality and high price is acquired with a large antenna, its owner can intercept and reproduce all the first, second, and third data streams of the signal.

If the first receiver is a home-use digital satellite broadcast receiver of low price, it will overwhelmingly be welcome by a majority of viewers. The second receiver accompanied with the medium antenna costs more and will be accepted by not common viewers but particular people who want to enjoy HDTV services. The third receiver accompanied with the large antenna at least before the satellite output is increased, is not appropriate for home use and will possibly be used in relevant industries. For example, the third data stream carrying super HDTV signals is transmitted via a satellite to subscriber cinemas which can thus play video tapes rather than traditional movie films and run movies at low cost.

When the present invention is applied to a TV signal transmission service, three different quality pictures are carried on one signal channel wave and will offer compatibility with each other. Although the first embodiment refers to a 4 PSK, a modified 8 QAM, a modified 16 QAM, and a modified 64 QAM signal, other signals will also be employed with equal success including a 32 QAM, a 256 QAM, an 8 PSK, and a 16 PSK, and a 32 PSK signal. It would be understood that the present invention is not limited to a satellite transmission system and can be applied to a terrestrial communications system or a cable transmission system.

The transmission method of the invention can also be applied to a 4-level or 8-level ASK signal as shown in FIG. 58 and FIGS. 68(a) and (b), respectively.

EMBODIMENT 2

A second embodiment of the present invention is featured in which the physical multi-level arrangement of the first embodiment is divided into small levels through e.g. discrimination in error correction capability, thus forming a logic multi-level construction. In the first embodiment, each multi-level channel has different levels in the electrical electric signal amplitude or physical demodulating capability. The second embodiment offers different levels in the logic reproduction capability such as error correction. For example, the data D1 in a multi-level channel is divided into two, D1-1 and D1-2, components and D1-1 is more increased in the error correction capability than D1-2 for discrimination. Accordingly, as the error detection and correction capability is different between D1-1 and D1-2 at demodulation, D1-1 can successfully be reproduced within a given error rate when the C/N level of an original transmitting signal is as low as disenabling the reproduction of D1-2. This will be implemented using the logic multi-level arrangement.

More specifically, the logic multi-level arrangement consists of dividing data of a modulated multi-level channel and discriminating distances between error correction codes by mixing error correction codes with product codes for varying error correction capability. Hence, a more multi-level signal can be transmitted.

In fact, a D1-1 channel is divided into two sub channels D1 and D1-2 and a D2 channel is divided into two sub channels D2-1 and D2-2.

This will be explained in more detail referring to FIG. 87 85 in which D1-1 is reproduced from a lowest C/N signal. If the C/N rate is d at minimum, three components D1-2, D2-1 and D2-2 cannot be reproduced while D1-1 is reproduced. If C/N is not less than c, D1-2 can also be reproduced. Equally, when C/N is b, D2-1 is reproduced and when C/N is a, D2-2 is reproduced. As the C/N rate increases, the reproducible signal levels are increased in number. The lower the C/N, the fewer the reproducible signal levels. This will be explained in the form of relationship between transmitting distance and reproducible C/N value referring to FIG. 86. In common, the C/N value of a received signal is decreased in proportion to the distance of transmission as expressed by the real line 861 in FIG. 86. It is now assumed that the distance from a transmitter antenna to a receiver antenna is La when C/N=a, Lb when C/N=b, Lc when C/N=c, Ld when C/N=d, and Le when C/N=e. If the distance from the transmitter antenna is greater than Ld, D1-1 can be reproduced as shown in FIG. 85 where the receivable area 862 is denoted by the hatching. In other words, D1-1 can be reproduced within a most extended area. Similarly, D1-2 can be reproduced in an area 863 when the distance is not more than Lc. In this area 863 containing the area 862, D1-1 can with no doubt be reproduced. In a small area 854 864, D2-1 can be reproduced and in a smallest area 865, D2-2 can be reproduced. As understood, the different data levels of a channel can be reproduced corresponding to degrees of declination in the C/N rate. The logic multi-level arrangement of the signal transmission system of the present invention can provide the same effect as of a traditional analogue transmission system in which the amount of receivable data is gradually lowered as the C/N rate decreases.

The construction of the logic multi-level arrangement will be described in which there are provided two physical levels and two logic levels. FIG. 87 is a block diagram of a transmitter 1 which is substantially identical in construction to that shown in FIG. 2 and described previously in the first embodiment and will not be further explained in detail. The only difference is that error correction code encoders are added as abbreviated to ECC encoders. The divider circuit 3 has four outputs 1-1, 1-2, 2-1, and 2-2 through which four signals D1-1, D1-2, D2-1, and D2-2 divided from an input signal are delivered. The two signals D1-1 and D1-2 are fed to two, main and sub, ECC encoders 872a and 873a of a first ECC encoder 871a respectively for converting to error correction code forms.

The main ECC encoder 872a has a higher error correction capability than that of the sub ECC encoder 873a. Hence, D1-1 can be reproduced at a lower rate of C/N than D1-2 as apparent from the CN-level diagram of FIG. 85. More particularly, the logic level of D1-1 is less affected by declination of the C/N than that of D1-2. After error correction code encoding, D1-1 and D2-2 D1-2 are summed by a summer 874a to a D1 signal which is then transferred to the modulator 4. The other two signals D2-1 and D2-2 of the divider circuit 3 are error correction encoded by two, main and sub, ECC encoders 872 b 872b and 873b of a second ECC encoder 871b respectively and then, summed by a summer 874b to a D2 signal which is transmitted to the modulator 4. The main ECC encoder 872b is higher in the error correction capability than the sub ECC encoder 873b. The modulator 4 in turn produces from the two, D1 and D2, input signals a multi-level modulated signal which is further transmitted from the transmitter unit 5. As understood, the output signal from the transmitter 1 has two physical levels D1 and D2 and also, four logic levels D1-1, D1-2, D2-1, and D2-2 based on the two physical levels for providing different error correction capabilities.

The reception of such a multi-level signal will be expelained explained. FIG. 88 is a block diagram of a second receiver 33 which is almost identical in construction to that shown in FIG. 21 and described in the first embodiment. The second receiver 33 arranged for intercepting multi-level signals from the transmitter 1 shown in FIG. 87 further comprises first and second ECC decoder 876a 876b, in which the demodulation of QAM, or any of ASK, PSK, and FSK if desired, is executed.

As shown in FIG. 88, a receiver signal is demodulated by the demodulator 35 to the two, D1 and D2, signals which are then fed to two dividers 3a and 3b respectively where they are divided into four logic levels D1-1, D1-2, D2-1, and D2-2. The four signals are transferred to the first and second ECC decoders 876a and 876b in which D1-1 is error corrected by a main ECC decoder 877a, D1-2 by a sub ECC decoder 878a, D2-1 by a main ECC decoder 877b, D2-2 by a sub ECC decoder 878b before all being sent to the summer 37. In the mixer 37, the four, D1-1, D1-2, D2-1, and D2-2, error corrected signals are combined into a signal which is then delivered from the output unit 36.

Since D1-1 and D2-2 are higher in the error correction capability than D1-2 and D2-2 respectively, the error rate remains less than a given value although C/N is fairly low as shown in FIG. 85 and thus, an original signal will be reproduced successfully.

The action of discriminating the error correction capability between the main ECC decoders 877a and 877b of high code gain and the sub ECC decoders 878a and 878b of low code gain will now be described in more detail. It is a good idea for having a difference in the error correction capability, i.e., in the code gain, to use in the sub ECC decoder a common coding technique, e.g. Reed-Solomon or BCH method, as shown in FIG. 165(b) for the ECC decoder, having a standard code distance and in %he the main ECC decoder, another encoding technique in which distance between correction codes is increased using Reed-Solomon codes, their product codes, or other long-length codes or a trellis decoder 744p, 744q, and 744r shown in FIGS. 128(d), 128(e), 128(f). A variety of known techniques for increasing the error correction code distance have been introduced and will not be explained in detail. The present invention can be associated with any known technique for having the logic multi-level arrangement.

Also, as shown in the block diagram of FIGS. 160 and 167, the transmitter further has an interleaver 744k and the receiver further has de-interleavers 759k and 936b. The interleave process is carried out by the use of the Interleave Table 954 shown in FIG. 168(a). De-interleave RAM 936x in the de-interleaver 936b is used for decoding the data. By this arrangement, the data transmission system having high reliability with respect to the burst error can be realized, resulting in stable transmitted images.

The logic multi-level arrangement will be explained in conjuction conjunction with a diagram of FIG. 89 showing the relationship between C/N and error race rate after error correction. As shown, the straight line 881 represents D1-1 at the C/N and error rate relation and the line 882 represents D1-2 at same.

As the C/N rate of an input signal decreases, the error rate increases after error correction. If C/N is lower than a given value, the error rate exceeds a reference value Eth determined by the system design standards and no original data will normally be reconstructed. When C/N is lowered to less than e, the D1 signal fails to be reproduced as expressed by the line 881 of D1-1 in FIG. 89. When e≦C/N<d, D1-1 of the D1 signal exhibits a higher error rate than Eth and will not be reproduced.

When C/N is d at the point 885d, D1-1 having a higher error correction capability than D1-2 becomes not higher in the error rate than Eth and can be reproduced. At the time, the error rate of D1-2 remains higher than Eth after error correction and will no longer be reproduced.

When C/N is increased up to c at the point 885c, D1-2 becomes not higher in the error rate than Eth and can be reproduced. At the time, D2-1 and D2-2 remain in no demodulation state. After the C/N rate is increased further to b′, the D2 signal becomes ready to be demodulated.

When C/N is increased to b at the point 885b, D2-1 of the D2 signal becomes not higher in the error rate than Eth and can be reproduced. At the time, the error rate of D2-2 remains higher than Eth and will not be reproduced. When C/N is increased up to a at the point 885a, D2-2 becomes not higher than Eth and can be reproduced.

As described above, the four different signal logic levels divided from two, D1 and D2, physical levels through discrimination of the error correction capability between the levels, can be transmitted simultaneously.

Using the logic multi-level arrangement of the present invention with a multi-level construction in which at least a part of the original signal is reproduced even if data in a higher level is lost, digital signal transmission will successfully be executed without losing the advantageous effect of an analogue signal transmission in which transmitting data is gradually decreased as the C/N rate becomes low.

ThankingThanks to up-to-date compression techniques, compressed image data can be transmitted in the logic multi-level arrangement for enabling a receiver station to reproduce a higher quality image than that of an analogue system and also, with not sharply but at steps declining the signal level for ensuring signal interception in a wider area. The present invention can provide an extra effect of the multi-layer arrangement which is hardly implemented by a known digital signal transmission system without deteriorating high quality image data.

In addition, the address data of the image segment data, the base image data for image compression, the scramble cancellation data shown in the descrambler (FIG. 66), and high priority (HP) data, i.e., the data (e.g., the frame synchronization signal and header) that is most essential to image expansion of the HDTV signal, is transmitted as D1-1 by the high code gain ECC encoder 743a (FIGS. 88, 133, 170, and 172), and is received by the high code gain ECC decoder 758 of the receiver 43.

This high priority data is protected because the error rate of priority data D1-1 does not increase noticeably. Fatal deterioration of the characteristic image quality of digital video transmissions is thus avoided, and a “graceful degradation” effect whereby image quality gradually deteriorates is obtained. The modulator 749 and demodulator 760 of FIGS. 133 and 170, respectively, can achieve this graceful degradation effect with 16-level QAM and 32-level QAM described above, 4-level VSB (FIG. 57) and 8-level VSB (FIG. 68) described below in the description of the fourth embodiment, and 8-level PSK.

Furthermore, as shown in the block diagrams of FIGS. 133 and 156, a big difference in the error rate of high priority data and low priority data can be created during signal reception by applying high code gain error correction coding of the high priority data by means of the ECC encoder 744a and trellis encoder 744b in the 2nd data stream input 744, while error correction encoding the low priority data with low code gain by the ECC encoder 743a only.

As a result, even if the C/N ratio of the transmission system deteriorates significantly, the high priority data can be received. Therefore, while the image quality deteriorates with the deterioration of the low priority data, the high priority data can also be reproduced in applications subject to severe C/N ratio deterioration, as found in the reception conditions encountered with mobile television receivers, and the pixel block positioning information is also reproduced. Because image block destruction is thus prevented, viewers are still able to receive and view broadcast programing under extremely poor reception conditions.

EMBODIMENT 3

A third embodiment of the present invention will be described referring to the relevant drawings.

FIG. 29 is a schematic total view illustrating the third embodiment in the form of a digital TV broadcasting system. An input video signal 402 of super high resolution TV image is fed to an input unit 403 of a first video encoder 401. Then, the signal is divided by a divider circuit 404 into three, first, second, and third, data streams which are transmitted to a compressing circuit 405 for data compression before being further delivered.

Equally, other three input video signals 406, 407, and 408 are fed to a second 409, a third 410, and a fourth video encoder 411 respectively which all are arranged identical in construction to the first video encoder 401 for data compression.

The four first data streams from their respective encoders 401, 409, 410, and 411 are transferred to a first multiplexer 413 of a multiplexer 412 where they are time multiplexed by a TDM process into a first data stream multiplex signal which is fed to a transmitter 1.

A part or all of the four second data streams from their respective encoders 401, 409, 410, and 411 are transferred to a second multiplexer 414 of the multiplexer 412 where they are time multiplexed to a second data stream multiplex signal which is then fed to transmitter 1. Also, a part or all of the four third data streams are transferred to a third multiplexer 415 where they are time multiplexed to a data stream multiplex signal which is then fed to the transmitter 1.

The transmitter 1 performs modulation of the three data stream signals with its modulator 4 by the same manner as described in the first embodiment. The modulated signals are sent from a transmitter unit 5 through an antenna 6 and an uplink 7 to a transponder 12 of a satellite 10 which in turn transmits it to three different receivers including a first receiver 23.

The modulated signal transmitted through a downlink 21 is intercepted by a small antenna 22 having a radius r1 and fed to a first data stream reproducing unit 232 of the first receiver 23 where its first data stream only is demodulated. The demodulated first data stream is then converted by a first video decoder 421 to a traditional 425 or wide-picture NTSC or video output signal 426 of low image resolution.

Also, the modulated signal transmitted through a downlink 31 is intercepted by a medium antenna 32 having a radius r2 and fed to a first 232 and a second data stream reproducing unit 233 of a second receiver 33 where its first and second data streams are demodulated respectively. The demodulated first and second data streams are then summed and converted by a second video decoder 422 to an HDTV or video output signal 427 of high image resolution and/or to the video output signals 425 and 426.

Also, the modulated signal transmitted through a downlink 41 is intercepted by a large antenna 42 having a radius r3 and fed to a first 232, a second 233, and a third data steam reproducing unit 234 of a third receiver 43 where its first, second, and third data streams are demodulated respectively. The demodulated first, second, and third data streams are then summed and converted by a third video decoder 423 to a super HDTV or video output signal 428 of super high image resolution for use in a video theater or cinema. The video output signals 425, 426, and 427 can also be reproduced if desired. A common digital TV signal is transmitted from a conventional digital transmitter 51 and when intercepted by the first-receiver 23, will be converted to the video output signal 426 such as a low resolution NTSC TV signal.

The first video encoder 401 will now be explained in more detail referring to the block diagram of FIG. 30. An input video signal of super high resolution is fed through the input unit 403 to the divider circuit 404 where it is divided into four components by sub-band coding process. In particular, the input video signal is separated by passing through a horizontal lowpass filter 451 and a horizontal highpass filter 452 of e.g. QAM mode to two, low and high, horizontal frequency components which are then subsampled into half of their quantities by two subsamplers 453 and 454 respectively. The low horizontal component is filtered by a vertical lowpass filter 455 and a vertical highpass filter 456 into a low horizontal low vertical component or HLVL signal and a low horizontal high vertical component or HLVH signal respectively. The two, HLVL and HLVH, signals are then subsampled into one half by two subsamblers 457 and 458 respectively and transferred to the compressing circuit 405.

The high horizontal component is filtered by a vertical lowpass filter 459 and a vertical highpass filter 460 into a high horizontal low vertical highpass component or HHVL signal and a high horizontal high vertical component or HHVH signal respectively. The two, HHVL and HHVH, signals are then subsampled into one half by two subsamplers 461 and 462 respectively and transferred to the compressing circuit 405.

The HLVL signal is preferably DCT compressed by a first compressor 471 of the compressing circuit 405 and fed to a first output circuit 472 as the first data stream.

Also, the HLVH signal is compressed by a second compressor 473 and fed to a second output circuit 464. The HHVL signal is compressed by a third compressor 463 and fed to the second output circuit 464.

The HHVH signal is divided by a divider 465 into two high resolution (HHVH 1) and super high resolution (HHVH 2) video signals which are then transferred to the second output circuit 464 and a third output circuit 468 respectively.

The first video decoder 421 will now be explained in more detail referring to FIG. 31. The first data stream or D1 signal of the first receiver 23 is fed through an input unit 501 to a descrambler 502 of the first video decoder 421 where it is descrambled. The descrambled D1 signal is expanded by an expander 503 to HLVL which is then fed to an aspect ratio changing circuit 504. Thus, the HLVL signal can be delivered through an output unit 505 as a standard 500, letterbox format 507, wide-screen 508, or sidepanel format NTSC signal 509. The scanning format may be of non-interlace or interlace type and its NTSC mode lines may be 525 or doubled to 1050 by double tracing. When the received signal from the digital transmitter 51 is a digital TV signal of 4 PSK mode, it can also be converted by the first receiver 23 and the first video decoder 421 to a TV picture. The second video decoder 422 will be explained in more detail referring to the block diagram of FIG. 32. The D1 signal of the second receiver 33 is fed through a first input 521 to a first expander 522 for data expansion and then, transferred to an oversampler 523 where it is sampled at 2×. The oversampled signal is filtered by a vertical lowpass filter 524 into HLVL. Also, the D2 signal of the second receiver 33 is fed through a second input 530 to a divider 531 where it is divided into three components which are then transferred to second, third, and fourth expanders 532-534 respectively for data expansion. The three expanded components are sampled at 2× by three oversamplers 535, 536, and 537 and filtered by a vertical highpass 538, a vertical lowpass 539, and a vertical highpass filter 540 respectively. Then, HLVL from the vertical lowpass filter 524 and HLVH from the vertical highpass filter 538 are summed by an adder 525, sampled by an oversampler 541, and filtered by a horizontal lowpass filter 542 into a low frequency horizontal video signal. HHVL from the vertical lowpass filter 539 and HHVH 1 from the vertical highpass filter 540 are summed by an adder 526, sampled by an oversampler 544, and filtered by a horizontal highpass filter 545 to a high frequency horizontal video signal. The two, high and low frequency, horizontal video signals are then summed by an adder 543 into a high resolution video signal HD which is further transmitted through an output unit 546 as a video output 547 of e.g. HDTV format. If desired a traditional NTSC video output can be reconstructed with equal success.

FIG. 33 is a block diagram of the third video decoder 423 in which the D1 and D2 signals are fed through a first 521 and a second input 530 respectively to a high frequency band video decoder circuit 527 where they are converted to an HD signal in the same manner as described above. The D3 signal is fed through a third input 551 to a super high frequency band video decoder circuit 552 where it is expanded, descrambled, and composed into HHVH 2 signal. The HD signal of the high frequency band video decoder circuit 527 and the HHVH 2 signal of the super high frequency band video decoder circuit 552 are summed by a summer 553 to a super high resolution TV or S-HD signal which is then delivered through an output unit 554 as a super resolution video output 555.

The action of multiplexing in the multiplexer 412 shown in FIG. 29 will be explained in more detail. FIG. 34 illustrates a data assignment in which the three, first, second, and third, data streams D1, D2, D3 contain in a period of T six NTSC channel data L1, L2, L3, L4, L5, L6, six HDTV channel data M1, M2, M3, M4, M5, M6 and six S-HDTV channel data H1, H2, H3, H4, H5, H6 respectively. In operation, the NTSC or D1 signal data L1 to L6 are time multiplexed by TDM process during the period T. More particularly, HLVL of D1 is assigned to a domain 601 for the first channel. Then, a difference data M1 between HDTV and NTSC or a sum of HLVH, HHVL, and HHVH 1 is assigned to a domain 602 for the first channel. Also, a difference data HI H1 between HDTV and super HDTV or HHVH 2 (See FIG. 30) is assigned to a domain 603 for the first channel.

The selection of the first channel TV signal will now be described. When intercepted by the first receiver 23 with a small antenna coupled to the first video decoder 421, the first channel signal is converted to a standard or widescreen NTSC TV signal as shown in FIG. 31. When intercepted by the second receiver 33 with a medium antenna coupled to the second video decoder 422, the signal is converted by summing L1 of the first data stream D1 assigned into the domain 601 and M1 of the second data stream D2 assigned to the domain 602 to an HDTV signal of the first channel equivalent in program to the NTSC signal.

When intercepted by the third receiver 43 with a large antenna coupled to the third video decoder 423, the signal is converted by summing L1 of D1 assigned to the domain 601, M1 of D2 assigned to the domain 602, and H1 of D3 assigned to the domain 603 into a super HDTV signal of the first channel equivalent in program to the NTSC signal. The other channel signals can be reproduced in an equal manner.

FIG. 35 shows another data assignment L1 of a first channel NTSC signal is assigned to a fistfirst domain 601. The domain 601 which is allocated at the front end of the first data stream D1, also contains at front a data S11 including a descrambling data and the demodulation data described in the first embodiment. A first channel HDTV signal is transmitted as L1 and M1. M1, which is thus a difference data between NTSC and HDTV, is assigned to two domains 602 and 611 of D2. If L1 is a compressed NTSC component of 6 Mbps, M1 is two times higher, that is, 12 Mbps. Hence, the total of L1 and M1 can be demodulated at 18 Mbps with the second receiver 33 and the second video decoder 423. According to current data compression techniques, HDTV compressed signals can be reproduced at about 15 Mbps. This allows the data assignment shown in FIG. 35 to enable simultaneous reproduction of an NTSC and HDTV first channel signal. However, this assignment allows no second channel HDTV signal to be carried. S21 is a descrambling data in the HDTV signal. A first channel super HDTV signal component comprises L1, M1, and H1. The difference data H1 is assigned to three domains 603, 612, and 613 of D3. If the NTSC signal is 6 Mbps, the super HDTV is as high as 36 Mbps. When a compressed rate is increased, super HDTV video data of about 2000 scanning line for reproduction of a cinema size picture for commercial use can be transmitted in an equal manner.

FIG. 36 shows a further data assignment in which H1 of a super HDTV signal is assigned to six time domains. If a NTSC compressed signal is 6 Mbps, this assignment can be nine times higher, that is, 54 Mbps of D3 data. Accordingly, super HDTV data of higher picture quality can be transmitted.

The foregoing data assignment makes the use of one of two, horizontal and vertical, polarization planes of a transmission wave. When both the horizontal and vertical polarization planes are used, the frequency utilization will be doubled. This will be explained below.

FIG. 49 shows a data assignment in which DV1 and DH1 are a vertical and a horizontal polarization signal of the first data stream respectively, DV2 and DH2 are a vertical and a horizontal polarization signal of the second data stream respectively, and DV3 and DH3 are a vertical and a horizontal polarization signal of the third data stream respectively. The vertical polarization signal DV1 of the first data stream carries a low frequency band or NTSC TV data and the horizontal polarization signal DH1 carries a high frequency band or HDTV data. When the first receiver 23 is equipped with a vertical polarization signal DH1 carries a high frequency band or HDTV data. When the first receiver 23 is equipped with an antenna for both horizontally and vertically polarized waves, it can reproduce the HDTV signal through summing L1 and M1. More specifically, the first receiver 23 can provide compatibility between NTSC and HDTV with the use of a particular type antenna.

FIG. 50 illustrates a TDMA method in which each data burst 721 is accompanied at front a sync data 731 and a card data 741. Also, a frame sync data 720 is provided at the front of a frame. Like channels are assigned to like time slots. For example, a first time slot 750 carries NTSC, HDTV, and super HDTV data of the first channel simultaneously. The six time slots 750, 750a, 750b, 750c, 750d, 750e, are arranged independent from each other. Hence, each station can offer NTSC, HDTV, and/or super HDTV services independently of the other stations through selecting a particular channel of the time slots. Also, the first receiver 23 can reproduce an NTSC signal when equipped with a horizontal polarization antenna and both NTSC and HDTV signals when equipped with a compatible polarization antenna. In this respect, the second receiver 33 can reproduce a super HDTV at lower resolution while the third receiver 43 can reproduce a full super HDTV signal. According to the third embodiment, a compatible signal transmission system will be constructed. It is understood that the data assignment is not limited to the burst mode TDMA method shown in FIG. 50 and another method such as time division multiplexing of continuous signals as shown in FIG. 49 will be employed with equal success. Also, a data assignment shown in FIG. 51 will permit a HDTV signal to be reproduced at high resolution.

As set forth above, the compatible digital TV signal transmission system of the third embodiment can offer three, super HDTV, HDTV, and conventional NTSC, TV broadcast services simultaneously. In addition, a video signal intercepted by a commercial station or cinema can be electronized.

The modified QAM of the embodiments is now termed as SRQAM and its error rate will be examined.

First, the error rate in 16 SRQAM will be calculated. FIG. 99 shows a vector diagram of 16 SRQAM signal points. As apparent from the first quadrant, the 16 signal points of standard 16 QAM including 83a, 83b 85, 84a, 83a 86a are allocated at equal intervals of 2δ.

The signal point 83a is spaced δ from both the I-axis and the Q-axis of the coordinate. It is now assumed that n is a shift value of the 16 SRQAM. In 16 SRQAM, the signal point 83a of 16 QAM is shifted to a signal point 83 where the distance from each axis is nδ. The shift value n is thus expressed as:
0<n<3.

The other signal points 84a and 86a are also shifted to two points 84 and 86 respectively.

If the error rate of the first data stream is Pe1, it is obtained from: Pe 1 - 16 = 1 4 erfc ( n δ 2 σ ) + 1 4 erfc ( 3 δ 2 σ ) = 1 8 erfc ( n p 9 + n 2 )
Also, the error rate Pe2 of the second data stream is obtained from: Pe 2 - 16 = 1 2 erfc ( 3 - n 2 δ 2 σ ) = 1 4 erfc ( 3 - n 2 δ 2 9 + n 2 p )

The error rate of 36 or 32 SRQAM will be calculated. FIG. 100 is a vector diagram of a 36 SRQAM signal in which the distance between any two 36 QAM signal points is 2δ.

The signal point 83a of 36 QAM is spaced δ from each axis of the coordinate. It is now assumed that n is a shift value of the 16 SRQAM. In 36 SRQAM, the signal point 83a is shifted to a signal point 83 where the distance from each axis is nδ. Similarly, the nine 36 QAM signal points in the first quadrant are shifted to points 83, 84, 85, 86, 97, 98, 99, 100, 101 respectively. If a signal point group 90 comprising the nine signal points is regarded as a single signal point, the error rate Pe1 in reproduction of only the first data stream D1 with a modified 4 PSK receiver and the error rate Pe2 in reproduction of the second data stream D2 after discriminating the nine signal points of the group 90 from each other, are obtained respectively from: Pe 1 - 32 = 1 6 erfc ( n δ 2 σ ) = 1 6 erfc ( 6 p 5 × n n 2 + 2 n + 25 ) Pe 2 - 32 = 2 3 erfc ( 5 - n 4 22 δ p ) = 2 3 erfc ( 3 p 40 × 5 - n n 2 + 2 n + 25 )

FIG. 101 shows the relationship between error rate Pe and C/N rate in transmission in which the curve 900 represents a conventional or not modified 32 QAM signal. The straight line 905 represents a signal having 10−1.5 of the error rate. The curve 901a represents a D1 level 32 SRQAM signal of the present invention at the shift rate n of 1.5. As shown, the C/N rate of the 32 SRQAM signal is 5 dB lower at the error rate of 10−1.5 than that of the conventional 32 QAM. This means that the present invention allows a D1 signal to be reproduced at a given error rate when its C/N rate is relatively low.

The curve 902a represents a D2 level SRQAM signal at n=1.5 which can be reproduced at the error rate of 10−1.5 only when its C/N rate is 2.5 dB higher than that of the conventional 32 QAM of the curve 900. Also, the curves 901b and 902b represent D1 and D2 SRQAM signals at n=2.0 respectively. The curves curve 902c represents a D2 SRQAM signal at n=2.5. It is apparent that the C/N rate of the SRQAM signal at the error data of 10−1.5 is 5 dB, 8 dB, and 10 dB higher at n=1.5, 2.0, and 2.5 respectively in the D1 level and 2.5 dB lower in the D2 level than that of a common 32 QAM signal.

Shown in FIG. 103 is the C/N rate of the first and second data streams D1, D2 of a 32 SRQAM signal which is needed for maintaining a constant error rate against variation of the shift n. As apparent, when the shift n is more than 0.8, there is developed a clear difference between two C/N rates of their respective D1 and D2 levels so that the multi-level signal, namely first and second data, transmission can be implemented successfully. In brief, n>0.85 is essential for multi-level data transmission of the 32 SRQAM signal of the present invention.

FIG. 102 shows the relationship between the C/N rate and the error rate for 16 SRQAM signals. The curve 900 represents a common 16 QAM signal. The curves 901a, 901b, 901c and D1 level or first data stream 16 SRQAM signals at n=1.2, 1.5, and 1.8 respectively. The curves 902a, 902b, 902c are D2 level or second data stream 16 SRQAM signals at n=1.2, 1.5, and 1.8 respectively.

The C/N rate of the first and second data streams D1, D2 of a 16 SRQAM signal is shown in FIG. 104, which is needed for maintaining a constant error rate against variation of the shift n. As apparent, when the shift n is more than 0.9 (n>0.9), the multi-level data transmission of the 16 SRQAM signal will be executed.

One example of propagation of SRQAM signals of the present invention will now be described for use with a digital TV terrestrial broadcast service. FIG. 105 shows the relationship between the signal level and the distance between a transmitter antenna and a receiver antenna in the terrestrial broadcast service. The curve 911 represents a transmitted signal from the transmitter antenna which is 1250 feet high. It is assumed that the error rate essential for reproduction of an applicable digital TV signal is 10−1.5. The hatching area 912 represents a noise interruption. The point 910 represents a signal reception limit of a conventional 32 QAM signal at C/N=15 dB where the distance L is 60 miles and a digital HDTV signal can be intercepted at minimum.

The C/N rate varies 5 dB under a worst case receiving condition such as bad weather. If a change in the relevant condition, e.g. weather, attenuates the C/N rate, the interception of an HDTV signal will hardly be ensured. Also, geographical conditions largely affect the propagation of signals and a decrease of about 10 dB at least will be unavoidable. Hence, successful signal interception within 60 miles will never be guaranteed and above all, a digital signal will be harder to propagate than an analogue signal. It would be understood that the service area of a conventional digital TV broadcast service is less dependable.

In case of the 32 SRQAM signal of the present invention or the 8-VSB shown in FIG. 68, a three-level signal transmission system is constituted as shown in FIGS. 133 and 137. This permits a low resolution NTSC signal of MPEG level to be carried on the 1-1 data stream D1-1, a medium resolution TV data of e.g. NTSC system to be carried on the 1-2 data stream D1-2, and a high frequency component of HDTV data to be carried on the second data stream D2. Accordingly, the service area of the 1-2 data stream of the SRQAM signal is increased to a 70 mile point 910a while that of the second data stream remains within a 55 mile point 910b, as shown in FIG. 105. FIG. 106 illustrates a computer simulation result of the service area of the 32 SRQAM signal of the present invention, which is similar to FIG. 53 but explains it in more detail. As shown, the regions 708, 703c, 703a, 703b, and 712 represent a conventional 32 QAM receivable area, a 1-1 data level D1-1 receivable area, a 1-2 data level D1-2 receivable area, a second data level D2 receivable area, and a service area of a neighbor analogue TV station respectively. The conventional 32 QAM signal data used in this drawing is based on a conventionally disclosed one.

For common 32 QAM signal, the 60-mile-radius service area can be established theoretically. The signal level will however be attenuated by geographical or weather conditions and particularly, considerably declined at near the limit of the service area.

If the low frequency band TV component of MPEG1 grade is carried on the 1-1 level D1-1 data and the medium frequency band TV component of NTSC grade on the 1-2 level D1-2 data and high frequency band TV component of HDTV on the second level D2 data, the service area of the 32 SRQAM signal of the present invention is increased by 10 miles in radius for reception of an EDTV signal of medium resolution grade and 18 miles for reception of an LDTV signal of low resolution grade although decreased by 5 miles for reception of an HDTV signal of high resolution grade, as shown in FIG. 106. FIG. 107 shows a service area in case of a shift factor n or s=1.8. FIG. 135 shows the service area of FIG. 107 in terms of area.

More particularly, the medium resolution component of a digital TV broadcast signal of the SRQAM mode of the present invention can successfully be