US9270311B2 - Methods and systems for calibrating an analog filter - Google Patents

Methods and systems for calibrating an analog filter Download PDF

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US9270311B2
US9270311B2 US14/560,827 US201414560827A US9270311B2 US 9270311 B2 US9270311 B2 US 9270311B2 US 201414560827 A US201414560827 A US 201414560827A US 9270311 B2 US9270311 B2 US 9270311B2
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code
capacitor
offset
low
pass filter
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US20150155898A1 (en
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Srinivas PINAGAPANY
Sergey Timofeev
Atul Salhotra
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Marvell International Ltd
Cavium International
Marvell Asia Pte Ltd
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Marvell World Trade Ltd
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H7/00—Multiple-port networks comprising only passive electrical elements as network components
    • H03H7/01—Frequency selective two-port networks
    • H03H7/0153—Electrical filters; Controlling thereof
    • H03H7/0161—Bandpass filters
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06—Receivers
    • H04B1/10—Means associated with receiver for limiting or suppressing noise or interference
    • H04B1/12—Neutralising, balancing, or compensation arrangements
    • H04B1/123—Neutralising, balancing, or compensation arrangements using adaptive balancing or compensation means
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H11/00—Networks using active elements
    • H03H11/02—Multiple-port networks
    • H03H11/04—Frequency selective two-port networks
    • H03H11/12—Frequency selective two-port networks using amplifiers with feedback
    • H03H11/1291—Current or voltage controlled filters
    • H04B17/0062—
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B17/00—Monitoring; Testing
    • H04B17/0082—Monitoring; Testing using service channels; using auxiliary channels
    • H04B17/0085—Monitoring; Testing using service channels; using auxiliary channels using test signal generators
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B17/00—Monitoring; Testing
    • H04B17/20—Monitoring; Testing of receivers
    • H04B17/21—Monitoring; Testing of receivers for calibration; for correcting measurements
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H2210/00—Indexing scheme relating to details of tunable filters
    • H03H2210/02—Variable filter component
    • H03H2210/025—Capacitor
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
    • H03H2210/00—Indexing scheme relating to details of tunable filters
    • H03H2210/04—Filter calibration method
    • H03H2210/046—Master -slave
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
    • H04B1/40—Circuits

Definitions

  • Wireless communication devices such as cellular telephones, contain sophisticated integrated electronics used to receive and transmit wireless data.
  • the analog electronics of such integrated electronics is subject to process variation from one wafer to the next. This can result in characteristics of various components—e.g., resistor values and capacitor values—varying to the point that it may be impossible to use a particular device without some form of individualized device compensation.
  • the issue of component variation can even extend to devices within a single chip. Thus, even two identically-designed devices in a single chip can and do exhibit substantial mismatch. This problem tends to increase in severity as integrated circuit geometries continue to shrink.
  • a method for compensating for non-idealities in a filter circuit that includes programmable filter circuitry including a first low-pass filter and a second low-pass filter both having a common desired cutoff frequency f 0 is disclosed.
  • the method includes, for a first desired bandwidth BW 0 corresponding to the common desired cutoff frequency f 0 , injecting a reference tone f R and a cutoff tone f C into the first low-pass filter, and measuring respective filter responses of the reference tone f R and the cutoff tone f C while changing capacitor codes that control a cutoff frequency f 0-I of the first low-pass filter until a first capacitor code I CODE is determined that most accurately causes the first low-pass filter to utilize the desired cutoff frequency f 0 ; for the first desired bandwidth BW 0 , injecting the reference tone f R and the cutoff tone f C into the second low-pass filter, and measuring respective filter responses of the reference tone f R and the cutoff tone f C while changing capacitor codes that control
  • a device for compensating for non-idealities in a filter circuit that includes programmable filter circuitry including a first low-pass filter and a second low-pass filter both having a common desired cutoff frequency f 0 corresponding to a first desired bandwidth BW 0 is disclosed.
  • the device includes code search circuitry that controls the first low-pass filter and the second low-pass filter; tone generation circuitry that injects a reference tone f R and a cutoff tone f C into both the first low-pass filter and the second low-pass filter; measurement circuitry that: (1) measures respective filter responses of the reference tone f R and the cutoff tone f C while the code search circuitry changes capacitor codes that control a cutoff frequency f 0-I of the first low-pass filter until a first capacitor code I CODE is determined that most accurately causes the first low-pass filter to utilize the desired cutoff frequency f 0 ; and (2) measures respective filter responses of the reference tone f R and the cutoff tone f C while the code search circuitry changes capacitor codes that control a cutoff frequency f 0-Q of the second low-pass filter until a second capacitor code Q CODE is determined that most accurately causes the second low-pass filter to utilize the desired cutoff frequency f 0 ; and calibration circuitry configured to calibrate for mismatch between the first low-pass filter and the second low
  • FIG. 1 is a block diagram of an example wireless communications device capable of transmitting and receiving wireless signals.
  • FIG. 2 depicts a block diagram of the down-converter of FIG. 1 .
  • FIG. 3 depicts the wireless communications device of FIG. 1 reconfigured so as to be capable of self-calibration.
  • FIG. 4 is a power response of an example low-pass filter used in the wireless communications device of FIG. 1 .
  • FIG. 5 depicts examples of phase mismatch that can occur between to identically-designed low-pass filters as a function of capacitor codes.
  • FIGS. 6A and 6B depict examples of how mismatch for low-pass filters for a particular bandwidth becomes worse at higher bandwidths.
  • FIG. 7 is a flowchart outlining a set of example operations for providing compensating for mismatched low-pass filters.
  • OFDM Orthogonal Frequency Division Modulation
  • Analog low-pass filters may contain banks of capacitors that can be programmably placed in and out of circuit such that a cutoff frequency may be fine-tuned.
  • a analog low-pass filter for an OFDM communication system operating for a bandwidth of 20 MHz will require an 8.75 MHz cutoff frequency while an 18.75 MHz cutoff frequency will be needed for a 40 MHz bandwidth, and a 38.75 MHz cutoff frequency will be needed for an 80 MHz bandwidth.
  • FIG. 1 is a block diagram of an example wireless communications device 100 capable of transmitting and receiving wireless signals.
  • the wireless communications device 100 includes a receive antenna 102 , a down-converter 104 , a first (I Channel) Analog-To-Digital Converter (I-ADC) 112 , a second (Q Channel) Analog-To-Digital Converter (Q-ADC) 114 , a transmit antenna 122 , an up-converter 124 , a first (I Channel) Digital-To-Analog Converter (I-DAC) 132 , a second (Q Channel) Digital-To-Analog Converter (Q-DAC) 134 , and a processor 150 .
  • I-DAC Digital-To-Digital Converter
  • Q-DAC Digital-To-Analog Converter
  • FIG. 2 depicts a block diagram of the down-converter 104 of FIG. 1 .
  • the down-converter 104 includes a low-noise amplifier (LNA) 210 , a first mixer 220 , an I-baseband filter 230 , a second mixer 222 , a Q-baseband filter 232 , a local oscillator (LO) 240 capable of producing a local oscillation signal cos( ⁇ LO t), where ⁇ LO is the local oscillation frequency, and a phase shift device 242 capable of shifting the local oscillation signal cos( ⁇ LO t) by ⁇ /2 radians.
  • LNA low-noise amplifier
  • LO local oscillator
  • FIG. 3 depicts the wireless communications device 100 of FIG. 1 reconfigured so as to be capable of self-calibration. Also shown in FIG. 3 , functional components of the processor 150 dedicated to filter calibration are displayed. Such functional components include tone generation circuitry 152 , code search circuitry 154 , power/phase measurement circuitry 156 and calibration circuitry 158 . In various embodiments, the embedded circuitries 152 - 158 may individually be made from dedicated logic, may exists as software/firmware routines located in a tangible, non-transitory memory and operated upon by one or more processors, or exist as combinations of software/firmware processors and dedicated logic.
  • each of the I-baseband (low-pass) filter 230 and the Q-baseband (low-pass) filter 232 are calibrated such that each will, to a practical extent possible, have a common desired cutoff frequency f 0 corresponding to a first desired bandwidth BW 0 . While there is no limitation as to the particular bandwidths or cutoff frequencies that may be used, for the purposes of explanation the first desired bandwidth BW 0 is 20 MHz, and the corresponding desired cutoff frequency f 0 is 8.75 MHz.
  • the I-baseband filter 230 and Q-baseband filter 230 are both fifth-order Chebyshev Type-1 filters using switch-capacitor technology.
  • Initial calibration starts with the tone generation circuitry 152 (via the I-DAC 132 and the Q-DAC 134 ) injecting both a reference tone f R and a cutoff tone f C into each of the I-baseband filter 230 and the Q-baseband filter 232 .
  • the I-baseband filter 230 and the Q-baseband filter 232 provide a respective output response consistent with their respective non-ideal cutoff frequencies, f 0-I and f 0-Q , while the power/phase measurement circuitry 156 (via the I-ADC 112 and the Q-ADC 114 ) measures the respective filter responses.
  • the code search circuitry 154 will vary separate digital control codes (“capacitor codes” or “cap codes”) to the I-baseband filter 230 and the Q-baseband filter 232 until the respective non-ideal cutoff frequencies, f 0-I and f 0-Q , match the ideal cutoff frequency f 0 as close as possible given the available resolution of the capacitor codes.
  • the code search circuitry 154 can provide any number of search algorithms to provide capacitor codes within a range of [ ⁇ 128 to 127] until respective particular capacitor codes are selected that most accurately causes the baseband filters ⁇ 230 , 232 ⁇ to utilize the desired cutoff frequency f 0 .
  • These selected capacitor codes will be referred to below as the first capacitor code I CODE and the second capacitor code Q CODE .
  • FIG. 4 is a power response 400 of an example low-pass filter useable in the wireless communications device of FIG. 1 and useful to explain how the reference tone f R and the cutoff tone f C may be used to select an appropriate capacitor code and utilize an appropriate cutoff frequency.
  • the power response 400 is atypical of a fifth-order Type-1 Chebyshev filter.
  • the reference tone f R which is well within the pass-band region, is assigned a value of 1.25 MHz, and the cutoff tone f C is assigned a value of 10 MHz.
  • the power ratio of the responses for the reference tone f R and the cutoff tone f C will vary as a function of the cutoff frequency f 0 so as become larger as the cutoff frequency f 0 decreases, and become smaller as the cutoff frequency f C increases.
  • the power ratio for an ideal cutoff frequency f 0 of 8.75 MHz can be precisely determined, and a capacitor code can be adjusted until the power response 400 best reflects a known, predictable power ratio for the filter responses of the reference tone f R and the cutoff tone f C .
  • the calibration circuitry 158 performs further calculations so as to better calibrate the I-baseband filter 230 and the Q-baseband filter 232 to compensate for filter mismatch for one or more additional bandwidths greater than bandwidth BW 0 .
  • the one or more additional bandwidths will be a multiple of BW 0 .
  • a second desired bandwidth BW 1 will equal N ⁇ BW 0 , where N is a positive integer greater than 1.
  • the calibration circuitry 158 is configured to, for a respective second cutoff frequency f 1 for a second/higher bandwidth BW 1 , determine a capacitor code offsets ⁇ I OFFSET and ⁇ Q OFFSET commensurate with the frequency offset ⁇ f, add the capacitor code offset ⁇ I OFFSET to the first capacitor code I CODE to produce a first compensated capacitor code I C-CODE , and add the capacitor code offset ⁇ Q OFFSET to the second capacitor code Q CODE to produce a second compensated capacitor code Q C-CODE .
  • the capacitor code offsets must not just reflect the frequency offset ⁇ f, but must also take into consideration a “fractional capacitor code” CI FRAC corresponding to the first desired bandwidth BW 0 , the fractional capacitor code CI FRAC being a value that lies between two consecutive capacitor codes [I CODE , I CODE+1 ] on I rail, keeping Qcode unchanged, and that ideally corresponds to both a zero phase difference and a zero power difference between a first low-pass filter and a second low-pass filter.
  • FIG. 5 depicts a chart 500 showing examples of phase mismatch that can occur between two identically-designed low-pass filters as a function of capacitor codes and capacitor code offsets ⁇ I OFFSET / ⁇ Q OFFSET to be used for other bandwidths.
  • the X-axis is a dimension being a combined I-Q capacitor code [I CODE , Q CODE ], and the Y-axis is a second dimension representing respective measured phase offsets between a first low-pass filter and a second low-pass filter as a function of the respective combined I-Q capacitor codes.
  • the point 502 at which the dotted line displays zero phase mismatch occurs about half-way between I-Q capacitor code [71,6D] (signed hexadecimal notation representing a difference of 4) and I-Q capacitor code [70,6D] (signed hexadecimal notation representing a difference of 3).
  • the fractional capacitor code CI FRAC will be a real, non-integer, number, and as such is incompatible with programmable filter circuitry that relies on discrete switches to program/calibrate.
  • the capacitor code offset ⁇ I OFFSET / ⁇ Q OFFSET may be determined by rounding the fractional capacitor code CI FRAC to a nearest integer, adding the capacitor code offset ⁇ I OFFSET to the first capacitor code I CODE to produce the first compensated capacitor code I C-CODE , and adding the capacitor code offset ⁇ Q OFFSET to the second capacitor code Q CODE to produce the second compensated capacitor code Q C-CODE .
  • a scaling factor ⁇ may be determined in a number of ways, in a number of embodiments a scaling factor ⁇ is determined based on empirical data.
  • FIGS. 6 A and 6 B depict examples of how mismatch for low-pass filters for a particular bandwidth becomes worse at higher bandwidths. While FIGS. 6A and 6B are exemplary, conceptually they are based on real-world experience so as to demonstrate that filter mismatch will increase as a function of ⁇ fc and the magnitude of BW 1 . An appropriate scaling factor ⁇ will reflect desired compensation for different ⁇ fc and different magnitudes of BW 1 .
  • the processor 150 applies the first compensated capacitor code I C-CODE to the first/I-baseband (low-pass) filter 230 , and applies the second compensated capacitor code Q C-CODE to the second/Q-baseband (low-pass) filter 232 , where after the baseband filters 230 and 232 may be used for higher bandwidths.
  • FIG. 7 is a flowchart outlining a set of example operations for providing compensating for mismatched low-pass filters, such as the I-baseband filter 230 and Q-baseband filter 232 discussed above and with respect to FIGS. 1-6 .
  • Such operations compensate for non-idealities in a filter circuit that includes programmable filter circuitry including a first low-pass filter and a second low-pass filter both having a common desired cutoff frequency f 0 .
  • a reference tone f R and a cutoff tone f C are injected into both the first low-pass filter and the second low-pass filter using, for example, separate DACs under the control of some form of tone generation circuitry.
  • the responses of the first low-pass filter and the second low-pass filter are digitized using respective ADCs so as to measure power responses of the reference tone f R and cutoff tone f C .
  • a capacitor code that controls a cutoff frequency f 0-I of the first low-pass filter is varied until a first capacitor code I CODE is determined that most accurately causes the first low-pass filter to utilize the desired cutoff frequency f 0 .
  • a capacitor code that controls the second low-pass filter is varied until a second capacitor code Q CODE is determined that most accurately causes the second low-pass filter to utilize the desired cutoff frequency f 0 .
  • a fractional capacitor code CI FRAC is determined again noting that a fractional capacitor code CI FRAC is a non-integer value that lies between two consecutive capacitor codes [I CODE , I CODE+1 ], and that ideally corresponds to both a zero phase difference and a zero power difference between the first low-pass filter and the second low-pass filter. While the particular methodology may vary from embodiment to embodiment, one approach to determining the fractional capacitor code C FRA may be had by interpolating a line using a plurality of points with each point having (See, FIG.
  • a first dimension being a combined I-Q capacitor code [I CODE , Q CODE ], and a second dimension being a respective measured phase offset between the first low-pass filter and the second low-pass filter using a respective combined I-Q capacitor code, then selecting a combined I-Q capacitor code value that corresponds to a substantially zero phase difference between the first low-pass filter and the second low-pass filter.
  • a scaling factor ⁇ is derived, for example, from empirical data.
  • a first compensated capacitor code I C-CODE is calculated by adding the capacitor code offset ⁇ I OFFSET to the first capacitor code I CODE .
  • a second compensated capacitor code Q C-CODE is calculated by adding the capacitor code offset ⁇ Q OFFSET to the second capacitor code Q CODE .
  • an operating bandwidth is changed from BW 0 to BW 1 , the first compensated capacitor code I C-CODE is applied to the first/I low-pass filter, and the second compensated capacitor code Q C-CODE is applied to the second/Q low-pass filter.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
  • Networks Using Active Elements (AREA)
  • Analogue/Digital Conversion (AREA)
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US9806745B2 (en) * 2015-02-10 2017-10-31 Marvell World Trade Ltd. Systems and methods for low pass filter mismatch calibration
US11942974B2 (en) * 2021-07-21 2024-03-26 Pharrowtech Bv Millimeter wave radio calibration circuit
WO2024049131A1 (ko) * 2022-09-02 2024-03-07 삼성전자주식회사 멀티 수신 대역폭 특성화 테이블을 이용한 수신 정확도 향상을 위한 캘리브레이션 시스템 및 방법

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EP3078139A1 (en) 2016-10-12
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CN105814819A (zh) 2016-07-27
WO2015085059A1 (en) 2015-06-11

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