US7760517B2 - Converter for an ion propulsion system - Google Patents

Converter for an ion propulsion system Download PDF

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US7760517B2
US7760517B2 US12/055,732 US5573208A US7760517B2 US 7760517 B2 US7760517 B2 US 7760517B2 US 5573208 A US5573208 A US 5573208A US 7760517 B2 US7760517 B2 US 7760517B2
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switch
connection
converter
bridge circuit
opening
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US20080259645A1 (en
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Frank Herty
Jochen Haegele
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Airbus Defence and Space GmbH
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Astrium GmbH
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33573—Full-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/53—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/537—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters
    • H02M7/5387—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration
    • H02M7/53871—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current
    • H02M7/53878—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only, e.g. single switched pulse inverters in a bridge configuration with automatic control of output voltage or current by time shifting switching signals of one diagonal pair of the bridge with respect to the other diagonal pair
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of ac power input into dc power output; Conversion of dc power input into ac power output
    • H02M7/42—Conversion of dc power input into ac power output without possibility of reversal
    • H02M7/44—Conversion of dc power input into ac power output without possibility of reversal by static converters
    • H02M7/48—Conversion of dc power input into ac power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/4815—Resonant converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to a converter for an ion propulsion system.
  • Ion propulsion systems require electrical voltages in a range of several thousand volts in order to operate. Therefore, there is a need for power supplies that can achieve a high specific output and simultaneously exhibit high efficiency.
  • the power supplies comprise converters that should exhibit, if possible, only a single converter stage in order to achieve the requisite high specific output while simultaneously continuing to exhibit high efficiency. If, however, a pre-regulator is dispensed with, it must be possible to control the single stage converter in its entirety. That is, when running with no load as well as under maximum load, the output voltage has to stay within the allowable limits.
  • the propulsion systems do not constitute a purely resistive load.
  • plasma variations may lead to periodic changes in the impedance.
  • Gridded ion propulsion systems may develop discharges that result in transient shorts (beam out). Therefore, the anode current supply has to be insensitive to changes in the impedance of the propulsion system.
  • the high transformation makes it necessary to take the winding capacitance of said transformer into special consideration.
  • the prior art converters for ion propulsion systems employ two stages.
  • the first stage comprises a pre-regulator, which does not exhibit electrical isolation and which generates a variable intermediate circuit voltage and a variable intermediate circuit current.
  • the second stage is an unregulated stage for electrical isolation and for raising to the required voltage value.
  • the unregulated stage is designed as a full resonant mode bridge circuit or as a push-pull stage. In both cases zero voltage switches and zero current switches (ZVS: zero voltage switching and ZCS: zero current switching) are used.
  • Exemplary embodiments of the present invention provide an improved converter, which can be employed for ion propulsion systems.
  • the present invention converts the voltage with a converter.
  • the present invention permits a high transformation ratio and, nevertheless, offers a simple and, thus, easy as well as economical possibility for adjusting the load current without having to be concerned about a high ripple of the output voltage.
  • the present invention is based on the recognition that it is possible to achieve a conversion of the voltage using a precise, but simultaneously simple adjustment of the load current in that the necessary circuit engineering measures are implemented in the primary and secondary circuit of the converter, and that the (high) capacitance of the secondary winding, necessitated by the high transformation ratio, cannot exert any negative influence on the voltage conversion.
  • a time delayed opening and/or closing of the switches inside the bridge circuit allows the resulting current paths to divide cyclically and conjoin again and the coupling point to be adjusted using the time lag in opening and/or closing the switches of the bridge circuit. Since, however, this coupling point acts directly on the current amplitude, which occurs during the steady state mode of the voltage conversion, the amount of the high voltage generated during the voltage conversion can also be indirectly controlled by adjusting the coupling point.
  • the present invention offers the advantage that, in contrast to the state of the art, simple engineering means make possible a good controllability of the load current while simultaneously continuing to exhibit a high voltage transformation ratio. Moreover, at the same time, in addition, a reaction to the impedance variations in the supplied load can still ensue with an adequate matching to the power to be transmitted. Furthermore, the matching to the power to be transmitted is realized only in a recharging operation during a switch-over operation of the converter so that there is almost no need to change the pulse duty factor.
  • the required ripple of the current, which is consumed by the converter, and the required ripple of the load current can be achieved with relatively low filtering complexity, even though the switching frequency can be set low (a feature that helps to reduce the frequency dependent losses in the filtering components or storage chokes).
  • the converter includes a bridge circuit with a first and second bridge circuit connection and with two bridge branches, the bridge circuit comprising four switches, each having a first and second connection, the first connection of a first and third switch respectively being connected to the first bridge circuit connection; the second connection of the first switch being connected to the first connection of a second switch; the second connection of the third switch being connected to the first connection of a fourth switch; and the second connection of the second and fourth switch respectively being connected to the second bridge circuit connection.
  • the converter also includes a storage inductance and a high voltage transformer with a primary and a secondary winding, the primary winding and the storage inductance being connected in series; and this series connection being connected to the second connection of the first switch and the second connection of the third switch; and the secondary winding being connected to an output of the converter.
  • the converter further includes a switch control unit, which is designed to drive switches inside a bridge branch in such a manner that a delay time is inserted between the switch-over process in order to prevent a bridge short circuit and in order to be able to carry out the switching operations under zero voltage conditions or lowest voltage.
  • a switch control unit which is designed to drive switches inside a bridge branch in such a manner that a delay time is inserted between the switch-over process in order to prevent a bridge short circuit and in order to be able to carry out the switching operations under zero voltage conditions or lowest voltage.
  • this switch control unit is designed to open or close-so as to be time delayed in relation to a switch of the second bridge branch-a switch of the first bridge branch at a control time.
  • an auxiliary inductance is connected between the primary winding and the second connection of the third switch.
  • a further embodiment of the invention may provide the converter with a diode branch, comprising two diodes.
  • the cathode of the first diode is connected between the primary winding of the high voltage transformer of the auxiliary inductance; and the anode of the first diode is connected to the second bridge circuit connection.
  • the cathode of the second diode should be connected to the first bridge circuit connection; and the anode of the second diode should also be connected between the primary winding and the auxiliary inductance.
  • one capacitance respectively may also be connected between the first and second connections of the second and fourth switch.
  • the voltage load can be kept advantageously low at all four switches during the turn-on operation (zero voltage switching), so that the turn-on losses may be kept as low as possible.
  • one diode respectively may be connected between the first and second connections of all switches, if the switch design that is used does not exhibit any inherent diode, the anode of which is connected to the first connection and the cathode of which is connected to the second connection of the respective switch.
  • a capacitance may be connected in parallel to the secondary winding of the high voltage transformer. This feature makes it possible to avoid undershooting a secondary-sided minimum capacitance, which is necessary for a resonant recharging of the inductances and for the intermediate accumulation of energy.
  • the switch control unit may be designed to determine the control time in such a manner that it corresponds to no more than one-tenth of the period of time that elapses between an opening and closing of the same switch.
  • another embodiment of the invention also provides a switch control unit that is designed to use—in the event of a repeated opening of the fourth switch upon opening a first switch—a control time that is different from the preceding opening of the fourth switch.
  • FIG. 1 is a block diagram of one embodiment of the inventive converter, depicted as a design alternative with a bridge rectifier in the secondary branch.
  • FIG. 2 is a diagram of various current waveforms, depicted in FIG. 1 , over the time.
  • FIG. 3 is an equivalent circuit diagram of the output circuit of the embodiment of the inventive converter that is depicted in FIG. 1 .
  • FIG. 4 is a diagram of a commutation waveform for the embodiment of the inventive converter that is depicted in FIG. 1 .
  • FIG. 5 is a diagram of the family of characteristic control curves in connection with the embodiment of the present invention that is depicted in FIG. 1 .
  • FIG. 6 is a diagram illustrating the effect of the output voltage on the transformer current in connection with the embodiment that is depicted in FIG. 1 .
  • FIG. 7 is a control-related block diagram of the embodiment of the inventive converter that is depicted in FIG. 1 .
  • the converter includes an H bridge and an additional diode branch.
  • a high voltage transformer is connected between the left branch S A , S B and the diode branch D x , D Y .
  • the storage choke L S is situated in series with respect to the primary winding.
  • Another storage choke L K is connected between the diode branch and the right branch of the bridge S C , S D .
  • the high voltage transformer HVT raises the intermediate circuit voltage V ZK to the required value and provides for electrical isolation.
  • the transformation ratio of the transformer is chosen in such a manner that it corresponds to the ratio of the (nominal) intermediate circuit voltage V ZK and the output voltage V OUT .
  • the winding capacitance of the high voltage transformer is modeled by a concentrated component C R ′′. Since this capacitance may not drop below a minimum value, it may be necessary to add an external capacitance (C R ′), so that the result is the resulting capacitance C R .
  • the two branches of the bridge are controlled in such a manner that the upper switch and the lower switch are closed alternatingly and for a period of time that is approximately the same.
  • the switch control unit SCU is configured in such a way that it can switch the switches S A , S B , S C and S D individually and at defined time intervals in relation to each other.
  • the adjustment of the output current is achieved by controlling the right bridge branch C-D in such a manner that all of the edges of the control signal of both switches lag by a defined time t c in relation to the signal edges of the branch A-B.
  • the time shift is referred to below as the control time t c .
  • the control time t c is only a fraction of the total duration of one switching cycle—that is, the duration of a period that is defined by the switching frequency and, consequently, has only a slight effect on the pulse duty factor of the transformer current.
  • the properties of the converter are characterized in that in the steady state operation the amplitudes of the primary and secondary currents of the high voltage transformer change only slightly during the energy transmission periods. Therefore, the result is an almost square waveform of the current, as depicted in FIG. 2 . Furthermore, there is a clear correlation between the current amplitude and the control time t c —a feature that will be explained below in detail. Thus, the current amplitude can be re-adjusted for each switching cycle by changing the control time.
  • these functions could also be achieved without the additional diode branch, which is depicted in FIG. 1 , and without L K .
  • the situation would develop that the diodes of the branch (S C , S D ) would switch hard.
  • the sudden buildup of the blocking voltage may result in a high reverse current, especially if MOSFETS are used as the switches and their (slow) body diode is utilized. The reverse current may cause the component to fail.
  • Both the input current and the output current consist of consecutive square wave or square wave-like current pulses.
  • the filtering stages at the input and the output have to be configured only with respect to the comparatively narrow gaps between the pulses. For the same reason the amplitude as well as the ripple of the currents do not increase, even if the switching frequency is decreased.
  • the switching frequency becomes less, a larger transformer core has to be accepted, the filter components and the chokes L S and L K do not have to be redesigned. Therefore, it is desirable for the switching frequency to be as low as possible, because in this way the eddy current losses in the magnetic components (proximity effect) may be significantly reduced.
  • FIG. 1 assumes ideal switches and rectifier diodes for the purpose of simplifying the following description.
  • the energy source (intermediate circuit) is connected to the drain (load circuit) by way of one storage choke or a plurality of storage chokes that are connected in-between.
  • a current flows from the source to the drain.
  • a portion of the energy is stored in the chokes (i.e., the inductances).
  • the remaining energy is transmitted to the load.
  • a portion of the energy, which is stored in the chokes is transmitted into the load and the rest is transmitted back into the source.
  • the direction of the current is switched over twice per switching cycle by means of the bridge circuit.
  • the current path runs via the switch S A and via the series connection, consisting of the storage choke L S , the transformer primary winding and the auxiliary choke L K .
  • the diodes D X and D Y do not carry any current.
  • the switch S D closes the current circuit.
  • the path runs correspondingly via S C and S B .
  • the output voltage is mapped (reflected) on the primary side of the transformer.
  • the transformation ratio of the transformer is selected, if possible, in such a way that the reflected output voltage matches as precisely as possible the intermediate circuit voltage.
  • the design of the transformation ratio must also consider the voltage situations of the diodes, the switches and the feed lines.
  • the circuit states include the respective states of the power switches (S A to S D ), the charge states of the energy accumulator as well as the resulting current paths of the circuit.
  • the circuit states remain in existence for a defined period of time within a switching cycle. These time periods are referred to below as the sections.
  • FIG. 3 is an equivalent circuit diagram, by which the related differential equations can be derived from the circuit states. Therefore, only those parts of the circuit that lie in the respective current path have to be taken into consideration.
  • the current and voltage waveforms of all of the components of the converter may be generated from the section by section solutions of these differential equations.
  • the waveforms are assembled section by section, the end values of the preceding section serving as the initial values for the solution of the differential equation of the next section. For symmetry reasons only the first half of the switching cycle will be examined below. Furthermore, all values relate to the primary side.
  • the ohmic resistances of the transformer windings, the secondary-sided scattering and the core losses have only a negligible effect and are, therefore, not considered.
  • the section 1a that is depicted in FIG. 4 starts with the opening of the switch S A , an operation that is indicated with the symbol S A ⁇ . This operation initiates the change-over from the positive to the negative half-cycle of the primary transformer current and, thus, also of the choke current I S .
  • the choke current I S which existed immediately before the beginning of this section, is used as the first initial value.
  • I s ⁇ ( t ) V ZK - V out ⁇ ⁇ L S ⁇ sin ⁇ ( ⁇ ⁇ ⁇ t ) + I ⁇ ( t 0 ) + cos ⁇ ( ⁇ ⁇ ⁇ t ) Voltage:
  • V B ⁇ ( t ) V ZK - ( V ZK - V OUT ) ⁇ ( 1 - cos ⁇ ( ⁇ ⁇ ⁇ t ) ) - I ⁇ ( t 0 ) ⁇ ⁇ C B ⁇ sin ⁇ ( ⁇ ⁇ ⁇ t ) Parameter:
  • Section 2a follows section 1a and describes the entire freewheeling of L S .
  • the current path for the freewheeling current runs by way of the primary winding of the transformer to the diode D X and continues to run to the diode of the switch S B . Therefore, the diode D X is kept in a conducting manner by the larger current I K . Only the reflected output voltage V OUT appears as the counter-voltage. The result is a linear decrease in the choke current.
  • the transmission of energy to the secondary side also terminates with this section.
  • I s ⁇ ( t ) I s ⁇ ( t 1 ⁇ a ) - V OUT L S ⁇ t
  • Section 3a begins with the zero passage of the current I S .
  • the diodes of the output rectifier block; and the auxiliary capacitor C R and/or C R ′ is/are charged up to the output voltage.
  • a resonant reversal process in which the elements L S and C R are involved, starts.
  • the current path for I S runs by way of the diode D X , which is forward conducting owing to the still positive current I K . Furthermore, the current I S flows by way of the switch S B , which is already closed at the beginning of this section—i.e., automatically closes following a lag time t d after opening the switch S A (indicated with the symbol S B ⁇ in FIG. 4 ). As a result, the resonance circuit consists of L S and the capacitance C R , converted to the primary side.
  • I s ⁇ ( t ) V OUT ⁇ ⁇ L S ⁇ sin ⁇ ( ⁇ ⁇ ⁇ t )
  • V R ( t ) V OUT ⁇ cos( ⁇ t )
  • section 4 starts. Even though the switches S C and S B are already closed, no energy has been transmitted yet to the output, because the diodes of the output rectifier cannot conduct yet. Then the energy flows from the voltage source V ZK into the capacitor C R , which could not be totally recharged yet owing to the interrupted reversal process. At this point the recharging process is continued.
  • Section 4 ends with the capacitor C R being completely charged up to the negative value of the output voltage.
  • I s ⁇ ( t ) - V ZK + V R ⁇ ( t 3 ) ⁇ ⁇ ( L S + L K ) ⁇ sin ⁇ ( ⁇ ⁇ ⁇ t ) + I S ⁇ ( t 3 ) ⁇ cos ⁇ ( ⁇ ⁇ ⁇ t )
  • V R ⁇ ( t ) - ( V ZK + V R ⁇ ( t 3 ) ) ⁇ ( 1 - cos ⁇ ( ⁇ ⁇ ⁇ t ) ) + I S ⁇ ( t 3 ) ⁇ ⁇ C R ⁇ sin ⁇ ( ⁇ ⁇ ⁇ t ) + V R ⁇ ( t 3 )
  • Section 1b also begins with the opening of the switch S A (an event that is indicated with the symbol S A ⁇ in FIG. 4 ).
  • L S has no counter-voltage and, consequently, holds the current constant.
  • the current path runs by way of the diode D X and the switch S D . Therefore, section 1b ends with the opening of the switch S D (an event that is indicated with the symbol S D ⁇ ).
  • Section 2b follows section 1b. It begins with the elapse of the control time—that is, with the opening of the switch S D . Consequently at this point then the choke current I K flows through the capacitor C D . This capacitor charges itself up until the capacitor voltage V D has reached the value of the intermediate circuit voltage. At this point section 2b ends.
  • I K ⁇ ( t ) I S ⁇ ( t 0 ) ⁇ cos ⁇ ( ⁇ ⁇ ⁇ t )
  • V D ⁇ ( t ) I S ⁇ ( t 0 ) ⁇ ⁇ C D ⁇ sin ⁇ ( ⁇ ⁇ ⁇ t )
  • section 3b begins.
  • the freewheeling path of I K consists of the diode of the switch S C , the voltage source V ZK as well as the diode D X .
  • the freewheeling of the choke L K counter to the voltage source V ZK effects a linear decrease in the choke current.
  • the actuation of the switches is configured in such a manner that the switch S C is closed no later than at zero passage of the choke current I K .
  • the closing of the switch S C after a lag time t d upon opening the switch S D is indicated with the symbol S C ⁇ ). Consequently I K can build up again linearly in the reverse direction. In this case the current path does not change.
  • the choke current L S is already negative (see section 3a). As long as I S stays more negative than I K , the difference between the currents has to flow through D X . In this way the diode D X remains conducting in the forward direction.
  • I K ⁇ ( t ) I K ⁇ ( t 2 ⁇ b ) - V ZK L K ⁇ t
  • the composite waveform of the choke currents I S and I K is obtained by placing the individual segments of the waveforms in a row, as depicted in FIG. 4 .
  • Both currents are identical up to the separation point. Due to the control time, IK decreases no later than IS. The difference between both currents flows through the diode branch. Both currents conjoin again at the coupling point. The commutation is terminated, when the transmission of energy ensues—that is, when the output rectifier become conductive. This occurs at the time t 4 .
  • the new initial value of the transformer current I S may be selected at the control time t c .
  • This initial value is called the knee point current I ⁇ since it marks the transition from the steep current edge of the commutation process to the flat segment of the transformer current.
  • control time t c has the effect that the coupling of the currents I S and I K also takes place at a later time. Therefore, the resonant reversal process (section 3a in FIG. 4 ) is not terminated until later. If one imagines that the waveform from section 3a continues, then the coupling point shifts closer to the zero line. In so doing, the transmission of energy also begins with a knee point current that is lower in terms of amount. If, in contrast, the control time is selected so as to be smaller, the initial value increases.
  • control time and the initial value I ⁇ is obtained numerically by repeatedly calculating the commutation current waveform for the various control times tc.
  • the time functions of the individual sections show that the following parameters may influence the initial value.
  • the above description relates in particular to the change-over of the currents I S and I K from the positive to the negative half-cycle.
  • the procedure is in the inverse direction—that is, during the change-over to the positive half-cycle analog.
  • FIG. 6 a illustrates the case, where the output voltage and the intermediate circuit voltage are the same size.
  • the medium sized current through the rectifier within an energy transmission intervals can be calculated as:
  • T is the duration of a complete switching cycle at the switching frequency f s ; and D is the pulse duty factor of the current pulse, where the duration of the negative and positive pulses are considered cumulatively.
  • V OUT 1 s ⁇ C F ⁇ [ I _ R - V OUT R L ]
  • V OUT 1 s ⁇ C F ⁇ [ D ⁇ I a + V ZK R M - V OUT R M - V OUT R L ]
  • the model that is, the pattern image for the dynamic behavior of the converter—can be derived, as depicted in FIG. 7 . Since the model is based on the values which have been averaged within an energy transmission interval, it applies to frequencies of up to about one-fourth of the switching frequency.
  • the resistance R M acts in such a manner that it is connected parallel to the load resistance R L .
  • L S , L K and f S it is possible to achieve that R M is significantly less than the minimum allowable load resistance, which is established by the power limits of the converter.
  • R M may be configured, for example, one order of magnitude less than R L .
  • the storage chokes L S and L K do not generate any poles or zero points in the controlled system, since their charge state is reset with every transmission of energy. Thus, there is no need to eliminate the poles and zero points with the aid of an underlying current mode control.
  • the converter is insensitive to load variations
  • the converter passes any change in the intermediate circuit voltage to the output
  • the present invention describes a converter principle that may be tailored, in particular, to electrical ion propulsion systems, which require a high voltage in a magnitude of several thousand volts in order to operate. (That is, the high voltage transformer exhibits a transformation ratio of, for example, 1:20).
  • the converter principle permits the output voltage to be held constant over a wide range of loads. This feature is made possible in that the amplitudes of the consecutive current pulses may be adjusted directly and independently of each other. Hence, additional equipment for pre-control (of the intermediate circuit voltage) is not necessary.
  • the converter operates with approximately square wave currents and a defined pulse duty factor.
  • the converter can be employed as an anode current supply for electrical ion propulsion systems in systems with regulated (that is, in the steady state case almost constant) bus voltage.
  • This converter is insensitive to impedance variations of the propulsion system.
  • the waveform of the input current and the output current consists of current pulses that are lined up in close succession. Thus, one can manage with very low capacitance and inductance values for the input and output filters.
  • the ripple of the input current and the output current does not increase as the switching frequency decreases. Therefore, the switching frequency can be set low without having to increase the size of the filter components and the storage chokes. Thus, the frequency dependent losses can be significantly reduced.
  • the transmission function of the converter consists of a low pass filter of first order, the cut-off frequency of which is only slightly load dependent.

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US20120014151A1 (en) * 2009-08-17 2012-01-19 Ideal Power Converters, Inc. Power Conversion with Added Pseudo-Phase
US8432711B1 (en) 2009-06-29 2013-04-30 Ideal Power Converters, Inc. Power transfer devices, methods, and systems with crowbar switch shunting energy-transfer reactance
US8446043B1 (en) 2010-11-30 2013-05-21 Ideal Power Converters, Inc. Photovoltaic array systems, methods, and devices and improved diagnostics and monitoring
US8514601B2 (en) 2009-08-17 2013-08-20 Ideal Power Converters, Inc. Power conversion with added pseudo-phase
US8531858B2 (en) 2011-02-18 2013-09-10 Ideal Power, Inc. Power conversion with current sensing coupled through saturating element
US9130461B2 (en) 2006-06-06 2015-09-08 Ideal Power Inc. Universal power conversion methods with disconnect after driving
US10622906B2 (en) 2016-12-19 2020-04-14 Mitsubishi Electric Corporation Power conversion apparatus and electric propulsion system

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EP2747270B1 (de) * 2012-12-21 2014-10-15 Inmotion Technologies AB Brückenzweig
WO2015105795A1 (en) * 2014-01-07 2015-07-16 Arizona Board Of Regents On Behalf Of Arizona State University Zero-voltage transition in power converters with an auxiliary circuit
EP3104508B1 (de) * 2014-02-05 2018-09-26 Mitsubishi Electric Corporation Ladevorrichtung an bord eines fahrzeuges und überspannungsunterdrückungsverfahren für eine ladevorrichtung an bord eines fahrzeuges
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US9130461B2 (en) 2006-06-06 2015-09-08 Ideal Power Inc. Universal power conversion methods with disconnect after driving
US8432711B1 (en) 2009-06-29 2013-04-30 Ideal Power Converters, Inc. Power transfer devices, methods, and systems with crowbar switch shunting energy-transfer reactance
US8441819B2 (en) 2009-06-29 2013-05-14 Ideal Power Converters, Inc. Power transfer devices, methods, and systems with crowbar switch shunting energy-transfer reactance
US8446745B1 (en) 2009-06-29 2013-05-21 Ideal Power Converters, Inc. Power transfer devices, methods, and systems with crowbar switch shunting energy-transfer reactance
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US8446042B2 (en) 2010-11-30 2013-05-21 Ideal Power Converters, Inc. Photovoltaic array systems, methods, and devices with improved diagnostics and monitoring
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US8531858B2 (en) 2011-02-18 2013-09-10 Ideal Power, Inc. Power conversion with current sensing coupled through saturating element
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EP1976108A2 (de) 2008-10-01
JP2008245516A (ja) 2008-10-09
JP5311857B2 (ja) 2013-10-09
CA2627254A1 (en) 2008-09-27
EP1976108A3 (de) 2016-01-27
CA2627254C (en) 2017-02-28
DE102007015302B4 (de) 2013-01-10
DE102007015302A1 (de) 2008-10-02
EP1976108B1 (de) 2020-10-21
US20080259645A1 (en) 2008-10-23

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