US6366492B1 - Semiconductor memory device capable of automatically controlling bit-line recovery operation - Google Patents

Semiconductor memory device capable of automatically controlling bit-line recovery operation Download PDF

Info

Publication number
US6366492B1
US6366492B1 US09/595,118 US59511800A US6366492B1 US 6366492 B1 US6366492 B1 US 6366492B1 US 59511800 A US59511800 A US 59511800A US 6366492 B1 US6366492 B1 US 6366492B1
Authority
US
United States
Prior art keywords
bit line
bbl
sense amplifier
inverter
memory device
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Expired - Fee Related
Application number
US09/595,118
Inventor
Atsushi Kawasumi
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Toshiba Corp
Original Assignee
Toshiba Corp
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Toshiba Corp filed Critical Toshiba Corp
Assigned to KABUSHIKI KAISHA TOSHIBA reassignment KABUSHIKI KAISHA TOSHIBA ASSIGNMENT OF ASSIGNORS INTEREST (SEE DOCUMENT FOR DETAILS). Assignors: KAWASUMI, ATSUSHI
Application granted granted Critical
Publication of US6366492B1 publication Critical patent/US6366492B1/en
Anticipated expiration legal-status Critical
Expired - Fee Related legal-status Critical Current

Links

Images

Classifications

    • G—PHYSICS
    • G11—INFORMATION STORAGE
    • G11C—STATIC STORES
    • G11C11/00—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor
    • G11C11/21—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor using electric elements
    • G11C11/34—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor using electric elements using semiconductor devices
    • G11C11/40—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor using electric elements using semiconductor devices using transistors
    • G11C11/41—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor using electric elements using semiconductor devices using transistors forming static cells with positive feedback, i.e. cells not needing refreshing or charge regeneration, e.g. bistable multivibrator or Schmitt trigger
    • G11C11/413—Auxiliary circuits, e.g. for addressing, decoding, driving, writing, sensing, timing or power reduction
    • G—PHYSICS
    • G11—INFORMATION STORAGE
    • G11C—STATIC STORES
    • G11C11/00—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor
    • G11C11/21—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor using electric elements
    • G11C11/34—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor using electric elements using semiconductor devices
    • G11C11/40—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor using electric elements using semiconductor devices using transistors
    • G11C11/41—Digital stores characterised by the use of particular electric or magnetic storage elements; Storage elements therefor using electric elements using semiconductor devices using transistors forming static cells with positive feedback, i.e. cells not needing refreshing or charge regeneration, e.g. bistable multivibrator or Schmitt trigger
    • G11C11/413—Auxiliary circuits, e.g. for addressing, decoding, driving, writing, sensing, timing or power reduction
    • G11C11/417—Auxiliary circuits, e.g. for addressing, decoding, driving, writing, sensing, timing or power reduction for memory cells of the field-effect type
    • G11C11/419—Read-write [R-W] circuits

Definitions

  • the present invention relates to a semiconductor memory device such as a SRAM (a Static Random Access Memory), and more particularly, it relates to bit line loads for bit lines in a SRAM, which is capable of accelerating a recovery operation.
  • SRAM Static Random Access Memory
  • FIG. 1 is a block diagram showing a memory core circuit in a conventional static random access memory (SRAM).
  • SRAM static random access memory
  • each memory cell MC in a memory cell array 100 in the memory core circuit is placed at each cross point of a plurality of bit line pairs BL and bBL (BL 0 and bBL 0 , BL 1 and bBL 1 ,) and a plurality of word lines WL (WL 0 , WL 1 , . . . ).
  • PMOS transistors QP 1 and QP 2 as bit line loads are connected to one end of each bit line pair BL and bBL.
  • Other end of each bit line pair BL and bBL is connected to data line pair DL and bDL through each column gate 105 which is made up of NMOS transistors QN 1 and QN 2 .
  • the data line pair DL and bDL is connected to a write-in driver 103 for data write-in operation, and also connected to a sense amplifier (SA) 102 for data readout operation.
  • SA sense amplifier
  • FIG. 2 is a timing chart showing a data write operation and a write recovery operation in the conventional SRAM shown in FIG. 1 .
  • the level of a write enable signal /WE is switched to a L level.
  • One of the data line pair DL and bDL becomes the L level and other becomes a H level according to the value of the write-in data.
  • These levels of the data line pairs DL and bDL are transferred to the bit line pair which are selected by the column gate 105 . Further, the levels of the data line pair DL and bDL are written into the memory cell MC that is selected by a word line WL (WL 0 , WL 1 , . . . WLn).
  • the write enable signal /WE is switched to the H level, and a write recovery operation is performed.
  • the levels of both the bit lines BL and bBL in each bit line pair are set to the H level for the following data readout operation or the following data write-in operation.
  • the level of the precharge signal /PCH becomes the L level and the PMOS transistors QP and QP 2 as the bit line loads enter an ON state (an active state).
  • both the capacity of the bit lines and the capacity of the data lines become large in large scale semiconductor integrated circuits, it is difficult to execute the write recovery operation at a high speed rate only by using the write-in driver 103 .
  • both the PMOS transistors QP 1 and QP 2 as the bit line loads in the conventional SRAM shown in FIG. 2 enter the ON state during the write recovery operation by setting the L level to the precharge signal /PCH. Thereby, the bit line pair BL and bBL can be charged rapidly.
  • an object of the present invention is, with due consideration to the drawbacks of the conventional semiconductor memory device, to provide a semiconductor memory device of a low power consumption with an improved timing margin which is capable of performing the recovery operation for the bit lines automatically only by the control of the write-in driver without any elapse of the timings between the ON/OFF operation of the PMOS transistors as the bit line loads and the ON/OFF operation of the write-in driver.
  • a semiconductor memory device has a memory cell array, a write-in driver, a sense amplifier, and bit line loads.
  • the memory cell array includes memory cells which are located at cross points of word lines and bit lines.
  • the write-in driver performs data write-in operation to transfer write-in data to the memory cells through the bit lines, and performs a recovery operation by supplying a high level voltage to the bit lines when a voltage potential of the bit lines drops to a low voltage potential after a completion of the data write-in operation.
  • the sense amplifier detects and amplifies data read from the memory cells through the bit lines. The operation of the bit line loads is controlled by the voltage potential of the bit lines.
  • Each bit line load is placed between a first power source and each of the bit lines.
  • the bit line load enters the ON state in order to accelerate the recovery operation when the voltage potential of the bit lines reach to a predetermined voltage potential during the recovery operation performed by the write-in driver.
  • the bit line loads for the bit lines are controlled automatically by the voltage potential of the bit lines and enters the ON state when the voltage potential of the bit lines is over a predetermined voltage level in order to accelerate the speed of the recovery operation. Accordingly, it is not necessary to match the timings between the write-in driver and the bit line loads.
  • This feature of the present invention causes to increase a timing margin and to suppress the power consumption to be used for the control of the bit line loads.
  • each of the bit line loads includes a PMOS transistor whose drain is connected to the corresponding bit line and whose source is connected to the first power source, and an inverter connected between the corresponding bit line and a gate of the PMOS transistor for inverting a voltage potential of the corresponding bit line and transferring the inverted voltage potential to the gate of the PMOS transistor.
  • the entering to the ON state of the PMOS transistor causes to increase the voltage potential of the bit lines rapidly to the H level.
  • the PMOS transistor enters the OFF state when the voltage potential of the bit line to be shifted to the L level drops under a threshold voltage of the inverter, no waste current flows through the bit line and it is thereby possible to perform the data write-in operation at high speed.
  • each of the bit line loads further includes a NMOS transistor connected to the PMOS transistor in series.
  • a drain of the NMOS transistor is connected to both the corresponding bit line and the drain of the PMOS transistor, and a source of the NMOS transistor is connected to a second power source, and a gate of the NMOS transistor is connected to the inverter in order to supply an output of the inverter.
  • each bit line load includes a flip flop as one inverter made up of the PMOS transistor and the NMOS transistor. In each bit line load, two inverters are connected in parallel and in reverse direction.
  • the NMOS transistor In this configuration of each bit line load, the NMOS transistor enters the ON state which is connected to the bit line in the bit line pair whose voltage potential is shifted to the L level during the data write-in operation. This causes to accelerate the level shifting of the bit line to the L level. Therefore it is possible to perform the data write-in operation at high speed.
  • the inverter in each of the bit line loads has a type in which an amplitude of an output voltage is suppressed.
  • the inverter in each of the bit line loads is a type in which an amplitude of an output voltage is suppressed.
  • each of the bit line loads has a NMOS transistor whose drain is connected to a first power source, and whose source is connected to the corresponding bit line, a first inverter whose input is connected to the corresponding bit line, and a second inverter whose input is connected to an output of the first inverter and whose output is connected to a gate of the NMOS transistor.
  • each of the memory cells is a SRAM cell
  • the sense amplifier is a sense amplifier of a current amplifier type for detection a change of a current flow and amplifying the current flow.
  • FIG. 1 is a diagram showing a configuration of a conventional SRAM
  • FIG. 2 is a timing chart of the operation of the conventional SRAM shown in FIG. 1;
  • FIG. 3 is a block diagram showing a configuration of a core circuit of a SRAM as a semiconductor memory device according to the first embodiment of the present invention
  • FIG. 4 is a block diagram showing a configuration of each memory cell in the SRAM
  • FIG. 5 is a block diagram showing a configuration of a sense amplifier in the SRAM
  • FIG. 6 is a circuit diagram showing another configuration of the sense amplifier in the SRAM.
  • FIG. 7 is a timing chart of the operation of the SRAM shown in FIG. 3;
  • FIG. 8 is a circuit diagram showing mainly a configuration of each current source load for each pair of bit lines BL and bBL in a SRAM according to the second embodiment of the present invention.
  • FIG. 9 is a block diagram showing a core circuit of a configuration of a SRAM according to the third embodiment of the present invention.
  • FIG. 10 is a block diagram showing a configuration of a core circuit of a SRAM according to the fourth embodiment of the present invention.
  • FIG. 3 is a block diagram showing a configuration of a SRAM as a semiconductor memory device according to the first embodiment of the present invention.
  • FIG. 3 shows a configuration of a memory core circuit of the SRAM.
  • the reference number 1 designates a memory cell array in the memory core circuit of the SRAM.
  • the reference character MC designates each memory cell to which a data item is written and then stored, and from which the data item is readout.
  • the reference characters WL 0 , WL 1 , . . . , and WLn indicate word lines.
  • the reference characters BL and bBL denote each bit line pair (B 10 and bBL 0 , BL 1 and bBL 1 , . . . ).
  • the reference number 2 denotes a sense amplifier (SA) through which a data item stored in the memory cell MC is transferred to the outside of the memory core circuit.
  • the reference number 3 indicates a write-in driver through which a data item is written into the memory cell MC through a bit line pair BL and bBL.
  • This write-in driver is made up of an inverter I 3 , a NOR gate G 1 and an inverter I 1 , and a NOR gate G 2 and an inverter I 2 .
  • the reference number 4 designates a bit line load for a bit line BL (BL 0 , BL 1 , . . . ) in each bit line pair (BL and bBL).
  • the reference number 4 ′ designates a bit line load for a bit line bBL (bBL 0 , bBL 1 , . . . ) in each bit line pair (BL and bBL).
  • Each of the bit line loads 4 and 4 ′ is made up of a PMOS transistor and an inverter.
  • the bit line load 4 for the bit line BL 0 in the bit line pair (BL 0 and bBL 0 ) is made up of a PMOS transistor QP 1 and an inverter I 4 .
  • the bit line load 4 ′ for the bit line bb 10 in the bit line pair (BL 0 and bBL 0 ) is made up of a PMOS transistor QP 2 and an inverter I 4 .
  • the reference number 5 indicates a column gate which is placed for each bit line pair BL and bBL.
  • each memory cell MC is placed at each cross point between the word line WL (WL 0 , WL 1 , . . . , and WLn) and bit line pairs BL and bBL (BL 0 and bBL 0 , BL 1 and bBL 1 , . . . ), and one end of the bit line pair BL and bBL is connected to the PMOS transistors QP 1 and QP 2 in the bit line loads 4 and 4 ′, respectively. Other ends of which is connected to the data line pair DL and bDL through the corresponding column gate 5 .
  • the data line pair DL and bDL is connected to the write-in driver 3 for data write-in operation, and also connected to the sense amplifier 2 for data readout operation.
  • FIG. 4 is a block diagram showing a configuration of each memory cell MC.
  • each memory cell MC is a SRAM cell which comprises a flip flop in which inverters I 11 and I 12 are connected to each other.
  • Nodes N 1 and N 2 are connected to the bit lines bBL and BL, respectively, in the bit line pair through NMOS transistors QN 11 and QN 12 as transfer gates, respectively.
  • These NMOS transistors QN 11 and QN 12 are driven by the level of the word line WL.
  • each of the bit line loads 4 and 4 ′ comprises the PMOS transistor and the inverter, such as the PMOS transistor QP 1 and the inverter I 4 , and the PMOS transistor QP 2 and the inverter I 4 , for example.
  • the sources of the PMOS transistors QP 1 and QP 2 are connected to the power source Vcc and the drains of which are connected to the bit lines BL and bBL, respectively.
  • the inverter I 4 inverts the level of the voltage potential of the corresponding bit line BL or bBL and provides the inverted one to the gate of the corresponding PMOS transistor.
  • the bit line loads 4 and 4 ′ have no control terminal for its operation to which a control signal is supplied from a control means (not shown).
  • the ON/OFF operation of the PMOS transistor QP (QP 1 , QP 2 . . . . ) in the bit line loads 4 and 4 ′ is controlled automatically according to the voltage potential of each of the bit lines BL and bBL during the write recovery operation.
  • the operation of the bit line loads 4 and 4 ′ causes to increase the write recovery operation. This will be explained in detail.
  • the write-in driver 3 connected to both the data lines DL and bDL (DL 0 , bDL 0 , DL 1 , bDL 1 , . . . ) mainly comprises the NOR gates G 1 and G 2 whose operation is controlled by the write enable signal /WE.
  • the inverted data /DIN which is obtained through the inverter I 3 and the write-in data DIN are supplied to the NOR gates G 1 and G 2 , respectively. Accordingly, when the level of the write enable signal /WE is the L level, the data lines DL and bDL in the data line pair have opposite levels to each other according to the level (H level or L level) of the write-in data DIN. Thus, the complementary signals (for example, H level and L level, or L level and H level) are transferred to the bit lines BL and bBL in the bit line pair through the data lines DL and bDL.
  • SA 2 sense amplifier of a current amplifier type having a rapidly amplifying function
  • FIG. 5 is a block diagram showing a practical configuration of the sense amplifier (SA) 2 as the current amplifier type.
  • PMOS transistors QP 31 and QP 33 are connected in series between a sense input node SAIN 1 connected to one bit line BL in the bit line pair and a ground source Vss.
  • PMOS transistors QP 32 and QP 34 are connected in series between a sense input node SAIN 2 connected to other bit line bBL and a ground source Vss.
  • An activation signal /SA for the sense amplifier (SA) 2 is supplied to both the gates of the PMOS transistors QP 33 and QP 34 .
  • the gate of the PMOS transistor QP 31 is connected to the connection node N 12 through which the PMOS transistors QP 32 and QP 34 are connected to each other. Similar to this, the gate of the PMOS transistor QP 32 is connected to the connection node N 11 through which the PMOS transistors QP 31 and QP 33 are connected to each other. These connection nodes N 11 and N 12 become output nodes SAOUT 1 and SAOUT 2 of the sense amplifier (SA) 2 .
  • the difference of the currents supplied to the sense input nodes SAIN 1 and SAIN 2 according to a data item stored in a selected memory cell MC becomes the difference of the voltage potentials of the nodes N 11 and N 12 .
  • This voltage potential is feedback in positive direction to the PMOS transistors QP 31 and QP 32 in order to amplify the difference of the currents.
  • the outputs of the H level and the L level are transferred through the sense output nodes SAOUT 1 and SAOUT 2 .
  • FIG. 6 is a circuit diagram showing another configuration of the sense amplifier (SA) 2 as the current amplifier type.
  • SA sense amplifier
  • current detection circuits 41 and 42 which form two current detection systems are formed between the two sense input nodes SAIN 1 and SAIN 2 and the ground source Vss.
  • inverters 43 and 44 are also placed in order to invert the level of the output from the current detection circuits 41 and 42 .
  • One current detection circuit 41 comprises PMOS transistors QP 41 and QP 42 which form a current mirror circuit, and NMOS transistors QN 41 and QN 42 which are placed between the drains of both the PMOS transistors QP 41 and QP 42 and the ground source Vss.
  • the current detection circuit 42 comprises PMOS transistors QP 43 and QP 43 which form a current mirror circuit, and NMOS transistors QN 43 and QN 44 which are placed between the drains of both the PMOS transistors QP 43 and QP 44 and the ground source Vss.
  • both the gate and the drain of the PMOS transistor QP 41 which are connected to the sense input node SAIN 1 are connected to each other.
  • both the gate and the drain of the PMOS transistor QP 43 which are connected to the sense input node SAIN 2 are connected to each other.
  • both the gate and the drain of the NMOS transistor QN 42 in the current detection circuit 41 are connected to each other, and both the gate and the drain of the NMOS transistor QN 44 in the current detection circuit 42 are connected to each other.
  • the voltage potential of the node N 42 drops.
  • the change of the voltage potential at each of the nodes N 41 and N 42 becomes the change of the voltage potential of the drain in each of the NMOS transistors QP 41 and QP 43 , and then feedback to the PMOS transistors QP 42 and QP 44 .
  • one of the nodes N 41 and N 42 becomes the H level, and other becomes the L level according to the input data item. These levels are inverted by the inverters 43 and 44 and then output to the outside.
  • the write-in driver 3 When the level of the write enable signal /WE becomes the L level, the write-in driver 3 outputs the H level to one data line DL (or bDL) and the L level of the other data line bDL (or DL) according to the level of the write-in data item DIN. These complementary signals, the H level and the L level are transferred to the bit line pair including the bit lines BL and bBL selected by the corresponding column selector 5 , and then written into the memory cell MC selected by the word lines WL (WL 0 , WL 1 , . . . , WLn). These write-in operation is the same as that of the conventional semiconductor memory device.
  • the PMOS transistor QP 1 connected to the bit line BL in the bit line load keeps the ON state (the active state), so that the level of the bit line BL is kept at the H level. Then, the voltage potential of the other bit line bBL drops toward the L level from the H level.
  • the inverter 14 outputs the H level to the gate of the PMOS transistor QP 2 in the bit line load 4 ′. Thereby, the PMOS transistor QP 2 enters the OFF state. That is, the level of the bit line bBL drops approximately to the ground voltage Vss, because there is no current supply from the current source load 4 to the bit line bBL.
  • the write recovery operation can be performed by switching in level the write-in signal (/WE) to the H level.
  • both the invertors I 1 and I 2 in the write-in driver 3 output the H level, because the write-in signal (/WE) has been switched to the H level.
  • the H level of the data lines DL and bDL are transferred to the bit lines BL and bBL in each bit line pair.
  • the voltage potential of the bit line bBL at the L level side increases by the output from the write-in driver 3 .
  • the PMOS transistor QP 2 enters the ON state.
  • the current is supplied from the power source voltage Vcc to the bit line bBL through the PMOS transistor QP 2 .
  • the timing of the write recovery operation is determined by the timing control of the write-in driver 3 automatically. Therefore it is not necessary to adjust the time elapse between both the timing of the control of the bit line loads 4 and 4 ′ for the bit lines and the timing of the write-in driver 3 .
  • This feature provides a large timing margin in the write recovery operation in the SRAM.
  • levels of both the bit lines BL and bBL in each bit line pair are the H level at the initial state during the data readout operation, and the bit line loads 4 and 4 ′ are in the ON state.
  • the sense amplifier of the current amplify type having the configuration shown in FIG. 5 or FIG. 6 is used as the sense amplifier, it is possible to execute certainly the data sensing operation without setting the bit line loads 4 and 4 ′ to the OFF state.
  • FIG. 8 is a circuit diagram showing mainly a configuration of each bit line load for each pair of bit lines BL and bBL in a SRAM according to the second embodiment of the present invention.
  • the reference number 41 designates each bit line load for each bit line BL (BL 0 , BL 1 , . . . ), and the reference number 41 ′ denotes each bit line load for each bit line bBL (bBL 0 , bBL 1 , . . . ) in the SRAM of the second embodiment.
  • the bit line load 41 is made up of the inverter I 4 and an inverter I 5 .
  • the inverter I 5 is made up of a PMOS transistor QP 1 and a NMOS transistor QN 5 .
  • the bit line load 41 ′ is made up of the inverter I 4 and an inverter I 5 ′.
  • the inverter I 5 ′ is made up of a PMOS transistor QP 2 and a NMOS transistor QN 6 .
  • the bit line load 41 for the bit line BL (BL 0 , BL 1 , . . . ) includes the NMOS transistor QN 5 in addition to the configuration of the bit line load 4 shown in FIG. 3 .
  • the bit line load 41 ′ for the bit line bBL (bBL 0 , bBL 1 , . . . ) includes the NMOS transistor QN 6 in addition to the configuration of the bit line load 4 ′ shown in FIG. 3 .
  • the drains of the NMOS transistors QN 5 and QN 6 are connected to the bit line BL and the bit line bBL in each bit line pair, respectively.
  • the sources of the NMOS transistors QN 5 and QN 6 are connected to the ground voltage Vss.
  • the output of the inverter I 4 is supplied to the gates of the NMOS transistors QN 5 and QN 6 , and the PMOS transistors QP 1 and QP 2 .
  • the PMOS transistor QP 1 and NMOS transistor QN 5 forms the inverter I 5
  • the PMOS transistor QP 2 and NMOS transistor QN 6 forms the inverter I 5 ′.
  • the inverter I 5 (made up of the PMPS transistor QP 1 and the NMOS transistor QN 5 ) and the inverter I 4 are connected to each other, and the inverter I 5 ′ (made up of the PMPS transistor QP 2 and the NMOS transistor QN 6 ) and the inverter I 4 are also connected to each other.
  • This configuration forms a flip flop.
  • the operation of the bit line loads 41 and 41 ′ in the SRAM of the second embodiment is basically the same as that of the SRAM according to the first embodiment.
  • the difference between the first embodiment and the second embodiment is as follows:
  • the NMOS transistor QN 5 or QN 6 enters the ON state. That is, because the speed of the voltage transition of the bit line to the L level can be accelerated, it is possible to execute the data write-in operation at a high speed rate. Therefore the SRAM as the semiconductor memory device of the second embodiment is capable of performing the write-in operation at high speed in addition to the effect of the SRAM of the first embodiment.
  • FIG. 9 is a block diagram showing a configuration of a memory core circuit in a SRAM according to the third embodiment of the present invention.
  • the reference number 74 designates a bit line load for a bit line BL (BL 0 , BL 1 , . . . ) in each bit line pair (B 1 and bBL).
  • the reference number 74 ′ designates a bit line load for a bit line bBL (bB 10 , bBL 1 , . . . ) in the bit line pair.
  • Each of the bit line loads 74 and 74 ′ is the same in configuration as that of each of the bit line loads 4 and 4 ′ shown in FIG. 3, respectively.
  • the inverter I 4 in the bit line loads 4 and 4 ′ in the SRAM shown in FIG. 3 has a voltage amplitude (Vcc-Vss) of the power source voltage potential Vcc to the ground source voltage potential Vss.
  • a positive voltage Vss 1 in the bit line loads 74 and 74 ′ in the SRAM of the third embodiment shown in FIG. 9 is slightly higher than that of the voltage Vss of the ground source.
  • Other configuration components in the SRAM of the third embodiment are the same as those in the SRAM of the first embodiment shown in FIG. 3 .
  • the configuration of the bit line loads 74 and 74 ′ can improve the higher data readout operation. That is, during the data readout operation, the memory cell MC inputs the current from one of the bit lines BL and bBL according to the level of the data to be read out, and inputs no current from other bit line.
  • the current which flows into the activated sense amplifier (SA) 2 from the PMOS transistor QP 1 and QP 2 in the bit line loads 74 and 74 ′ through the bit lines BL and bBL in the bit line pair becomes i and (i ⁇ (i cell)) at the node in the bit lines BL and bBL in the bit line pair, respectively, where the input current to the memory cell MC is (i cell) only when the bit line loads 74 and 74 ′ are current sources to flow the constant current i.
  • the amount of the cell current (i cell) that flows through the PMOS transistor QP 1 in the bit line load 74 is increased by ⁇ i. That is, because a bias between the gate and the source of the PMOS transistor QP 1 is a constant, the amount of the current ⁇ i can be increased by increasing the voltage
  • Vss 1 the low level power source for the inverter I 4 in the bit line loads 41 and 41 ′ shown in FIG. 8, which is higher than that of the ground voltage Vss.
  • FIG. 10 is a block diagram showing a configuration of a memory score circuit of a SRAM according to the fourth embodiment of the present invention.
  • the reference number 84 designates a bit line load for the bit line BL (BL 0 , BL 1 , . . . ) in each bit line pair.
  • the bit line load 84 is made up of a NMOS transistor QN 01 , an inverter I 41 , and an inverter I 42 .
  • the NMOS transistor QN 01 is connected to the power source voltage Vcc, and the source of which is connected to the bit line BL (BL 0 , BL 1 , . . . ).
  • the gate of the NMOS transistor QN 01 is connected to the inverters I 41 and I 42 which are connected in series.
  • the input side of the inverter I 41 is connected to the bit line BL (BL 0 , BL 1 , . . . ).
  • the reference number 84 ′ designates a bit line load for the bit line bBL (bBL 0 , bBL 1 , . . . ) in each bit line pair.
  • the bit line load 84 ′ is made up of a NMOS transistor QN 02 , the inverter I 41 , and the inverter I 42 .
  • the NMOS transistor QN 02 is connected to the power source voltage Vcc, and the source of which is connected to the bit line bBL (bBL 0 , bBL 1 . . . )
  • the gate of the NMOS transistor QN 02 is connected to the inverters I 41 and I 42 which are connected in series.
  • the input side of the inverter I 41 is connected to the bit line bBL (bBL 0 , bBL 1 , . . . ).
  • the NMOS transistor QN 01 is driven through the gate thereof.
  • the gate of the NMOS transistor QN 1 is connected to the bit line BL (BL 10 , BL 1 , . . . ) through the two inverters I 41 and I 42 connected in series.
  • the voltage potential of the bit line BL (B 10 , BL 1 , . . . ) is transferred to the gate of the NMOS transistor QN 01 through the two inverters I 41 and I 42 connected in series.
  • the NMOS transistor QN 02 is driven through the gate thereof.
  • the gate of the NMOS transistor QN 2 is connected to the bit line bBL (bBL 0 , bBL 1 . . . ) through the two inverters I 41 and I 42 connected in series.
  • the voltage potential of the bit line bBL (bBL 0 , bBL 1 , . . . ) is transferred to the gate of the NMOS transistor QN 02 through the two inverters I 41 and I 42 connected in series.
  • the charged level of each of the bit lines BL and bBL is limited by the threshold voltage of the NMOS transistors QN 01 and QN 02 in the bit line loads 84 and 84 ′.
  • the present invention is not limited by these configuration described above.
  • the concept of the present invention can be applied to various semiconductor memories which are incorporate memory cells to store data based on the presence of a current through bit lines, such as a nonvolatile semiconductor memory device.

Landscapes

  • Engineering & Computer Science (AREA)
  • Microelectronics & Electronic Packaging (AREA)
  • Computer Hardware Design (AREA)
  • Static Random-Access Memory (AREA)

Abstract

A SRAM includes a memory cell array (1), a write-in driver (3), a sense amplifier (2), and bit line loads (4 and 4′). The write-in driver (3) transfers write-in data to the bit lines BL and bBL, and supplies a high voltage potential to the bit lines BL and bBL when the voltage potential of the bit lines BL and bBL drops to a low voltage potential in order to increase the voltage potential to a high voltage potential. The sense amplifier (2) detects data read out from the memory cells and amplifies the data. The bit line loads (4 and 4′) which are located at ends of the bit lines BL and bBL. The ON/OFF operation of the bit line loads (4 and 4′) is controlled automatically according to the voltage potential of the bit lines BL and bBL. Each bit line load (4 or 4′) has a PMOS transistor (QP1 or QP2) located between the bit line (BL or bBL) and a power source Vcc, and an inverter I4 for inverting the voltage potential of the bit line (Bl or bBL) and provides the inverted voltage potential to the gate of the PMOS transistor (QP1 or QP2).

Description

BACKGROUND OF THE INVENTION
1. Field of the Invention
The present invention relates to a semiconductor memory device such as a SRAM (a Static Random Access Memory), and more particularly, it relates to bit line loads for bit lines in a SRAM, which is capable of accelerating a recovery operation.
2. Description of the Prior Art
FIG. 1 is a block diagram showing a memory core circuit in a conventional static random access memory (SRAM). In FIG. 1, each memory cell MC in a memory cell array 100 in the memory core circuit is placed at each cross point of a plurality of bit line pairs BL and bBL (BL0 and bBL0, BL1 and bBL1,) and a plurality of word lines WL (WL0, WL1, . . . ).
PMOS transistors QP1 and QP2 as bit line loads are connected to one end of each bit line pair BL and bBL. Other end of each bit line pair BL and bBL is connected to data line pair DL and bDL through each column gate 105 which is made up of NMOS transistors QN1 and QN2.
The data line pair DL and bDL is connected to a write-in driver 103 for data write-in operation, and also connected to a sense amplifier (SA) 102 for data readout operation.
FIG. 2 is a timing chart showing a data write operation and a write recovery operation in the conventional SRAM shown in FIG. 1.
When a data item is written into one memory cell MC, as shown in FIG. 2, the level of a write enable signal /WE is switched to a L level. One of the data line pair DL and bDL becomes the L level and other becomes a H level according to the value of the write-in data. These levels of the data line pairs DL and bDL are transferred to the bit line pair which are selected by the column gate 105. Further, the levels of the data line pair DL and bDL are written into the memory cell MC that is selected by a word line WL (WL0, WL1, . . . WLn). After the completion of the data write-in operation to the memory cell MC, the write enable signal /WE is switched to the H level, and a write recovery operation is performed. In the write recovery operation, the levels of both the bit lines BL and bBL in each bit line pair are set to the H level for the following data readout operation or the following data write-in operation. In the write recovery operation, the level of the precharge signal /PCH becomes the L level and the PMOS transistors QP and QP2 as the bit line loads enter an ON state (an active state).
Because both the capacity of the bit lines and the capacity of the data lines become large in large scale semiconductor integrated circuits, it is difficult to execute the write recovery operation at a high speed rate only by using the write-in driver 103. In order to avoid this conventional drawback, both the PMOS transistors QP1 and QP2 as the bit line loads in the conventional SRAM shown in FIG. 2 enter the ON state during the write recovery operation by setting the L level to the precharge signal /PCH. Thereby, the bit line pair BL and bBL can be charged rapidly.
During both the data write-in operation and the data readout operation, the level of this precharge signal is kept at the H level (namely, /PCH=H).
However, such a conventional control method of the bit line recovery operation which is performed in the DRAM having the configuration shown in FIG. 1, it must be necessary to adjust both the timing of the ON/OFF operation of the write-in driver 103 and the timing of the ON/OFF of the PMOS transistors QP1 and QP2 as the bit line loads for the bit lines BL and bBL. When both the timings are not matched to each other, it happens that a write-in data item changes to a wrong data item, or a waste power consumption increases. Furthermore, because the recovery operation for the bit lines by the PMOS transistors QP1 and QP2 is performed for all of the bit lines BL and bBL (BL0, bBL0, BL1, bBL1, . . . ) which are selected by the word lines, the large power consumption for the gate capacitances of the PMOS transistors QP1 and QP2 (as the bit line loads) increases.
SUMMARY OF THE INVENTION
Accordingly, an object of the present invention is, with due consideration to the drawbacks of the conventional semiconductor memory device, to provide a semiconductor memory device of a low power consumption with an improved timing margin which is capable of performing the recovery operation for the bit lines automatically only by the control of the write-in driver without any elapse of the timings between the ON/OFF operation of the PMOS transistors as the bit line loads and the ON/OFF operation of the write-in driver.
In accordance with a preferred embodiment of the present invention, a semiconductor memory device has a memory cell array, a write-in driver, a sense amplifier, and bit line loads. The memory cell array includes memory cells which are located at cross points of word lines and bit lines. The write-in driver performs data write-in operation to transfer write-in data to the memory cells through the bit lines, and performs a recovery operation by supplying a high level voltage to the bit lines when a voltage potential of the bit lines drops to a low voltage potential after a completion of the data write-in operation. The sense amplifier detects and amplifies data read from the memory cells through the bit lines. The operation of the bit line loads is controlled by the voltage potential of the bit lines. Each bit line load is placed between a first power source and each of the bit lines. The bit line load enters the ON state in order to accelerate the recovery operation when the voltage potential of the bit lines reach to a predetermined voltage potential during the recovery operation performed by the write-in driver. Thereby, the bit line loads for the bit lines are controlled automatically by the voltage potential of the bit lines and enters the ON state when the voltage potential of the bit lines is over a predetermined voltage level in order to accelerate the speed of the recovery operation. Accordingly, it is not necessary to match the timings between the write-in driver and the bit line loads. This feature of the present invention causes to increase a timing margin and to suppress the power consumption to be used for the control of the bit line loads.
In addition, in a semiconductor memory device as another preferred embodiment of the present invention, each of the bit line loads includes a PMOS transistor whose drain is connected to the corresponding bit line and whose source is connected to the first power source, and an inverter connected between the corresponding bit line and a gate of the PMOS transistor for inverting a voltage potential of the corresponding bit line and transferring the inverted voltage potential to the gate of the PMOS transistor. Thereby, during the recovery operation performed by the write-in driver, the PMOS transistor enters the ON state automatically when the voltage of the bit line, whose voltage potential has dropped to a low voltage potential, reaches to and over a threshold voltage of the PMOS transistor. The entering to the ON state of the PMOS transistor causes to increase the voltage potential of the bit lines rapidly to the H level. On the other hand, during a data write-in operation, because the PMOS transistor enters the OFF state when the voltage potential of the bit line to be shifted to the L level drops under a threshold voltage of the inverter, no waste current flows through the bit line and it is thereby possible to perform the data write-in operation at high speed.
Furthermore, in a semiconductor memory device as another preferred embodiment of the present invention, each of the bit line loads further includes a NMOS transistor connected to the PMOS transistor in series. In the semiconductor memory device, a drain of the NMOS transistor is connected to both the corresponding bit line and the drain of the PMOS transistor, and a source of the NMOS transistor is connected to a second power source, and a gate of the NMOS transistor is connected to the inverter in order to supply an output of the inverter. Thereby, each bit line load includes a flip flop as one inverter made up of the PMOS transistor and the NMOS transistor. In each bit line load, two inverters are connected in parallel and in reverse direction. In this configuration of each bit line load, the NMOS transistor enters the ON state which is connected to the bit line in the bit line pair whose voltage potential is shifted to the L level during the data write-in operation. This causes to accelerate the level shifting of the bit line to the L level. Therefore it is possible to perform the data write-in operation at high speed.
Moreover, in a semiconductor memory device as another preferred embodiment of the present invention, the inverter in each of the bit line loads has a type in which an amplitude of an output voltage is suppressed.
In addition, in a semiconductor memory device as another preferred embodiment of the present invention, the inverter in each of the bit line loads is a type in which an amplitude of an output voltage is suppressed.
Furthermore, in a semiconductor memory device as another preferred embodiment of the present invention, each of the bit line loads has a NMOS transistor whose drain is connected to a first power source, and whose source is connected to the corresponding bit line, a first inverter whose input is connected to the corresponding bit line, and a second inverter whose input is connected to an output of the first inverter and whose output is connected to a gate of the NMOS transistor.
In addition, in a semiconductor memory device as another preferred embodiment of the present invention, each of the memory cells is a SRAM cell, and the sense amplifier is a sense amplifier of a current amplifier type for detection a change of a current flow and amplifying the current flow.
BRIEF DESCRIPTION OF THE DRAWINGS
These and other objects, features, aspects and advantages of the present invention will become more apparent from the following detailed description of the present invention when taken in conjunction with the accompanying drawings, in which:
FIG. 1 is a diagram showing a configuration of a conventional SRAM;
FIG. 2 is a timing chart of the operation of the conventional SRAM shown in FIG. 1;
FIG. 3 is a block diagram showing a configuration of a core circuit of a SRAM as a semiconductor memory device according to the first embodiment of the present invention;
FIG. 4 is a block diagram showing a configuration of each memory cell in the SRAM;
FIG. 5 is a block diagram showing a configuration of a sense amplifier in the SRAM;
FIG. 6 is a circuit diagram showing another configuration of the sense amplifier in the SRAM;
FIG. 7 is a timing chart of the operation of the SRAM shown in FIG. 3;
FIG. 8 is a circuit diagram showing mainly a configuration of each current source load for each pair of bit lines BL and bBL in a SRAM according to the second embodiment of the present invention;
FIG. 9 is a block diagram showing a core circuit of a configuration of a SRAM according to the third embodiment of the present invention; and
FIG. 10 is a block diagram showing a configuration of a core circuit of a SRAM according to the fourth embodiment of the present invention.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Other features of this invention will become apparent through the following description of preferred embodiments which are given for illustration of the invention and are not intended to be limiting thereof.
First Embodiment
FIG. 3 is a block diagram showing a configuration of a SRAM as a semiconductor memory device according to the first embodiment of the present invention. In particularly, FIG. 3 shows a configuration of a memory core circuit of the SRAM. In FIG. 3, the reference number 1 designates a memory cell array in the memory core circuit of the SRAM. The reference character MC designates each memory cell to which a data item is written and then stored, and from which the data item is readout. The reference characters WL0, WL1, . . . , and WLn indicate word lines. The reference characters BL and bBL denote each bit line pair (B10 and bBL0, BL1 and bBL1, . . . ).
The reference number 2 denotes a sense amplifier (SA) through which a data item stored in the memory cell MC is transferred to the outside of the memory core circuit. The reference number 3 indicates a write-in driver through which a data item is written into the memory cell MC through a bit line pair BL and bBL. This write-in driver is made up of an inverter I3, a NOR gate G1 and an inverter I1, and a NOR gate G2 and an inverter I2.
The reference number 4 designates a bit line load for a bit line BL (BL0, BL1, . . . ) in each bit line pair (BL and bBL). The reference number 4′ designates a bit line load for a bit line bBL (bBL0, bBL1, . . . ) in each bit line pair (BL and bBL). Each of the bit line loads 4 and 4′ is made up of a PMOS transistor and an inverter. For example, the bit line load 4 for the bit line BL0 in the bit line pair (BL0 and bBL0) is made up of a PMOS transistor QP1 and an inverter I4. Similarly, the bit line load 4′ for the bit line bb10 in the bit line pair (BL0 and bBL0) is made up of a PMOS transistor QP2 and an inverter I4.
The reference number 5 indicates a column gate which is placed for each bit line pair BL and bBL. In the configuration shown in FIG. 1, each memory cell MC is placed at each cross point between the word line WL (WL0, WL1, . . . , and WLn) and bit line pairs BL and bBL (BL0 and bBL0, BL1 and bBL1, . . . ), and one end of the bit line pair BL and bBL is connected to the PMOS transistors QP1 and QP2 in the bit line loads 4 and 4′, respectively. Other ends of which is connected to the data line pair DL and bDL through the corresponding column gate 5.
The data line pair DL and bDL is connected to the write-in driver 3 for data write-in operation, and also connected to the sense amplifier 2 for data readout operation.
A description will be given of the operation of the semiconductor memory device according to the first embodiment.
FIG. 4 is a block diagram showing a configuration of each memory cell MC.
As shown in FIG. 4, each memory cell MC is a SRAM cell which comprises a flip flop in which inverters I11 and I12 are connected to each other. Nodes N1 and N2 are connected to the bit lines bBL and BL, respectively, in the bit line pair through NMOS transistors QN11 and QN12 as transfer gates, respectively. These NMOS transistors QN11 and QN12 are driven by the level of the word line WL.
In the configuration of the SRAM according to the first embodiment shown in FIG. 3, each of the bit line loads 4 and 4′ comprises the PMOS transistor and the inverter, such as the PMOS transistor QP1 and the inverter I4, and the PMOS transistor QP2 and the inverter I4, for example. The sources of the PMOS transistors QP1 and QP2 are connected to the power source Vcc and the drains of which are connected to the bit lines BL and bBL, respectively. The inverter I4 inverts the level of the voltage potential of the corresponding bit line BL or bBL and provides the inverted one to the gate of the corresponding PMOS transistor. Accordingly, the bit line loads 4 and 4′ have no control terminal for its operation to which a control signal is supplied from a control means (not shown). The ON/OFF operation of the PMOS transistor QP (QP1, QP2 . . . . ) in the bit line loads 4 and 4′ is controlled automatically according to the voltage potential of each of the bit lines BL and bBL during the write recovery operation. The operation of the bit line loads 4 and 4′ causes to increase the write recovery operation. This will be explained in detail.
The write-in driver 3 connected to both the data lines DL and bDL (DL0, bDL0, DL1, bDL1, . . . ) mainly comprises the NOR gates G1 and G2 whose operation is controlled by the write enable signal /WE.
The inverted data /DIN which is obtained through the inverter I3 and the write-in data DIN are supplied to the NOR gates G1 and G2, respectively. Accordingly, when the level of the write enable signal /WE is the L level, the data lines DL and bDL in the data line pair have opposite levels to each other according to the level (H level or L level) of the write-in data DIN. Thus, the complementary signals (for example, H level and L level, or L level and H level) are transferred to the bit lines BL and bBL in the bit line pair through the data lines DL and bDL.
It is preferred to use a sense amplifier of a current amplifier type having a rapidly amplifying function as the sense amplifier (SA) 2 which is connected to the data lines DL and bDL in the data line pair.
FIG. 5 is a block diagram showing a practical configuration of the sense amplifier (SA) 2 as the current amplifier type. In FIG. 5, PMOS transistors QP31 and QP33 are connected in series between a sense input node SAIN1 connected to one bit line BL in the bit line pair and a ground source Vss. In addition, PMOS transistors QP32 and QP34 are connected in series between a sense input node SAIN2 connected to other bit line bBL and a ground source Vss. An activation signal /SA for the sense amplifier (SA) 2 is supplied to both the gates of the PMOS transistors QP33 and QP34. The gate of the PMOS transistor QP31 is connected to the connection node N12 through which the PMOS transistors QP32 and QP34 are connected to each other. Similar to this, the gate of the PMOS transistor QP32 is connected to the connection node N11 through which the PMOS transistors QP31 and QP33 are connected to each other. These connection nodes N11 and N12 become output nodes SAOUT1 and SAOUT 2 of the sense amplifier (SA) 2.
In the sense amplifier (SA) 2 shown in FIG. 5, the difference of the currents supplied to the sense input nodes SAIN1 and SAIN2 according to a data item stored in a selected memory cell MC becomes the difference of the voltage potentials of the nodes N11 and N12. This voltage potential is feedback in positive direction to the PMOS transistors QP31 and QP32 in order to amplify the difference of the currents. Thereby, the outputs of the H level and the L level are transferred through the sense output nodes SAOUT1 and SAOUT2.
FIG. 6 is a circuit diagram showing another configuration of the sense amplifier (SA) 2 as the current amplifier type. In the configuration of the sense amplifier shown in FIG. 6, current detection circuits 41 and 42 which form two current detection systems are formed between the two sense input nodes SAIN1 and SAIN2 and the ground source Vss. In addition, inverters 43 and 44 are also placed in order to invert the level of the output from the current detection circuits 41 and 42. One current detection circuit 41 comprises PMOS transistors QP41 and QP42 which form a current mirror circuit, and NMOS transistors QN41 and QN42 which are placed between the drains of both the PMOS transistors QP41 and QP42 and the ground source Vss. Similar to this configuration, the current detection circuit 42 comprises PMOS transistors QP43 and QP43 which form a current mirror circuit, and NMOS transistors QN43 and QN44 which are placed between the drains of both the PMOS transistors QP43 and QP44 and the ground source Vss.
In the current detection circuit 41, both the gate and the drain of the PMOS transistor QP41 which are connected to the sense input node SAIN1 are connected to each other. On the contrary, in the current detection circuit 42, both the gate and the drain of the PMOS transistor QP43 which are connected to the sense input node SAIN2 are connected to each other.
In addition, both the gate and the drain of the NMOS transistor QN42 in the current detection circuit 41 are connected to each other, and both the gate and the drain of the NMOS transistor QN44 in the current detection circuit 42 are connected to each other.
In the sense amplifier circuit (SA) having the configuration shown in FIG. 6, when the magnitude of the input current to be supplied to the sense input node SAIN1 is large, the voltage between the source and the drain of the PMOS transistor QP42 becomes large and the voltage potential of the node N41 drops, because the POS transistor QP42 controls so that the current flowing through the PMOS transistor QP41 also flows through the PMOS transistor QP42 itself. Thereby, a voltage potential of the node N41 drops.
On the contrary, when the magnitude of the input node to be supplied to the sense input node SAIN2, the voltage potential of the node N42 drops. The change of the voltage potential at each of the nodes N41 and N42 becomes the change of the voltage potential of the drain in each of the NMOS transistors QP41 and QP43, and then feedback to the PMOS transistors QP42 and QP44. As a result, one of the nodes N41 and N42 becomes the H level, and other becomes the L level according to the input data item. These levels are inverted by the inverters 43 and 44 and then output to the outside.
Next, a description will be given of the data write-in operation and the data recovery operation in the SRAM as the first embodiment.
When the level of the write enable signal /WE becomes the L level, the write-in driver 3 outputs the H level to one data line DL (or bDL) and the L level of the other data line bDL (or DL) according to the level of the write-in data item DIN. These complementary signals, the H level and the L level are transferred to the bit line pair including the bit lines BL and bBL selected by the corresponding column selector 5, and then written into the memory cell MC selected by the word lines WL (WL0, WL1, . . . , WLn). These write-in operation is the same as that of the conventional semiconductor memory device.
In the data write-in operation described above, when the H level is transferred to the bit line BL and the L level is supplied to the other bit line bBL in the bit line pair, the PMOS transistor QP1 connected to the bit line BL in the bit line load keeps the ON state (the active state), so that the level of the bit line BL is kept at the H level. Then, the voltage potential of the other bit line bBL drops toward the L level from the H level. When the level of the bit line bBL is lower than that of a threshold voltage of the inverter 14, the inverter 14 outputs the H level to the gate of the PMOS transistor QP2 in the bit line load 4′. Thereby, the PMOS transistor QP2 enters the OFF state. That is, the level of the bit line bBL drops approximately to the ground voltage Vss, because there is no current supply from the current source load 4 to the bit line bBL.
The write recovery operation can be performed by switching in level the write-in signal (/WE) to the H level. At this time, both the invertors I1 and I2 in the write-in driver 3 output the H level, because the write-in signal (/WE) has been switched to the H level. The H level of the data lines DL and bDL are transferred to the bit lines BL and bBL in each bit line pair. The voltage potential of the bit line bBL at the L level side increases by the output from the write-in driver 3. When the level of the bit line bBL is over the threshold voltage of the inverter I4 corresponding to the bit line bBL, the PMOS transistor QP2 enters the ON state. After this, the current is supplied from the power source voltage Vcc to the bit line bBL through the PMOS transistor QP2. This increase in speed the write recovery operation. That is, the write recovery operation in the SRAM according to the first embodiment is executed by the combination of the write-in driver 3 and the current source load 4 shown in FIG. 1. Even if the column gate 5 is closed before the completion of the write recovery operation, the write recovery operation can be executed continuously by the PMOS transistor QP2 unless the level of the bit line bBL is over the threshold voltage of the inverter I4.
As described above, according to the SRAM of the first embodiment, the timing of the write recovery operation is determined by the timing control of the write-in driver 3 automatically. Therefore it is not necessary to adjust the time elapse between both the timing of the control of the bit line loads 4 and 4′ for the bit lines and the timing of the write-in driver 3. This feature provides a large timing margin in the write recovery operation in the SRAM. In addition, it is possible to reduce the total power consumption of the SRAM, because the bit line loads 4 and 4′ for the bit lines whose level is changed to the L level during the data write-in operation and no waste current thereby flows.
On the other hand, levels of both the bit lines BL and bBL in each bit line pair are the H level at the initial state during the data readout operation, and the bit line loads 4 and 4′ are in the ON state. However, when the sense amplifier of the current amplify type having the configuration shown in FIG. 5 or FIG. 6 is used as the sense amplifier, it is possible to execute certainly the data sensing operation without setting the bit line loads 4 and 4′ to the OFF state. That is, it is possible to sense the bit lines BL and bBL in each bit line pair at a high speed without setting the bit line loads 4 and 4′ to the OFF state by detecting and amplifying the difference of the currents flowing to the sense amplifier (SA) 2 through both the bit lines BL and bBL in the bit line pair from the bit line loads 4 and 4′ according to the magnitude of the current which flows into the selected memory cell MC.
Second Embodiment
FIG. 8 is a circuit diagram showing mainly a configuration of each bit line load for each pair of bit lines BL and bBL in a SRAM according to the second embodiment of the present invention. The reference number 41 designates each bit line load for each bit line BL (BL0, BL1, . . . ), and the reference number 41′ denotes each bit line load for each bit line bBL (bBL0, bBL1, . . . ) in the SRAM of the second embodiment.
The bit line load 41 is made up of the inverter I4 and an inverter I5. The inverter I5 is made up of a PMOS transistor QP1 and a NMOS transistor QN5. The bit line load 41′ is made up of the inverter I4 and an inverter I5′. The inverter I5′ is made up of a PMOS transistor QP2 and a NMOS transistor QN6.
Next, a description will be given of the operation of the SRAM according to the second embodiment.
The bit line load 41 for the bit line BL (BL0, BL1, . . . ) includes the NMOS transistor QN5 in addition to the configuration of the bit line load 4 shown in FIG. 3. Similarly, the bit line load 41′ for the bit line bBL (bBL0, bBL1, . . . ) includes the NMOS transistor QN6 in addition to the configuration of the bit line load 4′ shown in FIG. 3.
The drains of the NMOS transistors QN5 and QN6 are connected to the bit line BL and the bit line bBL in each bit line pair, respectively. The sources of the NMOS transistors QN5 and QN6 are connected to the ground voltage Vss. The output of the inverter I4 is supplied to the gates of the NMOS transistors QN5 and QN6, and the PMOS transistors QP1 and QP2.
In the bit line loads 4 and 4′ in the SRAM according to the second embodiment, the PMOS transistor QP1 and NMOS transistor QN5 forms the inverter I5, and the PMOS transistor QP2 and NMOS transistor QN6 forms the inverter I5′.
The inverter I5 (made up of the PMPS transistor QP1 and the NMOS transistor QN5) and the inverter I4 are connected to each other, and the inverter I5′ (made up of the PMPS transistor QP2 and the NMOS transistor QN6) and the inverter I4 are also connected to each other. This configuration forms a flip flop.
The operation of the bit line loads 41 and 41′ in the SRAM of the second embodiment is basically the same as that of the SRAM according to the first embodiment. The difference between the first embodiment and the second embodiment is as follows:
In the bit line loads 41 and 41′, when the voltage potential of the bit line which is changed to the L level during the data write-in operation is not more than the threshold voltage of the inverter I4, the NMOS transistor QN5 or QN6 enters the ON state. That is, because the speed of the voltage transition of the bit line to the L level can be accelerated, it is possible to execute the data write-in operation at a high speed rate. Therefore the SRAM as the semiconductor memory device of the second embodiment is capable of performing the write-in operation at high speed in addition to the effect of the SRAM of the first embodiment.
Third Embodiment
FIG. 9 is a block diagram showing a configuration of a memory core circuit in a SRAM according to the third embodiment of the present invention. In FIG. 9, the reference number 74 designates a bit line load for a bit line BL (BL0, BL1, . . . ) in each bit line pair (B1 and bBL). The reference number 74′ designates a bit line load for a bit line bBL (bB10, bBL1, . . . ) in the bit line pair. Each of the bit line loads 74 and 74′ is the same in configuration as that of each of the bit line loads 4 and 4′ shown in FIG. 3, respectively.
The inverter I4 in the bit line loads 4 and 4′ in the SRAM shown in FIG. 3 has a voltage amplitude (Vcc-Vss) of the power source voltage potential Vcc to the ground source voltage potential Vss. On the contrary, a positive voltage Vss1 in the bit line loads 74 and 74′ in the SRAM of the third embodiment shown in FIG. 9 is slightly higher than that of the voltage Vss of the ground source. Other configuration components in the SRAM of the third embodiment are the same as those in the SRAM of the first embodiment shown in FIG. 3.
Next, a description will be given of the operation of the SRAM according to the third embodiment.
The configuration of the bit line loads 74 and 74′ can improve the higher data readout operation. That is, during the data readout operation, the memory cell MC inputs the current from one of the bit lines BL and bBL according to the level of the data to be read out, and inputs no current from other bit line. At this time, the current which flows into the activated sense amplifier (SA) 2 from the PMOS transistor QP1 and QP2 in the bit line loads 74 and 74′ through the bit lines BL and bBL in the bit line pair becomes i and (i−(i cell)) at the node in the bit lines BL and bBL in the bit line pair, respectively, where the input current to the memory cell MC is (i cell) only when the bit line loads 74 and 74′ are current sources to flow the constant current i.
In a practical case, the amount of the cell current (i cell) that flows through the PMOS transistor QP1 in the bit line load 74 is increased by Δi. That is, because a bias between the gate and the source of the PMOS transistor QP1 is a constant, the amount of the current Δi can be increased by increasing the voltage |Vds| between the drain and the source. As a result, the currents flowing to the sense amplifier (SA) 2 are i and (i+Δi−(i cell)). Therefore the current difference i−(i+Δi−(i cell)) is smaller than that in an ideal current source load.
On the contrary, the gate voltage to enter both the PMOS transistors QP1 and QP2 into the ON state is the positive voltage Vss1 in the SRAM according to the third embodiment. That is, these PMOS transistors QP1 and QP2 in the ON state act a pentode=s operation in a small area of the voltage |Vgs| of the gate and the source.
As a result, it is possible to suppress the increasing Δi of the current caused by the PMOS transistor QP1 when the cell current (i cell) flows through the PMOS transistor QP1. This can keep a large difference between the currents flowing from the bit line pair (BL and bBL) to the sense amplifier (SA) 2. As a result, it is possible to read out a data item from a memory cell MC at a high speed rate.
In addition, it is possible to use the voltage Vss1, as the low level power source for the inverter I4 in the bit line loads 41 and 41′ shown in FIG. 8, which is higher than that of the ground voltage Vss.
Fourth Embodiment
FIG. 10 is a block diagram showing a configuration of a memory score circuit of a SRAM according to the fourth embodiment of the present invention. In FIG. 10, the reference number 84 designates a bit line load for the bit line BL (BL0, BL1, . . . ) in each bit line pair. The bit line load 84 is made up of a NMOS transistor QN01, an inverter I41, and an inverter I42. The NMOS transistor QN01 is connected to the power source voltage Vcc, and the source of which is connected to the bit line BL (BL0, BL1, . . . ). The gate of the NMOS transistor QN01 is connected to the inverters I41 and I42 which are connected in series. The input side of the inverter I41 is connected to the bit line BL (BL0, BL1, . . . ).
The reference number 84′ designates a bit line load for the bit line bBL (bBL0, bBL1, . . . ) in each bit line pair. The bit line load 84′ is made up of a NMOS transistor QN02, the inverter I41, and the inverter I42. The NMOS transistor QN02 is connected to the power source voltage Vcc, and the source of which is connected to the bit line bBL (bBL0, bBL1 . . . ) The gate of the NMOS transistor QN02 is connected to the inverters I41 and I42 which are connected in series. The input side of the inverter I41 is connected to the bit line bBL (bBL0, bBL1, . . . ).
Other configuration components in the SRAM of the fourth embodiment shown in FIG. 10 are the same as those in the SRAM of the first embodiment shown in FIG. 3.
Next, a description will be given of the operation of the SRAM according to the fourth embodiment is as follows.
The NMOS transistor QN01 is driven through the gate thereof. The gate of the NMOS transistor QN1 is connected to the bit line BL (BL10, BL1, . . . ) through the two inverters I41 and I42 connected in series. The voltage potential of the bit line BL (B10, BL1, . . . ) is transferred to the gate of the NMOS transistor QN01 through the two inverters I41 and I42 connected in series.
Similarly, the NMOS transistor QN02 is driven through the gate thereof. The gate of the NMOS transistor QN2 is connected to the bit line bBL (bBL0, bBL1 . . . ) through the two inverters I41 and I42 connected in series. The voltage potential of the bit line bBL (bBL0, bBL1, . . . ) is transferred to the gate of the NMOS transistor QN02 through the two inverters I41 and I42 connected in series.
Thus, according to the configuration shown in FIG. 10, it is also possible to perform the write recovery operation automatically after the completion of the data write-in operation based on the same principle of the first embodiment. However, the charged level of each of the bit lines BL and bBL is limited by the threshold voltage of the NMOS transistors QN01 and QN02 in the bit line loads 84 and 84′.
Although the above descriptions, the SRAM is used as the semiconductor memory, the present invention is not limited by these configuration described above. For example, the concept of the present invention can be applied to various semiconductor memories which are incorporate memory cells to store data based on the presence of a current through bit lines, such as a nonvolatile semiconductor memory device.
While the above provides a full and complete disclosure of the preferred embodiments of the present invention, various modifications, alternate constructions and equivalents may be employed without departing from the true spirit and scope of the invention. Therefore the above description and illustration should not be construed as limiting the scope of the invention, which is defined by the appended claims.

Claims (12)

What is claimed is:
1. A semiconductor memory device comprising:
a memory cell array having memory cells which are located at cross points of word lines and bit lines;
a write-in driver for performing data write-in operation to transfer write-in data to said memory cells through said bit lines, and for performing a recovery operation by supplying a high level voltage to said bit lines when a voltage potential of said bit lines drops to a low voltage potential after a completion of said data write-in operation;
a sense amplifier for detecting and amplifying data read from said memory cells through said bit lines; and
bit line loads whose operation being controlled by voltage potential of said bit lines, each bit line load being placed between a first power source and each of said bit lines, and said bit line load entering an ON state in order to accelerate said recovery operation when said voltage potential of said bit lines reach to a predetermined voltage potential during said recovery operation performed by said write-in driver.
2. The semiconductor memory device as claimed in claim 1, wherein each of said bit line loads comprises:
a PMOS transistor whose drain is connected to said corresponding bit line and whose source is connected to said first power source; and
an inverter connected between said corresponding bit line and a gate of said PMOS transistor for inverting a voltage potential of said corresponding bit line and transferring said inverted voltage potential to said gate of said PMOS transistor.
3. The semiconductor memory device as claimed in claim 2, wherein each of said bit memory is a SRAM cell, and said sense amplifier is a sense amplifier of a current amplifier type for detecting a change of a current flow and amplifying said current flow.
4. The semiconductor memory device as claimed in claim 2, each of said bit line loads further comprises a NMOS transistor connected to said PMOS transistor in series,
wherein a drain of said NMOS transistor is connected to both said corresponding bit line and said drain of said PMOS transistor, and a source of said NMOS transistor is connected to a second power source, and a gate of said NMOS transistor is connected to said inverter in order to supply an output of said inverter.
5. The semiconductor memory device as claimed in claim 4, said inverter in each of said bit line loads is a type in which an amplitude of an output voltage is suppressed.
6. The semiconductor memory device as claimed in claim 5, wherein each of said memory cells is a SRAM cell, and said sense amplifier is a sense amplifier of a current amplifier type for detecting a change of a current flow and amplifying said current flow.
7. The semiconductor memory device as claimed in claim 4, wherein each of said memory cells is a SRAM cell, and said sense amplifier is a sense amplifier of a current amplifier type for detecting a change of a current flow and amplifying said current flow.
8. The semiconductor memory device as claimed in claim 2, said inverter in each of said bit line has a type in which an amplitude of an output voltage is suppressed.
9. The semiconductor memory device as claimed in claim 8, wherein each of said memory cells is a SRAM cell, and said sense amplifier is a sense amplifier of a current amplifier type for detecting a change of a current flow and amplifying said current flow.
10. The semiconductor memory device as claimed in claim 1, wherein each of said bit line loads comprises:
a NMOS transistor whose drain is connected to a first power source, and whose source is connected to said corresponding bit line;
a first inverter whose input is connected to said corresponding bit line; and
a second inverter whose input is connected to an output of said first inverter and whose output is connected to a gate of said NMOS transistor.
11. The semiconductor memory device as claimed in claim 1, wherein each of said memory cells is a SRAM cell, and said sense amplifier is a sense amplifier of a current amplifier type for detection a change of a current flow and amplifying said current flow.
12. The semiconductor memory device as claimed in claim 10, wherein each of said memory cells is a SRAM cell, and said sense amplifier is a sense amplifier of a current amplifier type for detecting a change of a current flow and amplifying said current flow.
US09/595,118 1999-06-18 2000-06-16 Semiconductor memory device capable of automatically controlling bit-line recovery operation Expired - Fee Related US6366492B1 (en)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
JP17321899A JP3967493B2 (en) 1999-06-18 1999-06-18 Semiconductor memory device
JP11-173218 1999-06-18

Publications (1)

Publication Number Publication Date
US6366492B1 true US6366492B1 (en) 2002-04-02

Family

ID=15956330

Family Applications (1)

Application Number Title Priority Date Filing Date
US09/595,118 Expired - Fee Related US6366492B1 (en) 1999-06-18 2000-06-16 Semiconductor memory device capable of automatically controlling bit-line recovery operation

Country Status (3)

Country Link
US (1) US6366492B1 (en)
JP (1) JP3967493B2 (en)
KR (1) KR100378270B1 (en)

Cited By (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US6738301B2 (en) 2002-08-29 2004-05-18 Micron Technology, Inc. Method and system for accelerating coupling of digital signals
US20040240300A1 (en) * 2003-05-29 2004-12-02 Tae-Joong Song Semiconductor memory device with selectively connectable segmented bit line member and method of driving the same
US20060203583A1 (en) * 2005-03-09 2006-09-14 Kabushiki Kaisha Toshiba Semiconductor memory and data read method of the same
US20080192550A1 (en) * 2007-02-13 2008-08-14 Micron Technology, Inc. Circuits, systems and methods for driving high and low voltages on bit lines in non-volatile memory
US20100097857A1 (en) * 2007-04-10 2010-04-22 Raul-Adrian Cernea Predictive Programming in Non-Volatile Memory
US20100202218A1 (en) * 2009-02-06 2010-08-12 Thomas Nirschl System and Method for Level Shifter
US8379436B2 (en) 2011-01-28 2013-02-19 Kabushiki Kaisha Toshiba Semiconductor memory device

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4586166A (en) 1983-08-31 1986-04-29 Texas Instruments Incorporated SRAM with improved sensing circuit
US4951252A (en) * 1988-10-25 1990-08-21 Texas Instruments Incorporated Digital memory system
US5828614A (en) * 1997-04-07 1998-10-27 Cypress Semiconductor Corp. Memory cell sensing method and circuitry for bit line equalization
US5966319A (en) * 1997-07-15 1999-10-12 Mitsubishi Denki Kabushiki Kaisha Static memory device allowing correct data reading

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4586166A (en) 1983-08-31 1986-04-29 Texas Instruments Incorporated SRAM with improved sensing circuit
US4951252A (en) * 1988-10-25 1990-08-21 Texas Instruments Incorporated Digital memory system
US5828614A (en) * 1997-04-07 1998-10-27 Cypress Semiconductor Corp. Memory cell sensing method and circuitry for bit line equalization
US5966319A (en) * 1997-07-15 1999-10-12 Mitsubishi Denki Kabushiki Kaisha Static memory device allowing correct data reading

Cited By (18)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US6738301B2 (en) 2002-08-29 2004-05-18 Micron Technology, Inc. Method and system for accelerating coupling of digital signals
US20040196729A1 (en) * 2002-08-29 2004-10-07 Kirsch Howard C Method and system for accelerating coupling digital signals
US6925019B2 (en) 2002-08-29 2005-08-02 Micron Technology, Inc. Method and system for accelerating coupling of digital signals
US20040240300A1 (en) * 2003-05-29 2004-12-02 Tae-Joong Song Semiconductor memory device with selectively connectable segmented bit line member and method of driving the same
US6952363B2 (en) * 2003-05-29 2005-10-04 Samsung Electronics Co., Ltd. Semiconductor memory device with selectively connectable segmented bit line member and method of driving the same
US20060203583A1 (en) * 2005-03-09 2006-09-14 Kabushiki Kaisha Toshiba Semiconductor memory and data read method of the same
US7248520B2 (en) * 2005-03-09 2007-07-24 Kabushiki Kaisha Toshiba Semiconductor memory and data read method of the same
US8045395B2 (en) 2007-02-13 2011-10-25 Micron Technology, Inc. Circuits, systems and methods for driving high and low voltages on bit lines in non-volatile memory
US7719899B2 (en) 2007-02-13 2010-05-18 Micron Technology, Inc. Circuits, systems and methods for driving high and low voltages on bit lines in non-volatile memory
US20100220530A1 (en) * 2007-02-13 2010-09-02 Micron Technology, Inc. Circuits, systems and methods for driving high and low voltages on bit lines in non-volatile memory
US20080192550A1 (en) * 2007-02-13 2008-08-14 Micron Technology, Inc. Circuits, systems and methods for driving high and low voltages on bit lines in non-volatile memory
US8363490B2 (en) 2007-02-13 2013-01-29 Micron Technology, Inc. Circuits, systems and methods for driving high and low voltages on bit lines in non-volatile memory
US8760933B2 (en) 2007-02-13 2014-06-24 Micron Technology, Inc. Circuits, systems, and methods for driving high and low voltages on bit lines in non-volatile memory
US20100097857A1 (en) * 2007-04-10 2010-04-22 Raul-Adrian Cernea Predictive Programming in Non-Volatile Memory
US20100202218A1 (en) * 2009-02-06 2010-08-12 Thomas Nirschl System and Method for Level Shifter
US8130558B2 (en) * 2009-02-06 2012-03-06 Infineon Technologies Ag System and method for level shifter
US8437175B2 (en) 2009-02-06 2013-05-07 Infineon Technologies Ag System and method for level shifter
US8379436B2 (en) 2011-01-28 2013-02-19 Kabushiki Kaisha Toshiba Semiconductor memory device

Also Published As

Publication number Publication date
JP2001006372A (en) 2001-01-12
KR100378270B1 (en) 2003-03-29
JP3967493B2 (en) 2007-08-29
KR20010069208A (en) 2001-07-23

Similar Documents

Publication Publication Date Title
US7447058B2 (en) Write margin of SRAM cells improved by controlling power supply voltages to the inverters via corresponding bit lines
KR100373223B1 (en) Semiconductor device
US6380592B2 (en) Low power RAM memory cell using a precharge line pulse during write operation
KR100391020B1 (en) Data holding circuit
US5040146A (en) Static memory cell
US7254077B2 (en) Circuit and method for high speed sensing
US7023749B2 (en) Semiconductor integrated circuit device
US5966319A (en) Static memory device allowing correct data reading
JPH02201797A (en) Semiconductor memory device
US6426905B1 (en) High speed DRAM local bit line sense amplifier
US6282137B1 (en) SRAM method and apparatus
US20120257442A1 (en) Semiconductor memory device
US6865129B2 (en) Differential amplifier circuit with high amplification factor and semiconductor memory device using the differential amplifier circuit
US5742552A (en) Timing control for clocked sense amplifiers
EP0306519B1 (en) Current sensing differential amplifier
US6366492B1 (en) Semiconductor memory device capable of automatically controlling bit-line recovery operation
JP3188634B2 (en) Data holding circuit
US6707741B1 (en) Current steering reduced bitline voltage swing, sense amplifier
EP0827151B1 (en) Self cut-off type sense amplifier operable over a wide range of power supply voltages
KR950005171B1 (en) Current mirror amplifier and driving method
EP0411818B1 (en) Current sensing amplifier for a memory
US5777934A (en) Semiconductor memory device with variable plate voltage generator
US6316812B1 (en) Static semiconductor memory device with expanded operating voltage range
US5438551A (en) Semiconductor integrated circuit device
US8411490B2 (en) Sense amplifier for static random access memories

Legal Events

Date Code Title Description
AS Assignment

Owner name: KABUSHIKI KAISHA TOSHIBA, JAPAN

Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:KAWASUMI, ATSUSHI;REEL/FRAME:010888/0974

Effective date: 20000609

CC Certificate of correction
FPAY Fee payment

Year of fee payment: 4

REMI Maintenance fee reminder mailed
LAPS Lapse for failure to pay maintenance fees
STCH Information on status: patent discontinuation

Free format text: PATENT EXPIRED DUE TO NONPAYMENT OF MAINTENANCE FEES UNDER 37 CFR 1.362

FP Lapsed due to failure to pay maintenance fee

Effective date: 20100402