US5053650A

US5053650A - Monolithic semiconductor integrated circuit device having current adjusting circuit

Info

Publication number: US5053650A
Application number: US07/457,308
Authority: US
Inventors: Yuichi Ohkubo; Satoru Sekiguchi; Toshihiko Watanabe; Nobuaki Yoneya
Original assignee: Hitachi Ltd; Hitachi Microcomputer Engineering Ltd
Current assignee: Hitachi Microcomputer System Ltd; Hitachi Ltd
Priority date: 1988-12-27
Filing date: 1989-12-26
Publication date: 1991-10-01
Anticipated expiration: 2009-12-26
Also published as: KR900011002A; KR0137869B1; JPH02174414A

Abstract

A monolithic semiconductor integrated circuit device includes a differentially operative circuit section, an amplifying element connected to define a current flowing in the differentially operative circuit section and a circuit for adjusting a current flowing in the amplifying element to thereby compensate for variations of electric characteristics from one semiconductor device to another. The current adjusting circuit includes at least one amplifying element and a load resistance for the amplifying element in the current adjusting circuit. The load resistance has a structure suitable for a trimming operation to adjustably determine the resistance value of the load resistance. The amplifying elements are in a current mirror circuit connection with their control electrodes being concerned with each other so that the electric current flowing in the current path between the current receiving and delivering electrodes of the amplifying element connected to define the current flowing in the differentially operative circuit section is controlled by the adjustably determined resistance of the load resistance.

Description

BACKGROUND OF THE INVENTION

The present invention relates to a semiconductor integrated circuit device, and more particularly to a monolithic semiconductor integrated circuit device including an audio signal processing circuit.

When circuit elements of a circuit dealing with an audio signal, for example a filter circuit, are incorporated in a semiconductor integrated circuit, a time constant of the filter circuit has manufacturing variations. For example, the resistance value of a resistor element has a variation of about ±25% and the capacitance value of a capacitor has a variation of about ±30%. As a result, the time constant of a time constant circuit including such resistor element and capacitor has a large variation of about ±60%. Especially, in the case where two or more filter circuits having such large variations are involved in a semiconductor integrated circuit, the possession of the above-mentioned variation by each filter circuit does not give assurance that no reliability for the electric characteristic of the overall system is lost.

A basic circuit for a filter has been disclosed in, for example, "PRACTICAL ELECTRONIC CIRCUIT HANDBOOK (2)" published by CQ Publishing Co., Ltd. on Oct. 20, 1975, pp. 281-289. In the active filter circuit, time constant circuit elements including a capacitor C and a resistor R are constructed by external parts in order to suppress the variation of a cut-off frequency of the filter circuit.

On the other hand, in order to improve the high-frequency performance of a transistor included in an integrating circuit, JP-A-63-193710 (laid open on Aug. 11, 1988) has proposed to make an emitter current of the transistor as large as possible for the purpose of increasing a transition frequency of the transistor. The proposed integrated circuit includes a differential amplifier and is provided with a terminal for supplying a signal which controls the emitter current of a transistor forming the differential amplifier.

SUMMARY OF THE INVENTION

An object of the present invention is to provide a monolithic semiconductor integrated circuit device including an audio signal processing circuit, in which variations of electric characteristics from one chip to another can be compensated for with a simple construction.

Another object of the present invention is to provide a monolithic semiconductor integrated circuit device in which compensation for variations of filter characteristics can be made with a simple construction.

According to one aspect of the present invention, a monolithic semiconductor integrated circuit device includes a differentially operative circuit section, an amplifying element connected to define a current flowing in the differentially operative circuit section and a circuit for adjusting a current flowing in the amplifying element to thereby compensate for variations of electric characteristics from one semiconductor device to another. The current adjusting circuit includes at least one amplifying element and a load resistance for the amplifying element in the current adjusting circuit. The load resistance has a structure suitable for a trimming operation to adjustably determine the resistance value of the load resistance. The amplifying elements are in a current mirror circuit connection with their control electrodes being connected with each other so that the electric current flowing in the current path between the current receiving and delivering electrodes of the amplifying element connected to define the current flowing in the differentially operative circuit section is controlled by the adjustably determined resistance of the load resistance.

According to another aspect of the present invention, a monolithic semiconductor integrated circuit device includes a plurality of filter circuits. Each filter circuit includes a pair of differentially operative transistors and a cut-off frequency of the filter circuit is changed by controlling an emitter current of the transistor pair. Amplifying elements dealing with the emitter current are in a current mirror circuit connection which receives an adjustable control current through a resistor-trimming operation. The emitter currents of the differentially operative transistor pairs are adjusted en bloc through the resistor-trimming operation so that the cut-off frequencies of the respective filter circuits are corrected en bloc.

With the above construction, the time constants of the respective filter circuits can be readily made identical with each other by the resistor-trimming operation at one location since the characteristics of circuit elements formed in one chip have substantially the same deviation.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic circuit diagram of a monolithic semiconductor integrated circuit device according to an embodiment of the present invention.

FIGS. 2A to 2D are diagrams showing examples of the structure of load resistor means for amplifying means in a current adjusting circuit shown in FIG. 1.

FIGS. 3A and 3B are a circuit diagram and a block diagram of a monolithic semiconductor integrated circuit device including a filter circuit according to an embodiment of the present invention.

FIGS. 4A and 4B are a circuit diagram and a block diagram of a monolithic semiconductor integrated circuit device including another filter circuit according to another embodiment of the present invention.

FIG. 4C shows a gain versus frequency characteristic useful for explaining the operation of the device shown in FIGS. 4A and 4B.

FIGS. 5A and 5B are a circuit diagram and a block diagram of a monolithic semiconductor integrated circuit device including a further filter circuit according to a further embodiment of the present invention.

FIGS. 6A and 6B are a circuit diagram and a block diagram of a monolithic semiconductor integrated circuit device including a still further filter circuit according to a still further embodiment of the present invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 shows a circuit diagram of a main part of a monolithic semiconductor integrated circuit according to an embodiment of the present invention. Circuit blocks and circuit elements shown in the figure are formed by means of known semiconductor circuit fabrication techniques on a single semiconductor substrate which may be a monocrystalline silicon substrate.

In the present embodiment, the electric characteristics of filter circuits indicated by blocks (that is, a low-pass filter F₁, a high-pass filter F₂ and a notch filter F₃), for example, the cut-off frequencies (or time constants) thereof can be respectively adjusted by currents I₁, I₂ and I₃ which are set by a current adjusting circuit CA. Detailed explanation of such filter circuits will be made later.

The current adjusting circuit CA includes at least one amplifying element (first amplifying element) or NPN transistors Q₂₀ and Q₂₁ in the shown example, a collector load resistor R₉₀ for the transistor Q₂₀ and emitter load resistors R₅ to R₈ for the transistor Q₂₀. The collector and base of the transistor Q₂₀ are connected with the base and emitter of the transistor Q₂₁, respectively. The collector of the transistor Q₂₁ is connected with a terminal of a reference voltage V_ref. The collector resistor R₉₀ is connected between the collector of the transistor Q₂₀ and the terminal of the reference voltage V_ref, and the emitter resistors R₅ and R₈ are connected in parallel between the emitter of the transistor Q₂₀ and a common potential (for example, grounded) line.

The currents I₁ to I₃ flowing in the above-mentioned filter circuits are formed by transistors Q₂₂ to Q₂₄, respectively. The emitters of the transistors Q₂₂ to Q₂₄ are connected with emitter resistors R₁₀ to R₁₂, respectively. The bases of the transistors Q₂₂ to Q₂₄ are commonly connected and those transistors Q₂₂ to Q₂₄ are in a current mirror circuit connection together with the transistor Q₂₀. Thereby, the currents I₁ to I₃ flowing in the respective filter circuits are set by the current adjusting circuit CA. A field-effect transistor may be used in place of the bipolar transistor Q₂₀. In that case, the transistor Q₂₁ is not required.

In the above construction, the emitter of the transistor Q₂₀ is provided with the emitter resistors or trimming resistors R₅ to R₈ connected, for example, in parallel with each other for adjustably changing an emitter current flowing in the transistor Q₂₀, or in other words, for adjustably changing the currents I₁ to I₃ flowing in the above-mentioned filter circuits. The resistors R₅ to R₈ are not limited to the shown parallel connection form but may be provided with another connection form.

The parallel connection line of the trimming resistors R₅ to R₈ is selectively cut off by, for example, irradiation thereof with a laser beam. For example, a state in which all of the resistors R₅ to R₈ are connected in parallel with each other exhibits the smallest combined resistance, and the combined resistance value is made large by properly cutting off the connection line of the resistors R₅ to R₇. When the resistance value of the emitter resistance of the transistor Q₂₀ is thus made large, the voltage dividing ratio of the emitter resistance to the collector resistor R₉₀ is changed to increase a base potential V_c of the transistor Q₂₀, thereby increasing the currents I₁ to I₃ in a constant ratio.

Resistors R₉₁ to R₉₃ may be connected with the collector resistor R₉₀ of the transistor Q₂₀ in a parallel form or another form. In this case, when the combined resistance value is made large by selectively cutting off a connection line of the resistors R₉₀ to R₉₃, the resistance ratio of the combined resistance on the collector side of the transistor Q₂₀ to the combined resistance on the emitter side thereof is changed to lower the base potential V_c of the transistor Q₂₀, thereby decreasing the currents I₁ to I₃ in a constant ratio. Namely, it is possible not only to increase the potential V_c but also to lower the same.

FIGS. 2A and 2B show one example of that structure of the collector load resistor means and/or emitter load resistor means of the transistor Q₂₀ shown in FIG. 1 which is suitable for a trimming operation. In FIG. 2A, the load resistor means includes a plurality of resistor elements r₁₁ to r_1n (the resistance values of which may be the same or different) and two conductors l₁ and l₂ for making respective common connections of opposite ends of the resistor elements. For a trimming operation on the load resistor means, the conductor l₂ is cut off at one of a plurality of locations indicated by one-dotted chain lines by use of, for example, a laser beam. As a result, the resistance value of the load resistor means is adjusted. FIG. 2B is an electrical connection diagram of the resistor elements shown in FIG. 2A.

FIGS. 2C and 2D show an example of another structure of the load resistor means of the transistor Q₂₀. In FIG. 2C, the load resistor means includes a plurality of resistor elements r₂₁ to r_2n (the resistance values of which may be the same or different), a conductor l₃ for commonly connecting one-side ends of the resistor elements r₂₁ to r_2n, conductors l₄, l₅ and l₆ for connecting the other-side ends of two adjacent resistor elements. A trimming operation on the load resistor means is achieved by cutting off the conductor l₃ at one or more of a plurality of locations indicated by one-dotted chain lines by use of, for example, a laser beam. FIG. 2D is an electrical connection diagram of the resistor elements shown in FIG. 2C.

The structure shown in FIG. 2A and the structure shown in FIG. 2C may be used in combination with each other.

As is apparent from the foregoing, various configurations using a plurality of resistor elements and a plurality of conductors can be considered as the structure of the load resistor means of the transistor Q₂₀ which is suitable for the resistor-trimming operation.

FIG. 3A shows a circuit diagram of a monolithic semiconductor integrated circuit device including a low-pass filter and high-pass filter circuit, and FIG. 3B shows a block diagram of the device shown in FIG. 3A.

Constant current sources I₀ are provided between the emitters of PNP type input transistors Q₁ and Q₂ and a power source voltage C_cc, respectively. The bases of the transistors Q₁ and Q₂ are connected with first and second input terminals, respectively. A resistance R₁ for obtaining an input signal current is provided between the emitters of the input transistors Q₁ and Q₂. The collectors of the transistors Q₁ and Q₂ are connected with diode-connected NPN transistors Q₃ and Q₄ which form active load circuits. The common emitter (or cathode) of the transistors Q₃ and Q₄ is coupled with a grounded potential point through a diode-connected NPN transistor Q₅. The above-mentioned circuit elements form a differential input stage A₁ or a first differentially operative circuit.

Output voltages formed by the diode-connected transistors Q₃ and Q₄ in which collector currents of the transistors Q₁ and Q₂ flow are supplied to the bases of differentially operative NPN transistors Q₆ and Q₇ which form a controllable voltage-current conversion circuit VVIC or a second differentially operative circuit.

The controllable voltage-current conversion circuit VVIC includes the differentially operative transistors Q₆ and Q₇, PNP transistors Q₈ and Q₉ which are provided in a current mirror circuit connection on the collector sides of the differentially operative transistors Q₆ and Q₇ and form active load circuits, and an adjustable current source circuit, as mentioned above, which includes a transistor Q₂₂ forming a second amplifying element causing the above-mentioned current I₁ to flow in the common emitter of the differentially operative transistors Q₆ and Q₇ and an emitter resistor R₁₀ of the transistor Q₂₂. The base of the transistor Q₂₂ is connected with the base of the transistor Q₂₀ of the current adjusting circuit CA so that a current mirror circuit connection is provided as a whole. A signal current formed by the differentially operative transistors Q₆ and Q₇ of the controllable voltage-current conversion circuit VVIC serves as a charging/ discharging current for a capacitor C₁ which will be explained later, and the current signal is fed back to the base of the transistor Q₂ which is a feedback terminal (or the second input terminal) of the above-mentioned differential input stage A₁. The capacitor C₁ forming a filter circuit is provided between the feedback terminal and an earth potential. The capacitor C₁ is incorporated in the semiconductor integrated circuit. For example, the capacitor C₁ is constructed by an inter-layer insulating film which has no voltage dependency and two electrodes which have the interlayer insulating film sandwiched therebetween. The other various methods of forming a capacitor in a semiconductor integrated circuit are known and any one thereof suitable for the filter circuit in the present embodiment can be used.

The differential input stage A₁ operates such that an input signal V_in supplied to the base of the transistor Q₁ which is a non-inverted input (+) or first input terminal of the differential input stage and a DC potential of the base of the transistor Q₂ which is an inverted input (-) or second input terminal of the differential input stage become equal to each other. Since the capacitor C₁ is provided at the noninverted input (-) of the input stage A₁, the input signal V_in is AC-wise attenuated in its high frequency region. Accordingly, a low-pass filter circuit can be constructed by providing a proper buffer circuit (or a high impedance element) at a junction point of the capacitor C₁ and the collectors of the transistors Q₈ and Q₆ so that the inverted input (-) signal is outputted through the buffer circuit.

In the present embodiment, the output of the low-pass filter circuit is inputted to a buffer circuit A₃. An output of the buffer circuit A₃ and the input signal V_in are subtracted from each other by a subtracter circuit constructed by an operational amplifier circuit A₂ and resistors R_21a, R_21b, R_22a and R_22b, thereby forming a high-pass filter output signal V_out.

Provided that the ON-resistance value of each of the transistors Q₃ and Q₄ is r_d1 and the resistance value of each of the emitter resistances of the transistors Q₆ and Q₇ is r_e1, there are satisfied a relation of r_d1 =(kT/q)÷I₀ and a relation of r_e1 =(kT/q)÷I₁ /2. Here, kT/q is about 26 mV. Since the current I₁ can be changed by a resistor-trimming operation as mentioned above, it is possible to correspondingly change the resistance r_e1.

In the circuit shown in FIG. 3A, when the input signal V_in is inputted, a current of (V_in -V_x)/R₁ is outputted from the collector of the transistor Q₁ and a phase-inverted current thereof is outputted from the collector of the transistor Q₂. V_x is a voltage on a junction point of the capacitor C₁ and the base of the transistor Q₂. The diode-connected transistors Q₃ and Q₄ serve as loads so that signal voltages outputted from the collectors of the transistors Q₃ and Q₄ result in r_d1 ·(V_in -V_x)/R₁. These signal input voltages supplied to the bases of the differentially operative transistors Q₆ and Q₇ which form the controllable voltage-current conversion circuit VVIC.

Since the emitter resistance of each of the differentially operative transistors Q₆ and Q₇ is r_e1, as mentioned above, the following current is obtained from the collectors of the transistors Q₆ and Q₇. Namely, in the controllable voltage-current conversion circuit VVIC, a signal current is formed in such a manner that the above-mentioned signal voltages r_d1 ·(V_in -V_x)/R₁ are applied to the emitter resistances r_e1 of the differentially operative transistors Q₆ and Q₇. Accordingly, the signal current formed is [r_d1 ·(V_in -V_x)/R₁ ]÷r_e1. This current is supplied to the capacitor C₁. The capacitor C₁ serves to attenuate a high frequency component of the supplied current signal. Therefore, there is obtained the following signal voltage V_x :

V.sub.x =[(V.sub.in -V.sub.x)/R.sub.1 ]×(r.sub.d1 jωC.sub.1)(1)

On the other hand, in the subtracter circuit A₂, a relation of

V.sub.in -V.sub.x /2+V.sub.out -V.sub.x /2=0               (2)

is satisfied (on the assumption that R_21a =R_21b and R_22a =R_22b)

The equations (1) and (2) provide the following input/output transfer function: ##EQU1##

The equation (3) shows a high-pass filter and a cut-off frequency f_H of the filter is represented by ##EQU2##

By setting r_e1 and r_d1 so as to satisfy a relation of (r_e1 /r_d1)>1 for the time constant defined by C₁ ·R₁ (r_e1 /r_d1), the capacitance value of the capacitor C₁ is equivalently made large. Thereby, it is possible to obtain a filter circuit having a relatively large time constant or a relatively low cut-off frequency even if a capacitor incorporated in a semiconductor integrated circuit has a relatively small capacitance. Also, the resistance value of the resistor r_e1 can be corrected by adjusting the current I₁ through the resistor-trimming operation in the current adjusting circuit CA. Thereby, it is possible to compensate for variations of cut-off frequencies f_H from one semiconductor integrated circuit chip to another which may be caused by the deviations of electric characteristics of circuit elements in a semiconductor integrated circuit.

FIG. 4A shows a circuit diagram of a monolithic semiconductor integrated circuit device including another filter circuit, and FIG. 4B shows a block diagram of the device shown in FIG. 4A.

An input signal V_in and a feedback signal V_x are supplied to a differential input stage A₁ (or a first differentially operative circuit) similar to that in the device shown in FIG. 3A.

Output voltages of the differential input stage A₁ formed by diode-connected transistors Q₃ and Q₄ in which collector currents of the differentially operative transistors Q₁ and Q₂ flow are applied to the bases of differentially operative NPN transistors Q₆ and Q₇ which form a controllable voltage-current conversion circuit VVIC (or a second differentially operative circuit) and the bases of differentially operative NPN transistors Q₁₀ and Q₁₁ which form a voltage-current conversion circuit VIC (or a third differentially operative circuit).

Like the embodiment shown in FIGS. 3A and 3B, the controllable voltage-current conversion circuit VVIC (or the second differentially operative circuit) includes the differentially operative transistors Q₆ and Q₇, PNP transistors Q₈ and Q₉ which are provided in a current mirror circuit connection on the collector sides of the differentially operative transistors Q₆ and Q₇ and form active load circuits, and a transistor Q₂₂ and an emitter resistor R₁₀ thereof which are provided on the common emitter side of the differentially operative transistors Q₆ and Q₇ and form an adjustable current source circuit I₁₂. The transistor Q₂₂ and a current adjusting circuit CA are in a current mirror circuit connection, like the embodiment shown in FIG. 3A. A signal current (or a first current signal) formed by the differentially operative transistors Q₆ and Q₇ of the controllable voltage-current conversion circuit VVIC serves as a charging/discharging current for a capacitor C₁ and the current signal is fed back to the base of the transistor Q₂ which is a feedback terminal of the differential input stage A₁.

The voltage-current conversion circuit VIC (or the third differentially operative circuit) receiving the output signal of the differential input stage A₁ to convert it into a second current signal includes the differentially operative transistors Q₁₀ and Q₁₁, PNP transistors Q₁₂ and Q₁₃ which are provided in a current mirror circuit connection on the collector sides of the differentially operative transistors Q₁₀ and Q₁₁ and form active load circuits, and a constant current source I₀ which is provided to the common emitter of the differentially operative transistors Q₁₀ and Q₁₁.

An output signal of the controllable voltage-current conversion circuit VVIC and an output signal of the voltage-current conversion circuit VIC are supplied to a non-inverted input (+) and an inverted input (-) of an operational amplifier circuit A₂ which forms an adder circuit. The operational amplifier circuit A₂ includes a differentially operative circuit and an output circuit. The differentially operative circuit is composed of differentially operative NPN transistors Q₁₄ and Q₁₅ which receive the output signals of the voltage-current conversion circuits VVIC and VIC, PNP transistors Q₁₆ and Q₁₇ which are provided in a current mirror circuit connection on the collector sides of the differentially operative transistors Q₁₄ and Q₁₅ and serve as active loads, a capacitor C₂ for phase compensation which is provided between the base and collector of the transistor Q₁₇ on the output side, and a constant current source I₀ which is provided to the common emitter of the differentially operative transistors Q₁₄ and Q₁₅. The output circuit of the operational amplifier circuit A₂ is composed of an emitter-follower output transistor Q₁₈ and a constant current source I₀ provided to the emitter of the transistor Q₁₈.

The base of the transistor Q₁₄ which is the non-inverted input (+) of the operational amplifier circuit A₂ is connected with the commonly connected collectors of the transistors Q₆ and Q₈ which are an output terminal of the controllable voltage-current conversion circuit VVIC. The base of the transistor Q₁₅ which is the inverted input (-) of the operational amplifier circuit A₂ is connected with the commonly connected collectors of the transistors Q₁₀ and Q₁₃ which are an output terminal of the voltage-current conversion circuit VIC.

In order to allow the operational amplifier circuit A₂ to operate as an adder circuit, a feedback resistor R₂ is provided between an output terminal and the inverted input (-) of the operational amplifier circuit A₂. Namely, the resistor R₂ is inserted between the emitter of the output transistor Q₁₈ and the base of the differential transistor Q₁₅.

Explanation will now be made of the operation of the filter circuit of the present embodiment.

The differential input stage A₁ operates such that an input signal V_in supplied to the base of the transistor Q₁ which is a non-inverted input (+) or first input terminal of the differential input stage A₁ and a DC potential of the base of the transistor Q₂ which is an inverted input (-) or second input terminal of the differential input stage A₁ become equal to each other. Since the capacitor C₁ is provided at the inverted input (-) of the input stage A₁, the input signal V_in is AC-wise attenuated in its high frequency region. Accordingly, like the embodiment shown in FIG. 3A, a low-pass filter output is obtained from the inverted input (-) or the base of the transistor Q₂ with which the capacitor C₁ is connected.

When the input signal V_in is inputted, a current of (V_in -V_x)/R₁ is outputted from the collector of the transistor Q₁ and a phase-inverted current thereof is outputted from the collector of the transistor Q₂ . The diode-connected transistors Q₃ and Q₄ serve as loads so that signal voltages outputted from the collectors of the transistors Q₃ and Q₄ result in r_d1 ·(V_in -V_x)/R₁, provided that the ON-resistance of each of the transistors Q₃ and Q₄ both in a diode connection is r_d1. These signal voltages (or first and second output voltages) serve for input voltages of the differentially operative transistors which form the controllable voltage-current conversion circuit VVIC and the voltage-current conversion circuit VIC.

Provided that the resistance value of the emitter resistance of each of the differentially operative transistors Q₆ and Q₇ is r_e1, the following current is obtained from the collectors of the transistors Q₆ and Q₇.

In the controllable voltage-current conversion circuit VVIC, since a signal current is formed in such a manner that the above-mentioned signal voltages r_d1 ·(V_in -V_x)/R₁ are applied to the emitter resistances r_e1 of the differentially operative transistors Q₆ and Q₇, the signal current formed is [r_d1 ·(V_in -V_x)R₁ ]÷r_e1. This current is supplied to the capacitor C₁. Since the capacitor C₁ serves to attenuate a high frequency component of the supplied current signal, the signal V_x has a low-pass filter output characteristic. Thus, the signal fed back to the differential input stage A₁ has a low-pass filter characteristic. Accordingly, an output signal of the differential input stage A₁ has a high-pass filter characteristic in accordance with the decrease of feedback amount of the high frequency component.

In the present embodiment, the output signal of the differential input stage A₁ is converted into a current signal of [r_d1 ·(V_in -V_x)/R₁ ]÷r_e2 by the differentially operative transistors Q₁₀ and Q₁₁ of the voltage-current conversion circuit VIC and the current signal flows through the feedback resistor R₂ of the adder circuit A₂. Here, r_e2 is the resistance value of the emitter resistance of each of the transistors Q₁₀ and Q₁₁. From the above explanation of the operation, there are satisfied the following equations (4) and (5):

V.sub.x =(V.sub.in - V.sub.x)/R.sub.1 ]×r.sub.d1 /(r.sub.e1 ×jωC.sub.1)                                   (4)

V.sub.out =(V.sub.in - V.sub.x)/R.sub.1 ]×(r.sub.d1 /r.sub.e2)×R.sub.2 +V.sub.x.                        (5)

The equations (4) and (5) provide the following input/output transfer function: ##EQU3##

From the equation (6), it is apparent that there serves for not only a low-pass filter but also a high-pass filter, depending on the resistance values of the resistors R₁ and R₂.

In the case of R₁ r_e1 <R₂ r_e2, there is provided a high-pass filter the lower cut-off frequency f_L of which is C₁ ·R₂ (r_e1 /r_e2) and the higher cut-off frequency f_H of which is C₁ ·R₁ (r_e1 /r_e2), as shown by solid line in FIG. 4C. On the contrary, in the case of R₁ r_e1 >R₂ r_e2, there is provided a low-pass filter the lower cut-off frequency f_L of which is C₁ ·R₁ (r_e1 /r_e2) and the higher cut-off frequency f_H of which is C₁ ·R₂ (r_e1 /r_e2), as shown by dotted line in FIG. 4C. As a matter of design, r_e1 and r_e2 are made approximately equal to each other and hence the filter characteristic is determined in accordance with a relation in magnitude between R₁ and R₂.

Compensation for a variation of a time constant defined by CR(r_e1 /r_e2) can be made through such a resistor-trimming operation as explained in conjunction with FIG. 2A or 2B since the resistance r_e1 is changed by a constant current source I₁₂. Also, by setting r_e1 and r_e2 so as to satisfy a relation of (r_e1 /r_e2)>1, the time constant can be regarded as being equivalent to one in which the capacitance value of the capacitor C₁ is made large. Thereby, it is possible to obtain a relatively large time constant even if a capacitor incorporated in a semiconductor integrated circuit has a relatively small capacitance.

FIG. 5A shows a circuit diagram of a monolithic semiconductor integrated circuit device including a notch filter circuit, and FIG. 5B shows a block diagram of the device shown in FIG. 5A.

A differential input stage A₁, a first controllable voltage-current conversion circuit VVIC₁ and an operational amplifier A₂ shown in FIG. 5A have constructions similar to those of the circuits shown in FIG. 4A. A second controllable voltage-current conversion circuit VVIC₂ may have a construction similar to that of the first controllable voltage-current conversion circuit VVIC₁.

In the embodiment shown in FIGS. 5A and 5B, output signals of the differential input stage A₁ are supplied to the first and second controllable voltage-current conversion circuits VVIC₁ and VVIC₂ in order to obtain a notch filter characteristic. Second amplifying elements or transistors Q₂₃ and Q₂₄ in the first and second controllable voltage-current conversion circuits VVIC₁ and VVIC₂ are in a current mirror circuit connection together with a current adjusting circuit CA, like the embodiment shown in FIG. 3A. Accordingly, the transistors Q₂₃ and Q₂₄ form adjustable current sources I₃ and I₃ '. A load resistor R₂ is provided between an output terminal of the second controllable voltage-current conversion circuit VVIC₂ and a predetermined bias terminal V_B. A capacitor C₂ is provided between the outputs of the first and second controllable voltage-current conversion circuits VVIC₁ and VVIC₂. An output signal V_y of the second controllable voltage-current conversion circuit VVIC₂ and an input signal V_in are summed by an adder circuit which is constructed by the operational amplifier circuit A₂. Namely, the input signal V_in is supplied to a non-inverted input terminal (+) of the operational amplifier circuit A₂ while the signal V_y is supplied to an inverted input terminal (-) thereof through a resistor R_3a, and a resistor R_3b is provided between the inverted input (-) and an output of the operational amplifier circuit A₂.

Provided that the ON-resistance of each of the transistors Q₃ and Q₄ of the differential input stage A₁, the emitter resistance of each of differentially operative transistors Q₆ and Q₇ of the first controllable voltage-current conversion circuit VVIC₁ is r_e1 and the emitter resistance of each of differentially operative transistors Q₆ ' and Q₇ ' of the second controllable voltage-current conversion circuit VVIC₂ is r_e3, the following equations (7) and (9) are satisfied similarly to the above explained:

[(V.sub.in -V.sub.x)/R.sub.1 ]×2r.sub.d1 /r.sub.e1 -jωC.sub.1 C.sub.x +jωC.sub.2 (V.sub.y -V.sub.x)=0             (7)

[(V.sub.in -V.sub.x)/R.sub.1 ]×2r.sub.d1 /r.sub.e3 +(V.sub.y /R.sub.2)+jωC.sub.2 (V.sub.y -V.sub.x)=0            (8)

V.sub.out =V.sub.in +V.sub.y (under assumption of R.sub.3a =R.sub.3b).(9)

The notch frequency as represented by the corresponding angular frequency ω₀ and the gain G_vω.sbsb.o of the notch filter circuit at the notch angular frequencies will be determined from equations (7) to (9) as follows. ##EQU4##

From the above two equations, it can be seen that the notch frequency can be controlled by the control current I₃, i.e., by the resistance r_e1 in the first controllable voltage-current conversion circuit VVIC₁, while the gain at the notch frequency can be adjusted by the control current I₃ ', i.e., by the resistance r_e3 in the second controllable voltage-current conversion circuit VVIC₂ with the control current I₃ being first fixed, though the gain could be varied by the control current I₃ (the resistance r_e1).

FIG. 6A shows a circuit diagram of a monolithic semiconductor integrated circuit device including a high-pass filter circuit, and FIG. 6B shows a block diagram of the device shown in FIG. 6A.

A differential input stage A₁, a controllable voltage-current conversion circuit VVIC, an operational amplifier circuit A₃ and an adder circuit A₂ shown in FIG. 6A may have constructions similar to those of the circuits shown in FIG. 3A. A second amplifying element or transistor Q₂₃ in the controllable voltage-current conversion circuit VVIC is in a current mirror circuit connection together with a current adjusting circuit CA, like the embodiment shown in FIG. 3A. Accordingly, the transistor Q₂₃ forms an adjustable current source I₂.

However, in the embodiment shown in FIG. 6A, a feedback is applied to the operational amplifier circuit A₃ in order to reduce the DC offset and the signal distortion of an output V_out. Namely, a signal V_x in the controllable voltage-current conversion circuit VVIC having a low-pass filter characteristic is divided by resistors R_5a and R_5b through the operational amplifier A₃ in a voltage-follower configuration and is thereafter supplied to a non-inverted input (+) of the operational amplifier circuit A₂. An inverted input (-) of the operational amplifier circuit A₂ is applied with an input signal V_in through a resistor R_5a and a resistor R_5b is provided between the inverted input (-) and an output of the operational amplifier circuit A₂.

Briefly explaining the operation of the filter circuit of the embodiment shown in FIG. 6A, a high-pass filter characteristic is obtained subtracting the signal V_x having the low-pass filter characteristic from the input signal V_in. This can be quantitatively explained by equations (10) to (12) which will be shown just in below. Namely, since a relation of V_x =[(V_in -V_x)/R₁ ]×(r_d1 /r_e1)×1/jωC₁ is satisfied similarly to the above explained, there are satisfied the following equations (10) to (12): ##EQU5##

V.sub.y =V.sub.x /2                                        (11)

V.sub.in -V.sub.y +V.sub.out -V.sub.y =0.                  (12)

From the equations (10) to (12), a transfer function as shown by the equation (3) is obtained similarly to the case of the circuit shown in FIG. 3A (on the assumption that R_5a =R_5b and R_6a =R_6b).

In the embodiment shown in FIGS. 6A and 6B, since the feedback is applied also to the amplifier circuit A₃ to obtain the output signal V_out, it is possible to greatly reduce the DC offset voltage and the signal distortion. Also, a gain can be adjusted by setting the ratio of the resistance value of the resistor R_5a to that of the resistor R_5b and the ratio of the resistance value of the resistor T_6a to that of the resistor R_6b to desired values.

Functional effects obtained by the above-mentioned embodiments are as follows:

(1) Since the electric characteristics of circuit elements formed in a semiconductor integrated circuit have approximately uniform deviations in only one direction, it is possible to match the time constants of filter circuits with each other through a resistor-trimming operation at one location.

(2) By transmitting an output signal of a differential input stage to a controllable voltage-current conversion circuit and providing a capacitor charged/discharged by an output signal of the controllable voltage-current conversion circuit so that the output signal is fed back to a feedback terminal of the differential input stage, a signal having a low-pass filter characteristic and a signal having a high-pass filter characteristic can be obtained from the feedback terminal and an output terminal, respectively. A variety of filter circuits can be constructed by a signal processing of those signals including summation, subtraction and/or so on. A frequency response can be corrected by flowing a control current in the controllable voltage-current conversion circuit.

(3) The capacitance value of the capacitor provided to the feedback terminal of the differential input stage can be made equivalently large in accordance with the ratio of a load resistor of the differential input stage to the emitter resistance of transistors forming the voltage-current conversion circuit. Thereby, even in a semiconductor integrated circuit in which only a capacitor having a relatively small capacitance can be formed, a filter circuit having a relatively low cut-off frequency can be obtained.

In the foregoing, the invention made by the present inventors has been specifically explained on the basis of the embodiments thereof. However, of course, the present invention is not limited to the disclosed embodiments but various changes or modifications can be made without departing from the subject matter of the present invention. For example, a structure for changing a control current by a resistor-trimming operation can employ a variety of configurations inclusive of the trimming of patterns of resistor elements themselves instead of the disclosed selective cutting-off of the connection lines by which resistor elements are connected. Also, differentially operative amplifying elements forming a filter circuit may include MOSFETs (insulated-gate field-effect transistors), junction type FETs instead of bipolar transistors disclosed. Further, a specific construction for an adder or subtracter circuit for receiving the signal having the low-pass filter characteristic and the signal having the high-pass filter characteristic can employ a variety of configurations.

The present invention is widely applicable to a semiconductor integrated circuit device having a filter circuit incorporated therein.

Claims

We claim:

1. A monolithic semiconductor integrated circuit device, comprising:

a current adjusting circuit including at least one first amplifying element and first and second load resistor means for said amplifying element, said first amplifying element having a control electrode, a current receiving electrode connected with said first load resistor means and a current delivering electrode connection with said second load resistor means, at least one of said first and second load resistor means being means for trimming resistance to adjustably determine the resistance of said at least one of said first and second load resistor means for controlling the current flowing between said current receiving electrode and said current delivering electrode; and

an audio signal processing circuit including a differentially operative circuit section and a second amplifying element, said second amplifying element having a control electrode, a current receiving electrode and a current delivering electrode, one of said current receiving and delivering electrodes being connected with said differentially operative circuit section, said differentially operative circuit section having a frequency response depending on an electric current flowing in a current path between said current receiving and delivering electrodes of said second amplifying element, said first and second amplifying elements being in a current mirror circuit connection with their control electrodes being connected with each other so that said electric current flowing in the current path between said current receiving and delivering electrodes of said second amplifying element is controlled by said adjustably determined resistance of said at least one load resistor means to adjust the frequency response of said audio signal processing circuit, thereby compensating for variations of electric characteristics of the audio signal processing circuit.

2. A device according to claim 1, wherein the other of said first and second load resistor means is means for trimming resistance to adjustably determine the resistance of said other of said first and second load resistor means for controlling the current flowing between said current receiving electrode and said current delivering electrode.

3. A device according to claim 2, in which each of said first and second load resistor means includes a plurality of resistor elements arranged with connectors therebetween that may be cut during the trimming.

4. A device according to claim 1, in which said first and second amplifying elements are first and second bipolar transistors, and a third bipolar transistor is connected with said first bipolar transistor such that the base and emitter of said third bipolar transistor are connected with the collector and base of said first bipolar transistor, respectively.

5. A device according to claim 1, in which each of said first and second amplifying elements is a field-effect transistor.

6. A filter according to claim 2, wherein the other of said first and second load resistor means is means for trimming resistance to adjustably determine the resistance of said other of said first and second load resistor means for controlling the current flowing between said current receiving electrode and said current delivering electrode.

7. A filter according to claim 6, in which each of said first and second load resistor means includes a plurality of resistor elements arranged with connectors therebetween that may be cut curing the trimming.

8. A device according to claim 1, wherein the audio signal processing circuit has a time constant defined by a capacitance times a resistance times a value proportional to the adjustably determined resistance of said at least one load resistor means that directly determines the frequency response.

9. A device according to claim 8, wherein said means for trimming increases the time constant.

10. A filter in an integrated circuit form, comprising:

a current adjusting circuit including at least one first amplifying element and first and second load resistor means for said amplifying element, said first amplifying element having a control electrode, a current receiving electrode connected with said first load resistor means and a current delivering electrode connected with said second load resistor means, at least one of said first and second load resistor means being means for resistance to adjustably determine the resistance of said at least one of said first and second load resistor means for controlling the current flowing between said current receiving electrode and said current delivering electrode;

a first differentially operative circuit serving as an input stage and having a first input terminal for receiving an input signal to the filter and a second input terminal;

a capacitor connected with said second input terminal;

a second differentially operative circuit serving to convert an output voltage of said first differentially operative circuit to a corresponding current signal, said capacitor being also connected with said second differentially operative circuit to receive said current signal for a charge/discharge operation therewith, said second differentially operative circuit including a pair of differential amplifier elements and a second amplifying element, said second amplifying element having a control electrode, a current receiving electrode and a current delivering electrode, one of said current receiving and delivering electrodes being connected with said pair of differential amplifier elements, said second differentially operative circuit having a frequency response depending on an electric current receiving and delivering electrodes of said second amplifying element, said first and second amplifying elements being in a current mirror circuit connection with their control electrodes being connected with each other so that said electric current flowing in the current path between said current receiving and delivering electrodes of said second amplifying element is controlled by said adjustably determined resistance of said at least one load resistor means; and

an adder circuit for summing an output signal representative of a terminal voltage of said capacitor and said input signal.

11. A filter according to claim 10, wherein the other of said first and second load resistor means is means for trimming resistance to adjustably determine the resistance of said other of said first and second load resistor means for controlling the current flowing between said current receiving electrode and said current delivering electrode.

12. A filter according to claim 11, in which each of said first and second load resistor means includes a plurality of resistor elements arranged with connectors therebetween that may be cut curing the trimming.

13. A filter according to claim 10, wherein said second differentially operative circuit has a time constant defined by a capacitances times a resistance times a value proportional to the adjustably determined resistance times a value proportional to the adjustably determined resistance of said at least one load resistor means that directly determine the frequency response.

14. A device according to claim 13, wherein said means for trimming increases the time constant.

15. A filter according to claim 10, wherein said first differentially operative circuit, said capacitor and said second differentially operative circuit constitute one of a low-pass filter, high-pass filter and notch filter, and further comprising another of the low-pass filter, high-pass filter and notch filter including another first differentially operative circuit, another capacitor and another second differentially operative circuit, substantially identical to the first mentioned first differentially operative circuit, capacitor and second differentially operative circuit all connected so that the frequency response of at least two separate filters is commonly adjusted by a single means for trimming.

16. A filter according to claim 15, wherein the other of said first and second load resistor means is means for trimming resistance to adjustably determine the resistance of said other of said first and second load resistor means for controlling the current flowing between said current receiving electrode and said current delivering electrode.

17. A filter according to claim 16, wherein each of said first and second load resistor means includes a plurality of resistor elements arranged with connectors therebetween that may be cut curing the trimming.

18. A filter according to claim 15, wherein said second differentially operative circuit has a time constant defined by a capacitance times a resistance times a value proportional to the adjustably determined resistance of said at least one load resistor means that directly determine the frequency response.

19. A device according to claim 18, wherein said means for trimming increases the time constant.

20. A filter according to claim 15, wherein the other of said first and second load resistor means is means for trimming resistance to adjustably determine the resistance of said other of said first and second load resistor means for controlling the current flowing between said current receiving electrode and said current delivering electrode.

21. A filter according to claim 20, wherein each of said first and second load resistor means includes a plurality of resistor elements arranged with connectors therebetween that may be cut curing the trimming.

22. A filter in an integrated circuit form comprising:

a current adjusting circuit including at least one first amplifying element and first and second load resistor means for said amplifying element, said first amplifying element having a control electrode, a current receiving electrode connected with said first load resistor means and a current delivering electrode connected with said second load resistor means, at least one of said first and second load resistor means being means for trimming resistance to adjustably determine the resistance of said at least one of said first and second load resistor means for controlling the current flowing between said current receiving electrode and said current delivering electrode;

a capacitor connected with said second input terminal;

a second differentially operative circuit serving to convert an output voltage of said first differentially operative circuit to a corresponding current signal, said capacitor being also connected with said second differentially operative circuit to receive said current signal for a charge/discharge operation therewith, said second differentially operative circuit including a pair of differential amplifier elements and a second amplifying element, said second amplifying element having a control electrode, a current receiving electrode and a current delivering electrode, one of said current receiving and delivering electrodes being connected with said pair of differential amplifier elements, said second differentially operative circuit having a frequency response depending on an electric current flowing in a current path between said current receiving and delivering electrodes of said second amplifying element, said first and second amplifying elements being in a current mirror circuit connection with their control electrodes being connected with each other so that said electric current flowing in the current path between said current receiving and delivering electrodes of said second amplifying element is controlled by said adjustably determined resistance of said at least one load resistor means; and

a third differentially operative circuit serving to convert a second output voltage of said first differentially operative circuit to a corresponding second current signal; and

an adder circuit for summing an output signal of said second differentially operative circuit representative of a terminal voltage of said capacitor and an output signal of said third differentially operative circuit representative of said second current signal.

23. A device according to claim 22, wherein said second differentially operative circuit has a time constant defined by a capacitances times a resistance times a value proportional to the adjustably determined resistance of said at least one load resistor means that directly determine the frequency response.

24. A filter according to claim 23, wherein said means for trimming increases the time constant.

25. A filter according to claim 22, wherein said first differentially operative circuit, said capacitor and said second differentially operative circuit constitute one of a low-pass filter, high-pass filter and notch filter, and further comprising another of the low-pass filter, high-pass filter and notch filter including another first differentially operative circuit, another capacitor and another second differentially operative circuit, substantially identical to the first mentioned first differentially operative circuit, capacitor and second differentially operative circuit all connected so that the frequency response of at least two separate filters is commonly adjusted by a single means for trimming.

26. A filter according to claim 25, wherein said second differentially operative circuit has a time constant defined by a capacitance times a resistance times a value proportional to the adjustably determined resistance of said at least one load resistor means that directly determine the frequency response.

27. A device according to claim 26, wherein said means for trimming increases the time constant.