US4353112A - Switched-mode voltage converter - Google Patents

Switched-mode voltage converter Download PDF

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Publication number
US4353112A
US4353112A US06/072,261 US7226179A US4353112A US 4353112 A US4353112 A US 4353112A US 7226179 A US7226179 A US 7226179A US 4353112 A US4353112 A US 4353112A
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United States
Prior art keywords
voltage
input
capacitor
square
wave signal
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Expired - Lifetime
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US06/072,261
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English (en)
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Jan J. Rietveld
Alain Moreau
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US Philips Corp
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US Philips Corp
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Assigned to U.S. PHILIPS CORPORATION, A CORP.OF DE reassignment U.S. PHILIPS CORPORATION, A CORP.OF DE ASSIGNMENT OF ASSIGNORS INTEREST. Assignors: MOREAU, ALAIN, RIETVELD, JAN J.
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of dc power input into dc power output
    • H02M3/22—Conversion of dc power input into dc power output with intermediate conversion into ac
    • H02M3/24—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters
    • H02M3/28—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac
    • H02M3/325—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/337—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration
    • H02M3/3376—Conversion of dc power input into dc power output with intermediate conversion into ac by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate ac using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in push-pull configuration with automatic control of output voltage or current

Definitions

  • the invention relates to a switched-mode converter for converting an input d.c. voltage into an output d.c. voltage and comprising a generator for applying a periodic square-wave signal, whose amplitude depends on the input voltage, to an inductive network to which a rectifier and a smoothing capacitor are and wherein in operation a smoothed voltage is present across the capacitor.
  • a pulse generator comprises a plurality of switches by means of which a square-wave signal is applied to the series arrangement of a choke coil, the primary winding of a transformer and a separation capacitor.
  • the output d.c. voltage of the circuit is obtained by the rectification of the voltage present at the secondary side of the transformer.
  • the output voltage is stabilized by means of a pulse duration modulation of the control pulses which bring the switches alternately into the conducting and into the cutoff state, the duration of these pulses depending on the output signal to be controlled.
  • the choke stores energy during a portion of the period and transfers it in another portion, while voltage and current peaks are suppressed.
  • control is rather expensive. It requires, in addition to the modulator, a comparison stage in which a voltage derived from the output signal is compared with a reference voltage while the edges of the pulses must be sufficiently steep so that the output voltage must be properly smoothed and measures must be taken against an unwanted high frequency radiation. Such a control is therefore economically justified only for apparatus in which a substantially constant supply voltage is essential and whose consumed power is rather high. This applies to, for example, nonportable large-screen colour television receivers.
  • the invention is based on the recognition that the prior art converter is also usable without pulse duration control provided a simple and, consequently, cheap control is used, which control considerably improves the efficiency.
  • the switched-mode converter according to the invention is characterized in that means are provided for controlling the frequency of the square-wave signal in dependence on the input d.c. voltage.
  • Such a control is a forward control and has therefore all the known advantages thereof, namely the fact that changes in the input voltage are passed on without appreciable delay and that the circuit is stable owing to the absence of feedback.
  • This control can be of such a nature that the frequency of the square-wave signal is directly proportional to the input d.c. voltage or that the period of the square-wave signal decreases when the input d.c. voltage increases and decreases when the input voltage decreases, the relative variation of the period being greater than that of the input voltage.
  • the converter may be characterized in that a load is arranged in parallel with the smoothing capacitor so that the smoothed voltage is also the output voltage, and in that the current through the load is substantially constant.
  • the converter according to the invention may be characterized in that a series control transistor is arranged between the smoothing capacitor and the load with the output voltage present across the load being kept substantially constant by means of the series control transistor.
  • a series control transistor is arranged between the smoothing capacitor and the load with the output voltage present across the load being kept substantially constant by means of the series control transistor.
  • the duty cycle i.e. the ratio of the time interval in which the square-wave signal assumes a given value to the period, is substantially constant. The dissipation is still further reduced if said ratio is approximately equal to 0.5.
  • the square-wave signal can be produced by means of a sawtooth voltage and a threshold voltage, the sawtooth voltage being produced across a capacitor whose charging current originates from a current source and which is periodically discharged, the charging current flowing through a resistor connected to the input d.c. voltage and, in the above-mentioned second case, a substantially constant-voltage element, for example a Zener diode, being arranged in series with the resistor.
  • a substantially constant-voltage element for example a Zener diode
  • FIG. 1 shows the known converter but without control
  • FIG. 2 shows an equivalent circuit thereof
  • FIG. 3 shows waveforms occurring therein
  • FIG. 4 shows the output characteristic thereof
  • FIG. 5 shows the combination of the known converter with a series control circuit
  • FIG. 6 shows output characteristics of the converter according to the invention.
  • FIG. 7 shows an embodiment of the converter according to the invention.
  • the circuit of FIG. 1 is of a known type.
  • Two npn-switching transistors Tr 1 and Tr 2 are arranged in series between the terminals of a d.c. voltage source V 1 .
  • Diodes D 1 and D 2 are arranged in parallel with the collector-emitter path of each transistor Tr 1 and Tr 2 , and with the opposite conductivity direction with respect to that of said transistors.
  • the base of a driver transistor Tr 3 also of the npn-type, is supplied in operation with pulses produced by an oscillator OSC.
  • Oscillator OSC as well as transistor Tr 3 are provided with supply energy by source V 1 .
  • the primary winding L p1 of a driver transformer T 1 is included in the collector lead of transistor Tr 3 .
  • Secondary windings L s1 and L' s1 of transformer T 1 are respectively arranged between the base and the emitter of each of the transistors Tr 1 and Tr 2 , the winding senses of said windings having been chosen so that transistor Tr 1 is alternately in the conducting and then the cutoff state while at the same time transistor Tr 2 is alternately in the cutoff and then the conducting state.
  • Voltage V 1 is derived from the electric supply by means of a rectifier D 3 and a smoothing capacitor C 1 .
  • the emitter of transistor Tr 1 and the collector of transistor Tr 2 are interconnected.
  • Arranged between the junction A thus formed and the negative terminal of source V 1 is the series arrangement of an isolating capacitor C, an inductance L and the primary winding L p2 of a transformer T 2 .
  • One end of a secondary winding L s2 of transformer T 2 is connected to the anode of a rectifier D 4 whereas the other end of winding L s2 is connected to the anode of a rectifier D 5 .
  • a smoothing capacitor C 2 and a load, which may be considered as a resistor R, are included between the interconnected cathodes of rectifiers D 4 and D 5 and the centre tap of winding L s2 .
  • V 0 voltage V 0 is present across capacitor C 2 and load R and a direct current I 0 flows through load R.
  • the negative terminal of voltage source V 0 is connected to ground and can be connected to the negative terminal of source V 1 .
  • transformer T 2 provides d.c. isolation between ground and the electric supply.
  • FIG. 2 shows an equivalent circuit of the circuit of FIG. 1.
  • elements Tr 1 , D 1 and Tr 2 , D 2 have been replaced by two ideal switches S 1 and S 2 .
  • point A alternately assumes the potential O and the potential V 1 .
  • the capacitance of capacitor C is assumed to be infinitely large whereas transformer T 2 and inductance L are replaced by a parallel inductance L 2 having an infinitely high value and a finite series inductance L 1 , allowance having been made in this inductance L 1 for the leakage inductance of transformer T 2 .
  • the symmetry at the secondary side of transformer T 2 is restored in the equivalent circuit of FIG. 2 by substituting a Graetz bridge rectifier circuit D 4 ,D' 4 ,D 5 ,D' 5 for the full-wave rectifier D 4 , D 5 .
  • FIG. 3a shows schematically the variation as a function of time of the voltage V L across inductance L 2 in the steady state and
  • FIG. 3b shows schematically the variation of the current i through inductance L 1 to the same time scale.
  • Transistor Tr 1 conducts during a portion ⁇ T of the period T of the signal of oscillator OSC, whereas transistor Tr 2 conducts during the remaining portion (1- ⁇ )T of period T.
  • the variation of current i during one period T is represented by four joined straight lines, a time shift t 1 for the zero crossing of current i at the beginning of interval ⁇ T and a time shift t 2 for the zero crossing of current i at the beginning of interval (1- ⁇ )T occurring relative to voltage V L .
  • a direct current I L which cannot flow through capacitor C but which can flow through inductance L 2 flows through inductance L 1 .
  • Current I L is equal to the mean value of current i in FIG. 3b.
  • L 1 represents the value of inductance L 1 in FIG. 2.
  • FIG. 4 shows that the output voltage is subjected to a variation from a value V 01 to a value V 02 when the output current varies between a value I 01 and a value I 02 at an input voltage which has remained constant. From equation (3) it can be derived that the internal resistance of the circuit is equal to ##EQU5## This shows that the internal resistance increases with an increasing I 0 and becomes infinitely large for a short-circuited output. It will be clear that similar results will also be obtained for other values of ratio ⁇ .
  • FIG. 4 shows the variation of voltage V 0 for a value V' 1 of voltage V 1 which is higher than the value considered above. If the input voltage varies between the values V 1 and V' 1 , FIG. 4 shows that the output voltage varies between the values V' 01 and V' 02 at a varying output current. For many applications such a variation is impermissible, so that stabilisation is required.
  • FIG. 5 shows the converter of FIG. 1, the same reference symbols having been used for the same elements, in combination with a stabilisation circuit.
  • a series transistor Tr 4 of the pnp type whose internal resistance is controlled in known manner in dependence on the voltage V 0 across the load, is included between capacitor C 2 and load R.
  • a npn transistor Tr 5 compare a voltage derived from voltage V 0 by means of a resistance voltage divider R 1 , R 2 with the reference voltage of a zener diode D 6 .
  • the collector current of transistor Tr 5 which is at the same time the base current of transistor Tr 4 , depends on the difference between the compared voltages.
  • a smoothing capacitor C 3 is arranged in parallel with load R and in the same circumstances as in FIG. 1 substantially the same current I 0 flows through load R as it does in FIG. 1.
  • the frequency of the switching signal is constant and, consequently, independent of the input voltage.
  • the invention is based on the recognition that the efficiency can be considerably improved by the use of a forward control of the frequency such that the frequency varies in dependence on the input voltage.
  • FIG. 6 shows the characteristic of FIG. 4, but now includes values based on the above numerical example.
  • FIG. 6 also shows the curve a which is obtained for a constant product V 1 T of the input voltage V 1 by the period T of the switching signal. It appears from formula (4) that the short-circuit current remains constant, whereas the output voltage in the unloaded condition keeps the value 0.5 V' 1 for an input voltage of V' 1 .
  • FIG. 6 shows that the values of the output voltage for curve a are always below those for the broken line curve. So such a control achieves a constant short-circuit current, which provides circuit protection and is advantageous for transistors Tr 1 and Tr 2 , and a decrease of the dissipation.
  • An oscillator generating a signal whose period is inversely proportional to the input voltage and which is therefore suitable to control transistor Tr 3 can be implemented at the following simple manner.
  • a capacitor is charged by a current source, the current being directly derived from voltage V 1 .
  • the voltage increases linearly across the capacitor. As soon as this voltage reaches a predetermined value the capacitor is quickly discharged.
  • Oscillators operating on this principle are known from the literature.
  • the produced sawtooth signal is thereafter converted in known manner into a square-wave signal.
  • a further improvement with respect to the broken line curve of FIG. 6 is obtained by means of a circuit whose output characteristic is represented by curve b.
  • this circuit the frequency of the switching signal varies in such a manner that curve b passes through point P.
  • this curve is a parabola intersecting the vertical axis at the same point as the broken line parabola and curve a. From this it appears that the short circuit current I 0 max decreases versus an increasing input voltage and that the dissipation in the series control circuit is still further decreased relative to the case of curve a.
  • FIG. 7 shows a complete circuit wherein elements corresponding to those in FIGS. 1 and 5 have been given the same reference numerals and wherein the oscillator satisfies the above conditions.
  • a capacitor C 4 is charged by a current flowing through the series arrangement of a Zener diode D 7 and a high-value resistor R 3 , connected to voltage V 1 .
  • Resistor R 3 may be considered as a current source.
  • a switch constructed in known manner by means of two complementary transistors Tr 6 and Tr 7 , is arranged to operate as a thyristor.
  • the thyristor is rendered conductive when the voltage across capacitor C 4 reaches approximately the substantially constant value of voltage across the series arrangement of a resistor R 4 and a Zener diode D 8 and serves as the discharging element for capacitor C 4 .
  • the anode gate of thyristor Tr 6 , Tr 7 is connected to voltage V 1 through a resistor R 6 whereas the cathode gate is connected to the negative terminal of source V 1 through a resistor R 7 .
  • the discharge stops when the voltage across capacitor C 4 has been reduced to approximately the substantially constant value of the voltage across an RC parallel network R 5 , C 5 situated in the cathode lead of the thyristor.
  • a substantially constant voltage which is subtracted from voltage V 1 is present across diode D 7 , which could be replaced by a voltage dependent resistor.
  • the voltage drop across resistor R 3 and, consequently, also the charging current of capacitor C 4 , flowing therethrough, is subjected to a relative variation which is greater than would be the case if diode D 7 were absent.
  • the relative variation of the period of the sawtooth voltage having a constant amplitude, produced across capacitor C 4 is therefore greater than that of voltage V 1 .
  • the discharging time of capacitor C 4 is very short as the discharge current flows through the emitter of transistor Tr 6 , this being a low-ohmic path.
  • an emitter follower transistor Tr 8 which serves as a separating stage, the sawtooth voltage present across capacitor C 4 is applied to the base of driver transistor Tr 3 which converts the sawtooth into a square-wave.
  • the series arrangement of a number of diodes, for example two diodes D 9 and D 10 , and a resistor R 8 which series arrangement is shunted by a decoupling capacitor C 6 , is included in the emitter lead of transistor Tr 3 .
  • a substantially constant threshold voltage is present at the emitter.
  • Transistor Tr 3 is brought to the saturation state as soon as its base voltage becomes somewhat higher than this threshold voltage.
  • Resistor R 8 is adjustable so that the ratio ⁇ can be adjusted to a particular value with this resistor.
  • the square-wave signal thus obtained is applied to the bases of transistors Tr 1 and Tr 2 via transformer T 1 .
  • Transistor Tr 1 is cutoff, whereas transistor Tr 2 conducts in the interval in which transistor Tr 3 conducts.
  • Attenuation networks are provided at the primary as well as at the secondary side of transformer T 1 .
  • a capacitor C 7 is arranged parallel with diode D 2 , causing the slope of the voltage at point A to be reduced during the transitions, so that switching losses are somewhat reduced.
  • Inductance L is constituted by the leakage inductance of transformer T 2 . In the foregoing the capacitance of capacitor C was assumed to be very high.
  • Information which depends on the measured difference between the said voltages, may be used to control a transistor representing a variable resistor and included between the collector of transistor Tr 7 and the negative terminal of voltage source V 1 .

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Details Of Television Scanning (AREA)
US06/072,261 1978-09-11 1979-09-04 Switched-mode voltage converter Expired - Lifetime US4353112A (en)

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NL7809226 1978-09-11
NL7809226A NL7809226A (nl) 1978-09-11 1978-09-11 Geschakelde spanningsomzetter.

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AU (1) AU5067879A (nl)
BE (1) BE878700A (nl)
DE (1) DE2935811A1 (nl)
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Cited By (26)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4400767A (en) * 1981-06-30 1983-08-23 Honeywell Information Systems Inc. Self start flyback power supply
US4507722A (en) * 1981-11-30 1985-03-26 Park-Ohio Industries, Inc. Method and apparatus for controlling the power factor of a resonant inverter
US4511956A (en) * 1981-11-30 1985-04-16 Park-Ohio Industries, Inc. Power inverter using separate starting inverter
US4578744A (en) * 1982-07-01 1986-03-25 Jovan Antula A. C. power converter
US4595974A (en) * 1984-09-10 1986-06-17 Burroughs Corporation Base drive circuit for a switching power transistor
US4629971A (en) * 1985-04-11 1986-12-16 Mai Basic Four, Inc. Switch mode converter and improved primary switch drive therefor
US4679131A (en) * 1985-05-10 1987-07-07 Rca Corporation Regulating power supply for video display apparatus
US4810952A (en) * 1986-06-30 1989-03-07 Swingline Inc. Circuitry and method for controlling power to fastener machine solenoid
US4835410A (en) * 1988-02-26 1989-05-30 Black & Decker Inc. Dual-mode corded/cordless system for power-operated devices
US4835409A (en) * 1988-02-26 1989-05-30 Black & Decker Inc. Corded/cordless dual-mode power-operated device
US4847513A (en) * 1988-02-26 1989-07-11 Black & Decker Inc. Power-operated device with a cooling facility
US4896255A (en) * 1987-06-05 1990-01-23 Siemens Aktiengesellschaft Power pack comprising resonant converter
US4945467A (en) * 1988-02-26 1990-07-31 Black & Decker Inc. Multiple-mode voltage converter
US4947308A (en) * 1989-04-17 1990-08-07 Zdzislaw Gulczynski High power switching power supply
US5043650A (en) * 1988-02-26 1991-08-27 Black & Decker Inc. Battery charger
US5053937A (en) * 1989-10-12 1991-10-01 Siemens Aktiengesellschaft Method for controlling push-pull series-resonant converter switching power supplies with regulated output voltage
US5062031A (en) * 1988-12-16 1991-10-29 Erbe Elektromedizin Gmbh Self oscillating power stage for inverted rectifier power supply
US5063488A (en) * 1988-09-16 1991-11-05 Kyushu University Switching power source means
US5448465A (en) * 1992-08-25 1995-09-05 Matsushita Electric Industrial Co., Ltd. Switching power supply without switching induced spike voltages
US5490052A (en) * 1992-04-24 1996-02-06 Matsushita Electric Industrial Co., Ltd. Switching power supply
WO2001026207A2 (en) * 1999-10-01 2001-04-12 Online Power Supply, Inc. Non-saturating magnetic element(s) power converters and surge protection
US6493242B1 (en) 1999-10-01 2002-12-10 Online Power Supply, Inc. Power factor controller
US20040012986A1 (en) * 2002-07-22 2004-01-22 Riggio Christopher Allen Two-stage converter using low permeability magnetics
US20090262562A1 (en) * 2008-04-22 2009-10-22 Zaohong Yang Efficiency improvement in power factor correction
US20100027299A1 (en) * 2008-07-29 2010-02-04 On-Bright Electronics (Shanghai) Co., Ltd. Systems and methods for adaptive switching frequency control in switching-mode power conversion systems
US20200099308A1 (en) * 2018-09-26 2020-03-26 Siemens Aktiengesellschaft Electrical power conversion system

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US4484113A (en) * 1981-02-16 1984-11-20 Rca Corporation Regulated deflection circuit
EP0204369B1 (en) * 1981-02-16 1993-04-28 RCA Thomson Licensing Corporation Deflection circuit
JPS58192491A (ja) * 1982-04-30 1983-11-09 Tokyo Electric Co Ltd 電気シエ−バ−

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Cited By (38)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4400767A (en) * 1981-06-30 1983-08-23 Honeywell Information Systems Inc. Self start flyback power supply
US4507722A (en) * 1981-11-30 1985-03-26 Park-Ohio Industries, Inc. Method and apparatus for controlling the power factor of a resonant inverter
US4511956A (en) * 1981-11-30 1985-04-16 Park-Ohio Industries, Inc. Power inverter using separate starting inverter
US4578744A (en) * 1982-07-01 1986-03-25 Jovan Antula A. C. power converter
US4595974A (en) * 1984-09-10 1986-06-17 Burroughs Corporation Base drive circuit for a switching power transistor
US4629971A (en) * 1985-04-11 1986-12-16 Mai Basic Four, Inc. Switch mode converter and improved primary switch drive therefor
US4679131A (en) * 1985-05-10 1987-07-07 Rca Corporation Regulating power supply for video display apparatus
US4810952A (en) * 1986-06-30 1989-03-07 Swingline Inc. Circuitry and method for controlling power to fastener machine solenoid
US4896255A (en) * 1987-06-05 1990-01-23 Siemens Aktiengesellschaft Power pack comprising resonant converter
US4835410A (en) * 1988-02-26 1989-05-30 Black & Decker Inc. Dual-mode corded/cordless system for power-operated devices
US4847513A (en) * 1988-02-26 1989-07-11 Black & Decker Inc. Power-operated device with a cooling facility
US4835409A (en) * 1988-02-26 1989-05-30 Black & Decker Inc. Corded/cordless dual-mode power-operated device
US4945467A (en) * 1988-02-26 1990-07-31 Black & Decker Inc. Multiple-mode voltage converter
US5043650A (en) * 1988-02-26 1991-08-27 Black & Decker Inc. Battery charger
US5063488A (en) * 1988-09-16 1991-11-05 Kyushu University Switching power source means
US5062031A (en) * 1988-12-16 1991-10-29 Erbe Elektromedizin Gmbh Self oscillating power stage for inverted rectifier power supply
US4947308A (en) * 1989-04-17 1990-08-07 Zdzislaw Gulczynski High power switching power supply
US5053937A (en) * 1989-10-12 1991-10-01 Siemens Aktiengesellschaft Method for controlling push-pull series-resonant converter switching power supplies with regulated output voltage
US5490052A (en) * 1992-04-24 1996-02-06 Matsushita Electric Industrial Co., Ltd. Switching power supply
US5448465A (en) * 1992-08-25 1995-09-05 Matsushita Electric Industrial Co., Ltd. Switching power supply without switching induced spike voltages
WO2001026207A2 (en) * 1999-10-01 2001-04-12 Online Power Supply, Inc. Non-saturating magnetic element(s) power converters and surge protection
WO2001026207A3 (en) * 1999-10-01 2002-01-17 Online Power Supply Inc Non-saturating magnetic element(s) power converters and surge protection
US6493242B1 (en) 1999-10-01 2002-12-10 Online Power Supply, Inc. Power factor controller
US6504423B2 (en) 1999-10-01 2003-01-07 Online Power Supply, Inc. Solid state driving circuit
US6507501B2 (en) 1999-10-01 2003-01-14 Online Power Supply, Inc. Individual or distributed non-saturating magnetic element(s) (referenced herein as NSME) power converters
US6567281B2 (en) 1999-10-01 2003-05-20 Online Power Supply, Inc. Individual or distributed non-saturating magnetic element(s) power converters and multi-stage converters
US20040012986A1 (en) * 2002-07-22 2004-01-22 Riggio Christopher Allen Two-stage converter using low permeability magnetics
US6952355B2 (en) 2002-07-22 2005-10-04 Ops Power Llc Two-stage converter using low permeability magnetics
US20090262562A1 (en) * 2008-04-22 2009-10-22 Zaohong Yang Efficiency improvement in power factor correction
CN101626197A (zh) * 2008-04-22 2010-01-13 弗莱克斯电子有限责任公司 功率因数校正的效率改进
US8363439B2 (en) * 2008-04-22 2013-01-29 Flextronics Ap, Llc Efficiency improvement in power factor correction
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FR2435850A1 (fr) 1980-04-04
GB2029989A (en) 1980-03-26
IT7925590A0 (it) 1979-09-10
GB2029989B (en) 1983-03-02
JPS5541193A (en) 1980-03-22
FR2435850B1 (nl) 1984-06-08
IT1122540B (it) 1986-04-23
BE878700A (fr) 1980-03-10
ES483999A1 (es) 1980-04-01
DE2935811A1 (de) 1980-03-13
NL7809226A (nl) 1980-03-13
AU5067879A (en) 1980-03-20

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