US4019140A - Methods and apparatus for reducing intelligible crosstalk in single sideband radio systems - Google Patents

Methods and apparatus for reducing intelligible crosstalk in single sideband radio systems Download PDF

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US4019140A
US4019140A US05/625,491 US62549175A US4019140A US 4019140 A US4019140 A US 4019140A US 62549175 A US62549175 A US 62549175A US 4019140 A US4019140 A US 4019140A
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signal
amplitude
location
carrier wave
coding
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Richard Barry Swerdlow
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AT&T Corp
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Bell Telephone Laboratories Inc
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Priority to US05/625,491 priority Critical patent/US4019140A/en
Priority to CA262,044A priority patent/CA1066362A/en
Priority to GB43378/76A priority patent/GB1561699A/en
Priority to NL7611677A priority patent/NL7611677A/xx
Priority to JP51125575A priority patent/JPS5253611A/ja
Priority to SE7611708A priority patent/SE413452B/xx
Priority to FR7631915A priority patent/FR2329037A1/fr
Priority to ES452652A priority patent/ES452652A1/es
Priority to BE171736A priority patent/BE847569A/xx
Priority to DE2648273A priority patent/DE2648273C2/de
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B14/00—Transmission systems not characterised by the medium used for transmission
    • H04B14/002—Transmission systems not characterised by the medium used for transmission characterised by the use of a carrier modulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/68—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission for wholly or partially suppressing the carrier or one side band
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03B—GENERATION OF OSCILLATIONS, DIRECTLY OR BY FREQUENCY-CHANGING, BY CIRCUITS EMPLOYING ACTIVE ELEMENTS WHICH OPERATE IN A NON-SWITCHING MANNER; GENERATION OF NOISE BY SUCH CIRCUITS
    • H03B2200/00—Indexing scheme relating to details of oscillators covered by H03B
    • H03B2200/006—Functional aspects of oscillators
    • H03B2200/0092—Measures to linearise or reduce distortion of oscillator characteristics

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  • this invention relates to radio transmission. More particularly, in a preferred embodiment, this invention relates to methods and apparatus for eliminating crosstalk in radio systems of the type that employ single-sideband modulation.
  • the high index components are located well below the message band and their effect is to spread or smear the message load in the frequency domain.
  • burble the high index components
  • the interference remains smeared but the desired signal is properly demodulated.
  • the smearing of an FM signal has two basic effects; first, it reduces the amplitude of the signal in each interfering voice circuit; and second, it mixes several interfering circuits so that they appear together at one desired location in the receiver. The net result is that, on the average, the interference power into a desired circuit is the same as without burble but the noise is now close to thermal in nature and consequently, much less annoying.
  • tone interference for example, from multifrequency operator or customer dialing tones, or from supervisory signals transmitted on the voice channels, would remain a problem.
  • phase-modulation may be employed to perform the desired coding of the single-sideband signal.
  • This low-frequency modulation is advantageously applied to the SSB carrier wave at the transmitter, prior to its modulation by the baseband signal.
  • the signal employed to generate the phase-modulation may be periodic, for example, a sinusoidal wave or pseudo-random noise, or aperiodic, for example, a truly random noise. In the latter case, the random code which is used to encode the SSB signal must be transmitted to the receiving location, along with the phase-modulated SSB signal, to ensure proper demodulation.
  • FIG. 1 is a block schematic diagram of an illustrative radio transmission system employing coded phase-modulation according to the invention
  • FIG. 2 is a diagram which illustrates various frequency spectra generated by the circuit of FIG. 1;
  • FIG. 3 is a graph showing how the harmonic power in a signal coded by the apparatus shown in FIG. 1 varies with the modulation index used to generate the same;
  • FIG. 4 is a diagram which illustrates how the apparatus shown in FIG. 1 acts to render interference crosstalk unintelligible
  • FIG. 5 is a block schematic diagram of a second illustrative embodiment of the invention wherein a coding pilot tone is added to the message signal, prior to amplitude modulation thereof;
  • FIG. 6 is a block schematic diagram of a third illustrative embodiment of the invention wherein a pilot coding tone is transmitted from the encoder to the decoder;
  • FIG. 7 is a graph depicting the effect of delay in the repeatered transmission medium, as a function of frequency
  • FIG. 8 is a graph depicting the effect of uncompensated channel delay upon distortion.
  • FIG. 9 is a graph depicting the effect of the index of modulation on residual AM distortion.
  • FIG. 1 depicts an illustrative radio transmission system according to the invention.
  • the system includes an encoder 10, illustratively located at the transmitting terminal of the system, and a decoder 20 correspondingly located at the receiving terminal.
  • Coder 10 includes a carrier oscillator 11 which applies a carrier wave to the input of an amplitude modulator 12, the output of which is connected to a bandpass filter 13, thence to a repeatered transmission medium shown schematically as element 14.
  • a phase-modulator 16 is interposed between the output of carrier oscillator 11 and the input to modulator 12.
  • a code generator 17 is connected to the input of phase-modulator 16 to control the nature of the phase-modulation which is impressed on the output of oscillator 11.
  • the decoder 20 includes an amplitude demodulator 21 which, in addition to the signal from repeatered transmission medium 14, also receives the output of a local oscillator 22.
  • the output of demodulator 21 is passed through a bandpass filter 23, the filtered output of which comprises the output of the decoder.
  • a phase-modulator 24 is interposed between the output of local oscillator 22 and the input to demodulator 21.
  • a code generator 26 is connected to the input of phase-modulator 24 to control the nature of the phase-modulation which is impressed on the output of oscillator 22.
  • FIG. 1 is a conventional SSB radio system.
  • the input to modulator 12, designated as w(t), is the baseband signal to be transmitted to the distant end of the system.
  • This baseband signal may be a single voice channel or a single sinusoidal tone, but more typically comprises a mastergroup, or combination of mastergroups, of a communications multiplex system.
  • phase-modulator 16 modulates the phase of the output signal from oscillator 11 with the output of code generator 17, which output is designated as f(t).
  • f(t) is considered periodic although, as will be explained later, f(t) may also be aperiodic.
  • ⁇ o is a carrier in the IF frequency range, typically 74 MHz.
  • x(t) the output of filter 13, will then be a single sideband signal in the 55-85 MHz band.
  • This SSB signal is then heterodyned in the normal manner up to the 4-6 GHz common carrier microwave band for transmission over the repeatered transmission medium.
  • the RF upconverter at the transmitting location and the corresponding downconverter at the receiving location have been omitted from the drawing.
  • the form of f(t) can range from a pure sinusoid to a noiselike signal.
  • the output of modulator 12, designated u(t) comprises the carrier ⁇ o and the upper and lower AM sidebands generated by modulator 12.
  • Filter 13 is selected so that it removes the carrier frequency ⁇ o and one of the sidebands generated by modulator 12 and, thus, generates x(t) the desired SSB signal.
  • the output of encoder 10, x(t) comprises a single sideband signal having a bandwidth which is slightly expanded with respect to the bandwidth of the baseband signal, w(t).
  • the reason for this expansion is the phase-modulation of the carrier introduced by phase-modulator 16.
  • the spectrum of x(t) is the convolution of the spectrum of w(t) with the spectrum of cos[ ⁇ o t+f(t)], the output of modulator 16.
  • the spectrum of this latter signal depends on the index of modulation and the highest frequency of f(t). This relationship may be more clearly seen in FIG.
  • graph (a) represents the spectrum of the convoluted signal cos[ ⁇ o t+f(t)], that is, the output of phase-modulator 16;
  • graph (b) represents the spectrum of the baseband signal, w(t), for a relatively simple case;
  • graph (c) represents the spectrum of u(t), that is, the output of modulator 12;
  • graph (d) represents the spectrum of the signal x(t), that is, the output of filter 13 after the carrier and lower sideband have been suppressed.
  • transmission medium 14 will comprise several microwave repeater stations, the free space between the towers and the terminal equipment involved in amplifying and filtering the signal.
  • Transmission medium 14 is assumed linear but may include components tending to produce spurious signals of low amplitude, due to non-linear circuit elements in the medium.
  • the output of the repeatered transmission medium designated as y(t), is applied to the input of AM demodulator 21.
  • the output of phase-modulator 24, that is the demodulating signal for decoder 20 is given by the expression cos[ ⁇ o t+g(t)].
  • the spectrum of u(t) is represented by waveform (c) in FIG. 2.
  • waveform (c) the spectrum of cos[ ⁇ o t+f(t)] extends to infinity on both sides so that, in theory, each sideband of u(t) overlaps the other.
  • the spectral energy decreases so rapidly that deleting one of the cosine terms in the mathematical expression is essentially the same as dropping a sideband, for example, by filtering the signal in a bandpass filter. For example, if we assume that f(t), the input to phase modulator 16 in FIG.
  • FIG. 3 where the above power ratio is plotted on the ordinate against n, the harmonic number for several different values of X. Even though only values at integer values of n are defined, the points have been joined for convenience. Any particular curve was ended when either the next value was below -60 dB or beyond the fourteenth harmonic. As a rule of thumb, the bulk of the spectral power is contained in a frequency range X ⁇ c . There is no sensible power beyond the sixteenth harmonic of f(t). For example, if the baseband signal to be coded had a lower cut-off frequency of 564 kHz, the upper bound on the coding frequency would be 34 kHz for an index of 3 ⁇ since the sixteenth harmonic of 35 kHz is 560 kHz.
  • decoder 20 which is identical with the baseband input to the distant encoder, but reduced in amplitude.
  • encoder 10 may be viewed as AM modulation with a local oscillator which has a great deal of phase noise, except that the noise is accurately known and well controlled.
  • the decoder 20 behaves like a phase-locked loop which follows the noisy carrier, except that it has available to it an exact replica of the noise.
  • control tones be placed on the channel after coding to prevent their being spread by the encoding process.
  • the composite signal will then be operated on by any repeater equalization and linearization contained in the transmission medium 14 in the same manner as if no coding were present.
  • the only constraint is that control tones be far enough away from the signal to avoid the dispersed spectrum. Since dispersion will generally be less than 60 kHz and control signals can be placed in slots 100 kHz or more wide, this proves to be no problem in practice.
  • a co-channel interfering signal for example, from some undefined interference source 30 which injects an RF signal somewhere along the repeatered transmission medium 14, or at the input to decoder 20.
  • the input to the decoder now comprises two signals, Y 1 (t) and Y 2 (t), where Y 2 (t) represents the interfering signal which, for the moment, is assumed to have no coding.
  • decoder 20 Since the operation of decoder 20 is linear, superposition can be applied so that the response to the combined signal Y 1 (t) + Y 2 (t) is the sum of the separate response to each component signal.
  • the desired signal Y 1 (t) will be decoded in the normal manner but the interfering signal will, in effect, be coded by the decoder. Its spectrum at the decoder output will therefore be convolved with the spectrum of cos[ ⁇ o t+f(t)] and will produce in the decoder output, z(t), essentially the same effect as does burble in an FM system. That is, each interfering voice circuit will be attenuated and several different voice circuits will be mixed together.
  • graph (a) represents the positive frequency spectrum of Y 1 (t), the signal corresponding to the baseband signal w(t), here, for simplicity, considered to be a pure sinusoidal tone.
  • Graph (b) of FIG. 4 represents the spectrum of the interfering tone, Y 2 (t), again considered for simplicity to be a simple sinusoidal tone.
  • Graph (c) represents the linear superposition of these two signals at the output of filter 23 and it is evident from the drawing that in the real situation, where w(t) is more complex, the interfering tone will be smeared so that it will be audible, if at all, with reduced amplitude over one or more voice channels in the mastergroup comprising w(t).
  • the interference power with or without coding is the same.
  • the burble i.e., code
  • the annoyance is significantly less than for unburbled interference.
  • a proposed CCITT requirement on the tone interference level allowable in a 40 dBrncO circuit is -68 dBmO.
  • the same circuit is allowed up to -48 dBmO of thermal noise. Since burble renders a tone noise-like, it is clear that a 20 dB reduction in subscriber annoyance from tones is possible with FM coding.
  • tone codes are easy to implement, but their subjective effects are not as effective in reducing annoyance as pseudo-random or random codes.
  • Decoders can be built for tone codes that are self-identifying, that is, the craftsperson need not set the tone code at the receiver. The number of acceptable tone codes is limited to perhaps ten.
  • pseudo-random codes the subjective effects are as good as for random codes, which are the best from this standpoint, but require setting the code at the receiver for best results. There should be little difficulty in achieving several tens of pseudo-random codes.
  • the random code the subjective effects are best, and there is no administrative problem since all codes are different. The problem here is that the decoder must reproduce the code, which requires generally a high signal-to-noise on the pilot sending the code.
  • decoders There are two types of decoders that will be discussed. The one, used as example so far, is where the code is available at both transmit and receive ends. In this case, only a synchronization signal need be sent. This synchronization signal would need roughly the same signal-to-noise as pilots used for carrier regeneration.
  • the second type of decoder is one which detects the code that was sent and uses it to decode the signal. One way to accomplish this at the receiver is to use a phase-locked loop on a pilot that has been coded. The signal-to-noise required for such a pilot, however, is far larger than that for the first receiver, because additive system noise present near the pilot will be transferred by the loop onto the load. This second type of receiver, however, must be used for random codes. Either receiver may be used for the tone or pseudo-random codes.
  • ⁇ c the pilot frequency
  • This frequency must be greater than several hundred Hz if the intelligibility of the crosstalk is to be suppressed, but if it is made too small, several harmonics of ⁇ c will lie in a voice circuit and thereby raise the total interference power from one source.
  • only one harmonic per voice circuit should be present and this suggests a coding frequency of about 2 kHz, which is at least several hundred Hz from the nearest tone frequency customarily employed on voice circuits for supervision purposes.
  • ⁇ c may be increased until the dispersing spectrum becomes too large to meet minimal error on decoding.
  • a particularly attractive feature of a single-frequency code is its self-identification. All code frequencies could be designed to fit within the capture range of the synchronization circuitry associated with the decoder, but once lock is established, the decoder has both the code and its synchronization. Thus, no code selection is needed at the receiver which makes administration of the overall coding system considerably easier to implement. This is essentially the second type of receiver discussed above.
  • a receiver which employs a code differing from the code of an interfering signal should produce a flat spectrum which is down always by the same amount and over the same bandwidth for any tone in the interfering signal.
  • One way to closely approach the ideal flat spectrum mentioned above, but retaining the convenience of periodicity, is to use pseudo-random noise as the encoding signal.
  • this type of signal may be generated by filtering a broadband binary signal which is generated by conventional digital techniques.
  • a counter operating at a clock rate of from ten to twenty times that of the desired noise cut-off frequency, as calculated below to achieve a given degree of crosstalk suppression, will generate a binary sequence with a period in the millisecond range which sequence is then filtered to the proper bandwidth.
  • the resultant signal is close to Gaussian statistics (over a period) but is periodic at the counter period.
  • This signal becomes the coding signal, f(t), which is then used to phase-modulate the carrier frequency ⁇ o .
  • f(t) the carrier frequency
  • a sufficiently high index of modulation is used to place almost all of the power into the first order sidebands.
  • carrier power is reduced by e -D 11 where D 323 is the mean square phase deviation of the coding signal in square radians.
  • the attenuation of an interfering 3 kHz voice circuit is: ##EQU4## where A s is the spectral power per 3 kHz slot below unmodulated carrier, in dB
  • f t is the baseband noise cut-off frequency, in Hz.
  • a 15 kHz noise bandwidth assures a flat 10 dB suppression and a mean square phase deviation of 2.3 suppresses the carrier by 10 dB.
  • the second order sidebands are important, giving a total dispersion bandwidth of about 60 kHz.
  • an automatic gain circuit is necessary at the decoder to accurately control the modulation index.
  • the decoded spectrum for an interference signal is similar for all possible codes, uniform suppression and spreading of the interfering signal is possible for all codes.
  • code generation is fairly simple; the same synchronizing frequency is used for all codes but the sequence is changed. This amounts to changing the switch settings on the binary counter used to generate the pseudo-random noise and several tens of separate codes are easily obtainable, for example, with TTL logic.
  • the second type of decoder mentioned above and discussed more thoroughly below, could be used.
  • the noise-like signal has the advantage of uniformity of action on interference over a large number of codes.
  • Both the single-tone code and the pseudo-random code approximate the burble generated on FM systems because in either case the interference is modulated by the cosine of a large angle variation.
  • the pseudo-random signal alone comes as close as a periodic signal can come to the waveform present on FM systems.
  • Both waveforms suggested above are periodic so that the dispersing spectrum comprises a set of tones. For the case of psuedo-random noise these tones are closely spaced perhaps by as little as 50-100 Hz.
  • FIG. 5 illustrates an embodiment of the invention wherein, at the transmitting location, the output of the pilot oscillator, ⁇ c is added to the baseband signal w(t), prior to modulation thereof in modulator 12.
  • code generator 17 produces an appropriately band-limited f(t) signal which is not known, a priori, to the decoder; for example, a truly random noise-like signal.
  • the tone before coding is: cos ⁇ c tand the tone after coding is: cos[ ⁇ c t+f(t)].
  • this signal is precisely the one needed to perform the decoding operation!
  • the signal must first be shifted to center on the carrier frequency ⁇ o .
  • it will have on it the system noise which is added from the point at which coding was performed, assuming the ⁇ c tone was blocked and then reinserted at the originating end.
  • the decoding process itself will add noise which is proportional to the noise present with the ⁇ c tone.
  • the signal needed to demodulate (i.e., to decode) the coded waveform is
  • ⁇ o is the regenerated carrier.
  • a bandpass filter 51 is connected to the output of transmission medium 14 and the characteristics thereof are selected such that only a narrow band of frequencies centered around ⁇ o + ⁇ c are passed (assuming upper sideband selection).
  • This coding signal is then applied to phase-modulator 24 in the normal way.
  • a delay element 53 is inserted between the output of transmission medium 14 and the input to demodulator 21 to compensate for the delay introduced in f(t) by filter 51 and demodulator 52.
  • Bandpass filter 51 will also give system noise at its output. A simple analysis will be shown to indicate the magnitude of the problem.
  • the output of demodulator 21, after decoding will be, taking as a test signal a sinusoid of frequency ⁇ s ,
  • n(t) is system noise band-limited to the effective spectral width of the decoding signal.
  • the power in the resultant signal is ##EQU7## where p w and p n are the powers of the signal and noise, respectively.
  • the signal-to-noise ratio is then essentially 2/p.sub. n .2 /p n is numerically equal to the signal-to-noise ratio out of bandpass filter 51.
  • this embodiment puts at least as much noise on the signal as was present in the band centered around the ⁇ c pilot tone. However, if blocking and reinserting of tone ⁇ c is used, then this added noise is reduced, since the signal-to-noise would only be that existing between two terminals, less the terminal noise.
  • phase-demodulation of the signal prior to decoding signal reconstruction will yield a maximum signal-to-noise improvement to 10 log ⁇ 2 or about 10 dB. This improvement, coupled with a modest increase in transmitted tone level, keeps the noise contribution well below system noise.
  • a second bandpass filter 57 may be connected to the output of demodulator 21.
  • the output of filter 57 is connected to a phase-demodulator 58, thence to an error detector 59. If the code signal applied to phase-modulator 24 is the correct code, and in synchronism with the code employed at the encoder, the output of amplitude demodulator 21 will include a component which comprises a pure tone of ⁇ c .
  • phase-demodulator 58 will comprise an error signal which when applied to an error detector, for example a tuned circuit resonant at ⁇ c , will develop a voltage proportional to the disagreement between g(t) and f(t).
  • FIG. 6 is similar to FIG. 1, except that the means needed for synchronizing the coding and carrier are shown.
  • the output of a pilot generator 41 is applied to the code generator 17 to synchronize the same. If the code comprises a single-tone, the output of generator 41 may itself comprise the coding tone, and, in that event, function generator 17 may be eliminated. If, however, the code comprises pseudo-random noise, the output of pilot generator 41 acts as a clock signal to drive the pseudo-random generator in function generator 17.
  • pilot generator 41 is transmitted over the transmission medium 14 along with the message load.
  • a detection and phase-lock circuit 42 detects and locks the phase of the received pilot tone and applies it to a pilot generator 43 to synchronize the same.
  • Generator 43 generates a local pilot tone, also designated ⁇ c , which is synchronized in frequency and phase with the pilot tone employed at the transmitting location.
  • generator 43 includes an automatic gain circuit to maintain the amplitude of ⁇ c at a constant level even if the amplitude of the incoming pilot tone from transmission medium 14 should change. This AGC is needed if the code is a tone. If the code is pseudo-random, the AGC is needed on block 26.
  • a carrier regeneration circuit 46 amplifies and filters the received carrier signal, then applies the signal to local oscillator 22 to synchronize the same.
  • FIG. 6 separate paths are shown for the synchronizing signals ⁇ c and ⁇ o . In practice, these tones are sent along with the coded output of modulator 12.
  • Carrier regeneration is, of course, tightly linked to coding since the phase modulation shown in FIG. 2 must have a carrier. After carrier regeneration has occurred on the carrier frequency ⁇ o , a separate lock using the pilot signal ⁇ c is established for g(t), the phase modulation on the carrier.
  • FIG. 7 shows a representative phase characteristic for repeatered medium 14 with illustrative pilot frequency positions superimposed thereon.
  • the medium 14 has delay distortion. Then the delay is different at different frequencies within the band.
  • the pilot being used for decoding is at ⁇ c , and we consider a voice circuit at ⁇ c + ⁇ . If the differential delay at these two frequencies is ⁇ , then the results above for synchronization errors are directly applicable, and FIG. 8 may be used.
  • the voice circuit at ⁇ c + ⁇ had a data tone in it. Then for a code of ⁇ radians index and differential delay of 100 nanoseconds, the data tone will have sidebands of about 54 dB down. If these sidebands fall into adjacent circuits carrying voice, these tones from the phase coding would constitute a serious interference in themselves.
  • the medium must have low values of delay distortion, and the code index and highest frequency must be chosen for acceptable values of residual interferences.
  • the effects of amplitude distortions are mostly felt in how they distort the code being sent with the synchronization signal, and requirements are not nearly so demanding as for delay distortion.
  • each intermodulation component is coded in exactly the same manner as is the desired signal. This result will hold for all odd orders of inband intermodulation, by extension of the above argument. Thus, coding has no effect on pertinent intermodulation for a narrow-band system. This result may also be extended to non-linearities with memory.
  • the decoded baseband signal is, in the notation previously used,
  • FIG. 9 shows the ratio of residual AM power to desired power (in dB) as a function of modulation index.
  • shape distortions if the waveform of g(t) is distorted relative to f(t) (see FIG. 1), the effect is similar to the situation wherein g(t) is delayed relative to f(t).
  • shape distortions can be treated as a sum of amplitude and phase distortion.
  • f(t) is a cosine wave
  • the modulation index might not be identical at the encoder and decoder. This might occur, for example, if the AGC control circuit at the decoder were defective, causing g(t) to have too high or too low a level relative to f(t).
  • the dB error is ##EQU12## for 50 dB suppression with X - 2 ⁇ , then ⁇ ⁇ 0.1 percent. If the predominant error is amplitude distortion and not waveform distortion, this accuracy might be achieved by the use of an AGC circuit referenced to a voltage standard. The waveform distortion under these circumstances may be analyzed in detail using techniques previously considered.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Noise Elimination (AREA)
  • Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
  • Dc Digital Transmission (AREA)
  • Transmission Systems Not Characterized By The Medium Used For Transmission (AREA)
US05/625,491 1975-10-24 1975-10-24 Methods and apparatus for reducing intelligible crosstalk in single sideband radio systems Expired - Lifetime US4019140A (en)

Priority Applications (10)

Application Number Priority Date Filing Date Title
US05/625,491 US4019140A (en) 1975-10-24 1975-10-24 Methods and apparatus for reducing intelligible crosstalk in single sideband radio systems
CA262,044A CA1066362A (en) 1975-10-24 1976-09-24 Methods and apparatus for reducing intelligible crosstalk in single sideband radio systems
GB43378/76A GB1561699A (en) 1975-10-24 1976-10-19 Signal transmission systems
JP51125575A JPS5253611A (en) 1975-10-24 1976-10-21 Method of transmitting information
NL7611677A NL7611677A (nl) 1975-10-24 1976-10-21 Werkwijze en inrichting voor het verminderen van verstaanbaar overspreken in een-zijband- radiostelsels.
SE7611708A SE413452B (sv) 1975-10-24 1976-10-21 Sett och anordning for reducering av uppfattbar overhorning vid radioforbindelser med enkelt sidband
FR7631915A FR2329037A1 (fr) 1975-10-24 1976-10-22 Procede et appareil de transmission d'information
ES452652A ES452652A1 (es) 1975-10-24 1976-10-22 Perfeccionamientos en dispositivos para transmitir una in- formacion desde un primer a un segundo lugar.
BE171736A BE847569A (fr) 1975-10-24 1976-10-22 Procede et appareil de transmission d'information,
DE2648273A DE2648273C2 (de) 1975-10-24 1976-10-25 Einseitenband-Verfahren zur Informationsübertragung und Vorrichtung zur Durchführung des Verfahrens

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BE (1) BE847569A (cs)
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DE (1) DE2648273C2 (cs)
ES (1) ES452652A1 (cs)
FR (1) FR2329037A1 (cs)
GB (1) GB1561699A (cs)
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US4432079A (en) * 1981-11-02 1984-02-14 The United States Of America As Represented By The Secretary Of The Navy Synchronous/asynchronous independent single sideband acoustic telemetry
US4462114A (en) * 1980-07-02 1984-07-24 Motorola, Inc. Signum signal generator
US4479226A (en) * 1982-03-29 1984-10-23 At&T Bell Laboratories Frequency-hopped single sideband mobile radio system
US4541118A (en) * 1983-12-22 1985-09-10 Motorola, Inc. SSB System with pilot coded squelch
US4675880A (en) * 1985-05-02 1987-06-23 The United States Of America As Represented By The Administrator Of The National Aeronautics And Space Administration Antimultipath communication by injecting tone into null in signal spectrum
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CA1066362A (en) 1979-11-13
JPS5253611A (en) 1977-04-30
GB1561699A (en) 1980-02-27
ES452652A1 (es) 1977-10-16
DE2648273A1 (de) 1977-04-28
BE847569A (fr) 1977-02-14
SE413452B (sv) 1980-05-27
FR2329037A1 (fr) 1977-05-20
SE7611708L (sv) 1977-04-25
NL7611677A (nl) 1977-04-26
DE2648273C2 (de) 1982-06-09
FR2329037B1 (cs) 1981-05-22

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