US3769542A - Flyback eht and sawtooth current generator having a flyback period of at least sixth order - Google Patents

Flyback eht and sawtooth current generator having a flyback period of at least sixth order Download PDF

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US3769542A
US3769542A US00245144A US3769542DA US3769542A US 3769542 A US3769542 A US 3769542A US 00245144 A US00245144 A US 00245144A US 3769542D A US3769542D A US 3769542DA US 3769542 A US3769542 A US 3769542A
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network
flyback
eht
sawtooth current
period
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R Pieters
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US Philips Corp
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US Philips Corp
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    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K4/00—Generating pulses having essentially a finite slope or stepped portions
    • H03K4/06—Generating pulses having essentially a finite slope or stepped portions having triangular shape
    • H03K4/08—Generating pulses having essentially a finite slope or stepped portions having triangular shape having sawtooth shape
    • H03K4/48—Generating pulses having essentially a finite slope or stepped portions having triangular shape having sawtooth shape using as active elements semiconductor devices
    • H03K4/60—Generating pulses having essentially a finite slope or stepped portions having triangular shape having sawtooth shape using as active elements semiconductor devices in which a sawtooth current is produced through an inductor
    • H03K4/62—Generating pulses having essentially a finite slope or stepped portions having triangular shape having sawtooth shape using as active elements semiconductor devices in which a sawtooth current is produced through an inductor using a semiconductor device operating as a switching device
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
    • H04N3/00—Scanning details of television systems; Combination thereof with generation of supply voltages
    • H04N3/10—Scanning details of television systems; Combination thereof with generation of supply voltages by means not exclusively optical-mechanical
    • H04N3/16—Scanning details of television systems; Combination thereof with generation of supply voltages by means not exclusively optical-mechanical by deflecting electron beam in cathode-ray tube, e.g. scanning corrections
    • H04N3/18—Generation of supply voltages, in combination with electron beam deflecting

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  • ABSTRACT A flyback EHT and sawtooth current generator for television display apparatus including a transformer and switching means connected thereto which are conducting during a scan period and are nonconducting during a flyback period, and reactances whether parasitic or nuts connected to the transformer which together with the transformer constitute a network having a minimum of three prescribed resonant frequencies during the flyback so that a scan which is substantially without any oscillations and a flyback EHT pulse having a wide and flat peak is produced.
  • the invention relates to a flyback EHT and sawtooth current generator particularly for television display apparatus, including switching means which are periodically non-conducting during a flyback period T and are conducting during a scan period T 'r and a network having input terminals connected to the switching means, said network comprising a transformer having at least one primary winding and possibly one or more coils connected thereto, through which said sawtooth current flows during the scan period, and a secondary winding to which a rectifier circuit is connected for generating said EHT from the voltage pulses occurring during the flyback period at the secondary winding, said network having during the flyback period a first resonant frequency f which is at least substantially equal to the expression in which K l and S is a correction factor which is equal to the relative reduction of the slope of the sawtooth current at the end of the scan period relative to this slope at the centre of the
  • EHT and sawtooth current generators In addition to the requirement that the scan period must be without free oscillations a quite different requirement imposed for EHT and sawtooth current generators is that the EHT generated varies as little as possible when the EHT load varies.
  • the generated EHT is generally used in television display apparatus as an acceleration voltage for the beam current in the display tube in which the average magnitude of the beam current, which is dependent on the average luminance of the scene displayed, constitutes the EHT load.
  • the extent to which the generated EHT is dependent on its load is expressed by the so-called EHT R which is the quotient between an EHT variation and the beam current variation causing this EHT variation. As the EHT R, of a flyback EHT and sawtooth current generator is lower, the EHT is less dependent on its load.
  • the Applicant's Netherlands Patent application 6714750 proposes to proportion the network in such a manner that the second frequency of the said TGSOLWTI frequencies is at least substantially equal to the expression mentioned in the preamble for K 5 (fifth harmonic tuning) while in addition the inductance (partly leakage inductance) and the capacitance (partly para sitic capacitance) present between the primary windings and the secondary winding are chosen to be such that the flyback pulse applied to the EHT rectifier has a peak which is as wide and as flat as possible.
  • An object of the invention is to provide a flyback EHT and sawtooth current generator in which a considerable further reduction of the EHT R, (to approximately 200 K0.) is possible while maintaining substantially no free oscillations and to this end the flyback EHT and sawtooth current generator according to the invention is characterized in that said network includes further reactances increasing the order of the network during the flyback period to a minimum of 6 to such an extent that the network has a third resonant frequency f while is at least substantially equal to the abovementioned expression for K 7.
  • a flyback period network of at least 6th order having at least three resonant frequencies all of which substantially (i 10 percent) satisfy the given expression for K l, 5 and 7, respectively. Then substantial freedom of oscillations during the scan period is obtained. In practice, some oscillations are found to occur predominantly as a result of the inevitable losses present in the network.
  • Both the flyback pulse occurring across the switching means and the flyback pulse applied to the rectifier now include three components, one having a frequency f one having a frequency f and one having a fre quency f,,.
  • the mutual amplitude ratios between these components determine the shape of the two pulses.
  • a network having a plurality of resonant frequencies of fixed values may be built up in many different manners; some examples thereof will be given with reference to the Figures. It is found, however, that independent of the manner in which the network is built up the mutual amplitude ratios of the three components of the flyback pulse occurring across the switching means and hence the shape of this pulse is always dependent in the same manner on the three resonant frequencies f ,f and f, as well as on the frequencies f and f (lying between fa .and fs and betweenf and f,, respectively) at which the input im-
  • the circuit arrangements according to the invention provide a possibility of obtaining a secondary flyback pulse having a considerably wider and flatter peak than is the case in the above-mentioned known circuit arrangements.
  • the shape of the secondary flyback pulse is determined by the mutual amplitude ratios between its components. Furthermore it has been found that in whatever manner the network is built up, these mutual amplitude ratios and hence the shape of the secondary flyback pulse are always dependent in the same manner on the location of the possibly present zero transmission frequencies relative to the three resonant frequencies which are present and which are determined by the expression given in the preamble.
  • the zero transmission frequencies of the network are those frequencies at which there is no transfer of energy from the input terminals to the rectifier circuit in the network which is assumed to be free from losses.
  • the number and the location of the zero transmission frequencies is indefinite and is dependent on the structure of the network.
  • FIGS. 1, 4, 5 and 7 show different embodiments of a flyback EHT and sawtooth current generator according to the invention
  • FIGS. 1a, lb, 10, 5a, 5b and 7a show different equivalent circuit diagrams of the respective embodiments and FIGS. 2, 3, 6 and 8 show characteristic fields for further illustration of the proportioning of the different networks.
  • FIG. 1 shows a transformer having a primary winding 2, one or more auxiliary windings 3 rigidly coupled to the primary winding, a secondary winding 4 and a tertiary winding 5.
  • a tap 6 on the primary winding is connected to the positive terminal of a voltage supply source 7 the negative terminal of which is connected to ground.
  • the series arrangement of a plurality of deflection coils 9, a linearity corrector l0 and an S-correction capacitor 11 is arranged between a second tap 8 and the lower end of the primary winding.
  • the tap 8 and the lower end of the primary winding are located symmetrically relative to the tap 6 so that this series arrangement is fed symmetrically relative to ground.
  • a transistor 12 operating as a switch is provided between the upper end of the primary winding and ground and a capacitor 13 is connected in parallel with this transistor.
  • Said secondary winding 4 is connected to ground at one end and at the other end to a rectifier circuit consisting of a rectifier 14 and a smoothing capacitor 15; the EHT generated by the rectifier is applied to the acceleration anode of a television display tube not further shown.
  • the lower end of the tertiary winding 5 is connected to the tap 6 of the primary winding.
  • a parallel LC- circuit which consists of an inductor l6 and a capacitor 17 is arranged between the upper end of the primary winding and the upper end of the tertiary winding.
  • Switching pulses which periodically cut off the transistor 12 at the end of each scan period are applied between the base and emitter of transistor 12 through a separating transformer 18, a series inductance 19 and a parallel diode 20.
  • the transistor 12 is a so-called slow switching transistor and the elements 19 and 20 are included so as to accelerate the switching off of the transistor at the end of the scan period.
  • FIG. la shows a first equivalent circuit diagram.
  • a switch SW denotes the electronic switch constituted by transistor 12 and diode 20.
  • the portion of the primary winding between the upper end of this winding and the tap 6 of FIG. 1 is denoted by 2'.
  • the inductance of the deflection coils 9 and the linearity corrector 10 is denoted by 21, but transformed to said portion of the primary winding.
  • 11 denotes the S- correction capacitor 11 likewise transformed to said portion.
  • the parasitic capacitance of the secondary winding as well as the input capacitance of the rectifier circuit is denoted by 22.
  • the mutual magnetic couplings between the primary, secondary and tertiary transformer windings are denoted by M1, M2 and M3.
  • FIG. lb A further equivalent circuit diagram is shown in FIG. lb. in this Figure the S-correction capacitor 11 is omitted because it has such a high value that it exerts substantially no influence during the flyback period.
  • the capacitor 13 is directly provided across the inductor 21 which is admissible because the impedance of the source 7 is very low.
  • the circuit diagram shows the magnetizing inductance 22 of the primary winding, the magnetizing inductance 23 of the tertiary winding and the parasitic capacitance 24 of this winding, the inductor 16 and the capacitor 17, the leakage inductance 25 between the primary and the tertiary winding and in parallel therewith the parasitic capacitance 26 between these windings, the leakage inductance 27 between the tertiary and the secondary winding and in parallel therewith the parasitic capacitance 28 between these windings, the leakage inductance 29 between the primary and the secondary winding and in parallel therewith the parasitic capacitance 30 between these windings and finally the capacitance 22 which represents the transform of the capacitor 22 of FIG. la.
  • switch SW is closed.
  • the voltage E from voltage supply source 7 is therefore present across capacitor C1 and also across inductor L1.
  • a (sawtooth) current linearly varying with time will flow through the inductor Ll.
  • switch SW is rendered non-conducting, free oscillations will occur in the network as a result of the magnetic energy present in L1.
  • These oscillations produce pulsatory voltages V] and V2, the so-called flyback pulses across capacitors C1 and C2, respectively.
  • the sawtooth current in the circuit diagram of FIG. 1 flows during the first part of the scan period through the diode 20, the base-collector junction of the transistor and subsequently through the transformer and the deflection coils to the voltage supply source and thus feeds back energy to the voltage supply source.
  • the base-emitterjunction of the transistor is rendered conducting by means of the pulses applied to the base electrode of the transistor so that during the second part of the flyback period the sawtooth current now reversed in polarity can flow from the voltage supply source through the transformer and the deflection coils and subsequently through the collector electrode and the emitter electrode of the transistor to ground, while the voltage supply source supplies energy to the network.
  • n liol 3 1 n liol
  • 1' is the duration of the flyback period
  • K is each odd positive integer
  • i is the value of the saw tooth current at the commencement of the flyback period
  • 1" is the derivative with time of this current at the commencement of the flyback period
  • d is a phase angle. It is possible to eliminate qb from the two equations. Then a power series in 'r i'o/l ⁇ , is produced for a r. If this series is limited to the first two terms it is found that:
  • the flyback network is built up in such a manner that a network of the 6th order at a minimum is obtained having at least three resonant frequenciesf ,f 6 and f and the porportioning of the network values is chosen in such a manner thatf a satisfies equation III for K l,f for K 5 and f for K 7.
  • the equation for the free oscillations of the network is: p p +11 p 6 p y) O in which p is the differential operator transformed in accordance with the Laplace arithmetic and a, e and y represent the three resonant frequencies of the flyback network, expressed in rad/sec.
  • a further consideration of the equivalent circuit diagram of FIG. lb shows that seven inductors are present therein whose currents can all be given independently of each other; however, only three voltages of the seven capacitors present can be given independently of each other, for example the voltages of the capacitors 13, 24 and 22, then the voltages across capacitors 17, 26, 28 and 30 are determined.
  • the network therefore has the order 10.
  • there are four direct current meshes present namely a first mesh 21, 22, a second mesh 22, 16, 23, a third mesh 16, 25 and a fourth mesh 25, 27, 29.
  • the equation for the free oscillations therefore becomes:
  • this input impedance of the network When the input impedance of the network is determined as a function of the frequency, i.e. the impedance at the terminals to which the switching means SW are connected, this input impedance will be at a maximum at the three resonant frequencies a, e and y. As is known a frequency is present between a and e and between 6 and y at which the input impedance is at a minimum (in a network without resistors equal to 0). These two zero impedance frequencies are hereinafter denoted by f and f and the associated angular frequencies are denoted by ,8 and ,8 Thus there always applies that:
  • the flyback pulse which will be produced during the flyback period at the input of the network has three sine functions, one for each frPquency, as a result of the three resonant frequencies of the network.
  • the shape of the flyback pulse is dependent on the mutual amplitude ratios of the three respective sine oscillations.
  • the amplitudes are, respectively, (A a/sin (b a), (A e/sin d) e) and (A /sin da Since in practical cases d) a, d) e and 4),, are only small phase angles, sin (1)01, sin (1) e and sin 4),, may be replaced by lg d) a lg d) E and lg (1:, with which the amplitudes change to
  • equation ll applies to each resonant frequency, a lg (b a e tgd) 6 ylg i /i
  • P yAy/ a
  • FIG. 2 shows a characteristics field in which horizontally B, varies from a to e and vertically B varies from 6 to y.
  • the Figure shows lines of constant P, and constant P These lines clearly show that as the left top corner of the field is more and more approached the ratio P, increases; that is to say, the amplitude of the 6 component relative to the amplitude of the a component increases. Likewise the amplitude of the y-component will increase relative to that of the a-component as the left bottom corner of the field is approached.
  • the field furthermore shows a shaded area.
  • the characteristics field and the shaded area of FIG. 2 are determined by means of a computer.
  • a, e and y are determined in accordance with equation III with K l, 5 and 7, respectively, and at a flyback ratio 'r/T 0.18 and an S-correction factor S O. Modification of this flyback ratio and of the S-correction factor within limits occurring in practice results, however, hardly in a change of the picture shown in FIG. 2.
  • the arrangement in which the generator is included is switched off so that the voltage supply source 7 does not provide voltage and the base connection of transistor 112 is disconnected.
  • a tone generator which covers the relevant frequency range is connected to the collector electrode of the transistor through a sufficiently high impedance.
  • a voltage measuring instrument having a sufficiently high input impedance is connected to the collector electrode, for example, an oscilloscope or a valve voltmeter.
  • the flyback pulse V applied to the rectifier circuit during the flyback period has a peak which is as wide and as flat as possible.
  • the flyback pulse V consists of three components, one of the frequency a, one of the frequency e and one of the frequency y.
  • E is a constant which is proportional to the supply voltage E and which is equal to E in the circuit diagram of FIG. 10.
  • 8 and 6 are the zero transmission frequencies of the network, i.e. those frequencies at which there is no energy transmission in a network which is completely without resistors between the input terminals of the network and the output terminals to which the rectifier circuit is connected. Due to the inevitable losses some energy transmission will take place at these frequencies in practice.
  • each network having three resonant frequencies which has a scan period without oscillations in the manner described.
  • the frequencies ,8, and B both of which are always present and the lowest ([3,) of which is always between a and e and the highest ([3,) is always between 6 and y both zero transmission frequencies may not always be present.
  • only one zero transmission frequency (5,) will be present or the second zero transmission frequency (8,) is so high that it may be left out of consideration.
  • only one zero transmission frequency is present namely the frequency at which the two-pole formed by L2, L3, C3 and L4 is in parallel resonance, hence 8,, (L2 L3 L4)/C3L3 (L2 L4).
  • the zero transmission frequency 8 will lie between 6 and y in such a manner that S, is between 0.09 and 0.l7 while an optimum flyback pulse is obtained at an S of approximately O. 14.
  • FIG. 3 shows with the aid of the above-mentioned equation the field of FIG. 2, but in this case with lines of constant 8 Since the frequency 8,, also determines the ratios 8, and S and hence the shape of the flyback pulse V these are also lines where the flyback pulse V has a constant shape.
  • the reference 0 denotes the line at which the ratio S 0.l7 while S O. 14 is given on line b and S 0.09 on line 0.
  • This field also shows the shaded area already mentioned with reference to FIG. 2.
  • B will therefore be chosen between the lines a and c of FIG. 3 and preferably in the non-shaded area.
  • the primary winding consists of two identical halves 2a and 2b while the voltage supply source 7 is arranged between the two halves.
  • the series arrangement of S-capacitor l1, linearity corrector 10 and deflection coils 9 is symmetrical relative to ground between the high end of the upper half 2a and the low end of the lower half 2b.
  • the transistor 12 and the capacitor 13 are included between taps 32 and 33 of the two primary halves which are arranged symmetrically relative to ground.
  • the tertiary winding 5 is also connected between these taps and this through the LC-circuit l6 and 17 and through a direct voltage isolation capacitor 31 of large value.
  • the advantage of the network of FIG. 4 relative to that of FIG. 1 is that not only the series arrangement of the elements 9, 10 and 11 but also the primary winding, the transistor 12, the capacitor 13 and substantially the tertiary winding 5 are located symmetrically relative to ground; this yields a considerable reduction of the parasitic radiation of the generator.
  • the equivalent circuit diagram of the flyback network of this embodiment is the same as shown in FIG. 10 so that also the above described phenomena are the same.
  • FIG. 5 deviates from the embodiment of FIG. 1 in that the parallel arrangement of the inductor l6 and the capacitor 17 is located between the high end of the primary winding 2 at one end and the capacitor 13 and the collector of the transistor 12 at the other end.
  • An important advantage of this circuit relative to the circuit arrangement of FIGS. 1 and 4 is that the tertiary transformer winding 5 can be omitted.
  • FIG. 5a a simplified equivalent circuit diagram can be set up as is shown in FIG. 5a. In this FIG.
  • the network of FIG. 5a is a 6th order network having three resonant frequencies a, e and y.
  • equations I and II are satisfied, while i and 1" again represent the current through L1 and the derivative of this current, respectively at the end of the scan period.
  • the equations I and II and hence the equation III derived therefrom are found to be generally applicable as conditions for a scan without oscillations.
  • equations IV, VI and VII which relate to the shape of the flyback pulse occurring across switch SW are found to be generally applicable for a flyback network having three resonant frequencies.
  • equations VIII and IX for any such network are found to be applicable. Since the equivalent circuit diagram of FIG. 5a has only one zero transmis sion frequency, namely the frequency at which L3 and C3 are in resonance, the equations X, XI and XII also apply to this equivalent circuit diagram.
  • V represents the voltage across the secondary winding 4
  • the equivalent circuit diagram of FIG. 50 would be correct when the low end of the secondary winding were directly connected to ground. However, since this low end is connected to the high end of capacitor 13 the voltage V V NV is actually applied to the rectifier in which N is the lid transformation ratio of the transformer.
  • the equivalent circuit diagram of FIG. 5 then becomes actually the diagram shown in FIG. 5b in which T is an ideal transformer having a transformation factor N because the magnetizing inductance and the leakage inductance of the transformer are already accounted for in L1 and L2.
  • a drawback of the embodiment of FIG. 5 is the following. As a result of the losses present in the flyback network and tolerance deviations a scan which is completely without any oscillation cannot be realized in practice.
  • the series arrangement of deflection coils, linearity corrector and S-capacitor is arranged through the primary winding substantially directly across switching transistor 12. This is still more evident from the equivalent circuit diagram of FIG. 1c where the inductor Ll is directly connected to switch SW. Scan oscillations which occur in the section L2, C2, L3, C3 and L4 cannot reach the deflection coils (Ll) because Ll is short circuited through the now conducting switch SW and through the voltage supply source. In the embodiment of FIG. 5 this is, however, not the case; scan oscillations which occur across the resonant circuit L3, C3 are also present across L1 and hence across the deflection coils. This causes unwanted modulation of the deflection in the display tube of the television apparatus.
  • FIG. 7a shows the equivalent circuit diagram having the most important network elements.
  • the flyback network is of the 6th order because six inductors and capacitors are present and because all capacitor voltages and induction currents can be given independently of each other.
  • there are no direct current or direct voltage solutions present so that the 6th order network has again three resonant frequencies a, e and y for which all given relations I to IX inclusive apply.
  • the network has one zero transmission frequency 8 namely the frequency at which the twopole constituted by C1, C3 and L3 is in series resonance.
  • the circuit 16-17 may be arranged, for example, in series with the deflection coils in which the B ,6 field of FIG. 8 continues to apply.
  • Such a circuit arrangement has, however, the drawback that possible remaining scan oscillations occur across the deflection coils.
  • the parallel LC-circuit I6-I7 is short-circuited, a network is always obtained having only two relevant resonant frequencies and of which the first is substantially equal to a and the second lies between 6 and y.
  • the network having a shortcircuited LC-circuit will thus firstly be proportioned in such a manner that the first of the two occurring resonant frequencies is approximately equal to a and the second lies between e and y; subsequently the LC- circuit is provided which is controlled in such a manner that a scan without oscillations as well as a flyback pulse V having a wide and flat peak is produced.
  • the frequency a will generally be equal to l/ ⁇ l LI (C1 C2). This frequency is thus determined in a first approximation by the inductance of the deflection coils and the linearity corrector, by the location of the tap on the primary winding to which these elements are connected as well as by the sum of the primary tuning capacitor 13 and the parasitic capacitance of the secondary winding and of the rectifier circuit which capacitance is multiplied by the square value of the transformation ratio of the transformer.
  • the second resonant frequency mentioned under item I and occurring when the LC-circuit is shortcircuited is determined to a considerable extent by the series arrangement ofCl and C2 and by the leakage inductance (L2 parallel to L4 in FIG. 1c and L2 in FIG. 5a 5b and 7a). Since this second resonant frequency must be between the ultimately desired values of e and y and since these are comparatively high frequencies, this means that the leakage inductance between the primary and secondary winding (in FIGS. 1 and 4 mainly the leakage inductance between tertiary and secondary windings) and the capacitance of the secondary winding are to be comparatively low. In the embodiments of FIGS.
  • the secondary and tertiary windings therefore be preferably wound across each other on the transformer core while in the embodiments of FIGS. 5 and 7 this is the case with the primary and the secondary windings.
  • the parasitic capacitance of the secondary winding may be maintained low by winding this winding in a narrow and high manner; however, this measure increases the leakage inductance with the other windings so that a comprise is to be found.
  • the num ber of turns of the secondary winding and hence both the leakage inductance and the parasitic capacitance of this winding is lower. Proportioning then becomes considerably simpler.
  • a drawback of such a voltage multiplier is, however, that the EHT R, is increased thereby.
  • the three resonant frequencies a, e and y of the flyback network at least substantially satisfy equation III for K being equal to I, 5 and 7, respectively.
  • the lowest resonant frequency a will always satisfy equation III for K I.
  • the three resonant frequencies it remains possible for the three resonant frequencies to satisfy equation III for K being equal to, for example, I, 3 and 5, respectively, or I, 3 and 7, respectively.
  • a scan without oscillations can then also be realized. It is, however, found that for a wide and flat secondary flyback pulse in the cases I, 3, 5 and I, 3, 7 the lowest zero transmission frequency must be considerably much lower than in the case I, 5, 7, namely lower than the second resonant frequency.
  • a I, 3, 5 generator is, for example, possible with a scan which is sufficiently without oscillations but in which the zero transmission frequency (frequencies) is(are) anything but optimum.
  • the secondary flyback pulse consequently has a much less wide peak as in the circuit arrangements according to the invention.
  • the EHT R will hardly be lower in such a circuit arrangement than in the known circuit arrangements which have only two resonant frequencies.
  • a flyback EHT and sawtooth current generator particularly for television display apparatus including switching means which are periodically nonconducting during a flyback period 1- and are conducting during a scan period T 1- and a network having input terminals connected to the switching means, the network compricing a transformer having at least one primary winding and possibly one or more coils connected thereto through which said sawtooth current flows during the scan period, and a secondary winding to which a rectifier circuit is connected for generating said EHT from the voltage pulses occurring during the flyback period at the secondary winding, said network, during the flyback period, having a first resonant frequencyf which is at least substantially equal to the expression wherein K l and S is a correction factor which is equal to the relative reduction of the slope of the sawtooth current at the end of the scan period relative to this slope in the middle of the scan period, and a second resonant frequency f G which is at least substantially equal to the said expression for K 5, characterized in that in said network further reactances are present which increase the order of the network during
  • a flyback EHT and sawtooth current generator as claimed in claim 1 characterized in that for the two frequencies f and f located between f andf and between f and f,,, respectively, at which the impedance at the input terminals of the network is at a minimum there applies that 7 03-121 /f 'fa 1,46 (f2"fe fu fe 3.
  • a flyback EHT and sawtooth. current generator as claimed in claim l in which the transformer has a tertiary winding and in which the switching means and a tuning capacitor are connected to the primary winding, characterized in that the tertiary winding is connected through a parallel LC-circuit to the primary winding and that the network. at least due to the inductance of said coils, the capacitance of the tuning capacitor, the parasitic capacitance of the secondary winding and the rectifier circuit, the inductance and the capacitance of said parallel LC-circuit and the leakage inductance between the secondary and tertiary winding constitutes a network of a minimum 6th order with the said three resonant frequencies.
  • a flyback EHT and sawtooth current generator as claimed in claim 1 in which a tuning capacitor is connected through a connection to the primary winding of the transformer, characterized in that a parallel LC- circuit is included in said connection and that the network, at least due to the inductance of said coils, the capacitance of the tuning capacitor, the parasitic capacitance of the secondary winding and the rectifier circuit, the inductance and the capacitance of said parallel LC-circuit and the leakage inductance between the primary and the secondary winding constitutes a network ofa minimum of 6th order with the said three resonant frequencies.

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Cited By (8)

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US3965391A (en) * 1974-06-04 1976-06-22 General Electric Company Balanced drive horizontal deflection circuitry with centering
US3980927A (en) * 1974-12-20 1976-09-14 Rca Corporation Deflection circuit
US4112337A (en) * 1975-12-08 1978-09-05 Hitachi, Ltd. High voltage generator
US4162433A (en) * 1974-03-28 1979-07-24 U.S. Philips Corporation Circuit arrangement including a line deflection circuit
US4166237A (en) * 1975-10-20 1979-08-28 North American Philips Corporation Horizontal deflection circuit for television camera
US4334173A (en) * 1980-09-22 1982-06-08 Zenith Radio Corporation Horizontal width control circuit for image display apparatus
US4686431A (en) * 1984-10-19 1987-08-11 U.S. Philips Corporation Line output circuit for generating a line frequency sawtooth current
US4897581A (en) * 1988-01-22 1990-01-30 Hitachi, Ltd. Horizontal deflection circuit

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3500116A (en) * 1967-10-31 1970-03-10 Philips Corp Deflection circuit for regulating the high voltage load
US3673458A (en) * 1968-11-20 1972-06-27 Philips Corp Circuit arrangement comprising switching means for periodically interrupting a current supplied to an inducting coil
US3691422A (en) * 1969-02-21 1972-09-12 Philips Corp Circuit arrangement for generating a sawtooth current in a line deflection coil for a display tube conveying a beam current and for generating an eht

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US3500116A (en) * 1967-10-31 1970-03-10 Philips Corp Deflection circuit for regulating the high voltage load
US3673458A (en) * 1968-11-20 1972-06-27 Philips Corp Circuit arrangement comprising switching means for periodically interrupting a current supplied to an inducting coil
US3691422A (en) * 1969-02-21 1972-09-12 Philips Corp Circuit arrangement for generating a sawtooth current in a line deflection coil for a display tube conveying a beam current and for generating an eht

Cited By (8)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4162433A (en) * 1974-03-28 1979-07-24 U.S. Philips Corporation Circuit arrangement including a line deflection circuit
US3965391A (en) * 1974-06-04 1976-06-22 General Electric Company Balanced drive horizontal deflection circuitry with centering
US3980927A (en) * 1974-12-20 1976-09-14 Rca Corporation Deflection circuit
US4166237A (en) * 1975-10-20 1979-08-28 North American Philips Corporation Horizontal deflection circuit for television camera
US4112337A (en) * 1975-12-08 1978-09-05 Hitachi, Ltd. High voltage generator
US4334173A (en) * 1980-09-22 1982-06-08 Zenith Radio Corporation Horizontal width control circuit for image display apparatus
US4686431A (en) * 1984-10-19 1987-08-11 U.S. Philips Corporation Line output circuit for generating a line frequency sawtooth current
US4897581A (en) * 1988-01-22 1990-01-30 Hitachi, Ltd. Horizontal deflection circuit

Also Published As

Publication number Publication date
ES401855A1 (es) 1975-03-01
FR2136558A5 (de) 1972-12-22
OA03999A (fr) 1979-09-15
GB1392511A (en) 1975-04-30
CH546021A (de) 1974-02-15
ZA722504B (en) 1973-11-28
AR192768A1 (es) 1973-03-14
CA953035A (en) 1974-08-13
NL7105268A (de) 1972-10-24
SE368132B (de) 1974-06-17
AT320755B (de) 1975-02-25
BR7202458D0 (pt) 1973-05-31
DE2218702A1 (de) 1972-11-02
BE782337A (fr) 1972-10-19
IT958750B (it) 1973-10-30

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