US20010017529A1 - Starting of synchronous machine without rotor position or speed measurement - Google Patents
Starting of synchronous machine without rotor position or speed measurement Download PDFInfo
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- US20010017529A1 US20010017529A1 US09/134,021 US13402198A US2001017529A1 US 20010017529 A1 US20010017529 A1 US 20010017529A1 US 13402198 A US13402198 A US 13402198A US 2001017529 A1 US2001017529 A1 US 2001017529A1
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/14—Electronic commutators
- H02P6/16—Circuit arrangements for detecting position
- H02P6/18—Circuit arrangements for detecting position without separate position detecting elements
- H02P6/185—Circuit arrangements for detecting position without separate position detecting elements using inductance sensing, e.g. pulse excitation
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/34—Arrangements for starting
Definitions
- the present invention relates to a system and method for starting a synchronous motor from a static inverter without reliance upon rotor position and speed measurements.
- the proposed system provides a method for performing a start of an auxiliary power unit (APU) for jet aircraft.
- APU auxiliary power unit
- Prior art includes the use of a DC motor to start the APU from a battery.
- the proposed system utilizes a pulse-width-modulated static inverter to produce a controlled AC power to drive the generator of the APU as a synchronous motor for starting.
- An important feature of the present invention includes the method in which the output of the inverter is modified as the APU is started using only electrical system measurements. Rather than measuring the rotor speed and position directly in an attempt to optimize the starting torque, the supply current to the stator is measured, filtered and is used as a feedback to estimate the motor magnetizing flux. Then, by adjusting the supply voltage to hold the phase angle between the current and the estimated flux at the proper angle, the starting current can be optimized. An additional restriction is applied to avoid exceeding the maximum supply voltage.
- FIG. 1 is a schematic block diagram of a preferred embodiment of the present synchronous motor drive system
- FIG. 2 is a diagram illustrative of the desired phase angle for controlling and otimizing developed torque
- FIG. 3 is a block diagram showing signal processing for providing development of the motors electrical angular velocity ⁇ using phase-locked loop with electrical angular velocity of the voltage output of the motor drive inverter;
- FIGS. 4A and 4B are block diagrams showing stabilization of flux estimation equations, FIG. 4A being illustrative of a block diagram of original equations showing them as equivalent to an oscillator while FIG. 4B shows a block diagram of flux estimation equations with addition of stabilization loops;
- FIGS. 5 through 8 are graphs displaying voltage and current variables during initial starting of the synchronous machine, more specifically FIGS. 5 and 6 show the time history of the reference PMW voltage magnitude and the average PMW inverter output voltage magnitude while FIGS. 7 and 8 show the time history of the inverters reference and average current magnitudes;
- FIG. 9 is a graph illustrative of motor drag torque
- FIG. 10 is a graph plotting average electrical torque
- FIG. 11 is a graph showing rotor speed versus time
- FIG. 12 is a graph showing average electrical power
- FIGS. 13 through 16 show the reference and the actual PWM voltages, as well as the actual and filtered synchronous machine armature currents at about 10-seconds into the machine's start-up, respectively.
- FIG. 13 is the reference and input voltage to the PWM inverter, while FIG. 14 is the resulting inverter voltage.
- FIGS. 15 and 16 are the input and the output of the filter respectively;
- FIGS. 17 through 20 and 21 through 24 are similar to FIGS. 13 through 16, but at about 20 and 30 seconds into the machine's start-up, respectively;
- FIGS. 25 through 28 display the time history of the actual and estimated values of q- and d-axis magnetizing fluxes during the start-up;
- FIGS. 29 through 32 respectively show the time history of the field voltage, flux, and current in per-unit, as well as the inverter output power factor angle.
- the method described hereinafter relates to a system used to start a synchronous motor from a static inverter without relying on rotor position or speed measurements.
- Features of the control system used to govern this process are hereinafter described in detail.
- An overview of the system is provided in sections B.1 and B.2, followed by details of the individual functional blocks in sections C.1 through C.8.
- Section D describes a computer simulation of the synchronous motor starting scheme and verifies the proper operation.
- FIG. 1 shows the components of the present synchronous motor drive system.
- Power to the synchronous motor is provided by a three-phase, variable voltage and variable frequency pulse-width-modulated (PWM) static inverter.
- PWM pulse-width-modulated
- the inverter provides AC power to the motor; the inverter's output voltage magnitude, frequency and phase being determined by the control system.
- Motor excitation is provided by a separate power source to establish field poles in the machine.
- FIG. 1 This provides control of the PWM inverter.
- the phasors of primary importance are the airgap magnetizing flux, ⁇ s qdm , and the stator current, I s qd both represented in a stationary reference frame.
- the developed torque can be controlled and optimized.
- the optimum phase angle is 90 electrical degrees, that is, to have the stator current in the rotor reference frame, I r qd , entirely in the q-axis.
- the optimum phase angle depends on whether the machine is saturated or not.
- ⁇ ⁇ - tan - 1 ⁇ X mq ⁇ I qs r X md ⁇ ( I qs r + I f )
- angle ⁇ is determined by solving equation below and is the angle between I s qd and the d-axis as shown in FIG. 2.
- the control system will command the static inverter to maintain a phase angle near the optimum angular displacement by injecting the appropriate current I s qd into the motor.
- phase angle of the current vector, I s qd is established in relation to the estimated airgap magnetizing flux, ⁇ s qdm , developed by the “Magnetizing Flux Estimation” portion of FIG. 1.
- the magnitude of the current vector is determined by measuring the power being provided to the machine and comparing this to a reference power input. This function is provided in the “Power Control Loop”. The calculated current magnitude is then combined with the current phase angle to provide a current reference signal I ref qd as an input to the “Current Control Loop”. This loop then provides a reference voltage, V ref qd , for the PWM inverter so that the inverter's output current matches the reference value of I ref qd . As the speed of the motor increases, the counter-voltage developed by the motor requires that the inverter's output voltage increase in order to maintain the desired reference current, I ref qd . At a certain speed this counter-voltage may exceed the voltage ability of the inverter.
- V qs s R S ⁇ I qs s + 1 ⁇ b ⁇ p ⁇ ⁇ ⁇ qs s
- This filter has a unity gain and zero phase shift at its tuned frequency ⁇ r and a sharply reduced gain at all other frequencies, hence, it would provide the desired filtering. Also note that since the system fundamental frequency changes with motor speed, the filter's tuned frequency must also change with motor speed. The filter's tuned frequency is determined by measuring the fundamental frequency of the supplied PWM voltage as described in section C.3. below.
- the measured ⁇ is used as an input to the harmonic filter and flux estimator.
- the feedback loops in their simplest form in the rotor reference frame shown in FIG. 4, may be implemented in any reference frame. However, they must be implemented in the stationary reference frame to eliminate the need for rotor position information.
- K, T, ⁇ r qs , and ⁇ r ds are the washout gain, time constant, q- and d-axis state variables, respectively, and the caret ( ) denotes an estimated variable.
- the desired phase angle of the motor current, I s qd can be calculated.
- the angular position of I ref qd must be set ahead of the position of ⁇ s qdm by the angle ⁇ , which is determined as explained in Section B.2.
- this mathematically ellegant approach is impractical since the field circuit resistance could significantly change with temperature variations and the resulting field current and the machine saturation condition could dramatically change.
- the angle ⁇ is reasonably close to 90 electrical degrees for realistic conditions and does not need to be determined precisely for near optimum operation.
- the magnitude of the reference current, I ref qd for current control is obtained via the power control loop.
- Instantaneous motor voltage and the fundamental component of motor current are multiplied to obtain instantaneous motor input power.
- This value of power, P e is compared to a given reference power, P ref , and the resulting error would be processed by an appropriate controller involving low pass filtering and proportional-integral (PI) regulation to determine the magnitude for the reference current.
- PI proportional-integral
- the reference current, I ref qd is then compared with the filtered measured currents to form error signals which in turn drive the PI current regulator blocks to arrive at a reference value for the inverter output voltage, V ref qd . See FIG. 1.
- the V ref qd is then converted to phase values V ref abc which in turn are inputed to the PWM inverter for appropriate switching actions through triangularized pulse-width-modulation.
- the V ref qd is also inputted to the frequency measurement block to determine the fundamental frequency of the inverter output voltage which is proportional to the motor speed.
- This means is provided by the field weakening loop. Initially the field current is maintained at a maximum possible value to give a strong rotor field and thus a high starting torque.
- the voltage commanded by the current control loop is sensed during the motor startup process. When this commanded voltage, V ref qd , exceeds a given reference value, V max , the exciter current is reduced by the field weakening loop. This maintains the motor back-voltage constant as the motor speed increases during the start cycle. This excitation control will extend the speed range that the converter maintains control of the motor start.
- the flux estimator is itself a dynamic system, it takes some time (about six seconds for the example in Section D) to overcome initial transients and correctly estimate the machine fluxes. Therefore, early in the startup, another control means should be used to provide starting of the synchronous machine while the flux estimator is reaching the correct estimating condition. Afterward, the flux estimator can be engaged in the preferred startup mode to provide a near optimum startup characteristic as described previously.
- FIGS. 5 through 32 A computer simulation of the system as described above was performed for starting an auxiliary power unit with a 90 kVA generator typically installed in mid to large size commercial transport jetliners. The results are shown in FIGS. 5 through 32.
- FIGS. 5 through 9 display voltage and current parameters during the initial starting of the synchronous machine. Note the discontinuities at about six seconds due to switching from early open loop start-up method to the preferred closed loop method.
- FIGS. 5 and 6 show the time history of the reference PWM voltage magnitude and the average PWM investor output voltage magnitude. It can be seen that the magnitude of the required PWM voltage is small at the beginning of the APU start-up and rises gradually as the APU speeds up. While the voltage magnitude is less than the PWM maximum limit (that is the reference value of the Field Weakening Loop) the Field Weakening proportional-integral (PI) regulator is driven to its upper limit, hence the motor excitation is at its maximum allowable level (see FIG. 21).
- the PWM maximum limit that is the reference value of the Field Weakening Loop
- PI Field Weakening proportional-integral
- FIGS. 7 and 8 show the time history of the inverter's reference and average current magnitudes. It is noted that early in the start up the motor power is small due to its small speed, hence the power controller PI regulator is driven to its upper limit resulting in the maximum allowable motor current. As the motor speeds up, the input power to the motor gradually rises. Upon switching from the early start-up method to the preferred start-up method, the motor input power suddenly increases to the desired vale of 0.1-per unit due to the increased motor torque (see FIG. 12). Thereafter, the PT regulator maintains the desired motor input power by reducing the motor current.
- FIGS. 9 through 12 shows the drag torque and the average generator electrical torque during the start up, as well as the generator speed and electrical power.
- the torque values in both these figures have a positive reference for generator produced torques and therefore have negative values for the motor that was simulated.
- the electrical torque, during the early start-up overcomes the drag torque, as well as resulting in a modest acceleration.
- the start-up torque is substantially higher, thereby resulting in an increased acceleration.
- the power controller loop starts to reduce the generator armature current, the electrical torque is also reduced.
- FIGS. 25 through 29 display the time history of the actual and estimated values of a q- and d-axis magnetizing fluxes during the start-up.
- the estimated values are obtained from the “Magnetizing Flux Estimation” block in FIG. 1.
- the flux estimator which takes about five seconds, the estimated and actual magnetizing flux values are in close agreement. They continue to track each other throughout the rest of the motor start time period.
- FIGS. 29 through 32 show the time history of the field voltage, flux and current in per-unit. The power factor input to the motor is also shown. The “Field Weakening Loop” is seen to take effect at approximately 21-seconds into the motor start. At this time the field voltage, flux, and current start to decrease as the motor speed increases.
- the computer simulation verifies the proper operation of the generator/start motor drive system and its ability to quickly bring the APU to starting speed.
- the hereinbefore described system utilizes a narrowband filter tuned to pass only the motor current's fundamental component and attenuates all other components. Since the motor speed changes with time, and fundamental frequency of the motor current is proportioned to speed, the filter's tuned frequency must change with motor speed. The motor's speed is indirectly measured using a phase-locked loop technique, this speed signal is then used to tune the filter. Use of such a filter to measure the synchronous machine's fundamental current is considered a significant feature.
- a second unique feature of the present invention is the use of wash-out feedback to stabilize the inherently unstable flux estimator.
- the wash-out feedback stabilizes the flux estimator and allows it to operate during motor starting, as described in section C.1.
- the hereinbefore described system does not use mechanical rotor position sensing to determine spatial positioning of the various vectors. Instead, the method relies on the “Magnetizing Flux Estimation” block, and the “Frequency Measurement” block to obtain required information..
- Frequency for control of the synchronous machine starting is obtained from the output voltage of the inverter and not from a mechanical measurement of the machine's shaft speed. This is a unique feature of the proposed synchronous machine startup method.
- the field of the synchronous machine is reduced at a set point in the start cycle. This addition of a field weakening loop during the synchronous machine startup is considered unique.
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Abstract
A system for starting synchronous motor from a pulse-width-modulated static inverter without using rotor position or speed measurements. The system monitors supply current to the stator and derives a relationship between that current and the magnetizing flux. The system utilizes a narrowband filter that is tuned to pass only the motor currents fundamental component and whose tuned frequency changes with variations in motor speed. Furthermore, the system reduces the excitation provided to the motor after a certain value of line voltage is sensed.
Description
- This application is a continuation of prior application Ser. No. 08/802,292, filed Feb. 18, 1997, now pending, which is a divisional of application Ser. No. 08/511,173, filed Aug. 4, 1995, now pending.
- The present invention relates to a system and method for starting a synchronous motor from a static inverter without reliance upon rotor position and speed measurements. The proposed system provides a method for performing a start of an auxiliary power unit (APU) for jet aircraft. Prior art includes the use of a DC motor to start the APU from a battery.
- Exemplary of the prior art is U.S. Pat. No. 4,361,791 (Allan B. Plunkett) which describes an apparatus for controlling a permanent magnet synchronous motor driven by a pulse width modulated inverter. The method forms a modified flux vector by phase shifting the measured flux vector, and using this modified flux vector as a feedback signal for inverter control. Further exemplary of the prior art is U.S. Pat. No. 4,855,519 (Heinrich-Karl Vogelmann) which describes an apparatus for determining the flux vector of a machine without using a mechanical shaft position indicator. This is done by injecting a component of stator current from which the error between the computed flux vector and the actual flux vector can be calculated. The error is used to modify the computed flux vector. In contrast, the proposed system utilizes a flux estimation system using well known synchronous machine equations with feedback added to ensure stability of the estimated flux.
- The proposed system utilizes a pulse-width-modulated static inverter to produce a controlled AC power to drive the generator of the APU as a synchronous motor for starting. An important feature of the present invention includes the method in which the output of the inverter is modified as the APU is started using only electrical system measurements. Rather than measuring the rotor speed and position directly in an attempt to optimize the starting torque, the supply current to the stator is measured, filtered and is used as a feedback to estimate the motor magnetizing flux. Then, by adjusting the supply voltage to hold the phase angle between the current and the estimated flux at the proper angle, the starting current can be optimized. An additional restriction is applied to avoid exceeding the maximum supply voltage.
- Further advantages and uses of the invention will become more apparent when considered in view of the following detailed description of a preferred embodiment of the invention when taken in conjunction with the accompanying drawings in which:
- FIG. 1 is a schematic block diagram of a preferred embodiment of the present synchronous motor drive system;
- FIG. 2 is a diagram illustrative of the desired phase angle for controlling and otimizing developed torque;
- FIG. 3 is a block diagram showing signal processing for providing development of the motors electrical angular velocity ω using phase-locked loop with electrical angular velocity of the voltage output of the motor drive inverter;
- FIGS. 4A and 4B are block diagrams showing stabilization of flux estimation equations, FIG. 4A being illustrative of a block diagram of original equations showing them as equivalent to an oscillator while FIG. 4B shows a block diagram of flux estimation equations with addition of stabilization loops;
- FIGS. 5 through 8 are graphs displaying voltage and current variables during initial starting of the synchronous machine, more specifically FIGS. 5 and 6 show the time history of the reference PMW voltage magnitude and the average PMW inverter output voltage magnitude while FIGS. 7 and 8 show the time history of the inverters reference and average current magnitudes;
- FIG. 9 is a graph illustrative of motor drag torque;
- FIG. 10 is a graph plotting average electrical torque;
- FIG. 11 is a graph showing rotor speed versus time;
- FIG. 12 is a graph showing average electrical power;
- FIGS. 13 through 16 show the reference and the actual PWM voltages, as well as the actual and filtered synchronous machine armature currents at about 10-seconds into the machine's start-up, respectively. FIG. 13 is the reference and input voltage to the PWM inverter, while FIG. 14 is the resulting inverter voltage. FIGS. 15 and 16 are the input and the output of the filter respectively;
- FIGS. 17 through 20 and21 through 24 are similar to FIGS. 13 through 16, but at about 20 and 30 seconds into the machine's start-up, respectively;
- FIGS. 25 through 28 display the time history of the actual and estimated values of q- and d-axis magnetizing fluxes during the start-up; and,
- FIGS. 29 through 32 respectively show the time history of the field voltage, flux, and current in per-unit, as well as the inverter output power factor angle.
- A. Introduction
- The method described hereinafter relates to a system used to start a synchronous motor from a static inverter without relying on rotor position or speed measurements. Features of the control system used to govern this process are hereinafter described in detail. An overview of the system is provided in sections B.1 and B.2, followed by details of the individual functional blocks in sections C.1 through C.8. Section D describes a computer simulation of the synchronous motor starting scheme and verifies the proper operation.
- B.1 System Overview
- FIG. 1 shows the components of the present synchronous motor drive system. Power to the synchronous motor is provided by a three-phase, variable voltage and variable frequency pulse-width-modulated (PWM) static inverter. Applicability to other forms of static inverter, e.g. six or twelve step inverters is expected although not studied and described hereinafter. The inverter provides AC power to the motor; the inverter's output voltage magnitude, frequency and phase being determined by the control system. Motor excitation is provided by a separate power source to establish field poles in the machine. Hereinafter follows a detailed description of the control system shown in FIG. 1. This provides control of the PWM inverter.
- B.2 Control System Description
-
- where the angle α is determined by solving equation below and is the angle between Is qd and the d-axis as shown in FIG. 2.
- (X md −X mq)I s qd 2 cos 2α+X md I s qd I f cos α=0
-
- where Ei is the machine internal voltage and the angle α is determined by solving the following equation
- E i cos α−X mq I s qd 2 cos 2α=0
- The control system will command the static inverter to maintain a phase angle near the optimum angular displacement by injecting the appropriate current Is qd into the motor.
- The phase angle of the current vector, Is qd, is established in relation to the estimated airgap magnetizing flux, Ψs qdm, developed by the “Magnetizing Flux Estimation” portion of FIG. 1.
- The magnitude of the current vector is determined by measuring the power being provided to the machine and comparing this to a reference power input. This function is provided in the “Power Control Loop”. The calculated current magnitude is then combined with the current phase angle to provide a current reference signal Iref qd as an input to the “Current Control Loop”. This loop then provides a reference voltage, Vref qd, for the PWM inverter so that the inverter's output current matches the reference value of Iref qd. As the speed of the motor increases, the counter-voltage developed by the motor requires that the inverter's output voltage increase in order to maintain the desired reference current, Iref qd. At a certain speed this counter-voltage may exceed the voltage ability of the inverter. This would cause the inverter to lose the capability to control the motor in the desired fashion. To avoid this loss of control, the excitation to the motor is reduced after a certain value of line voltage is sensed. This decrease in motor excitation reduces the motor's counter-voltage allowing the inverter to retain control of the motor start at high speeds. The decrease in motor excitation is accomplished by the “Field Weakening Loop”.
- Sections C.1 through C.8 provide more detailed data on the individual blocks of the system schematic of FIG. 1.
- C.1 Magnetizing Flux Estimation
- One of the most important functions of the control system is to determine the magnitude and angle of the vector Ψs qdm. This function is accomplished within the “Magnetizing Flux Estimation” block.
-
- Determination of the magnetizing flux in the fashion described is marginally stable and would often become unstable due to other system dynamics. A method to stabilize flux estimation is described in section C.4. In addition, the above equations require knowing the fundamental frequency current components Is qs and Is ds for determination of Ψs qdm.Finding these current components is complicated by the fact that the actual current waveform contains a significant amount of harmonic components. Direct use of the actual current waveform without filtering the harmonics would result in degradation of system operation, since, the control system would then try to respond to the harmonic components in addition to the fundamental. Extraction of this fundamental component is accomplished by the current measurement filter.
- C.2 Current Measurement Filter
- As discussed above, the filtering of the motor input current is an important element. Inadequate filtering may result in unacceptable current control and PWM converter operation. This is somewhat contrary to what has been observed in typical drive system applications using induction meters in which the motor's armature series inductance provides sufficient filtering of harmonic currents. Synchronous motors of high rating do not have as high a series inductance to provide inherent filtering action. Consequently, the filter is necessary for acceptable operation. The filter must effectively attenuate high order harmonic currents without introducing significant phase lag in the measured armature fundamental frequency current. In order to eliminate the harmonic components from the control system, a narrowband filter tuned to the system fundamental frequency ωr is used. The mathematical representation of this filter is:
- This filter has a unity gain and zero phase shift at its tuned frequency ωr and a sharply reduced gain at all other frequencies, hence, it would provide the desired filtering. Also note that since the system fundamental frequency changes with motor speed, the filter's tuned frequency must also change with motor speed. The filter's tuned frequency is determined by measuring the fundamental frequency of the supplied PWM voltage as described in section C.3. below.
- C.3 Determination of Electrical Frequency of the Motor
- As the above description indicates, determination of the motor's electrical angular velocity, ωr, is critical for operation of the control system. In the proposed scheme this is accomplished without sensing the motor's shaft speed. Instead it is determined mathematically using a well known phase-locked loop approach to measure the electrical angular velocity of the voltage output of the motor drive inverter. The block diagram is shown in FIG. 3. Signals proportional to cos θr and sin θr, derived from the PWM reference voltages, are combined with cos θ and sin θ terms generated by a local phase-locked oscillator. This multiplication and subsequent subtraction results in a signal proportional to sin (θr-θ). If θr-θ is very small, the sin (θr-θ) term represents a very slowly varying sine wave. Feeding this input to a proportional-integral controller results in a change in the controller's output, ω which is the frequency of the internal oscillator, until θr-θ becomes zero. At this point the sin (θr-θ) term equals zero, and the integrator's output remains locked onto ωr (=dθr/dt). This measurement approach is accurate when the value of a ω being reasonably close to ωr so that the argument of the sine term is small. Otherwise, the sin (θr-θ) may become oscillatory and the error that this term represents may not be decreased by the rest of the loop to zero. In the motor start system described herein we know that initially ωr=0. This may be used to initialize the internal oscillator (i.e., ω(0)=0). Afterward, it would track the electrical angular velocity continually as the machine speeds up.
- The measured ω is used as an input to the harmonic filter and flux estimator.
- C.4 Stabilization of the Flux Estimation Loop
- As was mentioned in section C.1, the flux estimation loop is marginally stable and would often become unstable due to other system dynamics. In order to see this marginal stability, it is instructive to transform the flux equations to the rotor reference frame as follows (p denotes differentiation, d/dt):
- The fact that the eigenvalues (poles) of these equations lie on the imaginary axis means that the system is marginally stable. Other system dynamics may push these poles to the right-half plane thus making the overall system unstable. It would be desirable to move these poles into the left half plane to stabilize these equations without impacting the steady state and low frequency values of flux vector components Ψr qs and Ψr ds.
- To solve this instability problem, feedback loops of wash-out form have been included in the flux estimation portion. These feedback loops are shown in FIG. 4 and result in movement of the poles to the left half plane. This shift in the poles causes damping of high frequency transient components, but it does not impact the steady-state and low frequency response of the flux estimator. The steady-state value of the flux estimator will continue to show the correct value.
- Note that the feedback loops, in their simplest form in the rotor reference frame shown in FIG. 4, may be implemented in any reference frame. However, they must be implemented in the stationary reference frame to eliminate the need for rotor position information. The stabilized flux estimator equations in the stationary reference frames are then:
-
- C.5 Current Control Loop
- Once the phase angle of Ψs qdm is known the desired phase angle of the motor current, Is qd, can be calculated. To obtain maximum motor torque the angular position of Iref qd must be set ahead of the position of Ψs qdm by the angle δ, which is determined as explained in Section B.2. This requires knowledge of the synchronous machine parameters, in particular, the field circuit resistance to determine the field current If and the machine saturation condition. However, this mathematically ellegant approach is impractical since the field circuit resistance could significantly change with temperature variations and the resulting field current and the machine saturation condition could dramatically change. Fortunately, the angle δ is reasonably close to 90 electrical degrees for realistic conditions and does not need to be determined precisely for near optimum operation. In practice, selecting a fixed value of δ between 70 and 110 degrees would result in an acceptable, near optimum performance. Note that for an unsaturated machine, a δ of more than 90 electrical degrees would not reinforce the magnetizing flux, while a δ less than 90 electrical degrees would reinforce the magnetizing flux and possibly cause saturation. These factors should be considered in selecting a fixed value ofδ.
- C.6 Power Control Loop Function
- As described before, the magnitude of the reference current, Iref qd, for current control is obtained via the power control loop. Instantaneous motor voltage and the fundamental component of motor current (obtained from the current filter described earlier) are multiplied to obtain instantaneous motor input power. This value of power, Pe, is compared to a given reference power, Pref, and the resulting error would be processed by an appropriate controller involving low pass filtering and proportional-integral (PI) regulation to determine the magnitude for the reference current. This magnitude information is combined with the desired current phase angle to determine the reference current, Iref qd, for the current regulators. The reference current, Iref qd, is then compared with the filtered measured currents to form error signals which in turn drive the PI current regulator blocks to arrive at a reference value for the inverter output voltage, Vref qd. See FIG. 1. The Vref qd is then converted to phase values Vref abc which in turn are inputed to the PWM inverter for appropriate switching actions through triangularized pulse-width-modulation. The Vref qd is also inputted to the frequency measurement block to determine the fundamental frequency of the inverter output voltage which is proportional to the motor speed.
- C.7 Field Weakening Loop Function
- In order to provide torque control of the synchronous machine at high speeds it is necessary that the inverter be able to inject the required current into the machine windings. As the machine speed increases, the internal voltage of the synchronous motor, caused by field excitation, also increases. This acts as a back-voltage to the PWM inverter. In order to counteract this back voltage, the voltage supplied by the inverter must increase as the motor speed increases. With a constant field excitation, a speed would eventually be reached at which the inverter would be unable to satisfy the commanded current loop unless some means were in place to reduce the internally generated synchronous machine voltage.
- This means is provided by the field weakening loop. Initially the field current is maintained at a maximum possible value to give a strong rotor field and thus a high starting torque. The voltage commanded by the current control loop is sensed during the motor startup process. When this commanded voltage, Vref qd, exceeds a given reference value, Vmax, the exciter current is reduced by the field weakening loop. This maintains the motor back-voltage constant as the motor speed increases during the start cycle. This excitation control will extend the speed range that the converter maintains control of the motor start.
- C.8 System Startup
- Unfortunately, since the flux estimator is itself a dynamic system, it takes some time (about six seconds for the example in Section D) to overcome initial transients and correctly estimate the machine fluxes. Therefore, early in the startup, another control means should be used to provide starting of the synchronous machine while the flux estimator is reaching the correct estimating condition. Afterward, the flux estimator can be engaged in the preferred startup mode to provide a near optimum startup characteristic as described previously.
- The following early startup technique was found effective through computer simulation studies and laboratory tests. The technique is comprised of the following four steps:
- 1. Energize the synchronous machine's field circuit with a maximum possible field voltage while the stator winding is energized by a very low frequency voltage (about two hertz for the example) at a very small voltage magnitude determined by a current controller regulating the stator current to its maximum allowable level. The synchronous machine should start rotating at a speed corresponding to the supplied frequency as the machine's field flux increases.
- 2. After a couple of seconds, when the synchronous machine's field flux is nearly established, ramp-up the inverter output frequency at a fairly slow rate of a few hertz per second (four hertz per second for the example) to a suitable value (about 20 hertz for the example). The synchronous machine should follow this frequency ramp-up and continue rotating accordingly, in an open loop manner.
- 3. Allow a few seconds for such an open loop operation mode to provide enough time for the flux estimator to overcome startup transients and correctly estimate the machine fluxes.
- 4. Switch to the preferred startup mode using the flux estimator for near optimum start-up performance.
- D. Simulation Results
- A computer simulation of the system as described above was performed for starting an auxiliary power unit with a 90 kVA generator typically installed in mid to large size commercial transport jetliners. The results are shown in FIGS. 5 through 32.
- FIGS. 5 through 9 display voltage and current parameters during the initial starting of the synchronous machine. Note the discontinuities at about six seconds due to switching from early open loop start-up method to the preferred closed loop method. FIGS. 5 and 6 show the time history of the reference PWM voltage magnitude and the average PWM investor output voltage magnitude. It can be seen that the magnitude of the required PWM voltage is small at the beginning of the APU start-up and rises gradually as the APU speeds up. While the voltage magnitude is less than the PWM maximum limit (that is the reference value of the Field Weakening Loop) the Field Weakening proportional-integral (PI) regulator is driven to its upper limit, hence the motor excitation is at its maximum allowable level (see FIG. 21).
- FIGS. 7 and 8 show the time history of the inverter's reference and average current magnitudes. It is noted that early in the start up the motor power is small due to its small speed, hence the power controller PI regulator is driven to its upper limit resulting in the maximum allowable motor current. As the motor speeds up, the input power to the motor gradually rises. Upon switching from the early start-up method to the preferred start-up method, the motor input power suddenly increases to the desired vale of 0.1-per unit due to the increased motor torque (see FIG. 12). Thereafter, the PT regulator maintains the desired motor input power by reducing the motor current.
- FIGS. 9 through 12 shows the drag torque and the average generator electrical torque during the start up, as well as the generator speed and electrical power. (The torque values in both these figures have a positive reference for generator produced torques and therefore have negative values for the motor that was simulated.) The electrical torque, during the early start-up, overcomes the drag torque, as well as resulting in a modest acceleration. During the preferred start-up method, however, the start-up torque is substantially higher, thereby resulting in an increased acceleration. As the power controller loop starts to reduce the generator armature current, the electrical torque is also reduced.
- FIGS. 13 through 16 show the reference and the actual PWM voltages, as well as the actual and filtered synchronous machine armature currents at about 10-seconds into the machine's start-up, respectively. FIG. 13 is the reference input voltage to the PWM inverter, while FIG. 14 is the resulting inverter voltage. FIGS. 15 and 16 are the input and the output of the filter, respectively. FIGS. 17 through 20 and21 through 24 are similar to FIGS. 13 through 16, but at about 20 and 30 seconds into the machine's start-up, respectively. As these figures indicate the filter performance is excellent. The filter eliminates undesirable harmonics without introducing any phase lag into the fundamental component.
- Also note that as the motor speed increases, the field induced motor voltage increases. This requires that the PWM inverter reference and output voltages increase to maintain proper current regulation (See FIG. 5, as well as the PWM reference voltages in FIGS. 13, 17, and21). As the PWM reference voltage approaches the PWM maximum limit, the field weakening loop reduces the generator excitation level. The inverter can then maintain control of the motor start as the speed continues to increase.
- FIGS. 25 through 29 display the time history of the actual and estimated values of a q- and d-axis magnetizing fluxes during the start-up. The estimated values are obtained from the “Magnetizing Flux Estimation” block in FIG. 1. After attention of start-up translates in the flux estimator, which takes about five seconds, the estimated and actual magnetizing flux values are in close agreement. They continue to track each other throughout the rest of the motor start time period.
- FIGS. 29 through 32 show the time history of the field voltage, flux and current in per-unit. The power factor input to the motor is also shown. The “Field Weakening Loop” is seen to take effect at approximately 21-seconds into the motor start. At this time the field voltage, flux, and current start to decrease as the motor speed increases.
- The computer simulation verifies the proper operation of the generator/start motor drive system and its ability to quickly bring the APU to starting speed.
- E. Hardware Verification of Synchronous Machine Startup Schemes
- The synchronous machine starting method described herein has been implemented in hardware and further investigated to verify its operational features. The result indicated that the proposed method works as described, and is insensitive to parameter changes which typical synchronous machines can normally experience.
- F. System Features Included
- 1. Filtering of Measured Motor Current
- As indicated in section C.2, one of the problems with control of a relatively large synchronous motor is that the machine doesn't provide enough inherent filtering of the motor's current. The machine current contains significant high frequency components in addition to the fundamental component. The high frequency components adversely affect control of the motor current.
- The hereinbefore described system utilizes a narrowband filter tuned to pass only the motor current's fundamental component and attenuates all other components. Since the motor speed changes with time, and fundamental frequency of the motor current is proportioned to speed, the filter's tuned frequency must change with motor speed. The motor's speed is indirectly measured using a phase-locked loop technique, this speed signal is then used to tune the filter. Use of such a filter to measure the synchronous machine's fundamental current is considered a significant feature.
- 2. Stabilization of the Inherently Unstable Flux Estimator
- A second unique feature of the present invention is the use of wash-out feedback to stabilize the inherently unstable flux estimator. The wash-out feedback stabilizes the flux estimator and allows it to operate during motor starting, as described in section C.1.
- 3. Elimination of the Need for Rotor Position
- The hereinbefore described system does not use mechanical rotor position sensing to determine spatial positioning of the various vectors. Instead, the method relies on the “Magnetizing Flux Estimation” block, and the “Frequency Measurement” block to obtain required information..
- 4. Use of Measured Frequency Instead of the Rotor Speed
- Frequency for control of the synchronous machine starting is obtained from the output voltage of the inverter and not from a mechanical measurement of the machine's shaft speed. This is a unique feature of the proposed synchronous machine startup method.
- 5. Use of the Field Weakening Loop
- In order to provide a limit for power output of the start inverter, the field of the synchronous machine is reduced at a set point in the start cycle. This addition of a field weakening loop during the synchronous machine startup is considered unique.
- G. Abbreviations and Nomenclature
P1 Proportional-plus-Integral Controller PWM Pulse-Width-Modulation Ψs qdm Phasor representing magnetizing flux linkage in stationery reference frame Ψs mq q-axis component of magnetizing flux linkage in stationery reference frame Ψs md d-axis component of magnetizing flux linkage in stationery reference frame Ψr qs q-axis component of stator flux linkage in rotor reference frame Ψr ds d-axis component of stator flux linkage in rotor reference frame Ψs qs q-axis component of stator flux linkage in stationary reference frame Ψs ds d-axis component of stator flux linkage in stationary reference frame Is qd Phasor representing stator current in stationary reference frame Is qs q-axis component of stator current Is qd in stationary reference frame Is ds d-axis component of stator current Is qd in stationary reference frame Ir qs q-axis component of stator current Is qd in rotor reference frame Ir ds d-axis component of stator current Is qd in rotor reference frame Iref qd Phasor representing the reference (desired) value of stator current in stationary reference frame Vs qd Phasor representing stator voltage in the stationary reference frame Vs qs q-axis component of stator voltage in stationary reference frame Vs ds d-axis component of stator voltage in stationary reference frame Vr qs q-axis component of stator voltage in rotor reference frame Vr ds d-axis component of stator voltage in rotor reference frame Rs Stator resistance Xls Stator leakage reactance Xmd d-axis magnetizing reactance Xmq q-axis magnetizing reactance If Field current s Laplace transform operator ζ Filter damping ratio θr Instantaneous electrical angle of rotor position ωr Electrical angular velocity of the rotor θ Instantaneous phase angle of the phase-locked loop ω Electrical angular velocity of the phase-locked loop ωb Base electrical angular velocity Pe Developed electromagnetic power Pref Reference (desired) power p Differential operator {circumflex over ( )} Estimated variable indicator - Different embodiments of the present invention may be developed without departing from the spirit and scope of the invention, the present preferred embodiment being merely illustrative of the present invention defined only in the appended claims.
Claims (1)
1. In a synchronous machine drive that operates by utilizing an estimate of the stator flux, a method for stabilizing the flux estimation process by providing stabilizing feedback signals including a transfer function of:
K sτ/(1+sτ)
where K and τ are adjustable gain and time constants, respectively.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US09/134,021 US6362590B2 (en) | 1995-08-04 | 1998-08-14 | Starting of synchronous machine without rotor position or speed measurement |
Applications Claiming Priority (3)
Application Number | Priority Date | Filing Date | Title |
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US08/511,173 US5818192A (en) | 1995-08-04 | 1995-08-04 | Starting of synchronous machine without rotor position of speed measurement |
US08/802,292 US5877606A (en) | 1995-08-04 | 1997-02-18 | Starting of synchronous machine without rotor position or speed measurement |
US09/134,021 US6362590B2 (en) | 1995-08-04 | 1998-08-14 | Starting of synchronous machine without rotor position or speed measurement |
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US08/802,292 Continuation US5877606A (en) | 1995-08-04 | 1997-02-18 | Starting of synchronous machine without rotor position or speed measurement |
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US20010017529A1 true US20010017529A1 (en) | 2001-08-30 |
US6362590B2 US6362590B2 (en) | 2002-03-26 |
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US08/511,173 Expired - Fee Related US5818192A (en) | 1995-08-04 | 1995-08-04 | Starting of synchronous machine without rotor position of speed measurement |
US08/802,292 Expired - Fee Related US5877606A (en) | 1995-08-04 | 1997-02-18 | Starting of synchronous machine without rotor position or speed measurement |
US09/134,021 Expired - Fee Related US6362590B2 (en) | 1995-08-04 | 1998-08-14 | Starting of synchronous machine without rotor position or speed measurement |
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US08/802,292 Expired - Fee Related US5877606A (en) | 1995-08-04 | 1997-02-18 | Starting of synchronous machine without rotor position or speed measurement |
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EP (1) | EP0757432B1 (en) |
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Cited By (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20040100221A1 (en) * | 2002-11-25 | 2004-05-27 | Zhenxing Fu | Field weakening with full range torque control for synchronous machines |
US20070024232A1 (en) * | 2005-07-29 | 2007-02-01 | Takahiro Suzuki | Motor controller, washing machine, air conditioner and electric oil pump |
WO2010049412A1 (en) | 2008-10-27 | 2010-05-06 | Vestas Wind Systems A/S | Direct power and stator flux vector control of a generator for wind energy conversion system |
US20130289934A1 (en) * | 2012-04-27 | 2013-10-31 | The Board Of Trustees Of The University Of Illinois | Angle-based speed estimation of alternating current machines utilizing a median filter |
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Families Citing this family (46)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
FR2753319B1 (en) * | 1996-09-10 | 1998-12-04 | Soc D Mecanique Magnetique | ANGULAR POSITION DETECTION DEVICE FOR DRIVING A PERMANENT MAGNET-DRIVEN SYNCHRONOUS MOTOR |
FI112735B (en) * | 1997-12-03 | 2003-12-31 | Kone Corp | A method for controlling a synchronous permanent magnet motor |
US6281656B1 (en) * | 1998-09-30 | 2001-08-28 | Hitachi, Ltd. | Synchronous motor control device electric motor vehicle control device and method of controlling synchronous motor |
US6014006A (en) * | 1999-06-24 | 2000-01-11 | Ford Global Technologies, Inc. | Induction motor control system with speed and flux estimation |
US6163128A (en) * | 1999-08-20 | 2000-12-19 | General Motors Corporation | Method and drive system for controlling a permanent magnet synchronous machine |
DE10007120B4 (en) * | 2000-02-17 | 2007-04-12 | LFK Lenkflugkörpersysteme GmbH | Current regulation of permanent-magnet synchronous motors for guided missiles with electromechanical rudder actuator |
NL1015153C2 (en) * | 2000-05-10 | 2001-11-22 | Gti Electroproject B V | Method and device for sensorless estimation of the relative angular position between the stator and rotor of a synchronous three-phase motor. |
US6304052B1 (en) * | 2000-06-27 | 2001-10-16 | General Motors Corporation | Control system for a permanent magnet motor |
FI112299B (en) * | 2000-12-22 | 2003-11-14 | Abb Industry Oy | Method in connection with the drive |
US6401875B1 (en) * | 2001-02-12 | 2002-06-11 | Otis Elevator Company | Absolute position sensing method and apparatus for synchronous elevator machines by detection stator iron saturation |
DE10115873A1 (en) * | 2001-03-30 | 2002-10-17 | Bosch Gmbh Robert | Method for controlling an electronically commutated direct current motor |
US20030067230A1 (en) * | 2001-10-09 | 2003-04-10 | Stancu Constantin C. | Maximum torque-per-ampere control of a saturated surface-mounted permanent magnet machine |
US6844701B2 (en) * | 2002-01-16 | 2005-01-18 | Ballard Power Systems Corporation | Overmodulation systems and methods for induction motor control |
US6683428B2 (en) * | 2002-01-30 | 2004-01-27 | Ford Global Technologies, Llc | Method for controlling torque in a rotational sensorless induction motor control system with speed and rotor flux estimation |
US6791204B2 (en) * | 2002-09-20 | 2004-09-14 | Honeywell International Inc. | Torque generation for salient-pole synchronous machine for start-up of a prime mover |
US20050035768A1 (en) * | 2002-10-02 | 2005-02-17 | Germano Rabach | Method and electromagnetic sensor for measuring partial discharges in windings of electrical devices |
US20040100220A1 (en) * | 2002-11-25 | 2004-05-27 | Zhenxing Fu | Weighted higher-order proportional-integral current regulator for synchronous machines |
US6876169B2 (en) * | 2003-01-14 | 2005-04-05 | Delphi Technologies, Inc. | Method and controller for field weakening operation of AC machines |
US6982533B2 (en) * | 2003-09-17 | 2006-01-03 | Rockwell Automation Technologies, Inc. | Method and apparatus to regulate loads |
US7045986B2 (en) * | 2004-02-20 | 2006-05-16 | Honeywell International Inc. | Position sensing method and apparatus for synchronous motor generator system |
US7075264B2 (en) * | 2004-03-31 | 2006-07-11 | Honeywell International Inc. | Instantaneous power floating frame controller |
US6940251B1 (en) | 2004-04-30 | 2005-09-06 | Honeywell International Inc. | Decoupling of cross coupling for floating reference frame controllers for sensorless control of synchronous machines |
US7095209B2 (en) * | 2004-09-29 | 2006-08-22 | Rockwell Automation Technologies, Inc. | Method and apparatus to regulate torque provided to loads |
US7211984B2 (en) * | 2004-11-09 | 2007-05-01 | General Motors Corporation | Start-up and restart of interior permanent magnet machines |
US6965212B1 (en) | 2004-11-30 | 2005-11-15 | Honeywell International Inc. | Method and apparatus for field weakening control in an AC motor drive system |
US7190581B1 (en) * | 2005-01-11 | 2007-03-13 | Midwest Research Institute | Low thermal resistance power module assembly |
FR2889370B1 (en) * | 2005-07-29 | 2007-09-07 | Valeo Equip Electr Moteur | METHOD FOR CONTROLLING A POLYPHASE VOLTAGE INVERTER |
JP4764785B2 (en) * | 2006-08-23 | 2011-09-07 | ルネサスエレクトロニクス株式会社 | Control device for synchronous motor |
EP1909369B1 (en) * | 2006-10-06 | 2020-05-20 | Schmidhauser AG | Switching assembly and method for insulation monitoring for converter applications in operation |
US8633662B2 (en) * | 2009-06-12 | 2014-01-21 | Standard Microsystems Corporation | Drive method to minimize vibration and acoustics in three phase brushless DC (TPDC) motors |
US8378602B2 (en) * | 2009-11-18 | 2013-02-19 | Standard Microsystems Corporation | System and method for aligning a rotor to a known position |
US9385641B2 (en) * | 2009-11-18 | 2016-07-05 | Standard Microsystems Corporation | System and method for inducing rotation of a rotor in a sensorless motor |
US8519662B2 (en) * | 2010-05-26 | 2013-08-27 | Rockwell Technologies, Inc. | Method and apparatus for controlling motor torque |
US8915088B2 (en) | 2010-06-11 | 2014-12-23 | Hamilton Sundstrand Corporation | Fuel control method for starting a gas turbine engine |
DE102010030079A1 (en) * | 2010-06-15 | 2011-12-15 | Robert Bosch Gmbh | Method and device for monitoring the insulation resistance in an ungrounded electrical network |
US8698432B2 (en) | 2010-08-31 | 2014-04-15 | Standard Microsystems Corporation | Driving low voltage brushless direct current (BLDC) three phase motors from higher voltage sources |
US8436564B2 (en) | 2010-09-01 | 2013-05-07 | Standard Microsystems Corporation | Natural commutation for three phase brushless direct current (BLDC) motors |
CN101944841B (en) * | 2010-09-21 | 2012-10-03 | 电子科技大学 | Inversion control digital filter |
JP5697036B2 (en) * | 2011-04-22 | 2015-04-08 | サンデン株式会社 | Motor control device |
CA2884597C (en) | 2012-09-13 | 2022-04-05 | Moog Inc. | Methods and apparatae for controlling and providing a voltage converter with a pulse-width-modulated switch |
CN103199775B (en) * | 2013-03-26 | 2016-05-04 | 上海交通大学 | Simplex winding high-power explosion-proof synchronous motor frequency conversion speed-adjusting system based on IGCT five level |
US9845806B2 (en) * | 2015-06-04 | 2017-12-19 | United Technologies Corporation | Engine speed optimization as a method to reduce APU fuel consumption |
EP3334029B1 (en) * | 2015-08-04 | 2022-12-28 | Mitsubishi Electric Corporation | Synchronous motor control device, compressor drive device, and air-conditioner |
DE102015218567A1 (en) * | 2015-09-28 | 2017-03-30 | Siemens Aktiengesellschaft | Synchronous machine module, vehicle drive and vehicle |
FR3064840B1 (en) * | 2017-04-03 | 2019-12-27 | Mmt ag | THERMOREGULATED HIGH POWER ELECTRIC MACHINE |
FR3086473B1 (en) * | 2018-09-20 | 2020-10-02 | Ifp Energies Now | METHOD OF DETERMINING THE MAGNETIC FLOW OF AN ELECTRIC MACHINE |
Family Cites Families (47)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3630496A (en) * | 1968-01-26 | 1971-12-28 | Babcock & Wilcox Co | Gas-cleaning apparatus |
US3626414A (en) * | 1968-11-29 | 1971-12-07 | North American Rockwell | Doppler processing apparatus and method |
US3845372A (en) * | 1970-11-19 | 1974-10-29 | Allis Chalmers Mfg Co | Circuit for starting electric motor from standstill with maximum torque |
US4039909A (en) * | 1975-02-10 | 1977-08-02 | Massachusetts Institute Of Technology | Variable speed electronic motor and the like |
DE2635965C3 (en) * | 1976-08-10 | 1979-01-18 | Siemens Ag, 1000 Berlin Und 8000 Muenchen | Circuit arrangement and method for forming an electrical quantity which is proportional to a flux component in a rotating field machine |
US4123692A (en) * | 1976-10-26 | 1978-10-31 | Allis-Chalmers Corporation | Adjustable speed electric motor drive having constant harmonic content |
US4080559A (en) * | 1976-11-15 | 1978-03-21 | General Electric Company | Torsional protective device for power system stabilizer |
GB1596681A (en) * | 1977-01-19 | 1981-08-26 | Sony Corp | Drive circuits with speed control for brushless dc motors |
US4137489A (en) * | 1977-07-21 | 1979-01-30 | General Electric Company | Feedback control for reduction of cogging torque in controlled current AC motor drives |
DE2811123C2 (en) * | 1978-03-15 | 1983-09-29 | Barmag Barmer Maschinenfabrik Ag, 5630 Remscheid | Procedure for starting up and ramping up a synchronous motor to operating speed with the aid of a converter |
US4320331A (en) * | 1979-10-01 | 1982-03-16 | General Electric Company | Transistorized current controlled pulse width modulated inverter machine drive system |
US4321478A (en) * | 1979-11-13 | 1982-03-23 | General Electric Company | Auxiliary power supply with kinetic energy storage |
US4361791A (en) * | 1981-05-12 | 1982-11-30 | General Electric Company | Apparatus for controlling a PWM inverter-permanent magnet synchronous motor drive |
DE3126318A1 (en) * | 1981-06-26 | 1983-01-13 | Siemens AG, 1000 Berlin und 8000 München | ELECTRIC RECTIFIER MOTOR SYNCHRONOUS DESIGN |
US4394610A (en) * | 1981-08-07 | 1983-07-19 | The United States Of America As Represented By The Administrator Of The National Aeronautics And Space Administration | Adaptive reference voltage generator for firing angle control of line-commutated inverters |
US4445080A (en) * | 1981-11-25 | 1984-04-24 | The Charles Stark Draper Laboratory, Inc. | System for indirectly sensing flux in an induction motor |
US4427934A (en) * | 1982-01-29 | 1984-01-24 | General Electric Company | Current limiter for a load commutated inverter |
US4443747A (en) * | 1982-04-01 | 1984-04-17 | General Electric Company | Transitioning between multiple modes of inverter control in a load commutated inverter motor drive |
JPS5951235B2 (en) * | 1982-11-19 | 1984-12-12 | 株式会社東芝 | Induction motor control device |
US4511834A (en) * | 1982-12-23 | 1985-04-16 | Borg-Warner Corporation | Control and stabilizing system for damperless synchronous motor |
US4473790A (en) * | 1983-01-03 | 1984-09-25 | General Electric Company | Control circuit for suppression of line resonances in current feedback pulse width modulation control systems with a minimum d-c filter |
JPS60261386A (en) * | 1984-06-05 | 1985-12-24 | Toshiba Mach Co Ltd | Speed controller of ac motor |
JPH0763233B2 (en) * | 1984-08-31 | 1995-07-05 | 株式会社東芝 | Synchronous motor control method |
US4683411A (en) * | 1984-09-21 | 1987-07-28 | General Electric Company | Synchronous motor protection |
US4904710A (en) * | 1985-10-31 | 1990-02-27 | The Dow Chemical Company | Gamma radiation resistant carbonate polymer compositions |
JPH0683584B2 (en) * | 1985-11-13 | 1994-10-19 | 山本電気株式会社 | Synchronous motor controller |
EP0228535A1 (en) * | 1985-12-04 | 1987-07-15 | Siemens Aktiengesellschaft | Method and device to determine the flux angle of an induction machine i.e. to operate the machine according to position |
CA1284349C (en) * | 1986-08-27 | 1991-05-21 | Craig R. Conner | Flux profile control for startup of an induction motor |
EP0274716A1 (en) * | 1987-01-09 | 1988-07-20 | Siemens Aktiengesellschaft | Method and device to determine the flux vector of an induction machine |
JPH0650954B2 (en) * | 1987-05-26 | 1994-06-29 | 株式会社東芝 | Commutatorless motor controller |
JPH07108119B2 (en) * | 1987-08-08 | 1995-11-15 | 三菱電機株式会社 | Induction motor controller |
GB8805420D0 (en) * | 1988-03-08 | 1988-04-07 | Framo Dev Ltd | Electrically powered pump unit |
JPH01271382A (en) * | 1988-04-21 | 1989-10-30 | Nippon Otis Elevator Co | Elevator start compensating device |
US4949021A (en) * | 1988-11-14 | 1990-08-14 | Sunstrand Corporation | Variable speed constant frequency start system with selectable input power limiting |
US4912387A (en) * | 1988-12-27 | 1990-03-27 | Westinghouse Electric Corp. | Adaptive noise cancelling for magnetic bearing auto-balancing |
US5029263A (en) * | 1989-10-19 | 1991-07-02 | Sundstrand Corporation | Electric start control of a VSCF system |
US5334923A (en) * | 1990-10-01 | 1994-08-02 | Wisconsin Alumni Research Foundation | Motor torque control method and apparatus |
US5164651A (en) * | 1991-06-27 | 1992-11-17 | Industrial Technology Research Institute | Starting-current limiting device for single-phase induction motors used in household electrical equipment |
US5315225A (en) * | 1991-09-30 | 1994-05-24 | Electric Power Research Institute | Converter for synchronous motor starting |
US5363032A (en) * | 1993-05-12 | 1994-11-08 | Sundstrand Corporation | Sensorless start of synchronous machine |
US5451852A (en) * | 1993-08-02 | 1995-09-19 | Gusakov; Ignaty | Control system having signal tracking window filters |
US5386186A (en) * | 1993-08-04 | 1995-01-31 | Eaton Corporation | Stator flux oriented control |
US5402053A (en) * | 1993-08-26 | 1995-03-28 | Wisconsin Alumni Research Foundation | Single phase to three phase converter capable of variable speed motor operation |
US5559419A (en) * | 1993-12-22 | 1996-09-24 | Wisconsin Alumni Research Foundation | Method and apparatus for transducerless flux estimation in drives for induction machines |
US5585709A (en) * | 1993-12-22 | 1996-12-17 | Wisconsin Alumni Research Foundation | Method and apparatus for transducerless position and velocity estimation in drives for AC machines |
US5642461A (en) * | 1994-11-14 | 1997-06-24 | Seagate Technology, Inc. | Economical wide range speed control system |
US5847535A (en) * | 1996-01-31 | 1998-12-08 | Parker-Hannifin Corporation | Active electronic damping for step motor |
-
1995
- 1995-08-04 US US08/511,173 patent/US5818192A/en not_active Expired - Fee Related
-
1996
- 1996-07-26 EP EP96202129A patent/EP0757432B1/en not_active Expired - Lifetime
- 1996-07-26 DE DE69623888T patent/DE69623888T2/en not_active Expired - Fee Related
- 1996-08-01 JP JP20373696A patent/JP3786476B2/en not_active Expired - Lifetime
-
1997
- 1997-02-18 US US08/802,292 patent/US5877606A/en not_active Expired - Fee Related
-
1998
- 1998-08-14 US US09/134,021 patent/US6362590B2/en not_active Expired - Fee Related
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US20070024232A1 (en) * | 2005-07-29 | 2007-02-01 | Takahiro Suzuki | Motor controller, washing machine, air conditioner and electric oil pump |
US7619385B2 (en) * | 2005-07-29 | 2009-11-17 | Hitachi, Ltd. | Motor controller, washing machine, air conditioner and electric oil pump |
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CN102246409A (en) * | 2008-10-27 | 2011-11-16 | 维斯塔斯风力系统集团公司 | Direct power and stator flux vector control of a generator for wind energy conversion system |
US8686695B2 (en) | 2008-10-27 | 2014-04-01 | Vestas Wind Systems A/S | Direct power and stator flux vector control of a generator for wind energy conversion system |
US20130289934A1 (en) * | 2012-04-27 | 2013-10-31 | The Board Of Trustees Of The University Of Illinois | Angle-based speed estimation of alternating current machines utilizing a median filter |
US9954624B2 (en) * | 2012-04-27 | 2018-04-24 | The Board Of Trustees Of The University Of Illinois | Angle-based speed estimation of alternating current machines utilizing a median filter |
Also Published As
Publication number | Publication date |
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JP3786476B2 (en) | 2006-06-14 |
DE69623888T2 (en) | 2003-01-23 |
JPH09121577A (en) | 1997-05-06 |
EP0757432A3 (en) | 1997-04-09 |
EP0757432A2 (en) | 1997-02-05 |
EP0757432B1 (en) | 2002-09-25 |
US6362590B2 (en) | 2002-03-26 |
US5877606A (en) | 1999-03-02 |
US5818192A (en) | 1998-10-06 |
DE69623888D1 (en) | 2002-10-31 |
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