TWI504165B - Wireless sensor reader - Google Patents

Description

Wireless sensor reader

This case is part of the continuation application of US Patent Application No. 12/419,326, filed on April 7, 2009, which is part of the continuation application of US Patent Application No. 12/075,858, filed March 14, 2008. The latter claims the priority of the U.S. Patent Provisional Application No. 60/918,164 filed on March 15, 2007, which is incorporated herein by reference.

The present invention is generally related to the reading of passive wireless sensors, and in particular to a reader circuit and method thereof that can be excited and sensed by a passive wireless sensor.

Passive wireless sensor systems employing resonant circuit technology are well known. Such systems use an excitation and reader circuit to use a passive wireless sensor for remote communication. Typically the wireless sensor is placed in a specific location, such as in a human body, to detect and report the parameters it senses. These sensed parameters vary with the resonant circuit frequency of the wireless sensor. The reader device samples the resonant frequency of the wireless sensor to determine the parameters it senses.

Early researcher Haynes revealed a swallowable pill with a wireless pressure sensor that uses a large reading device that surrounds the target body and measures the frequency using a discriminating circuit (HE Haynes and AL Witchey, "Medical electronics, the pill that ‵talks‵", RCA Engineer, vol 5, pp. 52-54. 1960). Nagumo also revealed a similar system in which the sensor includes an energy storage capacitor that can drive the sensor during resonance (J. Nagumo, A. Uchiyama, S. Kimoto, T. Watanuki, M. Hori, K. Suma). , A. Ouchi, M. Kumano, and H. Watanabe, "Echo capsule for medical use (a batteryless radioendosonde)", IRE Transactions on Bio-Medical Electronics. vol. BME-9, pp. 195-199, 1962).

A sensor that can measure brain fluid pressure is disclosed in U. A similar sensor that can measure intra-cranial pressure is disclosed in U.S. Patent No. 4,206,762 to Cosman. In particular, the Cosman patent describes the wireless measurement of the resonant frequency of a sensor using a grid dip system.

Several methods of reading passive wireless sensors are also described in other prior patents. For example, the Cosman patent discloses an external oscillator circuit that uses an implanted sensor for tuning, and a freeze detection measurement system that measures the resonant frequency of the sensor. No. 6,015,386 to Kensey et al. discloses a reader that utilizes a transmit sweep frequency and uses a phase detector for the transmitted signal to identify the frequency it emits during the scan and the resonant frequency of the sensor. Match the frequency points to excite the passive sensor. U.S. Patent No. 6,206, 835 to the disclosure of U.S. Patent No. 5,581,248, the disclosure of which is incorporated herein by reference. The reader technique detects a variable impedance loading effect of the reader as a function of frequency using the detected parameters of the sensor. U.S. Patent No. 7,432,723 to the entire disclosure of U.S. Pat. frequency. Ellis uses the ring-down response of the appropriate charging circuit to determine the resonant frequency of the sensor. U.S. Patent No. 6,111,520 to Allen et al. discloses a method of detecting the ringing of a white noise in a sensor and detecting a ringing drop response.

Some readers use a phase-locked-loop (PLL) circuit to lock the resonant frequency of the sensor. No. 7,245,117 to Joy et al. discloses an active PLL circuit and signal processing circuit that adjusts the transmit PLL frequency until the phase of the received signal coincides with the phase of the transmitted PLL signal. When a coincidence occurs, the transmit PLL frequency is equal to the resonant frequency of the sensor.

The PLL circuit can use the sample and hold (S/H) function to sample the input frequency and fix the PLL at a given frequency. A PLL with S/H can be used in many different applications. For example, U.S. Patent No. 4,531,526 to the entire disclosure of the entire disclosure of the disclosure of the entire disclosure of the disclosure of the disclosure of the disclosure of the disclosure of the disclosure of the disclosure of the disclosure of the disclosure of the disclosure of . This is used to maximize the sensor response relative to the next shot and measure the decay rate of the sensor's resonant amplitude to extract the sensed parameter values. No. 4,644,420 to Buchan describes a PLL with S/H that is used to sample a stream of tape data streams and maintain an appropriate sampling frequency to estimate the digital data pulses on the tape. U.S. Patent No. 5,006,819 to Buchan et al. provides additional reinforcement to this design concept. No. 5,920,233 to Denny describes a high speed sampling technique that uses an S/H circuit with a PLL to reduce charge pump noise from a phase-frequency detector to improve the low jitter performance of a frequency synthesis circuit. A PLL having an S/H circuit that pre-positions a control voltage of a voltage controlled oscillator when the lock frequency of the PLL changes is disclosed in US Pat. No. 4,511,858. This is used to increase the response speed of the PLL when the desired composite frequency changes. A PLL with an S/H circuit that enhances the frequency stepping of the PLL, which also provides an offset correction function, is disclosed in U.S. Patent No. 6,570,457 to Fischer et al. No. 3,872,455 to Fuller et al. discloses a PLL having a digital S/H that freezes the display of the frequency when the phase of the PLL is locked and preloads a frequency counter.

Readers that perform direct sampling and frequency analysis techniques are also known. An example of this is the U.S. Patent No. 7,048,756 to Eggers et al., which uses a resonance sensor to measure the temperature in the body, which exhibits a change in the reaction within the temperature threshold within the Curie temperature.

In addition, readers that apply digital signal analysis to improve performance and response are also well known. US Patent No. 7,466,120 to Miller et al. describes the use of a digital signal processor (DSP) to estimate the response of a passive blood pressure sensor that has been excited by a frequency and to estimate the response of the three frequency excitation to calculate the relative phase. delay.

Current designs of passive sensor readers, such as those described above, are each limited by several drawbacks. Haynes and Nagumo's early "pulsed echo ringing systems" required the use of large, high-power reader devices. In addition, as Collins reveals, such systems suffer from inaccuracies and poor resolution due to the difficulty in measuring the frequency of short-term ringing signals, and cause them to be abandoned and instead use different kinds of Sweep method (see C. Collins, "Miniature Passive Pressure Transensor for Implanting in the Eye", IEEE Transactions on Bio-Medical Engineering, vol BME-14, no. 2, April 1967).

Sweep sensor readers similar to those described in Cosman, Kensey, Ellis, and Spillman, and the pulse wave method described by Allen, require a relatively wide band tolerance due to government radio transmission regulations. This limits other uses of the band and also makes interference a potential problem. Readers that use a variable frequency transmitter to track the resonant frequency of a passive resonant sensor, such as Genest, Ellis, and Joy, suffer from similar problems. The extra circuitry required for sweeping and/or digital pursuit is also significant, increasing the size, cost, and failure rate of the reader. In addition, sensors that use digitally controlled frequency-tracking or frequency-sweeping systems require significant amounts of power for transmission, signal processing, sampling, and recovery of resonant frequencies, and limit the use of batteries on the reader. The ability to limit battery life on battery-operated readers as well. Therefore, there is a great need in the art for a passive sensor reader system that enhances performance.

A reading device is available as an interface for a wireless sensor whose resonant frequency varies proportionally to the sensed parameter. The reader emits a short pulse of energy at a fixed frequency such that the wireless sensor immediately rings at its resonant frequency or near its resonant frequency at the end of the transmission. The reader receives and amplifies the ringing signal and measures its frequency. In one embodiment, the reader performs this measurement by sending a signal to a phase-locked loop ("PLL") that is locked to the sensor's ringing frequency. Once the PLL is locked at the ringing frequency, the PLL's voltage controlled oscillator ("VCO") is placed in a holding mode to maintain the VCO's frequency at the locked frequency. The VCO frequency is counted to determine the resonant frequency of the sensor. Alternatively, the VCO control voltage itself can be sampled and used to determine the resonant frequency of the sensor based on a known correlation. When the VCO control voltage is sampled, the VCO frequency may not need to be locked if the voltage sampling is fast enough. The present invention also discloses further frequency determination methods and systems involving digital spectrum analysis.

A passive wireless sensor system includes a reader 10 for remote communication with a sensor 12. The reader 10 can utilize a signal, such as a radio frequency ("RF") pulse, at or near the resonant frequency of the sensor 12 to excite the sensor 12 (see Figure 1). The sensor 12 can transmit a ringing signal in response to the excitation pulse from the reader 10 for a short period of time.

The sensor 12 can be a passive device that does not include its own power source and can transmit a signal ringing signal 16 at or near the resonant frequency of the sensor 12 in response to the excitation signal 14. Sensor 12 can be organized to sense a particular parameter. For example, sensor 12 can include a fixed inductor 13 and a capacitor 15 that can vary depending on the sensed parameters. A varying inductance or capacitance changes the resonant frequency of the sensor 12. However, it should be understood that the sensor 12 can be any wireless sensor known in the art that can communicate with the remote end of the reader 10. In addition, although the sensor 12 of the present description is described as an RF resonance sensor, it can be understood that the sensor 12 can also be any acoustic resonance sensor, optical resonance sensor, or Other similar sensors as are known in the art. The reader 10 can actuate the sensor 12 with a corresponding signal. Additionally, the sensor 12 can be an active sensor or a passive sensor.

In an embodiment, the sensor 12 includes at least one inductive component 13 and a capacitive component 15. To vary the resonant frequency of the sensor 12 in proportion to the sensed parameter, either the inductive element 13 or the capacitive element 15, or both, can be configured to vary the inductance or capacitance in proportion to the sensed parameter. In the embodiment shown in Figure 1, the capacitive element 15 is altered and the inductive element 13 is fixed. A typical example of such components is that the sensor changes its capacitance in response to a change in pressure. These capacitive pressure sensors are known in the art.

In one embodiment, at least one inductive element 13 in the sensor 12 also functions as an antenna for the sensor 12 that couples energy back and forth between the other antenna 26 in the sensor 12.

The reader 10 can activate the sensor 12 by transmitting an excitation pulse 14 in the vicinity of the sensor 12. For example, the reader can transmit an RF excitation pulse 14 at or near the resonant frequency of the sensor 12. The sensor 12 can transmit a ringing signal 16 in response to the excitation pulse wave 14. The reader 10 can determine the frequency of the ringing signal 16 to determine the value of the parameter it senses.

The flowchart of FIG. 2 illustrates one example of the steps involved in the reader 10 performing a read process on the sensor 12. Each step is composed of multiple sub-steps, and these steps can be further divided into multiple levels of sub-steps. However, only the most basic, top-level steps are shown in the figure to clarify the sequence of operations of the reader during the acquisition of the read. In the start state 202, the sensor 12 has been configured such that its resonant frequency is proportional to the parameter it senses. Examples of sensing parameters that can be measured using capacitive or inductive properties are pressure, temperature, acceleration, angular rate, pH, glucose level, salinity, concentration, dielectric constant, humidity, proximity, electrolyte level, and The degree of oxygenation. Other known parameters are also sensed.

The sensor 12 is placed away from the reader 10. In one embodiment, the sensor 12 is implanted in a human or animal body for physiological measurements. Potentially useful locations include, but are not limited to, blood vessels, cranial cavities, eyes, bladder, stomach, lungs, heart, muscle surfaces, bone surfaces, or any body cavity. The sensor 12 can be implanted for short-term acute or long-term chronic time periods. The sensor 12 can be self-contained or can be integrated with other devices such as catheters, stents, bypasses, filters, heart rate adjusters, heart rate adjuster leads, vascular closures, and the like.

The sensor 12 is designed to have an operating frequency range 220 (not shown in Figure 2) that can be reflected in the range of its sensed parameters. When there is a need to obtain a read, the reader 10 can emit an excitation pulse 14 in the vicinity of the sensor 12, as shown by block 204 in FIG. Pulse 14 can be a short burst of energy at a predetermined fixed frequency. The frequency of the pulse wave 14 can be selected to be at or near the center of the operating frequency range 220 of the sensor 12, while the frequency band of the pulse wave 14 can be narrow. One benefit of narrow-band pulse waves is that they are less likely to interfere with other devices in their vicinity. A further benefit of narrow-band pulse waves is that their use allows the system designer to select a pulse frequency within a tightly defined frequency band specified by the specification, which is easier to comply with relevant government and industry specifications for electromagnetic spectrum applications. . In one embodiment, the pulse wave 14 is a narrow frequency centered at 13.56 MHz, which is the so-called industry, science, and medical (International, Scientific and Medical) designated by the International Telecommunications Union (ITU) for commercial RF devices (Industrial, Scientific) , and Medical, ISM) one of the frequencies. A further benefit of narrowband pulse waves is that they only require lower power than equivalent equivalent continuous transmission, which makes the reader 10 more suitable for battery operation and allows for the use of smaller components. The heat required is usually lower than the control of its higher power. Finally, one advantage of transmitting the pulse wave 14 at a fixed frequency in step 204 of Figure 2 is that the transmit circuit of the reader 10 is simpler than the sweep or continuous transmit practice.

Since the sensor 12 is located in close proximity to the reader 10, step 206 of Figure 2 is followed by processing. The sensor 12 is energized via an inductance coupled between its antenna and the antenna of the reader 10. The pulse wave 14 causes current to flow within the antenna of the sensor 12 to charge the "LC slot" formed by the capacitor 15 and the inductor 13. The pulse wave 14 is typically short-lived, and in step 208, the reader 10 abruptly interrupts the pulse wave 14. Immediately, the energy stored in the LC tank circuit of the sensor 12 begins to diverge and thus oscillates at the resonant frequency of the sensor 12. The sensor 12 thus transmits the ringing signal 16 at this frequency. After the transmission is completed, the reader 10 must immediately enter the receive mode, as shown in step 210, to detect and amplify the ring signal 16.

Depending on the measurement, during the frequency measurement, the ringing signal may be weak, have noise, or only last for a short time, causing problems in accuracy and resolution. Thus, in step 212 the reader 10 may lock and hold its sampled ringing signal at a constant frequency and a large amplitude to allow sufficient time in step 214 to achieve a highly accurate frequency measurement.

Figure 3 illustrates in a frequency domain the nature of the signal exchange between the sensor 12 and the reader 10 in an embodiment in a qualitative manner. The sensor 12 senses the physical parameters it desires with a predetermined range of operational values. It maps this range of physical parameters to a corresponding operating frequency range 220. When the resonant frequency of the sensor 12 is the minimum of its operating frequency range 220, the curve 224 is the transmission function of the sensor 12. The peak of the sensor transmission function 224 is present at the resonant frequency of the sensor 12. As the sensed parameter changes within the operating range, the sensor transmission function also moves within the operating frequency range 220 accordingly. As such, the sensor transmission function can fall anywhere within the operating frequency range 220, depending on the physical parameters being sensed. Its resonant frequency (peak of the transmission function curve) also corresponds to the value of the sensed parameter. The sensor transmission function also becomes the maximum value of the sensor transmission function 222 when the sensed parameter is at the other extreme within its operating range.

The narrow band function 14 in FIG. 3 represents the excitation pulse wave 14 shown in FIG. Its frequency, represented by f _xmt , is typically fixed or close to the center of the operating frequency range 220. The pulse wave 14 is typically narrow frequency, short time, and is fixed at a predetermined frequency f _xmt . These pulse characteristics allow the reader 10 to have several advantages over other readers that must sweep or must change their transmitted frequency: a simpler circuit, a simpler control software/body , lower overall power consumption (making battery operation possible), lower power (smaller) components, less internal heat dissipation, reduced sensitivity to external sources of electromagnetic interference, and electromagnetic interference to external devices Lower likelihood, and easier to comply with government frequency allocation regulations.

Another important property shown in FIG. 3 is the horizontal line representing the lowest signal detection threshold 226 of the reader 10. After the excitation pulse 14 is turned off, the sensor 12 will dissipate the energy it receives from the excitation pulse 14. In the absence of forced excitation of the pulse wave 14, this energy causes the sensor 12 to oscillate at a resonant frequency, sending a ringing signal 16 (not shown in Figure 3). The signal strength of the ringing signal 16 is determined by the intersection between the excitation pulse 14 and the sensor transfer function: the amplitude of the ringing signal is limited by the product of the two functions at that point. This product amplitude, which must be greater than or equal to the signal detection threshold 226 of the reader 10 at the intersection, allows the ring signal 16 to be detected and measured by the reader 10.

4 provides an illustrative example of the nature of a typical handshake between reader 10 and sensor 12 in the frequency domain. The procedure shown in this figure is the same as the flowchart in Figure 2. Among the initial conditions shown in Figure 4a, the value of the sensed parameter may cause the sensor 12 transfer function 228 to be set at a frequency centered in the operating frequency range 220. It is noted that the sensed parameter (i.e., transfer function 228) varies with a slower time scale than the electronic signal that travels back and forth between the sensor 12 and the reader 10, thus, the transfer function 228 is relatively The signals are obviously pseudo-static. Since the sensed parameter is pseudo-static relative to the electronic signal, the reader 10 can take multiple samples in a short period of time and average the samples to achieve a more accurate measurement.

In Figure 4b, the excitation pulse 14 is produced by the reader 10. The pulse wave 14 is a narrow frequency to signal centered at a frequency _fxmt which is at or near the center of the operating frequency range 220. When the reader 10 is physically close to the sensor, the reader 10 generates an excitation pulse 14, which is transmitted by the reader 10 to the sensor 12. In one embodiment, this energy transfer occurs due to the coupling of inductances, with f _{xmt being} within the RF band. The intersection 230 between the reader pulse 14 and the sensor transfer function 228 is noted. The product of the two amplitudes at this point can determine the amplitude of the ringing signal of the ringing signal 16.

Next, in Figure 4c, the reader 10 emits an excitation pulse wave 14. When the excitation energy subsides, the sensor 12 is subjected to a forced driving characteristic at the transmitting frequency, which has a phase error due to the non-transmitting frequency resonance, and shifts to a frequency corresponding to the resonant frequency of the sensor and its surroundings. Passive resonance characteristics, approximately at the peak of curve 228. Due to the resonant energy within the inductance of the sensor 12, a magnetic field that varies with time is generated around the sensor 12 at this resonant frequency, which can be used by the reader 10 as the resonant frequency. One sends a signal to detect.

If the sensor 12 is exposed noted below has the transfer function 228 further moves in the right direction in FIG. 4b (toward a direction to increase f _res) of the sensed parameter, the curve 228 will be reduced in amplitude point f _xmt As a result, the cross level 230 is also reduced. As f _res further increases and reaches f _max , the amplitude 230 of the intersection is equal to the minimum detection threshold 226 of the reader 10. If the transfer function 228 moves further to the right, f _res exceeds f _max and the cross-point amplitude 230 falls below the detection threshold 226 of the reader 10. At this point, the reader 10 no longer detects the ringing signal 16, i.e., f _res has now exceeded the operating frequency range 220 of the system. It is noted that the sensor 12 must be designed such that its transfer function 228 has sufficient bandwidth to maintain the cross-point amplitude 230 above the detection threshold 226 of the entire operating frequency range 220 of the reader 10. However, designing the sensor 12 to have a wide transfer function 228 generally reduces the amplitude peaks of the transfer function 228, so a balance must be found between amplitude and bandwidth. In general, as can be seen from Figure 4, the ability of the reader 10 to detect and measure the ringing signal 16 will also follow the power level of the ringing signal after the excitation pulse 14 is stopped, along with the system Q, and the vibration. The time period of the adjacent signal 16 depends.

The transfer function 228 shown in Figure 4, the signals 14 and 16, and the shape of the operating range 220 are merely examples of illustrative properties. In some embodiments, transfer function 228 may have different characteristics and may not have symmetry with respect to fres of its peak. Furthermore, the operating frequency range 220 may not be _symmetric with respect to its f _xmt , ie the frequency of the excitation pulse 16 . The asymmetry of the operating range 220 may occur due to the characteristics of the sensor 12, or may be deliberately designed to be non-symmetric in order to shift the asymmetry of the signal 16 or the ringing signal 14 on the transmission function 228.

In another embodiment, the reader 10 can emit a pulse wave that is not at the center or near the operating range 220 of the sensor 12. In this case, the reader 10 transmits a frequency-emitted pulse wave whose frequency is associated with a frequency tuning within the operating range 220 of the sensor 12. That is, higher or lower harmonics from the transmitted pulse wave can be used as the excitation pulse wave 16, as shown in FIG.

In yet another embodiment, the reader 10 can transmit two or more excitation pulses at two different frequencies, whether simultaneously or at different times. Such multiple excitation pulses may excite different portions of the operating frequency range 220. Alternatively, the frequency generated by adding or subtracting a combination of such multiple pulse waves, or harmonics thereof, may be used as the excitation frequency in FIG.

Referring again to FIG. 1, the reader 10 can also incorporate circuitry for converting the ringing read values from the sensor 12 to a digital mode and storing it on its own memory. In addition to being measured by the sensor 12, the memory of the reader 10 can also store other relevant data. Examples include time stamp data, tuning coefficients, and the body, body upgrade, item number, serial number, usage record, historical data, fabric data, diagnostic data, and host location and sensing required to achieve system functions. The application of the device, as well as the data defined by the user.

The reader 10 can also incorporate a human interface such as a display screen, LED, or audible indication corresponding to certain portions of the frequency data. In addition, the reader 10 can also process the frequency data it receives, such as averaging, filtering, curve matching, threshold monitoring, time stamping, trend analysis, comparison with other data, and the like.

The reader 10 can also communicate with a data interface 17, as shown in FIG. The data interface 17 is external to the reader 10 and is configured to receive electronic signals from the reader 10 and also to transmit signals to the reader 10. In addition, the data interface 17 can provide power to the reader 10, such as a battery within the electrical reader 10 for charging. Examples of the data interface 17 include a host computer, a docking station, a telephone network, a cellular telephone network, a GPS network, a fiber network, a Bluetooth network, a storage area network, and an internet network. Road webpage, a remote database, a data input device, an audible sound, and a display screen.

The reader 10 and the data interface 17 can be directly connected to each other, or indirectly through an intermediate device, or communicate via a remote connection. Both can be located within the same housing. The reader 10 and data interface 17 can be connected via a cable or using a wireless link. The reader 10 can send information to the data interface 17. Examples thereof include information related to the sensor 12, measured values from the sensor 12, time stamp data, item number, serial number, carcass update information, usage record, diagnostic data, historical data, status data, organization Information, about the application of the host location and sensor, as well as the data defined by the user. The data interface 17 provides information and instructions to the reader 10. For example, the data interface 17 can provide the reader 10 with schedules and intervals for sampling the sensor 12, tuning coefficients or search tables, and the functions, functions, and structure settings required to complete the system functions. Information such as diagnostic commands, resets, reboots, user-defined information, and instructions issued by the user.

The data interface 17 is further operable to communicate with a remote data system 18 for exchanging status and control signals, as well as providing sensing data. The remote data system 18 can include a data collection module 19 for receiving data from the data interface 17, a data recording module 20 for storing the received data, and a data display 21 for displaying the sensed data. Similar to the data interface 17, the remote data system 18 can store and process data, issue commands, and distribute such data and instructions to allow communication with multiple users on a data network. Similar to the connection between the reader 10 and the data interface 17, the data interface 17 and the remote data system 18 can be connected by cable or wirelessly. The architecture shown in FIG. 5 is an example of an embodiment in which the reader 10 is connected to the data interface 17 via a cable and the data interface 17 is wirelessly coupled to the remote data system 18. Although the example of FIG. 5 utilizes the remote data system 18 to link the functions of data recording to display, it will be understood by those skilled in the art that such functions may also utilize an external data interface 17 or reader. 10 and executed.

The reader 10, sensor 12, and data interface 17 systems described above are particularly advantageous for embodiments of the field of biomedical telemetry. In this embodiment, the sensor 12 is implanted in a human body to sense physiological parameters such as intra-arterial blood pressure. The sensor 12 is extremely suitable for such use because it is made by using conventional techniques, and is particularly suitable for such use, and since it is a passive sensor, it does not require a built-in power supply that is gradually depleted, and thus is special. Suitable for this purpose. The reader 10, by its very nature, can be physically small enough to be hand-held, battery powered, cool in nature, and electromagnetically compatible with other electronic components in its vicinity. This is because of its simplicity, low-power circuits can generate the aforementioned narrow-band, fixed-frequency excitation pulse. Thus, the reader 10 can be comfortably worn on a person's clothing, in the vicinity of the implanted sensor 12, to frequently read and process/store its data. Periodically, for example, once a day, the user can place the reader 12 in a data interface 17 in the form of a docking station. The data interface 17 can include circuitry for charging the battery of the reader 10, which can update the settings and software of the reader 10 and download its data. The data interface 17 can also communicate this information to the user via the internet or telephone link, as well as other related personnel, such as the user's physician. Due to the low power activation procedure of the reader 10, such a system can perform frequent, accurate blood pressure readings with minimal inconvenience to the patient in order to deliver the data to the caregiver in an efficient manner. Obviously, this embodiment can also be applied to sensing any other internal physiological parameter that can cause a change in resonant frequency on a passive LC sensor.

In a variation of this embodiment, the sensor 12 is coupled to another implantable medical center that can perform different functions. For example, sensor 12 can be a blood pressure sensor in combination with a vascular occlusion device, such as Angio Seal from St. Jude Medical, Inc. of St. Paul, Minn. product. In yet another variation of this embodiment, the reader 10 can be combined with another device. For example, the reader 10 can be attached to a mobile phone, a pair of glasses, a handheld music player, an animated game machine, a fitting on a garment, or a watch.

A sensor 12, which includes a capacitor 15 and an inductor 13, can be assembled in a single package. Alternatively, in some applications it may be advantageous to have the capacitor 15 placed away from the inductor 13, but the two components are still connected together by wires. As an example, in embodiments where the sensor 12 is implanted in a human body, the pressure sensitive capacitor 15 may be placed where the measured target pressure occurs, and as the antenna for the inductor 13, it is possible It is placed closer to the skin surface to minimize the coupling distance between the sensor 12 and the reader 10. The wires to which they are connected may be of any of a variety of forms commonly known, including wires, wire cores, flexible circuit boards, rigid printed circuit boards, direct feedthrough connections, or rigid tips.

In an implantable embodiment, the design of the sensor 12 as the least invasive implant method may also have advantages, such as a catheter-based implant method. In addition, it may be desirable to make a portion of the implantable sensor that is radio-impermeable or ultrasonically reflective to aid in implantation and post-implantation diagnostics.

The sensor 12 can be manufactured using a variety of well-known techniques. The capacitive sensor 15 can be fabricated using microelectromechanical systems (MEMS) technology, lithography, or conventional processing techniques. Inductor 13 can be a wound coil; a FR4, Teflon, Rogers, or other printed circuit board; Low Temperature Cofired Ceramic (LTCC), green tape (greentape), or other A ceramic printed circuit board; or other inductive technique as is known in the art. The inductor 13 may or may not have a core, and may further use an electromagnetic material incorporated in the aforementioned printed circuit board or ceramic technology. Inductors and capacitors can be packaged together like a multi-chip module (MCM).

In another embodiment, the system of FIG. 1 may further include an intermediate antenna 240, as shown in FIG. The intermediate antenna 240 includes two antennas: a reader side antenna 242 and a sensor side antenna 244, which are connected in series. The intermediate antenna 240 can enhance the signal coupling between the reader 10 and the sensor 12, and may have its usefulness when there are multiple obstacles 246 and 248 between the reader 10 and the sensor 12, such obstacles. It may not be possible to penetrate through conductive wires. As an example, for one sensor 12 implanted in a blood vessel, barrier 2 (248) represents the vessel wall and barrier 1 (246) represents the skin surface. If there is an intermediate antenna 240, the signal coupling between the reader 10 and the sensor 12 will be more efficient because its coupling utilizes conduction through the wires rather than the use of radiation through the system where the medium is. . In addition, the dimensions of the antennas 242 and 244 can each be adjusted to conform to their respective antennas at the sensor 12 and the reader 10, and to increase the efficiency of coupling. Finally, the sensor side antenna 244 can be aligned accurately across the sensor inductance 13 and reduce the misalignment between the reader 10 and the sensor 12 that may be caused by the absence of the intermediate antenna 240. Error. The intermediate antenna 240 can be fabricated in a bendable circuit, a wire wound coil, or other widely used manner. It has also been noted that by adding more intermediate antennas 240 for each pair of obstacles, the concept can be extended to the use of more than two obstacles.

In another embodiment, the sensor 12 of FIG. 1 may further include a second LC tank circuit having separate inductors and capacitors, referred to as reference resonators. The reference resonator can be fabricated using the same materials, processes, and components as the inductor 13 and capacitor 15, but with two key differences. First, the value of the component of the reference resonator is fixed, which does not vary with the second parameter. Second, its fixed resonant frequency is designed to sense the operating frequency range of the resonator outside of 2200. The purpose of the reference resonator is to provide a background reading that can be utilized to correct the sensor readings taken by the reader 12. Certain factors that can cause inaccuracies, such as reader distance, intervening media changes, sensor orientation relative to the reader, component aging, mechanical stress, electrical stress, air leak, temperature, Cell growth, thrombus, etc., may affect the reference resonator in a manner similar to sensing a resonator. Using the relationship between the reference resonator deviation at its fixed frequency and the sense resonator deviation from its nominal frequency, the reader can provide a correction factor to the sensed frequency based on the reference reading. In this embodiment, an additional step is introduced between steps 202 and 204 of FIG. 2, wherein the reader 10 emits an excitation pulse at a nominal resonant frequency of the reference resonator, observing the reference ringing frequency. The deviation is calculated and (or obtained from a search table) an appropriate correction factor for the upcoming reading obtained in step 210. Alternatively, the reference reading can be taken after the sensing is read. While each change experienced by the sensing resonator may not affect the reference resonator in exactly the same way, this "self-tuning" approach can take advantage of eliminating or reducing some of the common ones found in both resonators. Inaccuracy can improve its performance. It may be, for example, related to distance, pointing, physiological response, changes in intermediate muscles, and other long-term variations of sensor 12, often referred to as "sensor drift." In addition, the choice of frequency, as well as other design points of the reference resonator, must be taken care of in order to avoid coupling with the original sensing resonator and the interaction with the reader.

The reader 10 includes circuitry (Fig. 7) that can transmit the excitation pulse 14, receive the ring signal 16, and process the ring signal 16. For example, reader 10 includes timing and control circuitry 22 to fabricate and actuate other circuitry in reader 10. The solid arrows drawn by timing and control circuit 22 represent control interfaces, such as digital or low frequency signals. The timing and control circuit 22 further generates an RF signal (shown in phantom with arrows) that is sent to the transmit circuit 24. Transmitting circuit 24 receives the RF signal and sends it to excitation pulse 14 and antenna 26 to excite sensor 12. Timing and control circuitry 22 may provide only RF signals to transmit circuitry 24 while the excitation pulse is being transmitted to avoid leakage or coupling to other nodes in the system.

The reader 10 further includes an antenna 26 coupled to the transmit circuit 24 and the receive circuit 28. Transmitting circuit 24 uses antenna 26 to transmit excitation pulse 14 and receiving circuit 28 uses antenna 26 to receive ringing signal 16. In one embodiment, antenna 26 is coupled to both transmit circuitry 24 and receive circuitry 28 in addition to switching between transmission and reception. The design of such a shared antenna requires special consideration to avoid damage to the receiving circuit 28. In particular, care must be taken not to overload the sensitive amplification stage of the receiving circuit 28. In addition, the reader 10 needs to be present in the case where the transmitting circuit 24 drives the antenna 26 to be extremely high-powered, and when the antenna is in the receiving and amplifying phase and low voltage occurs, a fast switching is made between the two. For example, when transmitting an excitation pulse, the voltage of antenna 26 may exceed 200 volts peak to peak, and when received immediately after excitation pulse 14, it may be a single digit millivolt and quickly Attenuated to microvolts. Although the reader 10 is described as having a shared antenna 26, it will be appreciated that the reader 10 may also incorporate more than one antenna to perform its function of transmitting the excitation pulse 14 and receiving the ring signal 16, respectively. .

The reader 10 further includes a phase locked loop (PLL) 30 for receiving and locking the ringing signal 16. The receiving circuit 28 can amplify and adjust the ringing signal 16 prior to being sent to the PLL 30. PLL 30 includes a voltage controlled oscillator ("VCO") 32 (not shown in Figure 7) that is operable to lock a frequency within the resonant frequency range of the sensor when no signal is present, or when receiving a sense When the detector resonant frequency is present and no signal is present to increase the lock time, it can be selected to select a frequency above or below the sensor's resonant frequency range to increase the lock time. In one embodiment, a PLL is selected, which can perform better when the no-signal PLL lock frequency is slightly above the sensor resonance frequency range. The VCO 32 generates an ac signal that is equal to the frequency of the ringing signal, referred to as the count signal 250. The PLL 30 adjusts the split down count signal to match the frequency of the ring signal 16, and sends the count signal 250 to a counter 340. The VCO 32 interfaces with a frequency counter 34 that determines the count signal 250 and provides a digital signal representative of the frequency to the external interface circuit 36 for transmission to the data interface 17. With the operation at a higher frequency than the ringing signal 16, the time required to count and record the VCO 32 count signal 250 frequency can be significantly reduced.

Each component of the reader 10 is designed to operate in an efficient manner and reduce its power consumption. To this end, the reader 10 includes a function to reduce power. Its timing and control circuitry 22 controls the power state of each of its components using a wake-up timer 38 coupled to each component. (Figure 8). In the reduced mode, some components may be completely powered down, while other components may operate in the sleep mode, where the power supply continues to maintain its fabric, but its circuitry enters a static state. In order to minimize power consumption.

The timing and control circuitry 22 can be placed in a sleep or reduced power mode when each component of the reader 10 is not in use. In addition, the entire reader 10 may be placed in a low power mode at the system level and for a time specified by an external controller. Timing and control circuitry 22 may include a set of configuration buffers 40 that may receive timing instructions from external interface circuitry 36. These instructions may establish a timing period and other timing periods of the wake-up timer 38 prior to entering the reduced power mode. In addition to timing commands from outside the reader 10, the entry/exit reduced power mode may also be triggered by the fact that the threshold of one or more signals on the reader is exceeded. The body of the reader 10 may include an algorithm that determines the entry/exit reduction power mode.

During the time the read is granted, the wake-up timer 38 can wake up each component of the reader 10 at the appropriate time to ensure that each component is in an operational state when needed. In particular, the wake-up timer 38 can communicate with a transmit timer 42, a receive timer 46, a PLL timer 48, and a frequency counter timer 50 to wake up and control the reader 10, respectively. Individual components. Once initiated, each of these timers can be controlled and each individual component can be started. When fabricating, when the start signal 52 is issued to initiate other timers, the wake-up timer 38 can be delayed for a specific time, which can be zero seconds. As shown in Figure 8, the initiation signal 52 is not shown as a continuous line from the wake-up timer 38 to the respective timers to avoid line crossings and minimize confusion.

Once initiated, the transmit timer 42 establishes the appropriate sequencing and period for the power control 54, mitigation control 56, Q control 58, and RF actuation signal 60 to properly sequence the transmit circuitry 24 and transmit frequency generator 44. . The power control signal 54 controls the power state and sleep state of the transmit circuit 24. The mitigation control signal 56 controls a damper circuit in the transmit circuit 24 to rapidly diverge the energy of the antenna 26 at the end of the transmit period. The Q control signal 58 controls a switching circuit in the transmitting circuit 24 to reduce Q and modify the bandwidth of the antenna 26 during reception of the ringing signal 16. The RF actuation signal allows the transmit frequency generator 44 to send an RF signal to the transmit circuit 24. In one embodiment, the transmit frequency generator 44 provides only the RF signal to the transmit circuit 24 during the time when the transmit circuit 24 transmits an excitation pulse 14.

The receive timer 46 is configured to establish an appropriate ordering and period with respect to the power control signal 62 to properly sequence the receive circuits 28.

The PLL timer 48 establishes appropriate sequencing and counting intervals for the power control 64 and S/H modal 66 signals to properly sequence the PLL 30. The power control 64 controls the power state and sleep state of the PLL 30. The S/H modal signal 66 controls a sample and hold circuit in the PLL 30 for locking the PLL at the transmitted frequency and then at the frequency of the ringing signal 16, followed by the frequency holding of the VCO 32 count signal 250. On the frequency until the frequency is measured by the counter 34.

The frequency counter timer 50 establishes an appropriate sequencing and counting interval for the power control 68 and the start/stop count 70 signals to properly sequence the frequency counters 34. The power control signal 68 controls the power state and sleep state of the frequency counter 34. The start/stop count signal 70 controls the start and stop times of the frequency of the VCO 32 count signal 250.

Note that although Figure 8 contains signals for shared names, such as "Start", "Composition", and "Power Control", these signals are unique for each circuit block to which they are connected. For example, power control signal 68 from frequency counter timer block 50 is not the same as power control signal 64 from PLL timer block 48, as explained above.

Transmitting circuit 24 is configured to utilize excitation antenna 14 to transmit excitation pulse 14 onto sensor 12 (Fig. 7). The excitation pulse 14 may be a fixed or rapidly varying frequency cluster at or near the resonant frequency of the sensor 12. For example, the excitation pulse 14 may be a fixed frequency bundle within a plurality of frequency bands of the resonant frequency of the sensor 12. Alternatively, the excitation pulse 14 may be a fixed or rapidly varying frequency cluster or scan for a very short period of time at or near a frequency tuned to the resonant frequency of the sensor 12. The excitation pulse wave 14 may also be an ultra-wideband pulse wave. Since the ringing signal 16 is received after the emission of the excitation pulse wave 14 has ceased, such diverse practices of the excitation pulse wave 14 are possible. Therefore, the emission of the excitation pulse wave 14 may be limited to the frequency band, amplitude, and modulation procedure that can be tolerated by government regulations. Since the sensor 12 is purely passive, the radio band regulations may not be applicable to the sensor 12.

The excitation pulse 14 causes a single and complete sampling of the ringing signal 16 due to a single short burst of energy, and thus does not require significant emission time. With a lower transmit duty cycle, power consumption can be reduced, thereby reducing the transmit, receive, count, and duty cycle of the digital processing circuitry. With reduced power consumption, battery driving has become a far more viable practice for driving the reader 10.

The excitation pulse 14 can be organized to maximize several system parameters. For example, if the pulse wave 14 is excited using a fixed frequency, the frequency of the pulse bundle can be configured to amplify the parameter to a maximum, such as a maximum allowable emission peak power, during which the PLL is locked to the ringing signal 16 during "receiving" The maximum degree of freedom of interference from in-band or near-band interference at that time is the maximum worldwide acceptability of a particular frequency at which the reader can transmit for its desired sensing purpose, or other such criteria.

FIG. 9 shows the transmitting circuit 24. A level shifter 72 of the transmit circuitry 24 receives the control signals 54, 56, 58 and RF signals from the timing and control circuitry 22. The level shifter 72 buffers the input and converts the control logic level to a circuit drive level. A transmit driver 74 amplifies the RF signal to provide sufficient power to drive the antenna 26. Q control circuit 76 is actuated during reception to reduce the sum Q of antenna 26 and tuning and D.C. block 82. A damping circuit 78 is briefly actuated shortly upon termination of the emission of the excitation pulse wave 14 to absorb energy in the antenna and to allow the antenna to react to the ringing signal 16. Damping circuit 78 can provide different Q factors to the antenna to enhance reception of ringing signal 16. The power control circuit 80 controls the power-on and sleep modes of the components in the transmit circuit 24. Tuning and D.C. block 82 adjusts the tuning of antenna 26 and avoids direct current biasing of damping circuit 78 improperly. The RF output of the excitation pulse 14 from the transmitting circuit is wound to both the antenna 26 and the receiving circuit 28.

Once the excitation pulse 14 is transmitted by the transmitting circuit 24, the receiving circuit 28 is configured to listen to the ringing signal 16. Referring to Figure 10, a high Z buffer/clamp 84 includes a high impedance ("high Z") input device that limits the effects of the receiving circuit 28 on tuning and tuning performed by D.C. The high Z buffer/clamp 84 further protects the amplification stage 86 from extreme voltages that appear on the antenna 26 during the excitation pulse 14 being emitted. During excitation pulse transmission, the voltage on antenna 26 may be as high as 200 volts peak to peak, and only about 60 pico-farad of capacitance is required to tune the antenna. In one embodiment, a 1 picofarad capacitor is utilized as a high impedance input current limiting device for a 13.56 MHz transmit circuit. Bypassing the overvoltage to the power supply and bypassing the low voltage to ground, the low capacitance diode junction can be placed on the receiver side of the 1 pF capacitor so that the capacitor can limit the current through the diode It protects the receiving amplifier from the high voltages during transmission through the antenna 26.

The amplification stage 86 amplifies the ring signal 16 to a sufficient level to drive the PLL 30 input. It is necessary to carefully design the amplification stage 86 so that the emitted excitation pulse 14 is removed and clamped, and a low level of ringing signal 16 is received to achieve an appropriate transient response. A common gate amplifier stage with a low Q-tuned reactive drain load can be used to regulate the output of the high Z buffer/clamper 84 followed by a number of filters interspersed between the high gain amplifier stages. These non-resistive capacitor ("RC") filters are inductor-capacitor ("LC") filters. In an embodiment, such filters may all be RC bandpass filters. Another common gate amplifier stage with low Q-tuned reactive drain load can perform final band pass adjustment before the signal is fed to the PLL 30 input. This design allows all of these amplifier types to operate from very low signal input levels to very high signal input levels, without signal distortion such as frequency doubling or halving due to stage saturation characteristics, and The excellent high input impedance achieved with a common gate amplifier stage and the superior transient response characteristics of an RC filter interspersed between high gain amplifier stages. In particular, it should be noted that the stage-to-stage power supply is isolated from the signal to avoid unfavorable oscillations due to the extreme gain associated with the amplification stage 86.

Power control circuit 88 can supply power to and remove power from the amplifiers in amplifier stage 86 and high Z buffer/clamp 84 to reduce power consumption. It should be noted that the high Z buffer/clamp 84 is designed to provide complete protection even when power has been removed, because excessive energy will activate the amplification stage 86 until the energy is dissipated. Its input impedance will be high enough to avoid over-supplying power to the amplifier stage 86. In an embodiment, the receiving circuit 28 is activated during the firing of the excitation pulse 14 to reduce the time required for the PLL 30 to lock on the ringing signal 16.

The PLL 30 receives the amplified and adjusted ringing signal 16 from the receiving circuit 28. Referring to Figures 10 and 11, the RF signal from amplifier stage 28 of amplifier circuit 28 is fed to an RF buffer 90 of PLL 30. The RF buffer 90 can feed the RF signal to a selective RF slicer 92 that splits the RF signal frequency by an integer value (Fig. 11). The RF slicer 92 then feeds the RF signal to a first input of a phase frequency detector 84. The output of frequency detector 84 is fed to a sample hold (S/H) error amplifier 96. This S/H error amplifier 96 controls the frequency of the VCO 32. The count signal 250 output of the VCO 32 is fed to a VCO slicer 98, which is then fed to a second input of the frequency detector 84. PLL 30 may include an output buffer 102 to reduce the load on VCO 32 while also forwarding the frequency of count signal 250 to frequency counter 34. The VCO slicer 98 allows the VCO 32 to operate at a frequency that is significantly higher than the ringing signal 16. As a result, the time required to count and record the frequency of the VCO signal can be significantly reduced. In addition, a shorter counting interval can also reduce the drift of the VCO during counting and allow for higher sampling rates.

The phase frequency detector 84 is configured to determine the frequency and phase error between the separated RF signal and the separated VCO signal. This can preferably be achieved by filtering and amplifying the signal fed to the S/H error amplifier 96. In addition, its S/H specificity can also selectively lead the filtered and amplified signal to control the VCO 32. In this manner, a closed control loop can be formed that causes the VCO 32 to count the frequency of the signal 250 equal to the frequency of the ringing signal 16 multiplied by the integer of the split VCO splitter 98, divided by the RF splitter 92. Value. PLL 30 may include an additional frequency divider to optimize the circuit design and increase the potential VCO 32 frequency range.

The PLL timer 48 sends an S/H modal control signal 66 to the S/H error amplifier 96 of the PLL 30. The S/H modal control signal 66 can place the VCO 32 in a sampling mode. In a preferred embodiment, the VCO 32 is placed in a sampling mode for a predetermined length of time. In the sampling mode, the frequency of the separated VCO count signal is adjusted to conform to the frequency of the ringing signal 16, as previously described. When the S/H modal control signal 66 is placed in the hold mode, the S/H error amplifier 96 will hold its output constant, causing the VCO 32 control signal to be sufficient to slice the VCO 32 count signal 250 frequency. Time is roughly constant.

From the power control signal 64 from the PLL timer 48 to the power control circuit 104, it can be determined that the PLL 30 is in a sleep/power off mode in which the power is turned on or the power can be saved. Depending on the particular PLL 30 used, a control and communication chain (not shown) may be required to set an integer of the RF slicer 92, an integer of the VCO slicer 98, and the output of the phase frequency detector 84 and Its output fabric. The communication chain may be specific to the particular PLL 30 used.

The frequency counter 34 includes a counting stage 106, a counting buffer 108, and a power control circuit 110, as shown in FIG. The frequency count timer 50 sends a start/stop control input 70 to the count stage 106 and the count buffer 108. Frequency count timer 50 also sends a power control input 68 to power control circuit 110. Count stage 106 counts the frequency of the VCO signal from PLL 30 output buffer 102. When the start/stop control command starts, the count stage 106 starts counting and terminates when the start/stop control command is stopped. When the start/stop control command is stopped, the count buffer 108 is loaded with the count value from the count stage 106. The power control circuit 110 controls the power-on and sleep modes of the various components within the frequency counter 34. The count buffer 108 output can supply a count input to the external interface circuitry 36. The ringing signal 16, as well as its connected sensed parameters, can be determined from the frequency count value.

In other embodiments, other methods of measuring the frequencies received and amplified are possible. Such methods include directly counting the ringing signal or using various frequency to voltage conversion circuits in the prior art.

In operation, the readers 10 are ordered in the following manner. When the sensor is not sampled, all components of the reader 10 are placed in a power reduction mode. The wake-up timer 38 in the timer and control circuit 22 is configured to perform a particular sampling delay or sampling interval. At a particular point in time, wake-up timer 38 initiates a sampling procedure. In particular, wake-up timer 38 turns on or wakes up each component of the reader at the appropriate time to ensure that each component is in an operational state when needed.

External interface circuitry 36 is typically not required in the sampling process, except for receiving the final data generated by it. Its entry/exit low power mode can be handled by an internal or external controller rather than the timing and control circuitry 22. The timing and control circuit 22 provides an RF signal to the transmit circuit 24 for a short period of time, such as about 20 microseconds. The RF signal from the timing and control circuit 22 is then terminated and the transmit circuit 24 is controlled to rapidly reduce the signal transmitted by the antenna 26. Transmitting circuit 24 is then placed in an appropriate mode to allow reception of ringing signal 16 on antenna 26. In an embodiment, when antenna 26 is configured to receive ringing signal 16, the attenuation of antenna 26 is greater than the attenuation of ringing signal 16.

During the firing of the excitation pulse 14, the receiving circuit 28 receives, adjusts, and clamps the RF signal transmitted by the antenna 26. Once the firing of the excitation pulse 14 is stopped and the antenna 26 is configured to receive the ringing signal 16, the receiving circuit 28 transitions into a high gain receiving mode to receive the ringing signal 16 from the antenna 26. The PLL 30 is in a sampling mode to allow the RF buffer 90 to receive the output of the received circuit 28 that has been adjusted. When the antenna 26 begins to receive the ringing signal 16, the PLL 30 is shifted from the state originally locked to the frequency of the transmitted excitation pulse wave 14 to the frequency locked to the ringing signal 16. After a time interval for the PLL 30 to lock the frequency of the ringing signal 16, the PLL 30 is moved to the holding mode to maintain the VCO 32 counting signal 250 frequency at the frequency of the ringing signal 16. The time required for the lock can be predetermined or can be an adaptive approach based on the detected PLL lock condition. After the lock, the receiving circuit 28 and the transmitting circuit 24 can be turned off or placed in the sleep mode as appropriate.

Once the PLL 30 enters the hold mode, the timing and control circuit 22 instructs the frequency counter 34 to perform a controlled interval count of the VCO 32 count signal 250 frequency. When the counting is completed, the components of the PLL 30 are turned off or placed in a sleep state as appropriate, and their count values are transferred to the external interface circuit 36. The components of frequency counter 34 are then turned off or placed in a sleep state as appropriate, and then the components of timing and control circuit 22 are then turned off or placed in a sleep state as appropriate. If it is planned to perform interval sampling, the timing and control circuit 22 wakes up the timer 38 to count until the next sampling is due. Otherwise, timing and control circuitry 22 waits for a wake up command and any other desired instructions from external interface circuitry 36. In the cluster sampling mode, the power-on time required for the component to be ready may exceed the time the power is turned off, in which case the components will remain powered until the sample bundle is complete.

As shown in FIG. 13, an embodiment of PLL circuit 30 of reader 10 includes a number of features that can be added to PLL 30 to achieve different but equivalent functions relative to the circuitry of PLL 30 described above. Some or all of the variations that can be seen in Figures 11 and 13 can be used to enhance the operational performance of the PLL 30 of Figure 11. The selective input RF buffer 111 allows an RF signal, whether from the amplification stage 86, or a reference signal generated elsewhere in the reader 10, to be selected as the input to the RF slicer 92. Its selection is determined by the reference/receive control input of the RF buffer 111. The error amplifier 112 has been simplified and does not directly provide the ability to sample and hold, as previously described by the S/H error amplifier 96 of FIG.

The circuit components include an analog to digital (A/D) converter 113, a digital to analog (D/A) converter 114, and a switch 115, as shown in FIG. These components can be used to achieve sampling and holding functions. In the configuration of FIG. 13, a reference frequency signal "Ref signal" may be selected as the input of the RF buffer 111 during the period in which the reader 10 excites the pulse wave 14 to be transmitted to the sensor 12, and the reference signal will It is maintained until the RF signal of In A of the selectable input RF buffer 111 becomes stable and can be obtained by the receiving circuit 28. This reference signal allows the PLL 30 to "pre-lock" on a stable reference signal to reduce the lock time when a ring signal is available from the receiving circuit 28. The output of the selectable input RF buffer is split by the RF slicer 92 to any value greater than or equal to one, and the split buffered signal is then fed to the phase frequency detector 94. The output of phase frequency detector 94 is fed to an error amplifier 112 which provides the appropriate gain and frequency response to the VCO 32 in PLL 30 as a control signal. The output of error amplifier 112 is fed to input A of switch 115. When selected as input A, switch 115 dispatches error amplifier 112 signals to both VCO 32 and A/D converter 113. A/D converter 113 is then used to sample the control voltage of the VCO to determine that VCO 32 is locked to a control voltage level relative to a frequency of input A of selectable input RF buffer 111. The signal of A/D converter 113 can be used to indirectly measure VCO 32 frequency, as will be explained below, and can be used to determine the appropriate settings of D/A converter 114 so that switch 115 can be set to Input B is maintained at the locked frequency input level for any length of time to achieve a sampling and hold function similar to that described by S/H error amplifier 96 of FIG.

Several minor modifications to the operation of the aforementioned circuit of Figure 13 may allow for functionally equivalent results. One such modification is to adjust the voltage of the A/D converter 113 to the frequency of the RF signal of the particular receiving circuit 28 using the input B of the selectively input RF buffer 111 fed with a known frequency. . Once the relationship between the signal input of the RF buffer and the digital output of the A/D converter 113 is clearly defined, the output of the A/D converter 113 can be used to represent the frequency of the ringing signal 16. The output of the A/D converter 113 becomes a PLL output. Operating in this manner allows the A/D converter 113 to partially or completely replace the functional translation of the output buffer 102 and the frequency counter 34.

Another "modification of the foregoing operation of the circuit of Figure 13 is to use the data from the A/D converter 113 to perform a lock analysis of the PLL 30 in order to reduce the lock time and improve the accuracy of the lock frequency. When the sensor 12 signal 16 is available on the output of 28, the output of the error amplifier 112 converges to the locked voltage value, and then when the level of the signal of the sensor 12 signal 16 can be maintained by the lock, The way the forecast is divergent, so this is possible.

Another modification of the foregoing actions of the circuit of Figure 13 utilizes a frequency counter 34 that uses a D/A converter 114 to generate a particular voltage at the input of the VCO 32, at which the output of the A/D converter is recorded, And determining the frequency of the signal appearing on the output of the output buffer 102. This may allow the frequency counter 34 to be used to calibrate the A/D converter on one or more frequencies.

For those skilled in the art of circuit design, it is apparent that some minor modifications to the circuit of Figure 13 include rearranging switch 115 and D/A converter 114 from the position shown in Figure 13 to the phase frequency detector. The position between 94 and error amplifier 112. This rearrangement requires an additional step of tuning the D/A converter 114 through the error amplifier 112 to determine the appropriate ratio of control voltages required to achieve the VCO 32, which may utilize either the A/D converter 113 or the frequency. Counter 34 or both are achieved. However, such an arrangement allows the D/A converter 114 to be used for pre-locking, rather than using the reference signal of the input B of the selectable RF input buffer 111. Such an arrangement, if combined with the aforementioned A/D converter 113 tuning procedure that eliminates the output buffer 102 and the frequency counter 34, allows for the time required to resolve the resonant frequency of the sensor 12 for each ringing period. While operating the reader 10 requires a moderate reduction in power. Other micro-modifications of the foregoing embodiments are system processing loads that are assigned to appropriate locations based on power constraints, computational complexity, time critical requirements, or other system related priorities. Such modifications may cause the processing or analysis of the designer's data to be placed in the remote data system 18, the reader 10, by the A/D converter 113, or the D/A converter 114, or the frequency counter 34. Or any of the external data interfaces 17.

In yet another embodiment of the reader 10 circuit, the digital spectrum analysis circuit replaces the PLL 30 and frequency counter 34 of Figure 7, the result of which is a modified version of the block diagram shown in Figure 14. The digital sampling circuit 260 replaces the PLL 30, and the spectrum analysis circuit 262 replaces the frequency counter 34. The analog count signal 250 is also replaced by a digital count signal 264.

Functionally, the digital sampling circuit 260 extracts information from the ringing signal 16 during the short ringing of the ringing signal 16 and digitizes it. Receive circuit 28 may amplify and adjust ring signal 16 prior to transmission to digital sampling circuit 260. The digital sampling circuit 260 can directly sample the radio frequency output of the receiving circuit 28 to obtain time domain based data for further analysis.

In one embodiment, the reader 10 further includes a spectrum analysis circuit 262 to convert the time domain data output from the digital sampling circuit 260 into frequency domain data and buffer the frequency domain data for forwarding to the external interface circuit 36. The spectrum analysis circuit 262 may also include a distinguishing function to determine the ringing frequency of the ringing signal. As will be appreciated by those skilled in the art, some or all of the functionality of the spectrum analysis circuit 262 can be directly performed by the reader 10 or the remote data system 18, the main difference of which is the data sent via the external interface circuit 36. The form and quantity, and the processing energy required for where it is processed.

Digital sampling circuit 260 and spectrum analysis circuit 262 are controlled by timing and control circuit 22 in a manner similar to the embodiment depicted in FIG. The block diagram of Figure 15 depicts another embodiment of timing and control circuitry 22 that may be adapted to control the circuitry of the alternative reader 10 shown in Figure 14. The PLL timer 48 of FIG. 8 is replaced by the digital sampling timer 274 of FIG. This timer can establish an appropriate ordering and period for the power control 270 and the sample start 272 signals to order the digital sampling circuit 260. The power control signal 270 controls the power state and sleep state of the digital sampling circuit 260. The sample start signal 272 causes the digital sampling circuit 260 to collect an appropriate number of samples in a cluster sampling mode for transmission to the spectrum analysis circuit 262.

Similarly, the frequency count sequencer 50 of FIG. 8 can be replaced by the spectrum analysis timer 280 of FIG. The spectrum analysis timer 280 establishes appropriate sequencing and timing for the power control 276 and the analysis start 278 signals to order the spectrum analysis circuit 262. The power control signal 276 controls the power state and sleep state of the spectrum analysis circuit 262. The analysis initiation signal 278 controls the time at which the spectrum analysis circuit 262 begins to estimate the sampled bundle 264 provided by the digital sampling circuit 260.

In another embodiment of FIG. 14, the receiving circuit 28 is functionally and structurally equivalent to the receiving circuit 28 of the PLL-based embodiment of FIGS. 7 and 10, the only difference being that in the digital position Sampling circuit 260, rather than the input to PLL 30, outputs the output signal of amplifier stage 86 to analog to digital converter 290.

The block diagram of Figure 16 illustrates an embodiment of a digital sampling circuit 260. The RF signal from amplifier stage 28 of amplifier circuit 28 is fed to the analog to digital converter 260 input to the input of digital converter (ADC) 290. The ADC 290 converts the RF signal into a set of time-correlated samples that are taken in close enough intervals and have a sufficient number of samples to enable the spectrum analysis circuit 262 to achieve its required frequency accuracy. This set of time-dependent samples is referred to herein as bundle 264 of a digital sample.

The digital sample bundle 264 output from ADC 290 is fed to a spectral analysis circuit 262 time to frequency domain conversion circuit 94, as shown in FIG. The internal details of the frequency domain conversion 94 are not described in detail herein, so the conversion may include fast or discrete Fourier transform, discrete or continuous wavelet transform, and any of several Laplace transforms. Any of several methods, such as any of a number of Z conversions, or other conversion methods known in the art. The internal details of frequency domain conversion 94 may utilize either hardware or software or any combination of the two to achieve the desired conversion. Since the output of the frequency domain conversion 94 is generated at the sampling interval and may contain a plurality of values for transmission to the external data interface 17, a result buffer 96 of the spectrum analysis circuit 262 is shown to hold the values. Until these values are transferred to the external data interface 17.

In this digital spectrum analysis embodiment, in addition to the functions performed by the digital sampling circuit 260 and the spectrum analysis circuit 262 in relation to the determination of the frequency of the ringing signal 16, the reader 10 operates on the sequence and the aforementioned "reader operation sequencing". The description is similar. When the antenna 26 begins to receive the ringing signal 16, the digital sampling circuit 260 quickly samples for a predetermined or a calculated period of time to obtain a digital sample bundle 264. After the digital sampling bundle 264 is completed, the receiving circuit 28 and the digital sampling circuit 260 are turned off or placed in a sleep mode as appropriate. The spectrum analysis circuit 262 converts the digital sample bundle 264 into the frequency domain and places the result in the result buffer 96, which is then shifted into the low power mode. Next, the components of the timing and control circuit 22 are turned off or placed in a sleep mode as appropriate. If it is planned to perform interval sampling, the timing and control circuit 22 wakes up the timer 38 to count until the next sample is reached. Otherwise, timing and control circuit 22 waits for a wake up command from external interface circuit 36 and any other desired instructions. Under communication interface control, the sample data in result buffer 96 remains available to external interface circuitry 36 for transmission to remote data system 18.

It will be apparent to those skilled in the art that the foregoing digital frequency analysis embodiments can be modified in various minor ways to achieve functionally equivalent results. One such modification is the use of zero-padding of the ADC data, which is common practice in the time domain to frequency domain conversion practice in which the signal bundle data is evaluated. Another modification is to actually move the spectrum analysis circuit 262 from the reader 10 to the remote data system 18, whose ADC 90 data is transmitted by the reader 10 to the remote data system 18 in time domain form. Yet another such modification is to utilize frequency multiplication, division, summing or differential circuitry at a certain point in the reader 10 for any reason related to frequency selection, bandwidth, sampling time, etc. The ringing signal 16 is frequency converted to change the ringing signal 16 to a frequency signal of intermediate nature. Yet another modification is the use of signal processing techniques to filter, shape, analyze, compare with other data, or process and evaluate frequency or time domain data.

Similarly, those skilled in the art will readily appreciate that a combination of various frequency determination methods disclosed herein is possible and may have advantages in various applications. For example, an analog sampling and holding circuit can be used in conjunction with digital spectrum analysis to hold the ringing signal 16 for a sufficient amount of time to obtain an appropriate sample for digitization.

In another embodiment, a standard RFID tag, which is known in the art, can also be integrated into the sensor 12. Such a tag can have a separate antenna and operate at a frequency outside the range of the sensor operating range 220 (Sensor Operational Range 220). It can be encoded using the fabric information on the sensor 12.

The embodiments of the present invention have been described in detail above, and it is obvious that modifications and changes can be made by those skilled in the art after reading and understanding the specification. All such modifications and variations are intended to be included within the scope of the invention as defined by the invention.

10. . . Reader

17. . . External data interface

19. . . Remote data collection

20. . . Remote data record

twenty one. . . Remote data display

twenty two. . . Timing and control circuit

twenty four. . . Transmitting circuit

26. . . To the antenna

28. . . Receiving circuit

30. . . Phase locked loop

32. . . Voltage controlled oscillator

34. . . Frequency counter

36. . . External data interface

38. . . Wake up timer

40. . . Fabric buffer

42. . . Launch timer

44. . . Transmit frequency generator

46. . . Receiving timer

48. . . PLL timer

50. . . Frequency counter timer

52. . . Start

54. . . power control

56. . . Mitigation control

58. . . Q control

60. . . Enable

62. . . power control

64. . . power control

66. . . S/H mode

68. . . power control

70. . . Start/stop counter

72. . . Level shifter

74. . . Transmitter driver

76. . . Q control circuit

78. . . Damping circuit

80. . . Power control circuit

82. . . Tuning and D.C.

84. . . High Z buffer/clamp

86. . . Amplifier stage

88. . . Power control circuit

90. . . RF buffer

92. . . RF slicer

94. . . Phase frequency detector

96. . . S/H error amplifier

98. . . VCO slicer

102. . . Output buffer

104. . . Power control circuit

106. . . Counter level

108. . . Count buffer

110. . . Power control circuit

111. . . Select input RF buffer

112. . . Error amplifier

113. . . Analog to digital converter

114. . . Digital to analog converter

115. . . switch

202. . . Start state: the resonant frequency of the sensor is proportional to the parameter being sensed

204. . . The reader emits an excitation pulse at a fixed frequency

206. . . The sensor is charged via inductive coupling, storing energy in the resonant tank circuit

208. . . The reader terminates the transmission; when the reader emits the stored energy, it sends the ringing signal at its resonant frequency

210. . . The reader receives the ringing signal and zooms in

212. . . The reader locks and holds the ring signal

214. . . The reader measures the frequency of the signal it holds

250. . . Counting signal

260. . . Digital sampling circuit

264. . . Digital sampling bundle

262. . . Spectrum analysis circuit

270. . . power control

272. . . Sampling begins

274. . . Digital sampling timer

276. . . power control

278. . . Analysis begins

280. . . Spectrum analysis timer

290. . . Analog to digital converter

292. . . Power control circuit

Among the drawings of the present invention:

Figure 1 is a block diagram of a passive wireless sensor system;

2 is a flow chart illustrating a method for reading from a sensor;

The graph of Figure 3 illustrates in nature the characteristics of the frequency at which the signal is exchanged between the sensor and the reader;

Figure 4 continues to include a three-curve diagram that characterizes the frequency of the signal exchange between the sensor and the reader during the acquisition read.

5 is a block diagram of the passive wireless sensor system of FIG. 1 expanded to accommodate an external data interface and remote data processing functions;

Figure 6 is a block diagram of the passive wireless sensor system of Figure 1, supplemented by an intermediate antenna;

Figure 7 is a top-level block diagram of the internal circuit of the reader;

Figure 8 is a block diagram of the timing control and control portion of the reader circuit;

Figure 9 is a block diagram of the transmitting portion of the reader circuit;

Figure 10 is a block diagram of a receiving portion of a reader circuit;

Figure 11 is a block diagram of a phase locked loop portion of the reader circuit;

Figure 12 is a block diagram of the frequency counter portion of the reader circuit;

13 is a block diagram of another embodiment of a phase locked loop portion of the reader circuit shown in FIG. 11, having a digital sampling timer and a generating function required to perform sampling and holding;

14 is a block diagram of another embodiment of the internal circuit of the reader of FIG. 7, wherein the PLL and the frequency counter are replaced by a digital sampling circuit and a spectrum analysis circuit;

Figure 15 is a block diagram of another embodiment of the timing and control circuit of Figure 8, wherein the PLL timer and the frequency counter timer are replaced by a digital sampling timer and a spectrum analysis timer, respectively;

Figure 16 is a block diagram showing the internal structure of the digital sampling circuit block of Figure 14;

Figure 17 is a block diagram showing the internal construction of the spectrum analysis circuit block of Figure 14.

202. . . Start state: the resonant frequency of the sensor is proportional to the parameter being sensed

204. . . The reader emits an excitation pulse at a fixed frequency

206. . . The sensor is charged via inductive coupling, storing energy in the resonant tank circuit

208. . . The reader terminates the transmission; when the reader emits the stored energy, it sends the ringing signal at its resonant frequency

210. . . The reader receives the ringing signal and zooms in

212. . . The reader locks and holds the ring signal

214. . . The reader measures the frequency of the signal it holds

Claims (23)

A method for obtaining a measured value from a remote location, the method comprising: transmitting at least one excitation pulse wave to a wireless sensor at a fixed frequency; and reacting to the at least one excitation pulse wave by the wireless sensor Receiving at least one signal; sampling the received signal; and maintaining a constant frequency of holding the received signal with a reader for a period of time sufficient to determine the received signal, wherein the wireless sensor It is configured to change its resonant frequency in a manner that is proportional to at least one sensed parameter. The method of claim 1, further comprising transmitting a plurality of the excitation pulses at the fixed frequency, receiving the plurality of the signals, and determining a frequency of the signals. The method of claim 2, wherein the plurality of determined frequencies are averaged. The method of claim 1, wherein the excitation pulse wave is a radio frequency (RF) signal. The method of claim 4, wherein the frequency of the excitation pulse is about 13.56 MHz. The method of claim 1, further comprising the step of combining the wireless sensor with a device providing a different function. The method of claim 1, wherein the signal is a ringing signal. The method of claim 1, further comprising the steps of: transmitting at least one reference excitation pulse wave at a reference frequency different from the fixed frequency; reacting with the at least one reference excitation pulse wave by the wireless sensor Receiving at least one reference reaction signal in combination with one of the fixed reference resonators; sampling and holding the reference reaction signal; determining a frequency of the reference reaction signal; and improving the received signal according to the determined frequency to the at least An association of parameters is sensed. A system for obtaining a measurement from a remote location, the system comprising: a wireless sensor configured to change a resonant frequency of the at least one sensed parameter; and a reader Correspondingly configured to transmit at least one excitation pulse wave to a wireless sensor at a fixed frequency to receive at least one signal from the wireless sensor in response to the excitation pulse wave, the received signal Sampling is performed and the frequency at which the received signal is held is constant and continues for a period of time sufficient to determine the frequency of the received signal. The system of claim 9 wherein the wireless sensor comprises at least one capacitor and at least one inductor, and wherein the at least one inductance varies with the at least one sensed parameter. The system of claim 9, wherein the wireless sensor comprises at least one capacitor and at least one inductor, and wherein the at least one capacitor varies with the at least one sensed parameter. The system of claim 9, wherein the wireless sensor is combined with a device that provides a different function. The system of claim 9 further includes an intermediate antenna disposed between the reader and the wireless sensor. The system of claim 13 wherein the intermediate antenna comprises two antennas that are connected together by electrically conductive wires. The system of claim 14 wherein the first of the two antennas is configured and designed to optimize communication with the reader. The system of claim 14 wherein the second of the two antennas is configured and designed to optimize communication with the wireless sensor. The system of claim 9 wherein the at least one excitation pulse wave is at least one of: a pulse included in a frequency band of plus or minus twenty percent of the frequency at which the signal is received. Frequency; a pulse comprising a frequency within a frequency band of plus or minus twenty percent of the frequency of the first harmonic of the received signal; an ultra-wideband pulse having a pulse that is less than twice the period of the signal a width and a frequency band not less than one third of the frequency of the received signal; and a pulse which is not less than eight tenths of the received signal frequency and not higher than the signal frequency At a frequency of twelve, not less than ten and no more than ten thousand The cycle consists of. The system of claim 9 wherein the signal is a ringing signal. The system of claim 9, wherein the reader is a handheld device. The system of claim 9, wherein the reader is driven by a battery. The system of claim 9 wherein the wireless sensor further comprises an additional resonant circuit having a resonant frequency different from a fixed resonant frequency of the wireless sensor. The system of claim 21, wherein the reader is further configured to transmit a second excitation pulse at the fixed frequency of the additional resonant circuit and to receive a response from the additional resonant circuit. The system of claim 22, wherein the reader is further configured to determine a frequency of the reaction from the additional resonant circuit for modulating the at least one received signal.