TWI497071B - Control circuit for use with a four terminal sensor, and measurement system including such a control circuit - Google Patents

Description

Control circuit for use with a four-terminal sensor and measurement system including the same

The present invention relates to a control circuit for use with a four-terminal sensor, a combination of a four-terminal sensor and a control circuit, and a method of improving the accuracy of a metrology system when used with a four-terminal sensor. For example, the sensor can be a biosensor, such as a glucose sensor.

According to a first aspect of the present invention, there is provided a control circuit for use with a four-terminal sensor having first and second drive terminals and first and second measurement terminals, the control circuit being configured Driving a voltage between the first and second measurement terminals by driving at least one of the first and second drive terminals to sense a voltage difference between the first and second measurement terminals and controlling an excitation signal The difference is within the target voltage range, and the control circuit contains N poles in its gain frequency transfer characteristic and N-1 zero points in its gain frequency transfer characteristic, so that when the loop gain falls to unity gain, the closed loop The phase shift around is basically not 2π radians or multiples thereof, where N 2 is an integer.

Therefore, stability is achieved only by reducing the open-loop gain at low frequencies until the closed-loop gain in the control circuit has dropped to unity gain while maintaining the gain as a function of frequency, substantially -20 dB per 10 octave variations. In comparison to the situation, a higher increase in the control circuit can be used. The impedance measurement of the impedance of the four-terminal sensor is performed at a plurality of frequencies.

The N poles and N-1 zero points mentioned above are the poles and zeros that occur at frequencies at which the control circuit has sufficient gain to cause oscillations to occur. This means, for example, that the gain in the forward path within the loop is greater than the unity gain. Preferably, the gain margin is included in the loop to allow for manufacturing variations or temperature variations, and thus the N poles and N-1 zero points mentioned in the Summary are at a gain greater than the unity gain modified by the gain margin. Occurs, for example, at a gain of 0.5 or less, such as 0.3.

Above the controller gain or loop gain has dropped to less than the unity gain (preferably modified by the gain margin) such that the poles occurring in the frequency space of the gain, for example, less than 0.5, do not cause instability and can be ignored.

The use of a higher gain in the control circuit means that the voltage difference between the first and second measurement terminals can be more tightly controlled to a target value or a target value range, and thus the amount of other parameters will be correspondingly improved. Measurement.

Advantageously, the control circuit has first and second reference voltage input terminals for receiving the differential reference voltage. The differential reference voltage sets a target voltage for a voltage difference between the first and second measurement terminals.

Advantageously, the four terminal sensor comprises a load whose impedance varies, inter alia, as a function of the concentration of the chemical, enzyme or biological material. Alternatively, the impedance of the load can be varied as a function of the reaction. It is known to produce sensors for electronic detection of biological parameters. An example of such a widely used electrically readable biosensor includes a blood glucose measuring strip for treating diabetes.

In accordance with a second aspect of the present invention, a control circuit is constructed in conjunction with a four terminal sensor to form an embodiment of the first aspect of the present invention.

According to a third aspect of the present invention, a method of operating an instrument circuit including a multi-terminal sensor and an excitation circuit is provided, wherein the multi-terminal sensor has at least one driving terminal different from the first sensing terminal, and wherein The terminal sensor has another terminal and its The medium excitation circuit is configured to measure a voltage difference between the first sensing terminal and the other terminal, and use the voltage difference to control an excitation signal applied to the at least one driving terminal to connect the first sensing terminal to the other terminal The voltage difference between them remains at a target value or remains within the target range, and wherein the excitation circuit is configured to have at least one zero point in its transfer characteristic such that it satisfies the Barkhausen stability criterion.

Preferably, the at least one zero point is positioned below the frequency at which the forward gain of the excitation circuit has decreased to less than the unity gain such that the closed loop containing the excitation circuit cannot undergo self-sustained oscillation.

2‧‧‧ four-terminal sensor

4‧‧‧Control circuit

6‧‧‧Control circuit

8‧‧‧ Current measurement circuit

10‧‧‧Load/Unit

20‧‧‧First drive terminal

22‧‧‧Second drive terminal

24‧‧‧Resistors

26‧‧‧second resistance

30‧‧‧First measuring terminal

32‧‧‧Second measurement terminal

34‧‧‧Resistors

36‧‧‧Resistors

41‧‧‧ first input

42‧‧‧second input

43‧‧‧ third input

44‧‧‧ fourth input

50‧‧‧ Output node

52‧‧‧reference voltage

60‧‧‧Small signal grounding

70‧‧‧Sensor Resistors

80‧‧‧ analog digital converter

90‧‧‧Operational Amplifier

100‧‧‧ turning point

110‧‧‧ pole

120‧‧‧Parasitic capacitors

122‧‧‧Parasitic capacitors

140‧‧‧ pole

150‧‧‧Area

160‧‧‧ Curve

170‧‧‧00

172‧‧‧ pole

200‧‧‧ first input

202‧‧‧second input

210‧‧‧ Impedance

212‧‧‧ Impedance

214‧‧‧Inverting input

216‧‧‧First operational amplifier

220‧‧‧ capacitor

222‧‧‧ output

224‧‧‧ Output node

240‧‧‧Circuit block

240a‧‧‧ Block

250‧‧‧Operational Amplifier

252‧‧‧Input resistor

254‧‧‧ capacitor

256‧‧‧Resistors

260‧‧‧ Output node

270‧‧‧First reference voltage generator

302‧‧‧Second control circuit

307‧‧‧Second reference voltage generator

320‧‧‧Operational Amplifier

322‧‧‧ capacitor

324‧‧‧Resistors

The invention will now be described, by way of non-limiting example, with reference to the accompanying drawings in which: FIG. 1 is a circuit diagram of a measuring circuit of an embodiment of the invention; FIG. 2 is a circuit diagram of a current measuring circuit; Circuit diagram of the measuring circuit; FIG. 4 is a diagram showing an excitation signal that can be applied to electrochemical analysis of a suitable measuring unit; FIG. 5 is a diagram showing an idealized evolution of current of the electrochemical glucose measuring unit with respect to time; Figure 6 shows the impedance and frequency of the sensor for the 6-series blood glucose test; Figure 7 is the gain-frequency transfer characteristic of the op amp that has been stabilized by its manufacturer to prevent self-oscillation; Figure 8 is a plot of the phase shift of the amplifier versus frequency, The gain frequency response is shown in Figure 7; Figure 9 shows a portion of the circuit in Figure 1, but contains parasitic components; Figure 10 is a graph of the gain frequency characteristics of the circuit of Figure 1, with no compensation zeros in the response characteristics. Figure 11 is a diagram of the phase frequency response of the circuit showing the gain frequency response in Figure 10. 12 is a graph of the gain frequency response of an embodiment of the present invention; FIG. 13 is a graph showing the gain frequency response to the phase frequency of the embodiment shown in FIG. 12; FIG. 14 is a circuit diagram of the circuit, the first two stages of the circuit Can be used to introduce two poles and one zero, or all three stages can be used to introduce three poles and two zeros; Figure 15 is a diagram of a differential instrument loop that can be stabilized in accordance with the present invention; and Figure 16 is for Introduce a circuit diagram of another circuit of the pole zero pair.

1 is a circuit diagram of a measurement circuit consisting of a four-terminal sensor (generally indicated as 2) combined with a control circuit (indicated as 4) and a current measuring circuit (indicated as 8). The four terminal sensor includes a load 10 whose impedance varies as a function of the measured variable. Thus, for example, the load can be a unit for biological measurement whose impedance varies as a function of analyte concentration. The analyte can be, for example, blood glucose. The unit can be attached to the substrate and connected to terminals on the substrate such that unit 10 can be electrically energized and monitor the current flowing through the unit. As part of this measurement, it may be desirable to know the voltage across cell 10 and the current through it very accurately. The connection to unit 10 and from unit 10 can be subject to manufacturing variations and can exhibit impedance, and the impedance does change, which will affect the accuracy of the voltage measurement. To overcome this impedance problem, the unit can be provided as part of a four-terminal sensor. The four terminal sensor includes a first drive terminal 20 that is theoretically coupled to one end of the unit and a second drive terminal 22 that is theoretically coupled to the opposite end of the unit. The impedance represented by resistor 24 may exist between the first drive terminal 20 and the first end of unit 10. This first impedance 24 may be intentional, or it may be merely a function of the properties of the unit 10 and the connections with which it is made, and thus may be considered a parasitic component. Similarly, the second resistor 26 can be present in the path between the second side of the unit 10 and the second drive terminal 22. The four terminal sensor overcomes the problems of the resistors 24 and 26 by having first and second measurement terminals 30 and 32 connected to the first and second ends of the unit 10, respectively. These connections are also Intentional or parasitic impedances as represented by resistors 34 and 36, respectively, may be exhibited. Although the term "terminal" has been used herein, it should be understood that it can be replaced by the term "node."

If there is no current or substantially no current in the measurement circuit connected to the first and second measurement terminals 30 and 32, the cell output voltages appearing at the first and second measurement terminals 30 and 32 will be accurately represented. The voltage difference across cell 10. In fact, such conditions can be achieved by an operational amplifier using a high impedance front end. Such high impedance front ends typically use insulated gate field effect transistors as input devices. As a result, such a circuit does not substantially draw current from the measurement terminals.

Control circuit 6 has been schematically represented as an operational amplifier. This is basically correct because, although it has first to fourth inputs 41 to 44, its role in the closed loop shown in Figure 1 is to drive the voltage at its output node 50 to appear at input 41. The sum of the voltage difference between the voltage and the voltage appearing at input 42 and the voltage difference between the signal appearing at signal input 44 and the signal appearing at reference voltage input 43 is minimized. Each of these differences can be formed by an operational amplifier (ie, the difference between the signals at inputs 41 and 42 and the difference between the signals at inputs 43 and 44), and then each of these differences can be Used as an input to another op amp.

It should be understood that there is a small voltage difference between the values of the first input 41 and the second input 42 and between the input 43 and the voltage at the input 44, but the magnitude of this voltage difference is dependent on the gain of the control circuit 6. Broadly speaking, the magnitude of the voltage difference decreases proportionally as the gain of the control circuit 6 increases. Thus, the high gain within the control circuit 6 causes the voltage difference across the cell 10 to be controlled such that it exactly matches the voltage difference generated by the reference voltage 52 and supplied to the reference inputs 42 and 43 of the control circuit 6. The effect of the input offset has been ignored and it is assumed that appropriate techniques, such as auto-zeroing, will be used to reduce these sources of error.

In order to control the voltage across the unit 10, current must flow through the unit, for example from the first drive terminal 20 to the second drive terminal 22. As part of the measurement of the biological material in response to the unit, it is necessary to know the magnitude of the current through the unit. To this end, provide current Measuring circuit 8. In the example shown in FIG. 1, the measurement circuit 8 has been positioned between the second drive terminal 22 and the small signal ground 60. However, current measurement circuit 8 may also be provided in a feedback loop between output node 50 of control circuit 6 and first drive node 20 of four terminal sensor 2. The technology is freely chosen to a certain extent by the current measurement technique or circuit that it considers to be the most convenient to implement.

The reference voltage 52 can be configured to generate a DC voltage pulse, in which case it is desirable to measure the evolution of the current with respect to time. However, for purposes of verification and calibration, it may also be desirable for the reference voltage 52 to generate a varying signal, such as an alternating sinusoid, and in this case it becomes desirable for the measurement circuit 8 to know the phase of the sinusoidal signal such that it can be measured The magnitude and phase change of the current, for example, to infer the complex impedance of unit 10. The complex impedance can be determined by comparing the magnitude and phase of the voltage difference between the first and second measurement terminals with the magnitude and phase of the current flowing through the sensor.

FIG. 2 schematically illustrates a first current measurement circuit including a sense resistor 70 disposed in series between a second drive node 22 and a small signal ground 60. The voltage appearing across the sense resistor 70 can be measured by an analog to digital converter 80. The analog digital converter 80 can be implemented in any suitable converter technology, such as sigma-delta, successive approximation or flash memory technology, depending on the speed and accuracy requirements required by the circuit designer.

3 shows a variation of FIG. 2 in which a current sense resistor is placed in a feedback loop of an operational amplifier 90 having an inverting input coupled to a second drive terminal 22 and a non-inverting input coupled to the small Signal ground 60. Such a configuration may be advantageous because it means that by virtue of the amplifier 90 forming the virtual ground, the voltage at the second drive terminal 22 remains substantially constant and the impedance of the sense resistor 70 can be selected to be given at the output of the amplifier 90. A larger output voltage range. Again, the output voltage can be digitized by analog to digital converter 80.

The load 10 can be, for example, an electrochemical strip of electronic measurements, a common example of which is a glucose strip. Figure 4 depicts an amperometric measurement protocol for such a strip. Ampere metering process of measurement, the time period T ₀ the DC voltage is applied to both ends of the bar and held constant until time T1. The difference between time T ₁ and T ₀ is substantially 1 second, and the magnitude of the voltage can be approximately 500 mV. During the measurement protocol, the current through the cell changes substantially as shown in FIG. Therefore, the current rapidly rises to the initial value I ₀ and decays to the value I ₁ . The shape of the curve is a Cottrellian curve (which follows the Cottrell equation), and its shape basically follows And change. The value of parameter K varies as a function of analyte concentration. However, the value of K can also be a function of other parameters. A common parameter is temperature, but it can also change in the presence of contaminants. In a more complex form of the Cottrell equation, the value of K varies as the square root of the diffusion coefficient of the measured substance, and is the diffusion coefficient, which is a function of temperature. Therefore, it is desirable to perform some calibration measurements before or after the primary test to infer the factors that can be used to modify the K value such that, for example, the glucose test becomes more accurate.

The source of such errors has been observed by the skilled artisan, such as the complex impedance of the measurement unit 10 to infer temperature and some interfering chemicals. Thus, for example, it has been observed that for a glucose measurement unit, the change in impedance versus frequency as shown in Figure 6 has a turning point generally indicated as 100. As is known to the skilled person, the position of the turning point can be used, for example, to derive a correction factor for temperature measurement. Therefore, measuring the impedance as a function of frequency enables the temperature of unit 10 to be inferred. It is expected that this method can be applied to many biosensors that respond to individual analytes.

For example, if you want to measure the temperature, you can think that by assembling a temperature sensor in the unit, the temperature measurement will be better, but this may not be as desirable as originally thought. First, the temperature sensor will almost certainly tend to measure the temperature of the substrate (on which the cells are formed) rather than the temperature of the cell. Therefore, when a biological sample (eg, blood) is introduced into the measurement unit, the temperature of the unit will be different from the temperature of the substrate, and an equalization time will be required during which the chemical (analyte) in the sample, such as glucose. Reactions can occur between the reagents used to test the chemicals in the unit. In addition, the formation of the temperature sensor will Additional processing steps are required, and the temperature sensor itself may suffer manufacturing errors, and thus may not actually improve temperature measurements, and thus estimates of relevant parameters (eg, diffusivity).

Typically, the complex impedance of a cell is measured by inducing, for example, a low voltage sine wave of the order of 15 mV across the cell at a range of frequencies (such as 1 kHz, 2 kHz, 10 kHz, and 20 kHz). This impedance can be used in a known manner to apply a correction factor to the temperature. However, this can cause its own measurement problems.

As mentioned earlier, the control circuit can be viewed as a pole-like operational amplifier. Electronic engineers often know that amplifiers connected in the feedback loop have the ability to enter self-sustaining oscillations. In addition, most engineers know that op amp manufacturers prevent self-oscillation by intentionally inserting low-frequency poles (ie, low-pass filters) into the amplifier to modify the gain and frequency response of the amplifier. Broadly speaking, and as depicted in Figure 7, the insertion of a single pole 110 in the frequency response characteristic reduces the gain (expressed in decibels) by -20 dB per decade. This introduces a phase shift of -90° as a function of frequency at a frequency above the pole 110 position, as depicted in FIG. Most electrical engineers leave the university with this kind of work "rule of thumb", that is, when the gain in the loop exceeds the unity gain, the amplifier or feedback loop will remain stable as long as the gain of the loop is only reduced by -20dB per decade. .

As a result, stability is ensured as long as there is only one pole below the unity gain frequency of the operational amplifier. However, in other words, it is difficult to ensure that there is only one pole.

Figure 9 illustrates in detail the four terminal sensor of Figure 1, but this time includes the effects of parasitic components, particularly parasitic capacitors 120 and 122. It can be seen that the parasitic capacitor 120 acts in conjunction with the resistor 34 to form a low pass filter whose break point will be determined by the resistance of the resistor 34 and the capacitance of the parasitic capacitor 120. This low pass filter places the other pole into the frequency response of the signal path before it is placed in the control circuit. Similarly, resistor 36 and parasitic capacitor 122 also form a low pass filter. For the sake of simplicity, it is assumed that these two poles are at the same frequency and can therefore be viewed as a single low pass filter. If it is not the pole, it can be regarded as forming a single In the case of a low-pass filter, usually one pole will be more cumbersome than the other pole, so as a preference, the measures described herein for restoring loop stability will be applied to the more troublesome one of the poles.

Figure 10 is a schematic illustration of the frequency response characteristics of the circuit shown in Figure 1, wherein since the main pole 110 is included by design to provide loop stability, the main pole 110 is present in the control circuit frequency response and due to the resistance The filter 34 and the parasitic capacitor 120 and the resistor 36 and the parasitic capacitor 122 are inadvertently formed between the filters, so there is another pole 140. If the parasitic pole 140 occurs at a frequency at which the open loop gain of the control circuit 6 does not fall below 0 dB (less than the unity gain), then the frequency response has a certain range (usually indicated by 150) that has reached unity gain. -40 dB per decade power. Similarly, the second pole will add an additional 90° phase shift in the frequency response, as shown in Figure 11, such that at unity gain frequencies, the phase shift in the control circuit is substantially 180°. Due to the negative feedback loop, this combined 180° phase shift gives a total phase shift of approximately 360° and thus places the circuit in a position where it can experience self-sustained oscillation. In general, engineers know that this problem can be solved by reducing the open-loop gain of the amplifier to a lower value, as indicated by curve 160 in Figure 10, so that the intercept point with the 0 dB line is Now -20 dB every decade. However, this technique reduces the open-loop gain while bringing stability, and thus increases the error voltage between the reference voltage and the measured voltage difference. In addition, the gain must decrease with frequency, so the gain at 2 kHz will be 6 dB less than the gain at 1 kHz, and the gain at 10 kHz will (by definition) be 20 dB less than the gain at 1 kHz. And the gain at 20 kHz will be 26 dB less than the gain at 1 kHz. The error between the target voltage difference and the reference voltage difference will increase accordingly. Therefore, it can be seen that such a method of introducing circuit stability has a great influence on the measurement accuracy.

The stability of the anti-self-sustained oscillation can be solved by, for example, introducing a zero point 170 in the frequency response below the unity gain frequency, such that the slope of the frequency response is modified from a frequency of -40 dB per decade below the zero point 170 to be higher than At a frequency of zero 170 and unity gain frequency The next -20 dB per decade. Figure 12 shows such a configuration. Figure 12 also shows another pole 172 that occurs at frequencies above the unity gain (0 dB) frequency, only to indicate that the poles here do not introduce self-sustained oscillations. Figure 13 shows a generalized phase curve for such a transfer function showing that the phase transition is substantially -180° below zero 170 and substantially rises to -90° between the position of zero 170 and pole 172 and above Return to -180° at the frequency of the pole 172.

It is worth noting that a gain change of -40 dB per decade at frequencies occurring at frequencies below zero 170 does not cause oscillation. This is counterintuitive for many electronics engineers. However, the Barkhausen stability criterion determines whether the circuit oscillates. The Barkhausen rule applies to linear circuits in the feedback loop. The criterion states that if A is the gain of the amplifying component in the forward path of the circuit and β(Jω) is the transfer function of the feedback path such that βA is the loop gain around the feedback loop of the circuit, then the circuit will only be below The steady state oscillation is maintained at the frequency: i) the loop gain is equal to the absolute value of the unity gain, ie |βA|=1; and ii) the phase shift around the loop is zero or an integer multiple of 2π.

The Barkhausen rule is a necessary condition for oscillation, but not a sufficient condition. Some circuits that meet this criterion will not oscillate. However, if the circuit does not meet this criterion, it will not oscillate. This confirmation: as shown in Figure 12, even a high gain at a relatively low frequency is accompanied by a phase shift of -180° in the forward path, plus an additional -180° phase shift due to the formation of a feedback loop ( Therefore, it is basically equivalent to 2π), and the high gain itself at a relatively low frequency will not create an oscillation condition.

The gain frequency characteristic of Figure 12 can be modified by introducing additional poles and zeros, but only when the frequency response exceeds the unity gain value and the frequency response is only reduced by 20 dB per decade. Therefore, the response towards the lower frequency end of the graph can depend on how many poles have been introduced into the frequency response characteristic and drop by -40 dB per decade, -60 dB per decade, -80 dB per decade or more. dB every ten times.

14 illustrates a circuit that provides three poles and two zeros in a frequency response. For purposes of illustration, the circuit is only depicted as a single-ended circuit that receives a first input 200 representing a negative amplitude of the unit output signal and representative of the reference signal. A second input 202 of magnitude. Each of these signals is input to the inverting input 214 of the first operational amplifier 216 via respective impedances 210 and 212. The first operational amplifier 216 has its non-inverting input coupled to the small signal ground, and the capacitor 220 is coupled between the output 222 of the first operational amplifier 216 and its inverting input 214. Those skilled in the art will recognize that the presence of capacitor 220 in the feedback loop of first operational amplifier 216 forms an integrator. The ideal integrator places the pole at 0 Hz. However, in view of the fact that the first operational amplifier 216 has a finite gain, the pole is actually positioned close to 0 Hz instead of actually at 0 Hz. Output node 224 can be provided to selectively allow operation of the circuit as described in the prior art.

However, in accordance with embodiments of the present invention, one or more poles are also provided along with associated zeros. The pole and zero pairs may be provided by a circuit block (generally indicated as 240), in the present example, two such blocks 240, 240a have been provided in series. However, the invention may be practiced by virtue of including only one circuit block 240, or indeed three or more such circuit blocks. Circuit block 240 includes another operational amplifier 250 with its non-inverting input coupled to the small signal ground. An input resistor 252 is provided between the inverting input of the other operational amplifier 250 and the circuit to which it is supplied, which in the present example is an integrator formed around the first operational amplifier 216. The feedback loop around the other operational amplifier 250 includes a capacitor 254 in series with a resistor 256. It can be seen that the impedance of capacitor 254 dominates at low frequencies, and thus the feedback loop behaves as an integrator. In this particular configuration, the other pole provided by circuit block 240 occurs substantially at 0 Hz. It can also be seen that as the frequency rises, the impedance of capacitor 254 decreases and begins to become less important than the impedance of the other resistor 256. In fact, the value of the capacitor C 254 of the capacitor ₂₅₄ and the resistor R ₂₅₆ of the resistor ₂₅₆ are Insert a zero below. It can additionally be observed by inspection that the gain of circuit block 240 is determined by the ratio of resistor 256 to resistor 252 at a frequency above the frequency at which zero is introduced in circuit block 240. Output node 260 is provided such that the signal picked up from this node 260 corresponds to output node 50 of FIG. If the zero formed by capacitor 254 and resistor 256 occurs at a unity gain frequency below control circuit 6, stability will be ensured.

Another block similar to the first block may also be provided (but similar portions are indicated by the addition of "a") to introduce a second pole zero pair. The zero introduced by another circuit block need not be located at the same frequency as the zero provided by the first block.

Figure 15 illustrates an additional variation of the circuit shown in Figure 1, in which two control circuits are provided. An upper control circuit (which includes additional pole zero compensation, as described above) receives a first reference voltage from the first reference voltage generator 270 and controls a voltage at the first measurement terminal 30 to match the first reference voltage generator The reference voltage provided by 270. The second control circuit 302 receives the second reference voltage from the second reference voltage generator 307 and controls the voltage at the second measurement terminal 32 to be equal to the voltage from the second reference voltage generator 307. Therefore, the upper and lower branches of the sensor are driven to their respective voltages in a double-ended manner. The current sense resistor 70 can be inserted into any of the control loops (because, by definition, the currents in each control loop must be the same, and the voltage appearing across the resistor 70 can be digitized by the differential input analog to digital converter 80.

Figure 16 is a circuit diagram of another circuit known to those skilled in the art that can be used to provide a pole zero pair. It includes an operational amplifier 320 having a non-inverting input as a signal input. Capacitor 322 is coupled between the output of operational amplifier 320 and the inverting input of operational amplifier 320. Resistor 324 connects the inverting input to the small signal ground.

It should be noted that the designer can also place the zero point in the transfer characteristic without forming the pole of the association. As known to the skilled person, such a zero can be performed as a high pass filter (active filter or passive filter).

Therefore, the frequency of the control circuit or each control circuit can be modified by appropriately inserting an additional pole (preferably as an integrator) and a zero point (preferably as a high-pass filter). It should, in order, be able to use (with a much higher loop gain than by simply using gain attenuation alone rather than introducing an additional pole-zero pair to reduce the open-loop gain of the control circuit to ensure stability).

The scope of the patent application presented here is written in a single subordinate format to suit the USPTO application. However, in the case of use in other regions where the scope of multiple subordinate patent applications is frequently used, except for those areas that are technically not feasible, the scope of subordinate patent applications will be deemed to be multiple times subject to all the aforementioned patents. Subordinate patent application scope.

2‧‧‧ four-terminal sensor

4‧‧‧Control circuit

6‧‧‧Control circuit

8‧‧‧ Current measurement circuit

10‧‧‧Load/Unit

20‧‧‧First drive terminal

22‧‧‧Second drive terminal

24‧‧‧Resistors

26‧‧‧second resistance

30‧‧‧First measuring terminal

32‧‧‧Second measurement terminal

34‧‧‧Resistors

36‧‧‧Resistors

41‧‧‧ first input

42‧‧‧second input

43‧‧‧ third input

44‧‧‧ fourth input

50‧‧‧ Output node

52‧‧‧reference voltage

60‧‧‧Small signal grounding

Claims (26)

A control circuit for use with a four-terminal sensor having first and second drive terminals and first and second measurement terminals, the control circuit being configured to drive the Waiting for at least one of the first and second driving terminals to sense a voltage difference between the first and second measuring terminals and controlling the excitation signal such that the first and second measuring terminals are between The voltage difference is within a target voltage range, and wherein the control circuit includes N poles in its transfer characteristic and N-1 zeros in its transfer characteristic such that when the loop gain drops to unity gain, a closure The phase shift around the loop is substantially not 2π radians or a multiple thereof, where N is greater than one. A control circuit as claimed in claim 1, wherein the or each zero point is positioned in an open loop return characteristic of the control circuit to exhibit a frequency greater than a unity gain. The control circuit of claim 1, wherein at least one of the poles is provided by an integrator. The control circuit of claim 3, wherein a pole and zero pair is provided by a series combination of a capacitor and a resistor in a feedback loop of one of the operational amplifiers. The control circuit of claim 1, further comprising at least one reference signal input for receiving a reference signal, and wherein the control circuit functions to cause the voltage difference between the first and second measurement terminals Basically matching the at least one reference signal. The control circuit of claim 5, further comprising a signal generator for generating the reference signal at a plurality of frequencies. The control circuit of claim 1, further comprising a current sensor for measuring a current flowing through the four-terminal sensor. The control circuit of claim 6, further comprising a current sensor for measuring a current flowing through one of the four-terminal sensors and detecting a voltage of the four-terminal sensor and flowing through the sensing A phase detector of the phase difference between the currents of the devices. The control circuit of claim 1 is coupled to a four-terminal sensor having an impedance that varies in response to a measured variable. The control circuit of claim 9, wherein the measured variable is a biological sample. The control circuit of claim 9, wherein the measured variable is a blood glucose level. A control circuit according to claim 6 in combination with a four-terminal sensor for measuring biological sample parameters, wherein a plurality of impedance measurements are performed at a plurality of frequencies to determine a correction factor for biometric measurement. The control circuit of claim 12, wherein the correction is applied to an attenuation constant of one of the cottrellian curves. The control circuit of claim 10, wherein the control circuit is operated to maintain a DC voltage between the first and second measurement terminals during a biological parameter test of a biological sample. The control circuit of claim 9, wherein the four-terminal sensor is driven in a single-ended manner. The control circuit of claim 9, wherein the four-terminal sensor is driven in a double-ended manner. A method of operating an instrument circuit including a multi-terminal sensor and an excitation circuit, wherein the multi-terminal sensor has at least one driving terminal different from a first sensing terminal, and wherein the multi-terminal sensor Having another terminal, and wherein the excitation circuit is configured to measure a voltage difference between the first sense terminal and the other terminal and use the voltage difference to control an excitation signal applied to one of the at least one drive terminal Maintaining the voltage difference between the first sensing terminal and the other terminal to a target value or maintaining within a target range, and wherein the excitation current The path is configured to have at least one zero in its transfer characteristic such that it satisfies the Barkhausen stability criterion. The method of claim 17, wherein the excitation circuit has N poles and N-1 zeros in its transfer characteristic. The method of claim 18, wherein the pole is provided as an integrator. The method of claim 17, further comprising providing an AC component to the excitation signal at a plurality of frequencies or providing the excitation signal as the AC component, and measuring the complex of the multi-terminal sensor at a plurality of frequencies impedance. The method of claim 20, further comprising using a measure of the complex impedance to compensate for parameters of the multi-terminal sensor. The method of claim 21, wherein the sensor comprises a chemical or biosensing unit, and the compensated parameter is a temperature within the sensing unit. The method of claim 17, wherein the N-1 zeros occur at a frequency at which the control loop has a gain greater than one of the unity gains. The method of claim 17, wherein the N-1 zeros occur at a frequency at which the control loop has a gain greater than one of 0.5. The method of claim 17, wherein the N poles occur at a frequency at which the control loop has a gain greater than one of the unity gains. The method of claim 17, wherein the sensor is a four-terminal glucose sensor.