TWI383568B - High efficiency step-up power converters - Google Patents

Description

High efficiency boost power converter

The invention relates to a power converter, in particular to a DC-to-DC high efficiency boost power converter.

As shown in FIG. 1, a conventional boosting device includes a coupling circuit 10, a switch 13, a first diode 121, a second diode 122, and an output. The diode 123, a first capacitor 141, a second capacitor 142, and an output capacitor 143.

The coupling circuit 10 includes a first winding 11 and a second winding 12. And each of the windings 11, 12 has a polarity point end and a non-polar point end. The polarity point end of the first winding 11 is electrically connected to the external power source.

The first diode 121 includes an anode electrically connected to a non-polar point end of the first winding 11, and a cathode.

The second diode 122 includes an anode electrically connected to the cathode of the first diode 121, and a cathode electrically connected to the non-polar point end of the second winding 12.

The output diode 123 includes an anode electrically connected to a non-polar point end of the second winding 12, and a cathode.

The first capacitor 141 is electrically connected between the cathode of the first diode 121 and the ground.

The second capacitor 142 is electrically connected between the non-polar point end of the first winding 11 and the polarity point end of the second winding 12.

The output capacitor 143 is electrically connected to the cathode of the output diode 123 and Between the ground.

The switch 13 is electrically connected between the non-polar point end of the first winding 11 and the ground, and is switchable between a conductive state and a non-conductive state.

By the switching of the switch 13, the boosted output voltage can be obtained at both ends of the output capacitor 143. For details of the operation of the circuit, refer to this patent, and no further details are provided herein.

A disadvantage of the conventional boosting device is that when the switch 13 is turned on, the current of the second winding 12 flows through the switch 13, and the smaller the turns ratio of the first winding 11 and the second winding 12, the current flow of the second winding 12 The conduction loss caused by the switch will be closer to the conduction loss generated by the first winding 11, and the switching procedure and the switching loss must be increased once, resulting in a decrease in energy conversion efficiency.

Accordingly, it is an object of the present invention to provide a high efficiency boost power converter that provides high conversion efficiency and avoids the above-mentioned conventional deficiencies.

The power converter is adapted to boost a DC input voltage of an external power source into a DC output voltage, and includes: a coupling circuit including a first winding and a second winding, and each winding has a first a first end of the first winding receives the input voltage; a switch electrically connected between the second end of the first winding and the ground, and switchable between a conductive state and a non-conductive state An output diode comprising an anode electrically connected to the second end of the second winding and a cathode; An output capacitor electrically connected between the cathode of the output diode and the ground, and the voltage across the output is the output voltage; a clamp capacitor includes a first end electrically connected to the first end of the second winding and a grounded second end; a switching circuit, the first end of the clamping capacitor is switchably electrically connected to the second end of the first winding, or the first end of the clamping capacitor is switchably electrically connected To the anode of the output diode.

The utility model has the advantages of distinguishing the low current side high current and the high voltage side low current characteristic under the non-isolated structure, and the low voltage side circuit adopts the low voltage low conduction loss component, and all components have flexible switching to achieve high conversion efficiency.

The above and other technical contents, features and advantages of the present invention will be apparent from the following detailed description of the preferred embodiments.

As shown in FIG. 2, the high efficiency step-up power converter of the present embodiment is adapted to boost a DC input voltage V _{IN of} an external power source into a DC output voltage V _O , and includes: a coupling circuit 2 and a switch Q , a boost capacitor C ₂ , an output capacitor C _H , an output diode D _H , a clamp capacitor C ₁ , and a switching circuit 3 .

The coupling circuit 2 includes two windings wound around a core (not shown), which are a first winding L ₁ and a second winding L ₂ , respectively, wherein the turns ratio is 1:N. And each winding has a first end (which is a polarity point end) and a second end (which is a non-polar point end).

The switch Q has a electrically connected to one end of the first winding L nonpolar point end of _1, a is electrically connected to the other end of the ground, and a receiver control terminal external control signal, and based on the external control signal, the switch Q It can be switched between the on state and the non-conduction state.

The boosting capacitor C ₂ includes a first end electrically connected to the non-polar point end of the second winding L ₂ and a second end.

The output diode D _H includes an anode electrically connected to the second end of the boost capacitor C ₂ and a cathode.

The output capacitor C _{H is} electrically coupled between the cathode of the output diode D _H and ground.

The clamp capacitor C _{1 is} electrically connected between the polarity end of the second winding L ₂ and the ground.

The switching circuit 3 further switchably electrically connects the first end of the clamping capacitor C ₁ to the non-polar point end of the first winding L ₁ or switchably electrically connects the first end of the clamping capacitor C ₁ To the anode of the output diode D _H , and the switching circuit 3 includes: a first diode D _C and a second diode D _M . The first diode D _C has an anode electrically connected to the non-polar point end of the first winding L ₁ and a cathode electrically connected to the first end of the clamping capacitor C ₁ . The second diode D _M has an anode electrically connected to the first end of the clamp capacitor C ₁ and a cathode electrically connected to the anode of the output diode D _H .

Referring to FIG. 3, according to the switching of the switch Q , the high efficiency boost power converter will operate in six modes, and each mode will be described below. And the v _QG parameter in FIGS. _3-9 represents the voltage of the control terminal of the switch Q , the i _Lm parameter represents the excitation current of the coupling circuit 2, and v _Lm and v _L2 represent the voltage of the first winding L ₁ and the second winding, respectively. The voltage of L ₂ , i _{L 1} , i _{L 2} represents the current flowing through the first winding L _{1 and} the current flowing through the second winding L ₂ , respectively, and the i _Q and V _Q parameters respectively represent the flow through the switch Q. The current, the voltage across the switch Q , the i _DC and V _DC parameters represent the current flowing through the first diode D _C , the voltage across the first diode D _C , and the i _DM and V _DM parameters respectively represent a second diode current flows through D _M, D _M of the voltage across the second diode, i _DH, V _DH parameters represent the current flows through the output diode D _H, the output of the diode D _H The voltage at both ends.

Mode 1 (time: t ₀ ~ t ₁ ): Referring to Figure 4, in this mode, the switch Q has been turned on for a while and the second diode D _{M is} turned on, and the remaining diodes are not turned on. And in Figure 4, the direction of the current path is shown in this mode. The L _{k 1} and L _{k 2} parameters in FIGS. 4 to 9 respectively represent the leakage inductances of the first winding L ₁ and the second winding L ₂ , and L _m is the magnetizing inductance (also called mutual inductance), and the following is turned on for convenience of explanation. The diode is blacked out and is indicated by a dashed line when the switch Q is not conducting, and the turn-on voltage of the diode is ignored.

The boosting capacitor C2 is charged: an external power source (voltage is V _IN ) supplies current through the switch Q to energize the first winding L ₁ to generate an induced voltage v _LM = V _IN (1), and according to the number of turns The ratio induces a voltage to the second winding L ₂ , so the induced voltage on the second winding L ₂ is v _{L 2} = NV _IN (2), and the voltages of all the windings are positive at the polarity point end.

At the same time, the second winding L ₂ (the induced voltage is equal to NV _IN ) charges the boosting capacitor C ₂ to NV _IN via the second diode D _M .

Mode 2 (time: t ₁ ~ t ₂ ): As shown in Figure 5, in mode 2, switch Q begins to conduct. The trend of the current path in this mode 2 is indicated in Figure 5, and in mode two, the second diode D _{M is} continuously turned on.

The parasitic capacitance of the switch Q is charged: at the time point t = t ₁ , the current i _{L 1 of the} first winding L ₁ first charges the parasitic capacitance of the switch Q.

Meanwhile, when the switch Q is non-conductive immediately, subject to leakage effects of energy release, the currents of the first winding L ₁ i _{L 1} Q gradually away from the switch, the clamp capacitor C flows through the first diode D _{C 1,} and the clamp capacitor C ₁ is an excellent high-frequency response of the high-capacity capacitor, thereby quickly guide the current flowing through the switch Q i of _Q to the clamp capacitor C _1, so that the voltage clamp capacitor C _{1 C 1} V It can be considered as a stable low chopping DC voltage to ensure the maximum value of the Q voltage of the switch ( v _Q ).

Q switch parameter D is turned on the duty cycle, the voltage clamp capacitor C ₁ v _{C 1} can be calculated as _{v C 1 = V IN / (} 1- D) = v Q (3)

The leakage inductance L _{k 2 of the} second winding L2 stores energy for release: at this time, the second diode D _M continues to be turned on to release the leakage inductance L _{K 2 of} the second winding L ₂ to the boosting capacitor C ₂ .

Mode 3 (time: t ₂ ~ t ₃ ): As shown in Figure 6, in this mode three, the switch Q continues to be non-conducting, and the trend of the current path in this mode is indicated in Figure 6, and only The first diode D _{C is} turned on.

Capacitor C ₁ is charged: when the voltage v _Q across the switch Q is higher than the voltage v _{C 1 of} the clamp capacitor C ₁ , the first diode D _{C is} turned on, causing leakage of the first winding L ₁ The sense L _{K 1 is} released to the clamp capacitor C ₁ for charging.

A current I _L2 of the second winding L2 steering: when releasing the second winding leakage inductance L _{2 K 2} L of the energy, the second winding L _{2 L} of the current i ₂ at time point T ₂ is reduced to zero, after which the first winding L _{a. 1} of the exciting current i _Lm release of energy, and to the second induction coil L _2, a voltage of all windings at a point non-polar positive, the current of the second winding L i _{2 L 2} turned to the slow and Rising out of the non-polar point.

A portion of the current i _{L 2} of the second winding L ₂ is supplied to the second diode D _M as a reverse recovery current required for non-conduction, and a reverse bias voltage v _{DM of} the second diode D _M is established. This voltage forces the reverse voltage v _{DH of} the output diode D _{H to} gradually release to zero. The output diode D _H D _M and the sum of the currents of the second diode current equal to the second winding L i _{2 L} of _2, while the second winding leakage inductance L _{2 K 2} L of the current change rate limit The second winding L _{2 has} a high number of turns and a low current characteristic, so the reverse recovery current and the forward conduction current are small.

In addition, since the second diode D _M and the output diode D _{H are} connected in series between the clamp capacitor C ₁ and the output capacitor C _H , the second diode D _M and the an output diode D _H and the voltage, output capacitor C _H is equal to the output voltage V _O of the clamp capacitor C deducted voltage ₁ v _{C 1,} thus having the effect of clamping the voltage, and its output is lower than the desired breakdown voltage specification Voltage.

Mode 4 (time: t ₃ ~ t ₄ ): As shown in Figure 7, in this mode, the switch Q continues to be non-conducting. And in Figure 7, the direction of the current path is indicated, and in mode four, only the output diode D _{H is} turned on.

Energy transfer to the output

When the voltage of the parasitic capacitance of the output diode D _H is released to zero ( t = t ₃ ), the output diode D _H starts to conduct, and the second diode D _M does not conduct. At this time, the voltage v _{Lm of the} first winding L ₁ (positive at the non-polar point end) is equal to v _Lm = D V _IN /(1- D ) (4)

Therefore, the voltage v _{L 2 of the} second winding L ₂ can be converted into v _{L 2} = ND V _IN /(1- D ) (5)

During this mode, the clamp voltage of the capacitor C ₁ v _{C 1,} L ₂ of the second coil winding voltage V _{L 2,} C _2, and the boost capacitor voltage V _{C 2} is formed of a series path, a low current The pattern is released to the output capacitor C _H , so the output voltage V _O is obtained as V _O = v _{C 1} + v _{C 2} + v _{L 2} = (1+ N ) V _IN /1- D (6)

Therefore, the voltage gain G _{V 1} G _{V 1} = V _O / V _IN = (1 + N ) / 1- D (7)

Mode 5 (time: t ₄ ~ t ₅ ): As shown in Figure 8, in this mode, the switch Q begins to conduct. And in Figure 8, the direction of the current path is indicated, and in mode five, only the output diode D _{H is} turned on.

When the switch Q is turned on ( t = t ₄ ), since the first diode D _C is a low-voltage Schottky diode having no reverse recovery current, it is immediately reverse biased. Moreover, since the leakage inductance L _{k 1 of} the first winding L ₁ limits the rising rate of the current i _{L 1} and there is no reverse recovery current when the first diode D _C reverses, the switch Q cannot be from the first winding L. ₁ and the first diode D _C two paths take any current, naturally forming Zero Current Switching (ZCS), so the architecture has flexible switching characteristics when turned on, which can reduce the switching loss.

Since the current i _{L 2 of} the second winding L ₂ must continuously release the energy of the leakage inductance L _{k 2} , the current path is still the same as that of the mode four.

Mode six (time: t ₅ - t ₀ ): As shown in Fig. 9, in this mode six, the switch Q is continuously turned on. And in Figure 9, the direction of the current path is indicated, and in mode six, no diode is turned on.

Second winding current i ₂ L ₂ turned to the _L: L when the second winding leakage inductance of energy released after ₂ (t = t _5), all of the windings and the voltage polarity is positive at the end point, the second winding L the current i _{2 L 2} turned to, provide a small reverse current required by the output diode recovery current D _H and D _M while the parasitic capacitance of the second diode discharge starts to guide the second two The polar body D _{M is} gradually turned on.

When the second diode D _{M is} turned on ( t = t ₀ ), a switching cycle is completed, and then the working mode returns to the mode one.

According to equations (3) and (7), the voltage across the switch Q can be obtained as v _Q = V _O /( N +1) (8)

According to equation (8), the voltage across the switch Q is related to the output voltage and the turns ratio, regardless of the input voltage.

Experimental results: As shown in Figures 10~13, when the input power is 12V battery and the voltage is boosted to the output voltage of 170V DC, and the output power is 150W, 320W, 620W and 920W, having a zero current switching effect of voltage and current waveforms of the switch Q, 2 is designed as a coupling circuit at low magnetizing inductance, so the current I _Q Q switch ripple ratio is not high, and the Q switch is turned on.

As shown in Figures 14 to 16, when the input voltage is 12V and the output voltage and power are 170V and 620W, respectively, the experimental waveforms of the components are shown.

Fig. 14 shows that the current i _{L 1 of} the first winding L ₁ has a characteristic of a large current on the low voltage side, and the current i _{L 2} of the second winding L ₂ is a characteristic of a low current on the high voltage side, and it is verified that the boosting capacitor C ₂ has a rise. Pressure function.

As shown in FIG. 15 and FIG. 16, respectively, the voltage of the second diode D _{M and} the voltage of the output diode D _H are clamped to each other, and are lower than the output voltage, so that the low-voltage low-conduction loss of the Xiaoji diode can be selected. body. From the experimental waveform observation, the reverse recovery current of the diode is fully suppressed, and the current paths of the high voltage side and the low voltage side are clearly distinguished, and the voltage clamping effect of all components is close to the theoretical analysis.

As shown in Fig. 17, the waveform affects the output voltage when the load current fluctuates.

As shown in FIGS. 18-21, the voltage and current waveforms of the switch Q are used to boost the input power supply having a high fluctuation range voltage to 170V and the output powers are 150W, 320W, 620W, and 920W, respectively.

Overall, the load is from light load to heavy load, the input voltage is from 12V to 40V, and the voltage clamp range of the low-voltage switch Q is maintained at about 10V. Therefore, it can be widely applied to various high-varying voltage DC power converters.

As shown in Fig. 22~23, when the device is outputting 460W, the input power is the transient effect of the switching between the battery and the fuel cell on the output voltage, wherein the v _H and i _H parameters respectively represent the output voltage and the output current.

As shown in FIG. 22, in order to switch from a 12V battery to a 33V fuel cell supply input voltage, FIG. 23 switches from a 33V fuel cell to a 12V battery supply input voltage. According to the experimental results of the two figures, the output voltage V _{H is} hardly affected.

As shown in Figure 24, boosting the 12V battery to the 170V section requires a higher conduction duty cycle. Under light load conditions, the 12V battery switch Q has a lower current ripple, and the switch Q conduction loss is better than the measured fuel cell. Low, so the maximum experimental conversion efficiency of the battery is close to 97% when the boost ratio exceeds 14 times, but in the heavy load power range, the conversion efficiency will decrease due to the increase of current and conduction loss.

As shown in Fig. 25, in order to provide the input voltage conversion efficiency of the fuel cell, in the heavy-duty output range, since the current flowing through the switch Q is small, the conversion efficiency of the fuel cell is significantly higher than that of the battery at the same 1 kW output. %.

In summary, the high efficiency boost power converter of the present invention has the following advantages:

(1) High boost ratio: The voltage gain is high by the equation (7) G _{V 1} = V _O / V _IN = (1+ N ) / 1- D , and since the coupling circuit 2 has dual energy transfer between the exciting current and the induced current, Therefore, the value of the magnetizing inductance and the number of winding turns can be reduced, and only a low turns ratio and a wide duty cycle control are required to obtain a high output voltage gain.

(B) Since the magnetizing inductance of a desired value can be reduced, the capacity of the core can be reduced to reduce production costs, and the coupling circuit of the first winding 2 L ₁ may also reduce the required number of turns, when a large current through the first winding At L ₁ , the copper loss caused by the skin effect can also be reduced.

(3) The clamp capacitor C ₁ can absorb the line inductance energy, making the wiring easy, and is beneficial to the industrial utilization, and the energy absorbed by the clamp capacitor C ₁ can be directly boosted, and there is no circulation problem.

(4) The voltage clamping function can be achieved by the switch Q and all the diodes, and there is no problem of the short-circuit current when the switch Q is turned on and the reverse high recovery current of the diode.

(5) High conversion efficiency: Under the non-isolated architecture, the high-voltage side high current, high-voltage side low-current characteristics, and low current chopping characteristics are rigorously selected, and low-cost high-efficiency components suitable for the voltage range can be selected respectively, and when the switch Q is turned on When the current of the second winding L ₂ does not flow through the switch Q , compared with the conventionally, there is a lower conduction loss, and the electric energy absorbed by the clamp capacitor C ₁ is directly transmitted to the output end, which does not require additional Switch programs and have lower switching losses.

(6) It is proved that the conversion efficiency is not directly related to the boost ratio, and it is related to the size of the duty cycle and whether the conduction current of the switch Q is a square wave, which overcomes the technical bottleneck that the higher the boost ratio is, the lower the efficiency is.

(7) The violently changing current can be limited to the low-voltage side circuit to facilitate the prevention and treatment of electromagnetic interference.

(8) According to the theoretical analysis and the actual verification, the voltage withstand by the switch Q is only related to the output voltage and the turns ratio of the coupling circuit 2. This feature is more suitable for the application of the power conversion device with a wide range of DC input voltage variation.

However, the above is only the preferred embodiment of the present invention, when not The scope of the invention is to be construed as being limited by the scope of the invention and the scope of the invention.

2‧‧‧Coupling circuit

3‧‧‧Switching circuit

Output capacitance C _H ‧‧‧

C ₁ ‧‧‧Clamping capacitor

C ₂ ‧‧‧Boost Capacitor

D _H ‧‧‧Output diode

D _C ‧‧‧First Diode

D _M ‧‧‧Secondary

Q ‧‧‧Switch

L ₁ ‧‧‧First winding

L ₂ ‧‧‧second winding

1 is a circuit diagram of a conventional boosting device; FIG. 2 is a circuit diagram of a preferred embodiment of the present invention; FIG. 3 is a timing chart of the preferred embodiment of the present invention; BRIEF DESCRIPTION OF THE DRAWINGS FIG. 5 is a circuit diagram of the preferred embodiment of the present invention, illustrating operation in mode 2; FIG. 6 is a circuit diagram of the preferred embodiment of the present invention, illustrating Figure 3 is a circuit diagram of the preferred embodiment of the present invention, illustrating operation in mode four; Figure 8 is a circuit diagram of the preferred embodiment of the present invention illustrating operation in mode five; Figure 9 is a circuit diagram of the preferred embodiment of the present invention, illustrating operation in mode six; Figure 10 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating that when the output power is 150 W, the switch Voltage and current waveforms; Figure 11 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating voltage and current waveforms of the switch when the output power is 320 W; Figure 12 is a preferred embodiment of the present invention. Experimental measurement chart, indicating when The voltage and current waveform of the switch when the output power is 620 W; FIG. 13 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating the voltage and current waveform of the switch when the output power is 920 W; FIG. The experimental measurement chart of the preferred embodiment of the present invention illustrates the current waveforms of the first winding and the second winding; FIG. 15 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating the second two poles FIG. 16 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating voltage and current waveforms of the output diode; FIG. 17 is an experimental amount of the preferred embodiment of the present invention. FIG. 18 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating voltage and current waveforms of the switch when the input voltage is 36V; FIG. 19 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating voltage and current waveforms of the switch when the input voltage is 33 V; FIG. 20 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating When the input voltage is 31V, the on FIG. 21 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating voltage and current waveforms of the switch when the input voltage is 29 V; FIG. 22 is a preferred embodiment of the present invention. The experimental measurement chart shows the output voltage when the 12V battery is switched to the input voltage of the 33V fuel cell. FIG. 23 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating that the 33V fuel cell is switched to the 12V battery supply input. Output voltage FIG. 24 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating the conversion efficiency of the battery; and FIG. 25 is an experimental measurement diagram of the preferred embodiment of the present invention, illustrating the conversion efficiency of the fuel cell. .

2‧‧‧Coupling circuit

3‧‧‧Switching circuit

Output capacitance C _H ‧‧‧

C ₁ ‧‧‧Clamping capacitor

C ₂ ‧‧‧Boost Capacitor

D _H ‧‧‧Output diode

D _C ‧‧‧First Diode

D _M ‧‧‧Secondary

Q ‧‧‧Switch

L ₁ ‧‧‧First winding

L ₂ ‧‧‧second winding

Claims (8)

A high efficiency step-up power converter is adapted to boost a DC input voltage of an external power source into a DC output voltage, and includes: a coupling circuit including a first winding, and a second winding, and each The winding has a first end and a second end, the first end of the first winding receives the input voltage; a switch electrically connected between the second end of the first winding and the ground, and is in a conducting state and Switching between non-conducting states; an output diode comprising an anode electrically connected to the second end of the second winding and a cathode; an output capacitor electrically connected between the cathode of the output diode and the ground, and The voltage across the capacitor is the output voltage; a clamp capacitor includes a first end electrically connected to the first end of the second winding and a grounded second end; a boost capacitor electrically connected to the second winding Between the second end and the anode of the output diode; and a switching circuit, the first end of the clamping capacitor is switchably electrically connected to the second end of the first winding, or the clamping capacitor is further The first end is switchably electrically connected to the output two The anode body. The high efficiency step-up power converter of claim 1, wherein the switching circuit comprises a first diode, the first diode having a second electrically connected to the first winding An anode at the end, and a cathode electrically connected to the first end of the clamp capacitor. High efficiency boost power conversion as described in item 1 of the patent application scope The switching circuit includes a second diode having an anode electrically connected to the first end of the clamp capacitor and a cathode electrically connected to the anode of the output diode . The high efficiency step-up power converter of claim 2, wherein the switching circuit further comprises a second diode, the second diode having a first electrical connection to the clamp capacitor The anode of the terminal, and a cathode electrically connected to the anode of the output diode. The high efficiency step-up power converter of claim 4, wherein the first diode is not conducting when the second diode is turned on. The high-efficiency step-up power converter of claim 5, wherein the second winding charges the boosting capacitor via the second diode when the second diode is turned on . The high-efficiency step-up power converter of claim 1, wherein the boosting capacitor also passes through the output diode when the second winding charges the output capacitor via the output diode The body charges the output capacitor. The high efficiency step-up power converter of claim 1, wherein the first end of the first winding is a polarity point end, and the second end of the first winding is a non-polar point end, and The first end of the second winding is a polarity point end, and the second end of the second winding is a non-polar point end.