1280754 九、發明說明： • 【發明所屬之技術領域】 ， 本發明係有關無線通信系統。更特別是，本發明係有 關接收器中多重無線通信服務處理方法及裝置。 【先前技術】 _ 軟體定義無線電(SDR)係為多重無線通信標準被支援 魏定義單元處理之方案。有了軟體定義無線電，單硬體 平台係不需更換硬體組件即可支援多重無線通信標準，而 被下載軟體可再配置硬體。以此法，無線傳送/接收單元可 被快速配置來支援最新發展無線通信標準及協定。 典型單模式蜂巢基地台及無線傳送/接收單元係包含一 外差式無線電接收器類比前端，一固定採樣速率類比數位 _ 轉換器(ADC)及連續數位處理單元。類比前端中，預期信號 係被瀘波且接著被向下轉換為固定中頻(IF)頻帶。類比數位 轉換器係以數位處理解調演算預期信號要求頻寬及其他因 子為基礎當場被選擇之固定採樣速率來操作。 目前，無線傳送/接收單元係被配置處理經由多頻道接 收之多服務。例如，無線傳送/接收單元可支援數位蜂巢系 統(DCS)及寬頻分碼多重存取(WCDMA)系統之通信。各服 務係經由無線傳送/接收單元中之對應接收器路徑被處理而 被分別輸入無線傳送/接收單元中之數據機以便處理。然 而，僅一服務於給定時間被支援於各接收器路徑中。 5 1280754 s 徑之多遽‘麵帶分聽料刊接收器路 2 Γ線接㈣㈣瞻務及/峨時=種 令頻且接於類__波及分顺下轉換為 接者以固疋採樣速率被分別轉換為數位樣本。 ^比數位轉_之採樣速耗為_接收器功率消耗 抬夕j °31常，類比數位轉換器及數據機中其他處理 鬼功率雜係與採樣速率成正比；較樣 較低採樣速率更多功率。 μ行而比 因此，先前技術無線傳送/接收單元需廣泛硬 援多服務及配置係翻期該無線傳送/接收單元電池 式0 【發明内容】 本發帰有關接收II巾多重絲通信服務處理方法及 裝置。依據本發明，—_上無線通信服務係被同時接收 及處理。該服務係經由不同載波頻帶被傳送，而該被接收 載波頻帶係被向下轉換為中頻帶。本地振盪器_頻率係 被设定使該多服務之被向下轉換中頻帶下降為單中頻帶。 替代實施例中，軟體定義無線電係使用一類比數位轉換器 及適應性選擇包含兩個或更多被接收於兩個或更多不同頻 帶之複數輸入信號類比數位轉換之採樣頻率及適應性選擇 該本地振盪器頻率來實施。各輸入信號係經由不同頻帶運 6 1280754 载不同服務。輸入信號係被同時接收。各服務係受到最小 信號雜訊及失真比(SINAD)測量。輸入信號係藉由以特定頻 率混合該輸入信號及多本地振盪器信號而被轉換為中頻帶 信號。本地振盪器頻率係被適應性選擇使該中頻帶某些程 度上彼此頻譜接近或重疊。該服務之信號雜訊及失真比係 以各複數頻譜重疊條件來測量。本地振盈器頻率及採樣頻 率接著被以該信號雜訊及失真比測量結果為基礎來調整。 該處理較佳被連續重複。 【實施方式】 此後’ ”無線傳送/接收單元"名詞係包含但不限於使用 者設備(UE)，行動台，固定或行動用戶單元 ，呼叫器，或 可操作於無線環境中之任何其他類型元件。此後，當被稱 為”基地台’’名詞者係包含但不限於B節點，位址控制器， 存取點(AP)或無線環境中之任何其他接介裝置。 本案的特徵可併入積體電路(1C)中或者可配置在包含 多個互連組件的電路中。 本發明係提供單接收器鏈結中支援同時接收多重無線 通信服務處理之方法及裝置。該硬體可藉由軟體來配置。 此後’本發明將參考數位蜂巢系統及寬頻分碼多重存取分 頻雙工(FDD)當作同時服務例來解釋。然而，應注意本發明 可應用至任何其他服務及任何數量同時服務。圖示中之數 值係被提供當作例證而非限制，而只要不背離本發明傳 授，任何其他數值均可被實施。 7 第一圖為依據本發明第一實施例之接收器100方塊 圖。第二A-D圖係為第一圖接收器100中各階處之信號頻 譜圖示。如第二A圖所示，雙工器102及循環器104係限 制輸入頻譜之頻帶及組合預期服務下鏈頻帶，而於低雜訊 放大器(LNA)l〇6之前最小化元件損失。只要低雜訊放大器 106具有充足增益(l〇_15dB)來最小化來自該接收器鏈結剩 餘者之第二階雜訊指數貢獻，則此亦建立主要包含低雜訊 放大為106加上低雜訊放大器1〇6之前任何損失之雜訊指 數之系統雜訊指數。雙工器1〇2係移除位於預期下鏈頻帶 間之任何中間上鏈頻帶(如第二A圖中之分頻雙工上鏈頻 ▼)’因而避免寬頻低雜訊放大器106之飽和。兩完全接收 頻帶中之任意頻道可被同時接收，而該服務選擇係軟體可 配置。 頻帶限制輸入頻譜係被低雜訊放大器1〇6放大且被第 一濾波器108濾波。被第一濾波器108濾波^复之輸入頻譜 係，顯示於第二B圖。該頻帶限制輸入頻譜係藉由具有一 =定本地振盪器1頻率之混合器被向下轉換為第一中頻頻 i第一中頻係再次被第二濾波器112濾波來移除鏡像頻 率及阻斷态；且接著藉由可變增益放大器(VGA)n4放大。 可變增盈放大器114所輸出之第一中頻頻譜係被顯示於第 二C圖。 、使用鏡像頻率轉換，第二向下轉換係藉由具有本地振 盪态2之混合器116指示。第二中頻頻譜係被顯示於第二d 圖。本地振盪器2頻率係被設定使該第二向下轉換促使該 1280754 夕服務下鏈頻帶被摺疊為如第二D圖所示之單第二中頻頻 見。數位蜂巢系統下鏈頻帶及寬頻分碼多重存取分頻雙工 下鏈頻帶係被摺疊為單第二中頻頻寬。此促成使用高Q濾 波裔以該第二中頻頻寬來表減頻外(〇ut-〇flband)阻斷器及 干擾裔。多本地振盪器頻率亦可被用來放置多服務下鏈頻 帶於被定義第二中頻頻寬内任何處。 第一圖之接收器100係執行兩向下轉換。然而，應注 意第一圖中之接收器100配置及稍後將被解釋之本發明其 他實施例僅為本發明較佳實施例，且一個或更多兩向下轉 換可被實施。當最小化第二中頻頻寬時，本地振盪器〗，2 係使用具有固定濾波器之適應性頻率方案被設定來摺疊接 收下鏈頻帶為第二中頻頻寬。 最終中頻"ί§號於猎由滤波器118，及可變增益放大器 120處理之後，係藉由類比數位轉換器124被進一步向下採 樣。藉由最小化第二中頻頻寬，類比數位轉換器124之採 樣速率可為適應性，因而最小化最終數位向下轉換為基帶 之功率消耗。 最終中頻頻寬係視接收器之信號及失真比測量而定。 信號雜訊及失真比測量係包含接收器處理頻寬内之失真產 物。通常僅一信號出現於此頻寬内而失真產物並不被產 生，所以僅需信號雜訊比(SNR)測量。因為接收器中出現多 信號，所以失真產物係被產生於該處理頻寬内且這些位準 於信號雜訊比測量中必須被解釋。依據本發明，當最高信 號雜訊及失真比被測量時，最小頻寬係被選擇，而相反地， 9 !28〇754 當最低信號雜訊及失真比被測量時，最大最終頻寬係被選 ' 擇。 第三圖為依據本發明第二實施例之接收器200方塊 圖。第四A-D圖係為第三圖接收器中各階處之信號頻譜圖 示。如第四Α圖所示，雙工器202及循環器204係限制輸 入頻譜之頻帶。頻帶限制輸入頻譜係被低雜訊放大器206 放大且被第一濾波器208滤波。被第一濾、波器2〇8濾、波後 鲁 之輸入頻譜係被顯示於第四B圖。 輸入信號接著藉由混合該輸入信號及本地振盪器1所 產生之信號而被向下轉換為中頻信號。第二實施例中，兩 下鏈頻帶係使用兩固定本地振盪器1頻率及兩固定本地振 盪器2頻率被轉換為最終中頻處之鄰近頻帶。各服務之輸 入信號係使用不同本地振盪器頻率而被向下轉換。此例 中，數位蜂巢系統下鏈頻帶係以本地振盪器1A及本地振盪 器2A頻率被向下轉換，而寬頻分碼多重存取系統係以本地 • 振盪器1B及本地振盪器2B頻率被向下轉換。 各服務之輸入信號頻帶限制係籍由分別具有本地振盪 器1A及本地振盪器1B之混合器210被向下轉換為第一中 頻頻寬’且再次被第二濾波器212濾、波來移除鏡像頻率及 藉由可變增益放大器214放大。可變增益放大器214所輸 出之第一中頻頻譜係被顯示於第四C圖。 第二向下轉換係藉由分別具有本地振盪器2A及本地 振盡器2B之之混合器216來指示。濾、波器218所輸出之第 二中頻頻譜係被顯示於第四D圖。本地振盪器1A，本地振 10 1280754 盡器IB，本地振盪器2A及本地振盪器2B頻率係被設定使 -該第二向下轉換促使該多服務下鏈頻帶被彼此鄰接放置於 • 如第四D圖所示之第二中頻頻寬中。此例中，數位蜂巢系 統下鏈頻帶及寬頻分碼多重存取分頻雙工下鏈頻帶係被轉 換為最終中頻頻帶中之鄰近頻帶。多本地振盪器頻率亦可 被用來放置多服務下鏈頻帶於被定義第二中頻頻寬内任何 處。最終中頻頻率係於藉由濾波器218，222及可變增益放 大器220處理之後被進一步向下採樣。藉由最小化第二中 頻頻寬，類比數位轉換器224之採樣速率可為適應性，因 而最小化最終數位向下轉換為基帶之功率消耗。 第五圖為依據本發明第三實施例之接收器300方塊 圖。第六A-D圖係為第五圖接收器300中各階處之信號頻 譜圖示。如第六A圖所示，雙工器302及循環器304係限 制輸入頻譜之頻帶。頻帶限制輸入頻譜係被低雜訊放大器 _ 306放大且被第一濾波器308濾波。被第一濾波器3〇8濾波 後之輸入頻譜係被顯示於第六B圖。 各服務之輸入信號頻帶限制係藉由分別具有本地振盪 器1A及本地振盪器1B之混合器31〇被向下轉換為第一中 頻頻寬，且再次被第二濾波器312濾波來移除影頻率及藉 由可變增益放大器314放大。可變增益放大器314所輸出 之第一中頻頻譜係被顯示於第六C圖。 苐二實施例中’來自下鍵頻帶之任何任意頻道均可使 用可配置本地振盪器2被向下轉換為射頻頻帶處之任意間 隔頻道。兩輸入信號之第二向下轉換係分別藉由具有本地 11 1280754 振盪器2A及本地振盪器況之混合器μα來指示。被濾波 器318濾波後之第二中頻頻譜係被顯示於第六D圖。如第 六D圖所示，本地振盪器2A及本地振盪器2β頻率係可為 調整，使該第二向下轉換可促使多服務下鏈頻帶被放置於 彼此相隔之第二中頻頻寬中。 、 可替代是’本地振盪器1A及本地振盪器2A係為可調 整，而本地振盪器2A及本地振盪器2B可為固定，或本地 振盪器兩者均為可調整。多本地振盪器頻率亦可被用來放 置多服務下鏈頻帶於被定義第二中頻頻寬内任何處。最終 中頻頻率係於藉由濾波器318，322及可變增益放大器32〇 處理之後被進一步向下採樣。藉由最小化第二中頻頻寬， 類比數位轉換器324之採樣速率可為適應性，因而最小化 最終數位向下轉換為基帶之功率消耗。 第七圖為依據本發明被用來實施適應頻率向下轉換之 接收器數據機中之查找表400方塊圖。預期服務，採樣頻 I及預期第二中頻係被當作對查找表400之輸入，而該查 找表400係輸出本地振盪器1及本地振盪器2頻率設定及 類比數位轉換器採樣頻率。查找表400係依據可得服務及 信號雜訊及失真比測量來最佳化頻率方案，採樣頻率及採 樣頻寬。查找表400可被用於本發明任何實施例中。 第八圖為依據本發明用於本地振盪器綜合頻率之本地 振盪器頻率綜合器500方塊圖。因為第二及第三實施例中 所示接收器係需多本地振盪器頻率，所以綜合器500必須 可產生這些頻率。本地振蘯器頻率綜合器500係包含一參 12 1280754 考振盪器502及一個或更多綜合器504。本地振盪器頻率綜 合器可選擇性進一步包含一個或更多絕緣器5〇6及一個或 更多循環器508。參考振盪器502可產生被輸入複數綜合器 504中之參考頻率。各綜合器504係依據查找表400所產生 之本地振盈益1及本地振靈器2頻率設定被調諧產生中頻 頻率。綜合器504所產生之中頻頻率係被傳送至混合器之 本地振盪器埠以向下轉換輸入信號。 _ 循環器508於可最小化綜合器功率消耗之低損失組合 方案中係較佳被用來組合兩綜合器之本地振盪器頻率。絕 緣器506係被提供於各綜合器5〇4輸出處以提供充分反向 絕緣來消除因其他綜合器所招引任一綜合器之頻率。可替 代疋，綜合器504中之緩衝放大器可被用來提供絕緣。此 促使綜合器方法藉由移除絕緣器5〇6而被進一步放大。 第九圖為依據本發明同時處理接收器中多重無線通信 鲁 服務處理6⑽之流賴。—偏±職係_絲介面被 同時接收(步驟5〇2)。各服務係經由不同載波頻帶來傳送。 被接收載波頻帶係使用本地振盪器被向下轉換為中頻頻 ▼，使该被向下轉換頻帶座落於單中頻頻帶中(步驟5〇4)。 替代實施例巾’軟體定義無線電係使㈣個或更多加 總本地振盡器同時接收兩個或更多服務及/或頻道來獨立控 制該兩個或更多服務及/或頻道之最終中頻及適應性選擇該 兩倾好本地㈣H解及採樣頻率。依據本發明此實 W狀軟體福無線電係適應性最小化該採樣頻率，因而 降低_數轉換1之辨雜聽據财之處理塊並增 13 1280754 雔 蕈 加總合電池壽命。本發明此實施例可被實施於基地台及無 線傳送/接收單元。 第十圖為依據本發明適應性選擇本地振盪器頻率及採 樣頻率用於被同時接收之複數輸入信號之類比數位轉換之 接收斋600方塊圖。接收器600係包含一天線602，一低雜 訊放大器604，一混合器606，兩本地振盪器608a及60813, 一相加器618，一類比數位轉換器610，一數位中頻處理單 元612 ’ 一基帶處理單元614，及一控制器616。兩個或更 多輸入信號係被天線602同時偵測用於兩個或更多服務及/ 或頻道。各服務及/或頻道係經由不同載波頻帶被傳送並符 合特定信號干擾雜訊及失真比(SINAD)的需求 <低雜訊放大 器604係放大該被接收輸入信號。 各本地振盪器608a，608b係產生各服務及/或頻道之對 應頻率本地振盪器信號。第十圖僅描繪兩本地振盪器當作 例證’但兩個以上本地振盪器亦可被用來放置多服務及/或 頻道下鏈頻帶於最終中頻頻寬内。本地振盪器信號頻率係 被控制器616控制。本地振盈器信號係被相加器618加總 並被轉送至混合器006 〇 混合器606係混合輸入信號及本地振盪器信號來轉換 各射頻輸入信號至中頻信號。僅一混合階被描繪於第一 圖。然而，應注意一個以上混合階可被實施來轉換各射頻 輸入信號至最終中頻信號。最終中頻頻帶係被選擇使服務 及/或頻道之中頻頻帶某些程度上彼此頻譜接近或重疊。頻 譜重疊可於接收器内產生對一個或兩者頻帶及/或頻道之干 14 1280754 r * 擾。 第十一 A-F圖為依據本發明描繪射頻輸入信號至最終 中頻帶之頻率轉換之中頻頻譜方塊圖。第十一 A-F圖中陰 ‘影區域係代表預期頻率頻道。 本地振盪裔頻率係被調整使向下轉換促使輸入信號被 轉換為如第十-A-F圖所示某些程度上彼此最終中頻帶接 近或重$。第十- A圖中’服務之中頻帶係彼此接近但不 籲重疊。因此，一頻帶至另一頻帶並不產生干擾。第十一B 圖中’兩中頻帶僅於非預期頻率頻道中彼此重疊。第十一 c 及D圖中’一預期頻道係獲得干擾器，而第十-E及F圖 中’兩者預細道皆獲得干擾器。第十一 F圖中，一服務 及/或頻道全部中頻帶均被與其他中頻帶重疊。 ▲，了避免中頻▼任何區域疊頻，採樣頻率應被設定為 高於最高中頻帶之最_率組成至少兩倍之值。採樣頻率 • 可低於該值’其中不位於預期頻道内之中頻帶區域疊頻係 可接受。因此’採樣頻率係藉由被同時處理之複數服務及/ 錢棚料最高頻雜紅服觀/或舰來紗。可避 ^預^頻道疊頻之最小採樣頻率-半係藉由第十-A-F圖 頭&示。可避免預期頻帶疊頻之最小所需採樣頻率 半係藉由第A-F圖中虛箭頭標示。若因預期頻道上 =成疊頻所造成之信號雜訊及失真比可容忍，則‘ 樣頻率可甚至低於虛箭頭所示者。 韓蓋程度從第十-A圖增加至第十—F圖時，採樣 僻下降但職頻道奴干擾增加。因此，覆蓋情況及採 Ϊ280754 樣頻率應被選擇考慮採樣頻率及干擾。 最終中頻帶處之^1選擇中頻頻寬及覆蓋情況係被適應 性調整為同時被測量預期服務及/或頻道之信號雜訊及失真 味函數。各服務及/或頻道係具有必須被滿足之最小信號雜 δί1及失真比準則。回去參考第十圖，基帶處理單元614係 測里各種覆蓋情況下之信號雜訊及失真比，而控制器616 係選擇具有滿足最小信號雜訊及失真比準則之最低採樣頻 率當作最佳採樣頻率之覆蓋情況。 \類比數位轉換器610係以控制器616所設定之採樣頻 率轉換中頻帶信號為數位信號。數位中頻處理單元及 土 f處理單元614係處理服務之數位信號。數位中頻處理 單元=12係執行從中頻至基帶之最終頻率轉換。數位中頻 處理單元612係分隔彼此之服務。 …藉由適應性控制服務及/或頻道之祕中頻帶，採樣頻 率，適雜最小化。最小化採樣鮮_鋪比數位轉 換器及數據機中處理塊之功率消耗，並增加總合電池壽命。 頻道狀況(如與包元之距離，鄰近頻道之改變等)係隨時 間改變。覆蓋情況及最佳採樣鮮之選擇細若干速率被 f平估。因為無線舰/接收單元並不知獅賴道出現 消失，且其可以較上述再評估預期為快之速率改變，所1 ϋ可接受之連接突然降級，最佳採樣頻率之頻= 重受及&擇枚储_為非連料閒置獅僅: 貧料被接收之週期。突瓣級不可接受之週期期間，姑^ 器係不需賴重疊而以支援此情況之最高採樣頻率1喿作收 16 1280754 無論職航及最絲樣_之_為何，該採樣頻 率均可猎由畜意引進不職之頻帶疊頻而被進—步降低。 第十二_依據本發明·顏性接收器中複數 輸入信號類比數位轉換之採樣鮮之處理驗流程圖。接 收器係同時接收兩個或更多服務及/或頻道之兩個或更多輸 入信號(步驟8〇2)。各服務及/或頻道係受到信號干擾及失真 比測量。輸入信號係藉由混合輸入信號及本地振盪器信鐃 而被轉換為中頻帶信號(步驟8〇4) 〇本地縫器頻率係被調 整使輸入信號之被轉換中頻帶信號某些程度上彼此頻譜接 近或重疊。該服務及/或頻道之信號雜訊及失真比係以各複 數頻譜重疊條件來測量(步驟806)。用於中頻信號類比數位 轉換之本地振盪器頻率及採樣頻率係以信號干擾雜訊及失 真比測量結果為基礎(步驟808)。步驟806及808係較佳被 定期或非定期重複。 雖然本發明之特性及元件被以特定組合說明於較佳實 施例中，但各特性及元件可不需較佳實施例之其他特性及 元件而被單獨使用，或有或無本發明其他特性及元件之各 種組合中。 17 1280754 【圖式簡單說明】 本發明可從以下實施例說明及附圖獲得更詳細了解，其 中： 第一圖為依據本發明第一實施例之接收器方塊圖; 第二A-D圖係為第一圖接收器中各階處之信號頻譜圖 示； 第二圖為依據本發明第二實施例之接收器方塊圖；1280754 IX. Description of the invention: • [Technical field to which the invention pertains] The present invention relates to a wireless communication system. More particularly, the present invention relates to a method and apparatus for processing multiple wireless communication services in a receiver. [Prior Art] _ Software Defined Radio (SDR) is a scheme that supports the definition of unit processing for multiple wireless communication standards. With a software-defined radio, a single-hardware platform supports multiple wireless communication standards without the need to replace hardware components, while the downloaded software can be reconfigured with hardware. In this way, the WTRU can be quickly configured to support the latest developments in wireless communication standards and protocols. A typical single mode cellular base station and wireless transmit/receive unit includes a heterodyne radio receiver analog front end, a fixed sample rate analog digital converter (ADC) and a continuous digital processing unit. In the analog front end, the expected signal is chopped and then downconverted to a fixed intermediate frequency (IF) band. The analog-to-digital converter operates with digital processing of the demodulation calculation expected signal required bandwidth and other factors based on the selected fixed sampling rate of the field. Currently, wireless transmit/receive units are configured to handle multiple services received via multiple channels. For example, the WTRU can support communication between a digital cellular system (DCS) and a wideband code division multiple access (WCDMA) system. Each service is processed via a corresponding receiver path in the WTRU and is separately input to the modem in the WTRU for processing. However, only one service is supported in each receiver path at a given time. 5 1280754 s 之 之 遽 面 面 面 面 面 面 面 面 面 接收 接收 接收 接收 接收 接收 接收 接收 接收 接收 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 四 = = = = = = = The rate is converted to a digital sample, respectively. ^The ratio of the sampling rate is _the receiver power consumption is increased by j °31. The analog digital converter and other processing power in the data machine are proportional to the sampling rate; the lower sampling rate is more power. Compared with the above, the prior art wireless transmitting/receiving unit needs to extensively support multiple services and configuration systems to revoke the wireless transmitting/receiving unit battery type 0 [Abstract] The present invention relates to a receiving II towel multi-wire communication service processing method And equipment. According to the present invention, the wireless communication service is received and processed simultaneously. The service is transmitted via different carrier frequency bands, and the received carrier frequency band is downconverted to the intermediate frequency band. The local oscillator_frequency system is set such that the down-converted mid-band of the multi-service is reduced to a single mid-band. In an alternative embodiment, the software-defined radio system uses an analog-to-digital converter and adaptively selects two or more analog input signals that are received in two or more different frequency bands for sampling frequency and adaptive selection. The local oscillator frequency is implemented. Each input signal is transported via different frequency bands 6 1280754 for different services. The input signal is received simultaneously. Each service is measured by the minimum signal noise and distortion ratio (SINAD). The input signal is converted to an intermediate frequency band signal by mixing the input signal and the multi-local oscillator signal at a specific frequency. The local oscillator frequency is adaptively selected such that the mid-bands are spectrally close to or overlapping each other to some extent. The signal noise and distortion ratio of the service is measured by complex spectral overlap conditions. The local oscillator frequency and sampling frequency are then adjusted based on the signal noise and distortion ratio measurements. This process is preferably repeated continuously. [Embodiment] The following 'wireless transmitting/receiving unit' includes no, but not limited to, user equipment (UE), mobile station, fixed or mobile subscriber unit, pager, or any other type operable in a wireless environment. Components. Hereinafter, what is referred to as a "base station" term includes, but is not limited to, a Node B, an address controller, an access point (AP), or any other interface device in a wireless environment. Features of the present invention may be incorporated into the integrated circuit (1C) or may be configured in a circuit comprising a plurality of interconnected components. SUMMARY OF THE INVENTION The present invention provides a method and apparatus for supporting simultaneous reception of multiple wireless communication service processing in a single receiver chain. The hardware can be configured by software. Hereinafter, the present invention will be explained with reference to a digital cellular system and wideband code division multiple access frequency division duplex (FDD) as simultaneous service examples. However, it should be noted that the present invention is applicable to any other service and any number of simultaneous services. The figures are provided as illustrations and not as limitations, and any other values can be implemented without departing from the teachings of the invention. The first figure is a block diagram of a receiver 100 in accordance with a first embodiment of the present invention. The second A-D diagram is a signal spectrum diagram at each stage of the receiver 100 of the first diagram. As shown in Figure 2A, duplexer 102 and circulator 104 limit the frequency band of the input spectrum and combine the expected service downlink frequency bands to minimize component loss prior to low noise amplifier (LNA) l6. As long as the low noise amplifier 106 has sufficient gain (10 〇 15 dB) to minimize the second order noise index contribution from the remainder of the receiver chain, this also establishes a low noise amplification of 106 plus low. The system noise index of any loss of noise index before the noise amplifier 1〇6. The duplexer 1〇2 removes any intermediate uplink frequency bands (e.g., the frequency division duplex upper frequency chain ▼ in the second A picture) located between the expected downlink frequency bands and thus avoids saturation of the wide frequency low noise amplifier 106. Any of the two fully received frequency bands can be received simultaneously, and the service selection software is configurable. The band limited input spectrum is amplified by the low noise amplifier 1〇6 and filtered by the first filter 108. The input spectrum system filtered by the first filter 108 is shown in the second B picture. The band limited input spectrum is downconverted to a first intermediate frequency by a mixer having a frequency of a local oscillator 1 and the first intermediate frequency is again filtered by the second filter 112 to remove the image frequency and resistance. Off state; and then amplified by a variable gain amplifier (VGA) n4. The first intermediate frequency spectrum output by the variable gain amplifier 114 is shown in the second C-picture. Using image frequency conversion, the second down conversion is indicated by a mixer 116 having a local oscillator state 2. The second intermediate frequency spectrum is shown in the second d-picture. The local oscillator 2 frequency is set such that the second down-conversion causes the 1280754 service downlink frequency band to be folded into a single second intermediate frequency as shown in the second D-picture. The digital honeycomb system downlink frequency band and the wide-band code division multiple access frequency division duplex downlink frequency band are folded into a single second intermediate frequency bandwidth. This facilitates the use of the high Q filter to demodulate the 〇ut-〇flband blocker and interferers with the second IF bandwidth. The multi-local oscillator frequency can also be used to place the multi-service downlink frequency band anywhere within the defined second IF bandwidth. The receiver 100 of the first figure performs two down conversions. However, it should be noted that the receiver 100 configuration in the first figure and other embodiments of the invention to be explained later are merely preferred embodiments of the present invention, and one or more two down conversions may be implemented. When the second intermediate frequency bandwidth is minimized, the local oscillator, 2 is configured to fold the receive downlink frequency band to the second intermediate frequency bandwidth using an adaptive frequency scheme with a fixed filter. The final intermediate frequency " ί § is further processed down by the analog digital converter 124 after being processed by the filter 118 and the variable gain amplifier 120. By minimizing the second intermediate frequency bandwidth, the sampling rate of the analog digital converter 124 can be adaptive, thereby minimizing the power consumption of the final digital down conversion to baseband. The final IF bandwidth depends on the receiver's signal and distortion ratio measurements. Signal noise and distortion ratio measurements include distortion products within the receiver's processing bandwidth. Usually only one signal appears in this bandwidth and distortion products are not generated, so only signal to noise ratio (SNR) measurements are required. Because of the presence of multiple signals in the receiver, distortion products are generated within the processing bandwidth and these levels must be interpreted in signal noise ratio measurements. According to the present invention, when the highest signal noise and distortion ratio are measured, the minimum bandwidth is selected, and conversely, 9 ! 28 〇 754 when the lowest signal noise and distortion ratio are measured, the maximum final bandwidth is Choose 'choose. The third figure is a block diagram of a receiver 200 in accordance with a second embodiment of the present invention. The fourth A-D diagram is the signal spectrum representation at each stage in the receiver of the third diagram. As shown in the fourth diagram, duplexer 202 and circulator 204 limit the frequency band of the input spectrum. The band limited input spectrum is amplified by low noise amplifier 206 and filtered by first filter 208. The input spectrum of the first filter, the filter 2〇8, and the post-wave Lu are displayed in the fourth B diagram. The input signal is then downconverted to an intermediate frequency signal by mixing the input signal with the signal generated by local oscillator 1. In the second embodiment, the two downlink frequency bands are converted to the adjacent frequency bands at the final intermediate frequency using two fixed local oscillator 1 frequencies and two fixed local oscillator 2 frequencies. The input signals for each service are downconverted using different local oscillator frequencies. In this example, the downlink frequency band of the digital cellular system is down-converted with the local oscillator 1A and the local oscillator 2A frequency, and the wide-band code division multiple access system is directed to the local oscillator 1B and the local oscillator 2B frequency. Down conversion. The input signal band limitation of each service is down-converted to the first intermediate frequency bandwidth by the mixer 210 having the local oscillator 1A and the local oscillator 1B, respectively, and is again filtered by the second filter 212 to remove the wave. The image frequency is amplified by a variable gain amplifier 214. The first intermediate frequency spectrum output by the variable gain amplifier 214 is shown in the fourth C-picture. The second down conversion is indicated by a mixer 216 having a local oscillator 2A and a local oscillator 2B, respectively. The second intermediate frequency spectrum output by the filter and waver 218 is shown in the fourth D picture. The local oscillator 1A, the local oscillator 10 1280754, the local oscillator 2A and the local oscillator 2B frequency are set such that the second down-conversion causes the multi-service downlink frequency band to be placed adjacent to each other. The second intermediate frequency bandwidth shown in Figure D. In this example, the digital cellular downlink frequency band and the wideband coded multiple access frequency division duplex downlink frequency band are converted to adjacent frequency bands in the final intermediate frequency band. The multi-local oscillator frequency can also be used to place the multi-service downlink frequency band anywhere within the defined second intermediate frequency bandwidth. The final intermediate frequency is further downsampled after being processed by filters 218, 222 and variable gain amplifier 220. By minimizing the second intermediate frequency bandwidth, the sampling rate of the analog digital converter 224 can be adaptive, thereby minimizing the power consumption of the final digital down conversion to baseband. Figure 5 is a block diagram of a receiver 300 in accordance with a third embodiment of the present invention. The sixth A-D diagram is a signal spectrum diagram at each stage of the receiver 300 of the fifth diagram. As shown in Figure 6A, duplexer 302 and circulator 304 limit the frequency band of the input spectrum. The band limited input spectrum is amplified by the low noise amplifier _306 and filtered by the first filter 308. The input spectrum filtered by the first filter 3〇8 is shown in Figure 6B. The input signal band limitation of each service is down-converted to a first intermediate frequency bandwidth by a mixer 31 having a local oscillator 1A and a local oscillator 1B, respectively, and is again filtered by the second filter 312 to remove the shadow. The frequency is amplified by a variable gain amplifier 314. The first intermediate frequency spectrum output by the variable gain amplifier 314 is shown in the sixth C-picture. Any of the arbitrary channels from the lower key band in the second embodiment can be down-converted to any of the spaced channels at the radio frequency band using the configurable local oscillator 2. The second down conversion of the two input signals is indicated by a mixer μα having a local 11 1280754 oscillator 2A and a local oscillator condition, respectively. The second intermediate frequency spectrum filtered by the filter 318 is shown in the sixth D picture. As shown in Fig. 6D, the local oscillator 2A and the local oscillator 2β frequency can be adjusted such that the second down-conversion can cause the multi-service downlink frequency band to be placed in the second intermediate frequency bandwidth that is spaced apart from each other. Alternatively, 'local oscillator 1A and local oscillator 2A are adjustable, while local oscillator 2A and local oscillator 2B can be fixed, or both local oscillators can be adjusted. The multi-local oscillator frequency can also be used to place the multi-service downlink frequency band anywhere within the defined second intermediate frequency bandwidth. The final IF frequency is further downsampled by processing by filters 318, 322 and variable gain amplifier 32 。. By minimizing the second intermediate frequency bandwidth, the sampling rate of the analog digital converter 324 can be adaptive, thereby minimizing the power consumption of the final digital down conversion to baseband. The seventh diagram is a block diagram of a lookup table 400 in a receiver modem that is used to implement adaptive frequency down conversion in accordance with the present invention. The expected service, the sampling frequency I and the expected second intermediate frequency system are treated as inputs to the lookup table 400, which outputs the local oscillator 1 and local oscillator 2 frequency settings and the analog digital converter sampling frequency. The lookup table 400 optimizes the frequency scheme, sampling frequency, and sampling bandwidth based on available service and signal noise and distortion ratio measurements. Lookup table 400 can be used in any embodiment of the invention. Figure 8 is a block diagram of a local oscillator frequency synthesizer 500 for a local oscillator integrated frequency in accordance with the present invention. Since the receivers shown in the second and third embodiments require multiple local oscillator frequencies, the synthesizer 500 must be able to generate these frequencies. The local oscillator frequency synthesizer 500 includes a reference 12 1280754 test oscillator 502 and one or more synthesizers 504. The local oscillator frequency synthesizer can optionally further include one or more insulators 5〇6 and one or more circulators 508. The reference oscillator 502 can generate a reference frequency that is input to the complex synthesizer 504. Each synthesizer 504 is tuned to generate an intermediate frequency based on the local oscillator 1 and local oscillator 2 frequency settings generated by lookup table 400. The intermediate frequency generated by synthesizer 504 is passed to the local oscillator of the mixer to downconvert the input signal. The circulator 508 is preferably used to combine the local oscillator frequencies of the two synthesizers in a low loss combination scheme that minimizes the power consumption of the synthesizer. Insulators 506 are provided at the output of each of the synthesizers 5〇4 to provide sufficient reverse insulation to eliminate the frequency of any of the synthesizers cited by other synthesizers. Alternatively, a buffer amplifier in synthesizer 504 can be used to provide insulation. This causes the synthesizer method to be further amplified by removing the insulator 5〇6. The ninth figure shows the multiplex of the multi-wireless communication service processing 6(10) in the receiver simultaneously processing according to the present invention. - The partial-grade _ silk interface is received simultaneously (step 5〇2). Each service is transmitted via a different carrier frequency band. The received carrier frequency band is down-converted to intermediate frequency ▼ using the local oscillator, so that the down-converted frequency band is located in the single intermediate frequency band (step 5〇4). Alternative Embodiments The 'software-defined radio system enables (four) or more total local oscillators to simultaneously receive two or more services and/or channels to independently control the finalization of the two or more services and/or channels. Frequency and adaptability select the local (four) H solution and sampling frequency. According to the present invention, the adaptability of the virtual system is minimized by the sampling frequency, thereby reducing the processing block of the _ number conversion 1 and increasing the battery life of 13 1280754 雔 。. This embodiment of the invention can be implemented in a base station and a wireless transmitting/receiving unit. The tenth diagram is a block diagram of the receiving satellite 600 for adaptively selecting the local oscillator frequency and the sampling frequency for analog digital conversion of the plurality of input signals received simultaneously in accordance with the present invention. The receiver 600 includes an antenna 602, a low noise amplifier 604, a mixer 606, two local oscillators 608a and 60813, an adder 618, an analog converter 610, and a digital intermediate frequency processing unit 612'. A baseband processing unit 614, and a controller 616. Two or more input signals are simultaneously detected by antenna 602 for two or more services and/or channels. Each service and/or channel is transmitted via a different carrier frequency band and meets the requirements of a particular signal interference noise and distortion ratio (SINAD) <low noise amplifier 604 amplifies the received input signal. Each local oscillator 608a, 608b generates a corresponding frequency local oscillator signal for each service and/or channel. The tenth graph depicts only two local oscillators as an illustration 'but more than two local oscillators can also be used to place multiple services and/or channel downlink bands within the final IF bandwidth. The local oscillator signal frequency is controlled by controller 616. The local oscillator signal is summed by adder 618 and forwarded to mixer 006. Mixer 606 is a mix input signal and a local oscillator signal to convert each RF input signal to an intermediate frequency signal. Only one mixing step is depicted in the first figure. However, it should be noted that more than one mixing stage can be implemented to convert each RF input signal to a final intermediate frequency signal. The final IF band is chosen such that the service and/or channel inter-frequency bands are somewhat close to or overlapping each other in frequency spectrum. The spectral overlap can produce a 14 1280754 r * disturbance to one or both of the frequency bands and/or channels within the receiver. The eleventh A-F diagram is a block diagram of a frequency converted intermediate frequency spectrum depicting an RF input signal to a final mid-band in accordance with the present invention. In the eleventh A-F diagram, the shaded area represents the expected frequency channel. The local oscillating frequency is adjusted such that the down-conversion causes the input signal to be converted to a degree close to or greater than the final intermediate frequency band to some extent as shown in the tenth-A-F diagram. In the tenth-A diagram, the bands in service are close to each other but do not overlap. Therefore, one frequency band to another frequency band does not cause interference. In the eleventh B-picture, the two mid-bands overlap each other only in the unintended frequency channel. In the eleventh c and D diagrams, an interfering device is obtained by an expected channel, and both of the tenth-E and F pictures are predisturbed to obtain a jammer. In the eleventh F picture, all the medium frequency bands of a service and/or channel are overlapped with other intermediate frequency bands. ▲, to avoid IF ▼ any area overlap frequency, the sampling frequency should be set to be at least twice the value of the highest _ rate of the highest mid-band. The sampling frequency • can be lower than this value' where the frequency band region overlap is not acceptable within the expected channel. Therefore, the sampling frequency is obtained by the multi-frequency service and/or the money shed. The minimum sampling frequency that can be avoided by pre-channel repetition frequency-half is indicated by the tenth-A-F header & The minimum required sampling frequency to avoid the expected band overlap is indicated by the dashed arrow in the A-F diagram. If the signal noise and distortion ratio caused by the expected frequency on the channel is tolerable, the ‘sample frequency can be even lower than the one shown by the dashed arrow. When the degree of Hangai increased from the tenth-A map to the tenth-F map, the sampling decreased, but the interference of the channel slaves increased. Therefore, the coverage and the 280754 sample frequency should be chosen to take into account the sampling frequency and interference. The final selection of the IF bandwidth and coverage at the mid-band is adaptively adjusted to simultaneously measure the signal noise and distortion function of the intended service and/or channel. Each service and/or channel has a minimum signal complexity δί1 and distortion ratio criteria that must be met. Referring back to the tenth figure, the baseband processing unit 614 measures the signal noise and distortion ratio under various coverage conditions, and the controller 616 selects the lowest sampling frequency that satisfies the minimum signal noise and distortion ratio criterion as the optimal sampling. Frequency coverage. The analog-to-digital converter 610 converts the mid-band signal into a digital signal at a sampling frequency set by the controller 616. The digital intermediate frequency processing unit and the soil f processing unit 614 are for processing digital signals of the service. The digital intermediate frequency processing unit = 12 performs the final frequency conversion from the intermediate frequency to the baseband. The digital intermediate frequency processing unit 612 separates the services of each other. ...with the adaptive control of the service and / or the secret frequency band of the channel, the sampling frequency is minimized. Minimize the power consumption of the sampled fresh-to-digital converter and the processing block in the modem and increase the total battery life. Channel conditions (such as distance from the package, changes in adjacent channels, etc.) are subject to change over time. The coverage and the best sampling options are fined by several estimates. Because the wireless ship/receiver unit does not know that the lion's mile has disappeared, and it can change at a faster rate than the above-mentioned re-evaluation expectation, the acceptable connection is suddenly degraded, and the frequency of the optimal sampling frequency = re-acceptance & Selecting the storage _ is the non-continuous idle lion only: the period in which the poor material is received. During the period of unacceptable period of the lobes, the system does not need to overlap to support the highest sampling frequency of this situation. 1 1280754 Regardless of the position of the job and the most silky _, the sampling frequency can be hunted. It was reduced by the introduction of the frequency band of the inactive position of the livestock. Twelfth _ According to the present invention, the sampling process of the sampling signal of the analog signal analog-to-digital conversion in the color receiver. The receiver simultaneously receives two or more input signals for two or more services and/or channels (step 8〇2). Each service and/or channel is subject to signal interference and distortion ratio measurements. The input signal is converted to a mid-band signal by mixing the input signal and the local oscillator signal (step 8〇4). The local taper frequency is adjusted so that the input signal is converted to the mid-band signal to some extent to each other. Close or overlap. The signal and distortion ratios for the service and/or channel are measured by complex spectral overlap conditions (step 806). The local oscillator frequency and sampling frequency for analog frequency conversion of the intermediate frequency signal are based on signal interference noise and distortion ratio measurements (step 808). Steps 806 and 808 are preferably repeated periodically or non-periodically. Although the features and elements of the present invention are described in the preferred embodiments in the preferred embodiments, the various features and elements may be used alone or without other features and elements of the present invention. Among the various combinations. 17 1280754 BRIEF DESCRIPTION OF THE DRAWINGS The present invention can be understood from the following description of the embodiments and the accompanying drawings, wherein: FIG. 1 is a block diagram of a receiver according to a first embodiment of the present invention; a signal spectrum diagram of each stage in a receiver; the second figure is a block diagram of a receiver according to a second embodiment of the present invention;
第四A-D圖係為第三圖接收器中各階處之信號頻譜圖 示； 第五圖為依據本發明第三實施例之接收器方塊圖； 第V、A-D圖係為第五圖接收器中各階處之信號頻譜圖 不， 第七圖為依據本發明被用來實施適應頻率向下轉換之查 找表(LUT)方塊圖「^ ^ ^ —The fourth AD diagram is a signal spectrum diagram at each stage in the receiver of the third diagram; the fifth diagram is a block diagram of the receiver according to the third embodiment of the present invention; the V and AD diagrams are in the receiver of the fifth diagram. The signal spectrum diagram at each stage is not, and the seventh figure is a look-up table (LUT) block diagram "^ ^ ^ — which is used to implement adaptive frequency down conversion according to the present invention.
『八圖為依據本發明用於本地振盪器綜合頻率方塊圖； 第九圖為依據本發明同時處理接收器中多重無線通 務處理之流程圖; σ k 」十圖為依據本發日㈣於適舰兩輸人信號 轉換之採樣頻率之接收器方塊圖； 、 第十-A_F圖為依據本發明描緣射頻帶至最終 率轉換方塊圖； 、頻之_ j十—圖為依據本發_於適應性選擇接㈣ 入域類比數轉換之採樣辭之處職糊。數輪 18 12807548 is a block diagram of the integrated frequency spectrum of the local oscillator according to the present invention; FIG. 9 is a flow chart of simultaneous processing of multiple wireless transceivers in the receiver according to the present invention; σ k "10 map is based on the present day (4) The receiver block diagram of the sampling frequency of the two-input signal conversion of the ship; the tenth-A_F diagram is a block diagram of the RF band to the final rate conversion according to the present invention; the frequency _ j ten-picture is based on the present invention _ In the adaptive selection (4), the sampling of the in-class analogy number conversion is the job. Number wheel 18 1280754
【主要元件符號說明】 100、200、300、600 接收器 106、206、306、604 低雜訊放大器 114、120、214、220、314、320 可變增益放大器 124、224、324、610 類比數位轉換器 400 查找表 500 本地振盪器頻率綜合器 502 參考振盪器 602 天線 608a、608b本地振盪器618 相加器 800 適應性選擇接收器中複數輸入信號類比數位轉換 之採樣頻率之處理流程圖 102、202、302 雙工器 104、204、304、508 循環器 108、208、308 第一濾波器 110、116、210 '216、310、316、 606混合器 112、212、312 第二濾波器 118、122、218、222、318、322 濾波器 504 綜合器 506 絕緣器 614 基帶處理單元 616 612 數位中頻處理單元 控制器 19[Main component symbol description] 100, 200, 300, 600 receivers 106, 206, 306, 604 low noise amplifiers 114, 120, 214, 220, 314, 320 variable gain amplifiers 124, 224, 324, 610 analog digital Converter 400 lookup table 500 local oscillator frequency synthesizer 502 reference oscillator 602 antenna 608a, 608b local oscillator 618 adder 800 adaptive selection of the sampling signal of the complex input signal analog digital conversion sampling frequency in the receiver 102, 202, 302 duplexer 104, 204, 304, 508 circulator 108, 208, 308 first filter 110, 116, 210 '216, 310, 316, 606 mixer 112, 212, 312 second filter 118, 122, 218, 222, 318, 322 Filter 504 Synthesizer 506 Insulator 614 Baseband Processing Unit 616 612 Digital Intermediate Frequency Processing Unit Controller 19