200931847 六、發明說明： 【發明所屬之技術領域】 哭由ίίΓΐ有關無線通信緖。更特別是，本發明係有關接收 器中多重無線通信服務處理方法及裝置。 【先前技術】 軟體定義無線電（SDR)係為多重無線通信標準被支 ο ^接古收中且射頻信號㈣信號藉由軟體定義C ^之方案。有了軟體定義無線電，單硬體平台財需更換硬體 、’ 4即可支援多重無線通信標準，而被下載軟體可再配置^體。 =此法’無線傳送/接收單元可被快速配置來支援最新發展盔 尨標準及協定。典型單模式蜂巢基地台及無線傳送/接收單先係包 ^二外差式無線電接收器類比前端，一固定採樣速率類比數位 鲁 且及連位處理單元°類比前端中’預期信號係被濾波 且接者被向下轉換為固定中頻（IF)頻帶。類比數位轉換器係以數 錢理解調演算預期信號要求頻寬及其他因子為基礎當場被選擇 固定採樣速率來操作。目前’無線傳送/接收單元係被配置處理 經由多頻道接收之多服務。例如’無線傳送/接收單元可支梭數位 蜂巢系統(DCS)及寬頻分碼多重存取(WCDMA)系統之通信。各服務 =經由無線傳送/接收單元中之對應接收器路徑被處理而被分別 輸士無線傳送/接收單元中之數據機以便處理。然而，僅一服務於 、’σ疋時間被支援於各接收器路徑中。目前無線傳送/接收單元設計 ，包含别端配置，其涉及開關或多工器及針對各服務頻帶分割信 號為不同、接收器路徑之多渡波器。當基地台或無線傳送/接收單元 以不同載波頻率支援單無線接收器中之多同時服務及/求頻道 時’各種服務及/或頻道係於類比前端中被濾波及分別向下轉換為 中巧且接著以固定採樣速率被分別轉換為數位樣本。類比數位轉 換器之採樣速率係為影響接收器功率消耗之因子之一。通常，類 比數位轉換器及數據機中其他處理塊之功率消耗係與採樣^率成 200931847 採樣速率係f比較低採樣速率更多功率。因此，先前 接收單元需廣泛硬體資敎援多服務及配皇係非 預期忒無線傳送/接收單元電池壽命型式。 【發明内容】 本發明係有關接收器中多重無線通信服務處理方法及裝置。 明’―個以上無線通信服務係被同時接收及處理。該服 =、f由不職波鮮被傳送，而該被接收触頻帶係被向下轉 ίϊί頻帶°本地振|器⑽頻率係被設定使該多服務之被向下 轉換中頻帶下降為單中㈣。替代實施例巾，軟體定義轉200931847 VI. Description of the invention: [Technical field to which the invention belongs] Cry by ίίΓΐ about wireless communication. More particularly, the present invention relates to a method and apparatus for processing multiple wireless communication services in a receiver. [Prior Art] The Software Defined Radio (SDR) is a scheme in which multiple wireless communication standards are supported and the RF signal (4) signal is defined by software. With the software-defined radio, the single-hardware platform needs to be replaced with hardware, and the multi-wireless communication standard can be supported, while the downloaded software can be reconfigured. = This method 'Wireless transmit/receive unit can be quickly configured to support the latest development of helmet standards and agreements. Typical single-mode cellular base station and wireless transmit/receive single-precision packet ^ two heterodyne radio receiver analog front end, a fixed sampling rate analog digital and parallel processing unit ° analog front end in the 'expected signal is filtered and The receiver is downconverted to a fixed intermediate frequency (IF) band. The analog-to-digital converter operates on the spot at a fixed sampling rate based on a cost-understanding calculation of the expected signal required bandwidth and other factors. Currently, the 'wireless transmitting/receiving unit is configured to handle multiple services received via multiple channels. For example, the 'wireless transmitting/receiving unit can communicate with the digital cellular system (DCS) and the wideband coded multiple access (WCDMA) system. Each service = is processed via a corresponding receiver path in the WTRU and is separately transferred to the modem in the WTRU for processing. However, only one service, 'σ time' is supported in each receiver path. The current wireless transmit/receive unit design, including the other end configuration, involves a switch or multiplexer and multiple ferrisers that split the signal for each service band and have different receiver paths. When the base station or the WTRU or the WTRU supports multiple simultaneous services and/or channels in a single wireless receiver at different carrier frequencies, the various services and/or channels are filtered and down-converted to the medium-sized front end in the analog front end. And then converted to digital samples separately at a fixed sampling rate. The sampling rate of the analog digital converter is one of the factors that affect the power consumption of the receiver. In general, the power consumption of analog-like digital converters and other processing blocks in the data machine is more than the sampling rate of the 200931847 sampling rate f. As a result, the previous receiving unit required extensive hardware support for multiple services and the undesired 忒 wireless transmit/receive unit battery life type. SUMMARY OF THE INVENTION The present invention relates to a method and apparatus for processing multiple wireless communication services in a receiver. More than one wireless communication service is received and processed simultaneously. The service =, f is transmitted by the inactive wave, and the received band is rotated downward. The local oscillator (10) frequency is set so that the multi-service is down-converted to a single frequency band. Medium (four). Alternative embodiment towel, software definition switch
使用一類比數位轉換器及適應性選擇包含兩個或更多被^收於兩 ΐ，更多不同頻帶之複數輸人信賴比數位轉換之採樣頻率及適 應=選擇該本地振i器頻率來實施。各輸人信號係經由不同頻帶 運載不同贿。輸人信麟翻時接收。各服務贼到最士信號 雜，，失真比(sINAD)測量。輸人信號係藉由以特定頻率混合‘輸 入#號及多本地振盪器信號而被轉換為中頻帶信號。本地振盪器 頻率係被適應性選擇使該中頻帶某些程度上彼此頻譜接近或重 疊。該服務之信號雜訊及失真比係以各複數頻譜重疊條件來測 ^本地振ϋ齡及採樣解接著被_域雜職失真比測 里結果為基礎來調整。該處理較佳被連續重複。 【實施方式】 此後，"無線傳送/接收單元"名詞係包含但不限於使用著設備 ⑽）’行動台，固定或行動用戶單元’，叫器，或可操作於無線 環境中之任何其他類型元件。此後，當被稱為”基地台"名詞^係 包含但不限於Β節點，位址控制器，存取點(处)或無線環境中之 何其他接介裝置。 本發明係提供單接收器鏈結中支援同時接收多重無線通信服 務處理之方法及裝置。該硬體可藉由軟體來配置。此後，本^明 將參考數位蜂巢系統及寬頻分碼多重存取分頻雙工(F加）當作同 時服務例來解釋。然而，應注意本發明可應用至任何其他服務及 200931847 任何數量同時服務。圖示中之數值係被提供當作例證而非限 而只要不背離本發明傳授，任何其他數值均可被實施。 第，一圖為依據本發明第一實施例之接收器1〇〇方塊圖。 A-D一圖係為第一圖接收器1〇〇中各階處之信號頻譜圖示。如第二a 所不，雙工器102及循環器1〇4係限制輸入頻譜之頻帶及組 服務下鏈頻帶，而於低雜訊放大器（LNA)1〇6之前最小化元 失。只要低雜訊放大器106具有充足增益（10_15dB)來最小化 該接收器鏈結剩餘者之第二階雜訊圖形貢獻，則此亦建立主 含低雜訊放大器106加上低雜訊放大器〗06之前任何損失之雜訊圖 形之系統雜訊圖形。雙工器1〇2係移除位於預期下鏈頻帶間之任何 〇 中間上鏈頻帶(如第二A圖中之分頻雙工上鏈頻帶），因而避免寬頻 低雜訊放大器106之飽和。兩完全接收頻帶中之任意頻道可被同時 接收，而該服務選擇係軟體可配置。頻帶限制輸入頻譜係被低雜 訊放大器106放大且被第一濾波器1〇8濾波。被第一濾波器1〇8濾波 後之輸入頻譜係被顯示於第二B圖。該頻帶限制輸入頻譜係藉由具 有一固定本地振盪器丨頻率之混合器11〇被向下轉換為第一中頻頻 寬。第一中頻係再次被第二濾波器112濾波來移除影像頻率灰阻斷 器；且接著藉由可變增益放大器（VGA)114放大。可變增益放大器 114所輸出之第一中頻頻譜係被顯示於第二⑽。使用影像齒率轉 換’第二向下轉換係藉由具有本地振盪器2之混合器116指示。第 ® 二中頻頻譜係被顯示於第二D圖。本地振盪器2頻率係被設定使該 第一向下轉換促使該多服務下鍵頻帶被摺疊為如第二D圖所示之 單第二中頻頻寬。數位蜂巢系統下鏈頻帶及寬頻分碼多重存取分 頻雙工下鏈頻帶係被摺疊為單第二中頻頻寬。此促成使用高Q濾波 益以該第二中頻頻寬來衰減頻外（out-of-band)阻斷器友干擾 器。多本地振盪器頻率亦可被用來放置多服務下鏈頻帶於被定義 第一中頻頻寬内任何處。 第一圖之接收器100係執行兩向下轉換。然而，應注意第一圖 中之接收器100配置及稍後將被解釋之本發明其他實施例僅為本 發明較佳實施例，且一個或更多兩向下轉換可被實施。當最小化 200931847 ί ΐ ’ 2係使用具有111定濾波器之適應 來摺疊接收下鍵頻帶為第二中頻頻寬。最終中 =纽藉由紐器118，122及可變戦放大㈤麟理之後，係 ^類比數位轉換器124被進-細下採樣。#由最小化第二令頻 ，類比數位轉換器124之採樣速率可為適應性，因而最 終數位向下轉換為基帶之功率消耗。 $ 最終中頻頻寬係視接收器之信號雜訊及失真比測量而定。信 號雜訊及失真比測量係包含接㈣處_寬内之失真產物。通常 僅-信號出現於此魏_失真絲並不被產生，所以僅需信號 雜訊比(SNR)測量。因為接收器中出現多舰，所以失真產物係被 產生於該處理頻寬内且這些位準於信號雜訊比測量中必須被解 ，、。依據本發明’當最高信號雜訊及失真比被測量時，最小頻寬 係被選擇，而相反地，當最低信號雜訊及失真比被測量時，最大 最終頻寬係被選擇。第三圖為依據本發明第二實施例之接收器2〇〇 方塊圖。第四A-D圖係為第三圖接收器中各階處之信號頻譜圖示。 如第四Α圖所示，雙工器2〇2及循環器2〇4係限制輸入頻譜之頻帶。 頻帶限制輸入頻譜係被低雜訊放大器2〇6放大且被第一濾硃器2〇8 濾波。被第一濾波器208遽波後之輸入頻譜係被顯示於第四β圖。 >輸入信號接著藉由混合該輸入信號及本地振盪器丨所產生之 t號而被向下轉換為中頻信號。第二實施例中，兩下鍵頻帶係使 用兩固定本地振盪器1頻率及兩固定本地振盪器2頻率被轉換為最 終中頻處之鄰近頻帶。各服務之輪入信號係使用不同本地振盪器 頻率而被向下轉換。此例中，數位蜂巢系統下鏈頻帶係以夺地振 盪器1A及本地振盪器2A頻率被向下轉換，而寬頻分碼多重;^取系 統係以本地振盪器1B及本地振盪器2B頻率被向下轉換。The use of a class of analog-to-digital converters and adaptive selection consists of two or more digital signals that are input to two, more complex frequency bands than the sampling frequency of the digital conversion and adaptation = select the local oscillator frequency to implement . Each input signal carries different bribes via different frequency bands. The input letter is received by Xinlin. Each service thief goes to the most signal, and the distortion ratio (sINAD) is measured. The input signal is converted to an intermediate frequency band signal by mixing the 'input # number and multi-local oscillator signals at a specific frequency. The local oscillator frequency is adaptively chosen such that the mid-bands are somewhat close to or overlapping each other in frequency spectrum. The signal noise and distortion ratio of the service is measured by the complex spectral overlap conditions. The local oscillator age and sampling solution are then adjusted based on the results of the _ domain miscellaneous distortion ratio measurement. This process is preferably repeated continuously. [Embodiment] Hereinafter, the "wireless transmitting/receiving unit" includes, but is not limited to, the device (10) using a mobile station, a fixed or mobile subscriber unit, a caller, or any other operable in a wireless environment. Type component. Hereinafter, what is referred to as a "base station" includes, but is not limited to, a node, an address controller, an access point (where), or any other mediation device in a wireless environment. The present invention provides a single receiver. The method and device for supporting simultaneous reception of multiple wireless communication services are supported in the chain. The hardware can be configured by software. Thereafter, the present invention will refer to the digital cellular system and the broadband code division multiple access frequency division duplex (F plus It is to be interpreted as a simultaneous service. However, it should be noted that the present invention is applicable to any other service and any number of simultaneous services of 200931847. The numerical values in the figures are provided by way of illustration and not limitation, Any other value can be implemented. The first figure is a block diagram of the receiver 1 according to the first embodiment of the present invention. The AD picture is a signal spectrum diagram of each stage in the receiver 1 of the first figure. If the second a does not, the duplexer 102 and the circulator 1〇4 limit the frequency band of the input spectrum and the group service downlink frequency band, and minimize the loss of the element before the low noise amplifier (LNA) 1〇6. Low noise amplifier 106 There is sufficient gain (10_15dB) to minimize the second-order noise pattern contribution of the remainder of the receiver chain, which also establishes any loss of noise before the main low-noise amplifier 106 plus the low noise amplifier 〖06 Graphical system noise pattern. The duplexer 1〇2 removes any 〇 intermediate upper frequency band between the expected downlink frequency bands (such as the frequency division duplex upper frequency band in Figure 2A), thus avoiding low bandwidth The saturation of the noise amplifier 106. Any of the two fully received frequency bands can be simultaneously received, and the service selection software is configurable. The band limited input spectrum is amplified by the low noise amplifier 106 and is amplified by the first filter 1〇8 Filtering. The input spectrum filtered by the first filter 1 〇 8 is shown in Figure 2. The band limited input spectrum is downconverted to a mixer 11 having a fixed local oscillator 丨 frequency. First IF bandwidth. The first IF system is again filtered by the second filter 112 to remove the image frequency gray blocker; and then amplified by a variable gain amplifier (VGA) 114. The output of the variable gain amplifier 114 First The IF spectrum is shown in the second (10). The image tonal conversion is used. The second down conversion is indicated by the mixer 116 with the local oscillator 2. The second IF spectrum is shown in the second D. Local The oscillator 2 frequency is set such that the first down-conversion causes the multi-service lower keyband to be folded into a single second intermediate frequency bandwidth as shown in the second D. Digital cellular system downlink frequency band and wideband code division multiple The access frequency division duplex downlink frequency band is folded into a single second intermediate frequency bandwidth. This facilitates the use of high Q filtering to mitigate the out-of-band blocker interference with the second intermediate frequency bandwidth. The multi-local oscillator frequency can also be used to place the multi-service downlink frequency band anywhere within the defined first intermediate frequency bandwidth. The receiver 100 of the first diagram performs two down conversions. However, it should be noted that the receiver 100 configuration in the first figure and other embodiments of the invention to be explained later are merely preferred embodiments of the present invention, and one or more two down conversions may be implemented. When the minimized 200931847 ί ΐ ' 2 series uses an adaptation with a 111 fixed filter to fold the receive lower key band to the second intermediate frequency bandwidth. In the end, the new analog device 118, 122 and the variable 戦 amplifier (five) are processed, and the analog-to-digital converter 124 is down-sampled. By minimizing the second frequency, the sampling rate of the analog digital converter 124 can be adaptive, and thus the final digits are down converted to baseband power consumption. The final IF bandwidth depends on the receiver's signal noise and distortion ratio measurements. The signal noise and distortion ratio measurement system contains distortion products within the width of (4). Usually only the -signal appears here. The distortion-filament is not generated, so only signal-to-noise ratio (SNR) measurements are required. Because multiple ships are present in the receiver, distortion products are generated within the processing bandwidth and these levels must be resolved in signal noise ratio measurements. According to the present invention, when the highest signal noise and distortion ratio are measured, the minimum bandwidth is selected, and conversely, when the lowest signal noise and distortion ratio are measured, the maximum final bandwidth is selected. The third figure is a block diagram of a receiver 2 according to a second embodiment of the present invention. The fourth A-D diagram is a signal spectrum diagram at each stage in the receiver of the third diagram. As shown in the fourth diagram, the duplexer 2〇2 and the circulator 2〇4 limit the frequency band of the input spectrum. The band limited input spectrum is amplified by the low noise amplifier 2〇6 and filtered by the first filter 2〇8. The input spectrum after being chopped by the first filter 208 is displayed in the fourth beta map. > The input signal is then downconverted to an intermediate frequency signal by mixing the input signal with the t number generated by the local oscillator 。. In the second embodiment, the two lower key bands are converted to the adjacent frequency band at the final intermediate frequency using two fixed local oscillator 1 frequencies and two fixed local oscillator 2 frequencies. The round-robin signals for each service are downconverted using different local oscillator frequencies. In this example, the downlink frequency band of the digital cellular system is down-converted with the frequency of the ground-shunt oscillator 1A and the local oscillator 2A, and the wide-band code division is multiple; the system is taken with the local oscillator 1B and the local oscillator 2B frequency. Down conversion.
各服務之輸入信號頻帶限制係藉由分別具有本地振盪器丨八及 本地振盪器1B之混合器210被向下轉換為第一中頻頻寬，且再次被 第二濾波器212濾波來移除影像頻率及藉由可變增益放大器214放 大。可變增益放大器214所輸出之第一中頻頻譜係被顯示於第四c 圖。第二向下轉換係藉由分別具有本地振盪器2人及本地振盏器2B 200931847 器f8所輸出之第二中頻頻譜係被顯 帶#；糸被玫疋使該第二向下轉換促使該多服務下鏈頻 ΪΪϋΐΐ置於如第四D圖所示之第二中頻頻寬中。此例中， 缺I鍵頻帶及寬頻分碼多重存取分頻雙工下鏈頻帶係 中頻頻帶中之鄰近頻帶。多本地振蘆器頻率赤可被 =服務下鏈頻帶於被定義第二中頻頻寬内任何處。最終 :頻率係於藉由濾、波器218，222及可變增益放大器22〇處理之後 向下採樣。藉由最小化第二中頻頻寬，類比數位轉換器The input signal band limits for each service are downconverted to a first intermediate frequency bandwidth by a mixer 210 having a local oscillator 及8 and a local oscillator 1B, respectively, and are again filtered by the second filter 212 to remove the image. The frequency is amplified by a variable gain amplifier 214. The first intermediate frequency spectrum output by the variable gain amplifier 214 is shown in the fourth c-picture. The second down conversion is performed by the second intermediate frequency spectrum system outputted by the local oscillator 2 and the local oscillator 2B 200931847 f8, respectively; The multi-service downlink frequency is placed in the second intermediate frequency bandwidth as shown in the fourth D-picture. In this example, the lack of I keyband and wideband code division multiple access frequency division duplex downlink frequency bands are adjacent frequency bands in the intermediate frequency band. The multi-local vibrator frequency red can be = service down-chain band anywhere within the defined second IF bandwidth. Finally, the frequency is downsampled by filtering, wavers 218, 222 and variable gain amplifier 22 〇. By minimizing the second IF bandwidth, analog to digital converter
❹ 似之採樣速率可為適應性，因而最小化最終數位向下轉換為基帶 之功率消耗。 第五圖為依據本發明第三實施例之接收器300方塊圖。第六 A-D圖係為第五圖接收器3〇〇中各階處之信號頻譜圖示。如第六八圖 所不，雙工器302及循環器304係限制輸入頻譜之頻帶。頻帶限制 ，入頻譜巧被低雜訊放大器306放大且被第一濾波器3〇8濾波。被 第一濾波器308濾波後之輸入頻譜係被顯示於第六B圖。 各服務之輸入信號頻帶限制係藉由分別具有本地振蘯器1A及 本地振盪器1B之混合器310被向下轉換為第一中頻頻寬，且再次被 第二濾波器312濾波來移除影像頻率及藉由可變增益放大器gw放 大。可變增益放大器314所輸出之第一中頻頻譜係被顯示於第六c 圖。 ’ 第三實施例中，來自下鏈頻帶之任何任意頻道均可使用可配 置本地振盪器2^1向下轉換為射頻頻帶處之任意間隔頻道。兩輸入 信號之第二向下轉換係分別藉由具有本地振盪器2A及本地振盪器 2B之混合器316來指示。被渡波器318遽波後之第二中頻頻譜係被 顯示於第六D圖。如第六D圖所示，本地振盪器2A及本地振盪器2B 頻率係可為調整，使該第二向下轉換可促使多服務下鏈頻帶被放 置於彼此相隔之第二中頻頻寬中。 可替代是’本地振盪器1A及本地振盪器2A係為可調整，而本 地振盪器2A及本地振盪器可為固定，或本地振盪器兩者均為可 200931847 調整。多本地振蘯器頻率亦可被用來放置多服務下鍵頻帶於被定 義第ΐ中ϊΐί，任何處。祕中頻頻率係於藉由濾波器318,322 及可變增i放大器32G處理之後被進-步向下採樣。藉由最小化第 二中頻頻寬’類崎位讎腿4之雜速衬騎雜 最 小化最終數位向下轉換為基帶之功率消耗。 第七圖為依據本發明被用來實施適應頻率向下轉換之接收器 數據機中之查找表400方塊圖。預期服務，採樣頻寬及預第二 頻係被當作触_4GG之輸人，喊麵表4_輸^本地振盪 器1及本地振盪器2頻率設定及類比數位轉換器採樣頻率。鲞找表 400係依據可付服務及彳§號雜訊及失真比測量來最佳化頻率方 〇 案，採樣頻率及採樣頻寬。查找表棚可被用財發明任何實施例 中。 第八圖為依據本發明用於本地振H絲鮮之本地振盡器 頻率綜合器500方塊圖。因為第二及第三實施例中所示接收器係需 多本地振盪器頻率，所以綜合器5〇〇必須可產生這些頻率。本地振 盪器頻率綜合器500係包含一參考振盪器5〇2及一個或更多緙合器 504。本地振盪器頻率綜合器可選擇性進一步包含一個或更’多'^緣 器506及一個或更多猶環器508。參考振盪器502可產生被輸入複數 綜合器504中之參考頻率。各綜合器504係依據查找表4〇〇所產生之 本地振盪器1及本地振盪器2頻率設定被調諧產生中頻頻率。綜合 ❹器504所產生之中麵率係被傳送至混合器之本地振璋^ 下轉換輸入信號。 ^循環器508於可最小化綜合器功率消耗之低損失組合方案中 係較佳被用來組合兩綜合器之本地振盈器頻率。絕緣器倍被接 巧於各綜合觀4輸出處以提供充分反向絕緣來消_其他^ 器所招引任一綜合器之頻率。可替代是，綜合器5〇4中之緩衝放大 器可被用來提供絕緣。此促使綜合器方法藉由移除絕緣器5〇6而被 進一步放大。 第九圖為依據本發明同時處理接收器中多重無線通信^艮務處 理600之流程圖。一個以上服務係經由無線介面被同時接收（步驟 200931847 =地傳送。被接收載波頻帶係使 於單中頻㈣頻解’使紐向下轉換頻帶座落 祕中頻及適雜選擇該兩個或更多本地 樣頻率。依據本發明此實施例之軟義無線電 採樣頻率’因而降低類比數位轉換器之功率消 命。本舰實施例可 〇 第十圖為依據本發明適應性選擇本地振盪器頻率及採樣頻率 用於被同時接收之複數輸入信號之類比數位轉換之接收器600方 ，圖。接收器600係包含一天線602，一低雜訊放大器6〇4，一混合 器606 ’兩本地振盪器6〇8a及608b，一相加器618，一類比數位轉 換器610 ’ 一數位中頻處理單元612 ’ 一基帶處理單元614，及一控 制器616。兩個或更多輸入信號係被天線6〇2同時偵測用於兩個或 更多服務及/或頻道。各服務及/或頻道係經由不同載波頻帶被傳 送並受到獨一信號干擾，雜訊及失真比^似的測量。低雜訊放大 器604係放大該被接收輸入信號。 各本地振盪器608a，608b係產生各服務及/或頻道之對應頻率 © 本地振蘯器信號。第一圖僅描緣兩本地振盡器當作例證，旦兩個 以上本地振盪器亦可被用來放置多服務及/或頻道下鏈頻帶於最 終中頻頻寬内。本地振盪器信號頻率係被控制器616控制。本地振 盪器信號係被相加器618加總並被轉送至混合器606。 混合器606係混合輸入信號及本地振盪器信號來轉換各射頻 輸入信號至中頻信號。僅一混合階被描繪於第一圖。然而，應注 意一個以上混合階可被實施來轉換各射頻輸入信號至最終中頻信 號。最終中頻頻帶係被選擇使服務及/或頻道之中頻頻帶某些程度 上彼此光譜接近或重疊。光譜重疊可於接收器内產生對一個或兩 者頻帶及/或頻道之干擾。 丨 200931847 Ο Ο 41 Ϊ十—A F圖為依據本發明描繪射頻輸入信號至最終t頻帶 轉換之中頻頻譜方塊圖。第十一 A_F圖中陰影區域係代表預 頻道。本地振盪器頻率係被調整使向下轉換促使輸人信號 換為如第十一 A-F圖所示某些程度上彼此最終中頻帶接 f豐。第十—A圖中，服務之中頻帶係彼此接近但不重疊。因此一， ，帶至另一頻帶並不產生干擾。第十_B圖中，兩中頻帶僅於 千^率頻道中彼此重疊。第十及D圖中，—翻頻道係獲得 器’而第十一圖中，兩者預期頻道皆獲得干擾器。第十 圖十，一服務及/或頻道全部中頻帶均被與其他中頻帶重疊。 ii避免ί頻帶任何區域疊頻，採樣頻率應被設定為高於最高中 ，帶之最高辭組成至少兩倍之值。採樣辭可低於該值，其中 期頻道内之中頻帶區域4頻係可接受。因此，採叙步員率 ==被同時處理之複數服務及/或頻道間具有最高頻 頻道來蚊。可避細娜道4狀最小採樣頻率一半 標示°可避免預期頻帶疊頻之最小所 上:二Si係藉由第十—A-F®中虛箭頭標示。若因預期頻道 率可號雜訊及失真比可容忍，則該挺樣頻 tLi虛箭頭所示者。#覆蓋程度從第十—镯增加至第 棒、w ’採樣解Ύ降但預期頻道中之干擾增加。因此，覆蓋 紐選料絲樣鮮及干擾。最終帽帶處之 =皮贼失真比錄。各歸及/或頻道係具有 係測量各讎情況下之信號雜訊及失真比,，而控 頻率當作;祕錢姐準狀最轉樣 頻帶係以㈣器616所設定之採樣頻率轉換中 俜數位中頻處理單元612及基帶處理單元614 ϋΐίίϊίΐ ί°數財贼理單元612係執行從中頻至基 帶之祕解轉換。數位中触理單元612係分紐此之 200931847 藉由賴性控舰務及/或_之最 性最小化。最小化採樣頻率係降低類比H 適應 理塊之功率雜，並增加總合f池壽命。轉轉及數據機中處 變。距離，鄰近頻道之改變等)係隨時間改 上述再評估翻為快之速率改變，n 了可以較 突然降級，最佳採樣頻率之頻譜重疊 ❹ 率1藉由蓄針料職之·独耐祕^；7低雜樣頻 轳騸ί依據本發明用於適應性選擇接收11中複數輸入信 2轉換之採樣頻率之處理800流程®。接收器係同，接收 多服務及/或頻道之兩個或更多輸入信號(步驟別2)。各 頻5係受到信號干擾雜訊及失真比測量。輸入信號係藉 8Π二^^入“號及本地振盪器信號而被轉換為中頻帶信號（步驟 〇地振盪器頻率係被調整使輸入信號之被轉換中頻帶信號 程度上彼此頻譜接近或重疊。該月艮務及/或頻道之信號雜訊及 〇 失真比係以各複數頻譜重疊條件來測量（步驟806)。用於中釋信號 類$數位，換之本地振盪器頻率及採樣頻率係以信號干擾雜訊及 失真比測量結果為基礎(步驟808)。步驟806及808係較佳被定期或 非定期重複。 雖然本發明之特性及元件被以特定組合說明於較佳實施例 中’但各特性及元件可不需較佳實施例之其他特性及元件而被單 獨使用’或有或無本發明其他特性及元件之各種組合中。： 【圖式簡單說明】 ，一圖為依據本發明第一實施例之接收器方塊圖；‘ 第二A-D圖係為第一圖接收器中各階處之信號頻譜圖示； 第三圖為依據本發明第二實施例之接收器方塊圖； 200931847 :圍D圖係為第三圖接收器中各階處之信號頻譜圖示； 圖為依據本發明第三實施例之接收器方塊圖；， 圖係為第五圖接收器中各階處之信號頻譜圖示； ®為依據本發明被用來實施適應頻率向下轉換之查找表 圖為依據本發明用於本地振盪器綜合頻率方塊圖；‘ 理之流程^為依據本發明同時處理接收器中多重無線通信服務處 .ί If為依據本發明用於適應性選擇兩輸入信賴比數位轉 換之採樣頻率之接收器方塊圖； Ο ❹ 她七圖為依據本發明描繪射頻帶至最終中頻之頻率轉 換方塊圖；以及 ， 缺栖為依據本發明用於適應性選擇接收器中複數輸入信 旒類比數位轉換之採樣頻率之處理流程圖。 【主要元件符號說明】 接收器 低雜訊放大器 可變增益放大器. 類比數位轉換器 查找表 本地振盪器頻率綜合器 參考振盪器 天線 丨 本地振盪器 相加器 適應性選擇接收蒸中複 數輸入信號類比數位轉 換之採樣頻率之處理流程圖 100、200、300、600 106、206、306、604 114 ' 120 ' 214 > 220 ' 314 ' 320 610 124、224、324 400 500 502 602 608a、608b 618 800 12The sampling rate can be adaptive, thus minimizing the power consumption of the final digit down conversion to baseband. Figure 5 is a block diagram of a receiver 300 in accordance with a third embodiment of the present invention. The sixth A-D diagram is a signal spectrum diagram at each stage of the receiver 3 of the fifth diagram. As shown in Figure 68, duplexer 302 and circulator 304 limit the frequency band of the input spectrum. The band limitation, the incoming spectrum is amplified by the low noise amplifier 306 and filtered by the first filter 3〇8. The input spectrum filtered by the first filter 308 is shown in Figure 6B. The input signal band limits for each service are downconverted to a first intermediate frequency bandwidth by a mixer 310 having a local oscillator 1A and a local oscillator 1B, respectively, and are again filtered by the second filter 312 to remove the image. The frequency is amplified by a variable gain amplifier gw. The first intermediate frequency spectrum output by the variable gain amplifier 314 is shown in the sixth c-picture. In the third embodiment, any arbitrary channel from the downlink frequency band can be down-converted to any spaced channel at the radio frequency band using the configurable local oscillator 2^1. The second down conversion of the two input signals is indicated by a mixer 316 having a local oscillator 2A and a local oscillator 2B, respectively. The second intermediate frequency spectrum after being chopped by the wave 318 is shown in the sixth D picture. As shown in the sixth diagram D, the local oscillator 2A and local oscillator 2B frequencies can be adjusted such that the second down-conversion can cause the multi-service downlink frequency bands to be placed in the second intermediate frequency bandwidth that is spaced apart from each other. Alternatively, 'local oscillator 1A and local oscillator 2A are adjustable, while local oscillator 2A and local oscillator can be fixed, or both local oscillators can be adjusted for 200931847. The multi-local oscillator frequency can also be used to place the multi-service lower keyband in the defined ΐί, anywhere. The secret intermediate frequency is processed down by the filters 318, 322 and the variable-increasing amplifier 32G and then down-sampled. By minimizing the second IF bandwidth, the spurs of the squad, the spurs of the spurs 4 minimize the final digits down-converted to baseband power consumption. Figure 7 is a block diagram of a lookup table 400 in a receiver modem that is used to implement adaptive frequency down conversion in accordance with the present invention. The expected service, sampling bandwidth and pre-second frequency are treated as the input of the _4GG, and the local oscillator 1 and the local oscillator 2 frequency setting and the analog digital converter sampling frequency are called.鲞 Lookup Table 400 optimizes the frequency scheme, sampling frequency and sampling bandwidth based on the payable service and 彳§ noise and distortion ratio measurements. The look-up table can be used in any embodiment of the invention. Figure 8 is a block diagram of a local synchronizer frequency synthesizer 500 for local oscillators in accordance with the present invention. Since the receivers shown in the second and third embodiments require multiple local oscillator frequencies, the synthesizer 5 must be able to generate these frequencies. Local oscillator frequency synthesizer 500 includes a reference oscillator 5〇2 and one or more couplers 504. The local oscillator frequency synthesizer can optionally further include one or more 'edges 506 and one or more loopers 508. The reference oscillator 502 can generate a reference frequency that is input into the complex synthesizer 504. Each synthesizer 504 is tuned to generate an intermediate frequency based on the local oscillator 1 and local oscillator 2 frequency settings generated by lookup table 4A. The face rate generated by the integrated buffer 504 is transmitted to the local oscillator of the mixer to down-convert the input signal. The circulator 508 is preferably used to combine the local oscillator frequencies of the two synthesizers in a low loss combination scheme that minimizes the power consumption of the synthesizer. The insulator is tuned to the output of each of the integrated views 4 to provide sufficient reverse insulation to eliminate the frequency of any of the combiners. Alternatively, a buffer amplifier in the synthesizer 5〇4 can be used to provide insulation. This causes the synthesizer method to be further amplified by removing the insulator 5〇6. Figure 9 is a flow diagram of simultaneous processing of multiple wireless communication processing 600 in a receiver in accordance with the present invention. More than one service is simultaneously received via the wireless interface (step 200931847 = ground transmission. The received carrier frequency band is such that the single intermediate frequency (four) frequency solution' makes the down-conversion frequency band locate the secret intermediate frequency and selects the two or More local sample frequency. The soft-sampling radio sampling frequency according to this embodiment of the invention thus reduces the power consumption of the analog-to-digital converter. The present embodiment can be adapted to select the local oscillator frequency in accordance with the present invention. And the sampling frequency is used for the analog-to-digital conversion of the plurality of input signals received by the receiver 600. The receiver 600 includes an antenna 602, a low noise amplifier 6〇4, and a mixer 606'. 6〇8a and 608b, an adder 618, an analog-to-digital converter 610', a digital intermediate frequency processing unit 612', a baseband processing unit 614, and a controller 616. Two or more input signals are antennas 6〇2 simultaneous detection for two or more services and/or channels. Each service and/or channel is transmitted via different carrier frequency bands and is subject to unique signal interference, noise and distortion ratio ^ The low noise amplifier 604 amplifies the received input signal. Each local oscillator 608a, 608b generates a corresponding frequency © local oscillator signal for each service and/or channel. As an example, more than two local oscillators can be used to place the multi-service and/or channel downlink frequency bands within the final intermediate frequency bandwidth. The local oscillator signal frequency is controlled by controller 616. Local oscillator The signal is summed by adder 618 and forwarded to mixer 606. Mixer 606 mixes the input signal and the local oscillator signal to convert each RF input signal to an intermediate frequency signal. Only one mixing stage is depicted in the first figure. However, it should be noted that more than one mixing stage can be implemented to convert each RF input signal to a final IF signal. The final IF band is selected such that the service and/or channel intermediate frequency bands are somewhat close to or overlapping each other in the spectrum. The spectral overlap can cause interference to one or both of the frequency bands and/or channels within the receiver. 丨200931847 Ο Ο 41 Ϊ10—AF map is a radio frequency input signal according to the present invention. Number to the final t-band conversion IF spectrum block diagram. The shaded area in the eleventh A_F diagram represents the pre-channel. The local oscillator frequency is adjusted so that the down-conversion causes the input signal to be replaced by the eleventh AF map. In some cases, the frequency bands in the service are close to each other but do not overlap each other. Therefore, the band is not interfered with another band. Tenth_B In the middle, the two mid-bands overlap each other only in the channel. In the tenth and D-pictures, the channel is obtained by the channel, and in the eleventh figure, both channels are expected to obtain the jammer. The entire intermediate frequency band of a service and/or channel is overlapped with other intermediate frequency bands. ii To avoid any area overlap frequency in the ί band, the sampling frequency should be set higher than the highest, and the highest frequency of the band is at least twice the value. The sampled word can be lower than this value, and the mid-band area 4 frequency system in the medium channel is acceptable. Therefore, the rate of adoption is == multiple services being serviced simultaneously and/or the highest frequency channel between mosquitoes. It can avoid the minimum sampling frequency of 4, and the minimum sampling frequency can be avoided. The minimum value of the expected frequency band can be avoided: the second Si is indicated by the dotted arrow in the tenth-A-F®. If the expected channel rate can be tolerated by the noise and distortion ratio, then the tricky frequency tLi is indicated by the dashed arrow. # Coverage increased from the tenth-bangle to the first, w' sample, but the expected interference in the channel increased. Therefore, covering the new selection of raw materials is fresh and disturbing. The final cap band = thief distortion ratio recorded. Each of the categorization and/or channel systems has the signal noise and distortion ratio measured under various conditions, and the control frequency is regarded as; the most variable frequency band of the secret money sister is converted by the sampling frequency set by the (four) 616. The digital intermediate frequency processing unit 612 and the baseband processing unit 614 ϋΐ ϊ ϊ ϊ 数 数 数 数 数 数 612 612 612 612 612 612 612 612 612 612 612 612 612 612 612 612 612 。 。 。 。 。 。 。 In the digital position unit 612, the 200931847 is minimized by the control of the ship and/or _. Minimizing the sampling frequency reduces the power mismatch of the analog H-adaptive block and increases the lifetime of the aggregate f-cell. Transfer and change in the data machine. The distance, the change of the adjacent channel, etc.) changes the rate of the above-mentioned re-evaluation over time, and can be more suddenly degraded, and the spectrum of the optimal sampling frequency overlaps with the rate 1 by the needle storage material. ^;7Low sample frequency ί According to the present invention, the process 800 for adaptively selecting the sampling frequency of the complex input 2 conversion in the reception 11 is performed. The receiver is the same, receiving two or more input signals of multiple services and/or channels (step 2). Each frequency 5 system is subject to signal interference noise and distortion ratio measurement. The input signal is converted to a mid-band signal by means of the "number" and the local oscillator signal (step 振荡器 the oscillator frequency is adjusted such that the converted mid-band signal of the input signal is spectrally close to or overlapping each other. The signal noise and 〇 distortion ratio of the service and/or channel are measured by each complex spectral overlap condition (step 806). For the intermediate release signal class, the digit is replaced by the local oscillator frequency and the sampling frequency. The signal interference noise and distortion ratio measurements are based on (step 808). Steps 806 and 808 are preferably repeated periodically or non-periodically. Although the features and elements of the present invention are described in a particular combination in the preferred embodiment, The various features and elements may be used alone or in combination with other features and elements of the invention without the additional features and elements of the preferred embodiments.: [Simple Description of the Drawings], Figure 1 is in accordance with the present invention. a receiver block diagram of an embodiment; 'the second AD picture is a signal spectrum diagram at each stage of the first picture receiver; the third figure is a receiver block according to the second embodiment of the present invention. Figure 31; 200931847: The surrounding D picture is a signal spectrum diagram of each stage in the receiver of the third figure; the figure is a block diagram of the receiver according to the third embodiment of the present invention; a signal spectrum diagram; ® is a look-up table for performing adaptive frequency down-conversion according to the present invention as a block diagram of a local oscillator integrated frequency according to the present invention; the process flow is based on the present invention simultaneously processing reception Multiple wireless communication service at the device. ί If is a block diagram of a receiver for adaptively selecting a sampling frequency of two-input confidence ratio digital conversion according to the present invention; Ο ❹ her seven diagrams depict the radio frequency band to the final intermediate frequency according to the present invention Frequency conversion block diagram; and, lack of habitat is a processing flow chart for sampling frequency of complex input signal analog-to-digital conversion in an adaptive selection receiver according to the present invention. [Main component symbol description] Receiver low noise amplifier can be Variable Gain Amplifier. Analog Digital Converter Lookup Table Local Oscillator Frequency Synthesizer Reference Oscillator Antenna 丨 Local Oscillator Adder Adaptation Process flow diagram 100, 200, 300, 600 106, 206, 306, 604 114 '120 ' 214 > 220 ' 314 ' 320 610 124, 224, 324 400 selected to receive the sampling frequency of the complex input signal analog to digital conversion 500 502 602 608a, 608b 618 800 12