201128912 六、發明說明： 【發明所屬之技術領域】201128912 VI. Description of the invention: [Technical field to which the invention belongs]
本發明係關於電源供應器，特別係關於一種應用間歇 式控制之諧振轉換器。 ~ B 【先前技術】 近年來，由於節能環保的目的，電源供應器係朝著高 效率（high efficiency)、高功率密度、高可靠性與低成本 方向發展。諧振型轉換器由於具有軟切換特性，且操作在 最大占空比（duty cycle)的狀態’因而在滿載時效率很高， 並受到很多人的青睞。然而’諧振型轉換器在輕載時= 並不理想。 >。 為了克服此問題’習知技術係將諧振型轉換写操作在 一間歇式工作模式(Burst Mode)，藉以降低單位時間内的切 換次數與總體損耗，使得效率能夠提升。舉例而言，如第 1圖中所示，於間歇式工作模式中，當誤差放大信號 等於或高於滯回比較電路之上限值Vref2時，壓頻轉換器 會被致能而用以產生振盪信號，使得半橋開關電路根據控 制信號(LVG與HVG)進行切換。相反地，當誤差放大信號 Vea低於來回比較電路之下限值vren時，壓頻轉換器會 被禁能而停止產生的振盪信號，使得半橋開關電路無驅動 信號。 然而，間歇式工作模式控制仍具有些許不足之處。舉 例而έ，由於誤差放大信號Vea會在滯回比較電路之上、 下限值Vref2與Vrefl之間波動，且誤差放大信號Vea與 201128912 振盪信號的頻率fosc成反比，所以在單次間歇模式工作週 期（Burst Mode Working Period ; BMWP)中，振蘯信號的頻 率fosc會隨著誤差放大信號Vea由高變低而由低變高。此 外，在單次間歇模式工作週期BMWP内的前幾個驅動脈衝 週期中，由於諧振網路阻抗降低所以會產生一個很大的諧 振電流（意即諧振電流不平衡），因而引發大的輸出電壓波 紋、音頻噪音以及最佳工作點變動（激磁電感電流磁偏以及 無法零電壓切換）···等問題。 【發明内容】 基於以上的考量，需要一種可降低輸出電壓波紋與音 頻噪音並提升效率的間歇式控制方式以及應用此控制方式 之諧振轉換器。 有鑑於此，本發明係提供一種諧振轉換器，包括一方 波產生器、一諧振電路、一輸出整流電路以及一控制器。 方波產生器用以提供一方波電壓，諧振電路用以根據方波 電壓進行諧振，而輸出整流電路用以根據諧振電路之諧 振，輸出一輸出電壓。控制器用以在一間歇模式工作週期 中提供一控制信號驅動方波產生器，其中控制信號包括至 少一第一脈衝組以及至少一第二脈衝組，並且第一脈衝組 包括至少一第一調節脈衝，而第二脈衝組包括複數驅動脈 衝。方波產生器係用以根據第一調節脈衝對諧振電路之一 激磁電感電流以及一諧振電容電壓進行預先調整，使得諧 振轉換器在第二脈衝組中之每個驅動脈衝的上升緣時，激 201128912 磁電感電流大抵上相等，並且諧振電容電壓也大抵上相等。 本發明亦提供另一種諧振轉換器，包括一方波產生 器、一諧振電路、一輸出整流電路以及一控制器。方波產 生器用以提供一方波電壓，諧振電路，用以根據方波電壓 進行諧振，而輸出整流電路，用以根據諧振電路之諧振， 輸出一輸出電壓。控制器用以在一間歇模式中提供一控制 信號驅動方波產生器，其中控制信號包括至少一第一脈衝 組以及至少兩個第二脈衝組，並且第一脈衝組係位於兩個 第二脈衝組之間，且包括至少一第一調節脈衝，而第二脈 衝組係位於第一調節脈衝之後，且包括複數驅動脈衝。方 波產生器係用以根據第一調節脈衝，調整諧振電路之一激 磁電感電流以及一諧振電容電壓，使得諧振轉換器在第二 脈衝組中之每個驅動脈衝的上升緣時，激磁電感電流大抵 上相等，並且諧振電容電壓也大抵上相等。 本發明亦提供一諧振轉換器的間歇模式控制方法，包 括在一間歇模式工作週期中，提供至少一第一調節脈衝， 用以對一諧振電路之一激磁電感電流以及一諧振電容電壓 進行預先調整，並且於上述第一調節脈衝之後，提供至少 一脈衝序列，用以間歇式地導通一方波產生器中之複數開 關元件。脈衝序列包括複數驅動脈衝，且第一調節脈衝係 用以調整諧振電路之激磁電感電流以及諧振電容電壓，使 得諧振轉換器在脈衝序列中之每個驅動脈衝的上升緣時， 激磁電感電流大抵上相等，並且諧振電容電壓也大抵上相 201128912 【實施方式】 為使本發明之上述目的、特徵和優點能更明顯易懂， 下文特舉較佳實施例，並配合所附圖式，作詳細說明如下。 、本發明係提供-觀振轉換賴由料振電路之激磁 電感電流以及諧振電容電壓進行預先調整，以便降低輸出 電壓波紋與音頻噪音並提升效率。 _ 第2圖係為本發明之諧振轉換器之一電路示意圖。如 圖所不，諧振轉換器100包括一主電路110以及一控制器 120，其中主電路110包括一方波產生器m、一諧振電路 113以及一輸出整流電路115。方波產生器iu用以提供一 方波電壓至諧振電路113。在此實施例中，方波產生器 係可為半橋式轉換器、全橋式轉換器、推挽式轉換器，但 不限定於此，方波產生器lu亦可為其它型式之轉換器。 牛例而σ方波產生器111係用以接收一輸入電壓vin， 並根據控制器120所產生之控制信號117，將輸入電壓vin φ 轉換成一方波電壓，並將方波電壓提供至諧振電路113。 一般而言，控制器12〇所產生之控制信號117係由複數脈 衝所構成。 5白振電路113係可由諧振元件(例如電感與電容)所構 成’用以接受來自方波產生器U1之方波電壓進行諧振， 而輸出整流電路H5搞接至諧振電路113，用以根據諧振 電路113之諧振產生一輸出電壓v〇。舉例而言，輸出整流 電路115係可為二極體整流電路或同步整流電路，但不限 定於此。控制器120係包括一正常模式控制器121、一間 201128912 :杈式控制态123以及一驅動器】25。控制器12〇用以根 轉換器_中用以反映輪出電流的信號(例如輸出電 f誤差放大信号），判斷贿轉換器之工作模式，並提 :對應之控制錢117至錢產生#⑴。在某些實施例 振轉換11⑽中用以反映輸出电流的信號亦可為譜 '"電谷上的電壓（亦稱為諧振電容電壓）、高頻變壓器中之 電流(：如激磁電感電流或諧振電流)中之一者或多者，但 而言’驅動器120係選擇性地根據正常 、工工制器121與間歇模式控制器！ 號’產^控制信號in輸出至方波產生器ln。振Μ 田。白振轉換器100在滿載或重載情況下，間歇模式控 =器123會被禁能，而正常模式控制器ΐ2ι會被致能二 Η·正常模式控制器121則會根據輸出電壓％的變動，產 ^振魅號，而驅動器12〇則根據正常模式控制器⑵所 生之振號’發出具有連續的驅動脈衝之控制信號m 至方波產生器in。換言之，諧振轉換器1〇〇此時操作在 一正常（工作）模式下。 當諧振轉換器100進入輕載或空载情況時，正常模式 二㈣121會被禁能，而間歇模式控制$ 123會被致能， =2一間歇（工作）模式。此時驅動器125則根據間歇 換式控制& 123所提供之振盪信號，發出具有至少一第一 =衝組以及至少一第二脈衝組之控制信號ιΐ7至方波產生 L/^例而言’第二脈衝組包括一個含有複數驅動脈 7脈衝序列’用以間歇式地導通方波產生器ιη中的開 關元件，但不限定於此。要注意的是，第-脈衝組係位於 201128912 第二脈衝组之前，或者是兩個相鄰的第二脈衝組之間，但 不限定於此。舉例而言，驅動器125可依序提供第一脈衝 組與第二脈衝組至方波產生器ηι。方波產生器iu用以 根據第脈衝組將諧振電路113之激磁電感電流與諧振電 容電壓預先調整至對應的預設值，使得在第二脈衝組中每 個脈衝的上升緣時，諧振電路113中激磁電感電流會大抵 上相等，並且諧振電容電壓也會大抵上相等。換言之，驅 動器125根據第二脈衝組使得諧振轉換器1〇〇如同操作於 正常（工作）模式之某種平衡狀態。須注意的是，諧振轉換 益100於正常模式下具有多種平衡工作狀態，而平衡工作 狀態與負載錢。此外，與第〗圖所示之習知技術不同的 是第二脈衝組是驅動器125藉由間歇模式控制器123根據 固定頻率之振盪信賴產生的。換言之，即使根據輸出電 壓Vo產生的誤差放大信號於滯回比較電路之上、下限值 之間波動，對第二脈衝組也不會有影響。 在某一實施例中，第一脈衝組係包括一個或多個第一 調節脈衝’而第-調節脈衝的寬度與個數係根據譜振電路 113中之諧振疋件的個數來決定，但不限定於此。在某些 實施例中，第-脈衝組係包括—個或多個第 : 及-個或多個第二調節脈衝，並且第-、第二調節脈二 寬度與個數亦根據諧振電路113中之譜振元件的個數來決 定，但不限定於此。在某些實施财，第―、第二調節脈 衝的寬度係可根據方程式計算得知。在某些實施例中，第 -、第二調節脈衝的寬度係可藉由偵_振電路之激磁電 感電》IL、伯振電流以及諧振電容電壓中之一者或多者而進 201128912 行即時地調整，但不限定於此。 由於第一脈衝組已將諧振電路之激磁電感電流與諧振 電容電壓預先調整至預設值，故可避免在第二脈衝組内的 前幾個驅動脈衝週期中產生很大的諧振電流。因此諧振電 流不平衡所引發大的輸出電壓波紋、音頻噪音以及最佳工 作點變動（激磁電感電流磁偏以及無法零電壓切換）.·.等 題亦可一併克服。 第3圖係為本發明之諧振轉換器之一實施例。如圖所 示，諳振轉換器包括一主電路210以及一控制器22〇， 主電路210包括一輸入電容211、一半橋式轉換器212、— 諧振電路213、一高頻變壓器214、_輸出整流電路215以 及一輸出電容216。在此實施例中，輸入電容221用以接 收並儲存輸入電壓Vin，半橋式轉換器212係作為一方波 產生器，用以根據控制器220所提供之控制信號217，將 輸入電壓Vin轉換成一方波電壓，並將方波電壓提供至諧 振電路213。在此實施例中，半橋式轉換器212係由開關 元件SW1與SW2所構成，但不限定於此。舉例而言，半 橋式轉換器212亦可由與二極體並聯連接之絕緣柵雙極性 電晶體(IGBT)、機電開關、微機械開關或是其他的主動半 導體開關所構成。高頻變壓器214、輸出整流電路215以 及輸出電容216係用以作為一輸出整流電路，以便提供輸 出電壓Vo。在此貫施例中，輸出整流電路21 $係由二極體 DSR1與DSR2所構成，但不限定於此，亦可由其它整流元 件所組成，例如同步整流管。 控制器220則包括一半橋驅動器221、一選擇開關 201128912 222、-正常模式控制器223、一間歇模式控制器224、一 時脈振盛器225、-滯回比較電路攻以及一電流檢測電 ，227。控制器220用以根據輸出電壓v〇，判斷諧振轉換 器2〇〇之工作模式，並提供對應之控制信號217至半橋 轉換器2U。正常模式控制器223係由1頻轉換電路咖 以及-回授誤差放大電路迎所構成。舉例而言，回授誤 差放大電路2232係用以根據輸出電壓¥〇與一既定電壓之 電壓差’產生-誤差放大信號Vea，而壓頻轉換電路咖 係則根據誤差放大信號Vea輸出對應之振盪信號。 在滿載或重載情況下，根據輸出電容21^的輸出電 f23vo舍、Γ模式控制器224會被禁能，而正常模式控制器 會破致能而發出連續的振盪信號。在此同時，由於 流檢測電阻227上的電壓會等於或高於滞回比較電路226 的上限值’滯回比較電路226之輸出信號以會控 =222將正常模式控制器223所產生的振盪信號輸出至 221。因此’半橋驅動器則根據正常模式控制 轉f生之振龍號而輸出控制信號217來驅動半橋式 常模；使得難轉換器之主電路2U)操作在正 的。譜振轉換器之主電路2ig在正常模式下 的動作係與習知技術相同，故於此不再累述。 在某些實施例中，滯回比較電路226亦可根據譜振電 上的電壓Vcr(亦稱為諧振電容電壓）、高 激磁電感電流或雜電流）以及回授誤差放 押制選摆Ϊ所產生的誤差放大信號^中之-者或多者來 工&擇開Μ 222’但不限定於此。譜振轉換旨議於正 201128912 常模式下具有多種平衡工作狀態，而平衡工作狀態與負載 有關。如第4圖所示’在正常模式下的某平衡狀態時，諧 振電路213之激磁電感電流1!^會具有一正峰值Immax以 及一負峰值lmmin，而锴振電路213之諧振電容電壓vcr 在激磁電感電流Ilm分別為正峰值Immax與負峰值immin 時具有對應之一第一電壓值Vcrmax以及一第二電壓值 Vermin。 在輕載或空載情況下，根據輸出電容上的輸出電壓 V〇,正常模式控制器223會被禁能，而間歇模式控制器224 會被致能，用以根據時脈振盪器225所發出之具有預設頻 率的時脈產生對應的振盪信號。在此同時，由於電流檢測 電阻227上的電壓會等於或低於滯回比較電路226的下限 脸Γ回比車又電路226之輸出信號228會控制選擇開關222 二:模式控制器224所產生之振盪信號輸出至半橋驅動 ° 。因此’半橋驅動器221則根據間歇模式控制器224 所產生之振i信號輸出控制信號217來驅 ⑽使得魏轉換器之主電路21G操作在間歇 224戶施例中，半橋驅動器221雜間歇模式控制器 ★ 產生之振盪信號而輸出的控制信號217包括至+ 一 組以及至少-第二脈衝組，用以控制半橋式轉換 ° 之開關元件SW1與SW2。第4圖係為譜振轉 之主電路於間歇模式時的工作波形示意圖。如 /SS2係為半橋式轉換器212之開關it件SW1之驅動作 旒’ Vgssl係為半橋式轉換器犯之開關元件 信號，W_振電路213中激磁電感Lm上的激== 12 201128912 電流，Ir係為諧振電路213的諧振電流（諧振電感上的電 流），Vcr為諧振電路213中之諧振電容電壓，ISR為輸出 整流電路215中二極體DSR1與DSR2的導通電流。 要注意的是，在此實施例中，控制信號217係由驅動 信號Vgss2與Vgssl所構成，控制信號217於時間t0至tl 具有一第一調節脈衝Atl，並於時間t2至t3具有一第二調 節脈衝At2，第一、第二調節脈衝Atl與At2係可視為前述 的第一脈衝組，而控制信號217中位於時間tl至t2間的多 • 個驅動脈衝所構成的脈衝序列係可視為前述的第二脈衝 組，但不限定於此。方波產生器212係在時間t0至tl，根 據第一調節脈衝Atl導通半橋式轉換器212之開關元件 SW1，將諧振電容電壓Vcr和激磁電感電流ILM分別由中 間值Vcrmid與0預先調整到第一電壓值Vcrmax與正峰值 Immax。接著，在時間tl至t2内，方波產生器212則會根 據控制信號217的第二脈衝組中之驅動脈衝，依序導通開 關元件SW2與SW1(意即間歇式地導通開關元件SW1與 • SW2)。再者，在時間t2至t3，方波產生器212則根據第二 調節脈衝At2，導通開關元件SW1，再將諧振電容電壓Vcr 和激磁電感電流Ilm分別預先調整到中間值Vcrmid與0。 第一調節脈衝係用以在時間t0至tl對諧振電容電壓 Vcr和激磁電感電流Ilm進行預先調整’使得讀振電路在時 間tl達到平衡的諧振工作狀態（亦稱為一種平衡狀態）。在 此實施例中，所謂平衡的諧振工作狀態即表示諧振電容電 壓Vcr和激磁電感電流ILM平衡，且諧振電容電壓Vcr和 激磁電感電流Ilm與在正常工作模式時相同，並同時保持 13 201128912 了谐振轉換器零電壓切換的特性。換言之，時間11至t2， 在開關元件SW1與SW2之控制信號217中每個驅動脈衝 的上升緣時，諧振電路213中電感電流Ilm大抵上會相等， 譜振電容電壓Vcr也大抵上會相等。 在本實施例中，第一、第二調節脈衝與At2的脈 衝寬度係由下列方程式所計算出的。 Δ/1 ωΐ %This invention relates to power supplies, and more particularly to a resonant converter that employs intermittent control. ~ B [Prior Art] In recent years, power supplies have been moving toward high efficiency, high power density, high reliability, and low cost for the purpose of energy saving and environmental protection. Resonant converters are highly efficient at full load due to their soft switching characteristics and operate in the state of the duty cycle, and are favored by many people. However, the 'resonant converter' is not ideal at light loads. >. In order to overcome this problem, the conventional technique converts the resonant type write operation into a batch operation mode (Burst Mode), thereby reducing the number of switching times and the total loss per unit time, so that the efficiency can be improved. For example, as shown in FIG. 1, in the intermittent mode of operation, when the error amplification signal is equal to or higher than the upper limit value Vref2 of the hysteresis comparison circuit, the voltage frequency converter is enabled to generate The oscillating signal causes the half bridge switching circuit to switch according to the control signals (LVG and HVG). Conversely, when the error amplifying signal Vea is lower than the lower limit value vren of the back-and-forth comparison circuit, the voltage-to-frequency converter is disabled to stop the generated oscillating signal, so that the half-bridge switching circuit has no driving signal. However, intermittent mode of operation control still has some deficiencies. For example, since the error amplification signal Vea fluctuates between the hysteresis comparison circuit and the lower limit values Vref2 and Vref1, and the error amplification signal Vea is inversely proportional to the frequency fosc of the 201128912 oscillation signal, it operates in a single pause mode. In the Burst Mode Working Period (BMWP), the frequency fosc of the vibrating signal changes from low to high as the error amplification signal Vea changes from high to low. In addition, in the first few driving pulse periods in the single intermittent mode duty cycle BMWP, a large resonant current (meaning that the resonant current is unbalanced) is generated due to the reduced impedance of the resonant network, thereby causing a large output voltage. Corrugation, audio noise, and optimum operating point variation (magnetic bias of the magnetizing inductor current and zero voltage switching). SUMMARY OF THE INVENTION Based on the above considerations, there is a need for an intermittent control method that reduces output voltage ripple and audio noise and improves efficiency, and a resonant converter to which the control method is applied. In view of the above, the present invention provides a resonant converter including a square wave generator, a resonant circuit, an output rectifying circuit, and a controller. The square wave generator is for providing a square wave voltage, the resonant circuit is for resonating according to the square wave voltage, and the output rectifier circuit is for outputting an output voltage according to the resonance of the resonant circuit. The controller is configured to provide a control signal to drive the square wave generator in an intermittent mode duty cycle, wherein the control signal comprises at least a first pulse group and at least a second pulse group, and the first pulse group includes at least one first adjustment pulse And the second pulse group includes a complex drive pulse. The square wave generator is configured to pre-adjust a magnetizing inductor current and a resonant capacitor voltage of the resonant circuit according to the first adjusting pulse, so that the resonant converter is excited at a rising edge of each of the driving pulses in the second pulse group 201128912 The magnetic inductor currents are roughly equal, and the resonant capacitor voltages are also roughly equal. The present invention also provides another resonant converter comprising a square wave generator, a resonant circuit, an output rectifier circuit and a controller. The square wave generator is used to provide a square wave voltage, a resonant circuit for resonating based on the square wave voltage, and an output rectifier circuit for outputting an output voltage according to the resonance of the resonant circuit. The controller is configured to provide a control signal to drive the square wave generator in an intermittent mode, wherein the control signal comprises at least a first pulse group and at least two second pulse groups, and the first pulse group is located in the two second pulse groups Between and including at least one first adjustment pulse, and the second pulse group is located after the first adjustment pulse and includes a plurality of drive pulses. The square wave generator is configured to adjust a magnetizing inductor current and a resonant capacitor voltage according to the first adjusting pulse so that the resonant converter has a magnetizing inductor current at a rising edge of each of the driving pulses in the second pulse group They are roughly equal, and the resonant capacitor voltages are also roughly equal. The present invention also provides an intermittent mode control method for a resonant converter, comprising: providing at least a first adjustment pulse for pre-adjusting a magnetizing inductor current and a resonant capacitor voltage of a resonant circuit during an intermittent mode duty cycle And after the first adjustment pulse, providing at least one pulse sequence for intermittently turning on the plurality of switching elements in the square wave generator. The pulse sequence includes a complex drive pulse, and the first adjustment pulse is used to adjust the magnetizing inductor current of the resonant circuit and the resonant capacitor voltage, so that the magnetizing inductor current is substantially greater when the resonant converter rises at the rising edge of each driving pulse in the pulse sequence. The above-mentioned objects, features and advantages of the present invention will become more apparent and obvious. as follows. The present invention provides that the (oscillation) conversion is based on the excitation inductor current of the material vibration circuit and the resonance capacitor voltage to be pre-adjusted to reduce output voltage ripple and audible noise and improve efficiency. _ Figure 2 is a circuit diagram of one of the resonant converters of the present invention. As shown, the resonant converter 100 includes a main circuit 110 and a controller 120. The main circuit 110 includes a square wave generator m, a resonant circuit 113, and an output rectifying circuit 115. The square wave generator iu is used to supply a square wave voltage to the resonance circuit 113. In this embodiment, the square wave generator can be a half bridge converter, a full bridge converter, a push-pull converter, but is not limited thereto, and the square wave generator lu can also be other types of converters. . The sigma square wave generator 111 is configured to receive an input voltage vin, convert the input voltage vin φ into a square wave voltage according to the control signal 117 generated by the controller 120, and provide the square wave voltage to the resonant circuit. 113. In general, the control signal 117 generated by the controller 12 is comprised of a plurality of pulses. 5 The white-vibration circuit 113 can be formed by a resonant element (such as an inductor and a capacitor) for receiving a square wave voltage from the square wave generator U1 for resonance, and the output rectifier circuit H5 is coupled to the resonant circuit 113 for resonance The resonance of circuit 113 produces an output voltage v〇. For example, the output rectifying circuit 115 can be a diode rectifying circuit or a synchronous rectifying circuit, but is not limited thereto. The controller 120 includes a normal mode controller 121, a 201128912: 杈 control state 123, and a driver. The controller 12 is used to reflect the signal of the wheel current in the root converter _ (for example, the output power f error amplification signal), to determine the working mode of the bribe converter, and to mention: corresponding control money 117 to money generation # (1) . The signal used to reflect the output current in some embodiments of the converter 11 (10) may also be the voltage on the spectrum '" the valley (also known as the resonant capacitor voltage), the current in the high frequency transformer (such as the magnetizing inductor current or One or more of the resonant currents, but the 'driver 120 is selectively based on the normal, the workmanship 121 and the intermittent mode controller! The number 'production control signal' is output to the square wave generator ln. Vibrate the field. When the white oscillator 100 is fully loaded or heavily loaded, the intermittent mode control = 123 will be disabled, and the normal mode controller ΐ 2 will be enabled. The normal mode controller 121 will change according to the output voltage %. The generator 12 is activated, and the driver 12 sends a control signal m with a continuous drive pulse to the square wave generator in according to the vibration signal generated by the normal mode controller (2). In other words, the resonant converter 1 is now operating in a normal (operating) mode. When the resonant converter 100 enters a light or no load condition, the normal mode two (four) 121 will be disabled, while the intermittent mode control $ 123 will be enabled, = 2 - intermittent (working) mode. At this time, the driver 125 sends a control signal ιΐ7 having at least a first = punch group and at least a second pulse group to the square wave generation L/^ according to the oscillation signal provided by the intermittent switching control & 123. The second pulse group includes a switching element including a complex driving pulse 7 pulse sequence 'to intermittently turn on the square wave generator i n, but is not limited thereto. It is to be noted that the first-pulse group is located before the second pulse group of 201128912 or between two adjacent second pulse groups, but is not limited thereto. For example, the driver 125 can sequentially provide the first pulse group and the second pulse group to the square wave generator ηι. The square wave generator iu is configured to pre-adjust the magnetizing inductor current and the resonant capacitor voltage of the resonant circuit 113 to corresponding preset values according to the first pulse group, so that the resonant circuit 113 is at the rising edge of each pulse in the second pulse group. The medium magnetizing inductor currents are roughly equal, and the resonant capacitor voltages are also roughly equal. In other words, the driver 125 causes the resonant converter 1 to operate in a certain equilibrium state of the normal (operating) mode in accordance with the second pulse group. It should be noted that the resonant conversion benefit 100 has a variety of balanced operating states in the normal mode, and balances the working state with the load. Further, unlike the conventional technique shown in Fig. 7, the second pulse group is generated by the driver 125 by the intermittent mode controller 123 based on the oscillation of the fixed frequency. In other words, even if the error amplification signal generated based on the output voltage Vo fluctuates between the lower limit value and the lower limit value, there is no influence on the second pulse group. In an embodiment, the first pulse group includes one or more first adjustment pulses 'the width and the number of the first adjustment pulses are determined according to the number of resonance elements in the spectral circuit 113, but It is not limited to this. In some embodiments, the first pulse group includes one or more first and/or one or more second adjustment pulses, and the first and second adjustment pulse widths and the number are also according to the resonance circuit 113. The number of spectral elements is determined, but is not limited thereto. In some implementations, the width of the first and second adjustment pulses can be calculated from the equation. In some embodiments, the widths of the first and second adjustment pulses can be instantaneously entered into the 201128912 by one or more of the magnetizing inductance "IL", the primary current, and the resonant capacitor voltage of the oscillator circuit. Adjustment, but not limited to this. Since the first pulse group has previously adjusted the magnetizing inductor current and the resonant capacitor voltage of the resonant circuit to a preset value, it is possible to avoid generating a large resonant current in the first few driving pulse periods in the second pulse group. Therefore, the large output voltage ripple, audible noise, and optimum operating point variation (magnetoresistance current magnetic bias and zero voltage switching) can be overcome by the resonant current imbalance. Figure 3 is an embodiment of a resonant converter of the present invention. As shown, the resonant converter includes a main circuit 210 and a controller 22, the main circuit 210 includes an input capacitor 211, a half bridge converter 212, a resonant circuit 213, a high frequency transformer 214, and an output. The rectifier circuit 215 and an output capacitor 216. In this embodiment, the input capacitor 221 is configured to receive and store the input voltage Vin, and the half bridge converter 212 is used as a square wave generator for converting the input voltage Vin into a control signal 217 provided by the controller 220. The square wave voltage is supplied to the resonance circuit 213. In this embodiment, the half bridge converter 212 is constituted by the switching elements SW1 and SW2, but is not limited thereto. For example, the half bridge converter 212 can also be constructed of an insulated gate bipolar transistor (IGBT), an electromechanical switch, a micromechanical switch, or other active semiconductor switch connected in parallel with the diode. The high frequency transformer 214, the output rectifying circuit 215, and the output capacitor 216 are used as an output rectifying circuit to provide an output voltage Vo. In this embodiment, the output rectifying circuit 21 $ is composed of the diodes DSR1 and DSR2, but is not limited thereto, and may be composed of other rectifying elements, such as a synchronous rectifier. The controller 220 includes a half bridge driver 221, a selection switch 201128912 222, a normal mode controller 223, an intermittent mode controller 224, a clock oscillating device 225, a hysteresis comparison circuit tap, and a current detecting circuit, 227. The controller 220 is configured to determine the operating mode of the resonant converter 2 according to the output voltage v, and provide a corresponding control signal 217 to the half bridge converter 2U. The normal mode controller 223 is composed of a 1-frequency conversion circuit and a feedback feedback amplifier circuit. For example, the feedback error amplifying circuit 2232 is configured to generate an error-amplifying signal Vea according to a voltage difference between the output voltage 〇 and a predetermined voltage, and the voltage-frequency converting circuit system outputs a corresponding oscillation according to the error amplification signal Vea. signal. In the case of full load or heavy load, according to the output voltage of the output capacitor 21^, the Γ mode controller 224 will be disabled, and the normal mode controller will break the energy and send a continuous oscillating signal. At the same time, since the voltage on the flow detecting resistor 227 is equal to or higher than the upper limit value of the hysteresis comparison circuit 226, the output signal of the hysteresis comparison circuit 226 is controlled = 222 to oscillate the normal mode controller 223. The signal is output to 221 . Therefore, the 'half-bridge driver' outputs the control signal 217 to drive the half-bridge normal mode according to the normal mode control and the main circuit 2U) of the hard-to-converter. The operation of the main circuit 2ig of the spectral converter in the normal mode is the same as that of the prior art, and therefore will not be described here. In some embodiments, the hysteresis comparison circuit 226 can also select a pendulum according to the voltage Vcr (also referred to as a resonant capacitor voltage), the high-excitation inductor current or the impurity current on the spectral vibration. The generated error amplification signal ^ - or more of the work & select Μ 222 ' but is not limited thereto. The spectrum conversion is intended to be positive. 201128912 There are many balanced working states in the normal mode, and the balanced working state is related to the load. As shown in Fig. 4, when in a certain equilibrium state in the normal mode, the magnetizing inductance current of the resonant circuit 213 has a positive peak Immax and a negative peak lmmin, and the resonant capacitor voltage vcr of the resonant circuit 213 is The magnetizing inductor current Ilm has a corresponding first voltage value Vcrmax and a second voltage value Vermin when the positive peak Immax and the negative peak immin are respectively. In the case of light or no load, the normal mode controller 223 is disabled based on the output voltage V〇 on the output capacitor, and the intermittent mode controller 224 is enabled for output from the clock oscillator 225. The clock with the preset frequency generates a corresponding oscillating signal. At the same time, since the voltage on the current detecting resistor 227 is equal to or lower than the lower limit of the hysteresis comparison circuit 226, the output signal 228 of the circuit 226 controls the selection switch 222: the mode controller 224 generates The oscillating signal is output to the half bridge drive °. Therefore, the 'half bridge driver 221 drives the control signal 217 according to the oscillation signal generated by the intermittent mode controller 224 to drive (10) the main circuit 21G of the Wei converter to operate in the intermittent 224 embodiment, and the half bridge driver 221 is in the intermittent mode. The control signal 217 output by the controller ★ generated oscillation signal includes a + group and at least a second pulse group for controlling the switching elements SW1 and SW2 of the half bridge conversion °. Fig. 4 is a schematic diagram showing the operation waveform of the main circuit of the spectral oscillation in the intermittent mode. For example, /SS2 is the driving of the switch SW1 of the half bridge converter 212. The Vgssl is the switching element signal of the half bridge converter, and the excitation of the magnetizing inductance Lm in the W_vibration circuit 213 == 12 201128912 Current, Ir is the resonant current of the resonant circuit 213 (current on the resonant inductor), Vcr is the resonant capacitor voltage in the resonant circuit 213, and ISR is the conducting current of the diodes DSR1 and DSR2 in the output rectifier circuit 215. It is to be noted that, in this embodiment, the control signal 217 is composed of the drive signals Vgss2 and Vgss1, and the control signal 217 has a first adjustment pulse At1 at times t0 to t1 and a second at times t2 to t3. Adjusting the pulse At2, the first and second adjustment pulses Atl and At2 can be regarded as the aforementioned first pulse group, and the pulse sequence formed by the plurality of driving pulses between the time t1 and the t2 in the control signal 217 can be regarded as the foregoing The second pulse group is not limited thereto. The square wave generator 212 is connected to the switching element SW1 of the half bridge converter 212 according to the first adjustment pulse At1, and adjusts the resonant capacitor voltage Vcr and the exciting inductor current ILM from the intermediate values Vcrmid and 0, respectively, to the time t0 to t1. The first voltage value Vcrmax and the positive peak Immax. Then, in time t1 to t2, the square wave generator 212 sequentially turns on the switching elements SW2 and SW1 according to the driving pulse in the second pulse group of the control signal 217 (that is, intermittently turns on the switching element SW1 and SW2). Further, at time t2 to t3, the square wave generator 212 turns on the switching element SW1 according to the second adjustment pulse At2, and adjusts the resonance capacitor voltage Vcr and the magnetizing inductor current Ilm to the intermediate values Vcrmid and 0, respectively. The first adjustment pulse is used to pre-adjust the resonant capacitor voltage Vcr and the magnetizing inductor current Ilm at time t0 to t1 such that the readout circuit reaches a balanced resonant operating state (also referred to as an equilibrium state) at time t1. In this embodiment, the so-called balanced resonant operating state means that the resonant capacitor voltage Vcr and the exciting inductor current ILM are balanced, and the resonant capacitor voltage Vcr and the exciting inductor current Ilm are the same as in the normal operating mode, while maintaining the resonance of 13 201128912. The zero-voltage switching characteristics of the converter. In other words, at time 11 to t2, when the rising edge of each of the driving pulses in the control signals 217 of the switching elements SW1 and SW2, the inductor current Ilm in the resonant circuit 213 is substantially equal, and the spectral capacitor voltage Vcr is also substantially equal. In the present embodiment, the pulse widths of the first and second adjustment pulses and At2 are calculated by the following equations. Δ/1 ωΐ %
Immax (Vin - Vcr max) x Cr x 〇)lImmax (Vin - Vcr max) x Cr x 〇)l
Vcr{tQ) = vin _ - Vcr max)2 -f ^ · ^(^0) = (nVo + Vcr max- Vin) x cos^O x At2)~ Lr xa〇xhn maxx sin(o,〇 x Δ/2) + Vin - nVo 其中，〇代表諧振電容 值，Lr代表諧振電感值，Lm代表激磁電感值，Vin代表輸 入電壓’ η代表高頻變壓器214之一、二次側的匝數比， Immax代表激磁電感電流iLM的正峰值，Vcrmax代表正常 工作模式中對應於激磁電感電流ILM為正峰值時譜振電容 Cr上的電壓值’Vermin代表正常工作模式中對應於激磁電 感電流Ilm為負峰值時错振電容Cr上的電壓值’ωι為第一 個驅動脈衝（第一調節脈衝)作用時諧振電路213的振盪角 頻率’ ω0為最後一個驅動脈衝（第二調節脈衝)作用時譜振 電路213的振盡角頻率，而Vcr(tO)為時間t0時|皆振電容 Cr上的電壓。 在此實施例中，由於時間tl至t2時間歇模式控器224 所提供之振盪信號的頻率不會隨著誤差放大信號Vea的波 14 201128912 動而變化，並且第一脈衝組已將譜振電路213之激磁電感 電流Ilm與諧振電容電壓Vcr預先調整至預設值，故可有 效地避免在第二脈衝組内的前幾個驅動脈衝週期中產生很 大的諧振電流。此外，由於整個工作週期諧振轉換器200 皆操作在平衡的工作狀態，因此可以滿足輸出電壓波紋、 音頻噪音以及輕載高效率的要求。 在一實施例中，由於配置了第一、第二調節脈衝Atl 與At2,在第二脈衝組的前幾個驅動脈衝週期中的諧振電流 • 尖峰會小於1.8倍平衡的諧振工作狀態時的諧振電流Ir， 藉以滿足低輸出電壓紋波和音頻噪音的一般要求。在另一 實施例中，第一、第二調節脈衝Δίΐ與At2係根據前述方 程式所配置，在第二脈衝組的前幾個驅動脈衝週期中的諧 振電流尖峰會小於1.4倍平衡的諧振工作狀態時的諧振電 流Ir，藉以滿足低輸出電壓紋波和音頻噪音的較高要求。 在一較佳實施例中，第一、第二調節脈衝Atl與At2進一 步微調，使得在第二脈衝組的前幾個驅動脈衝週期中的諧 • 振電流尖峰會小於1.2倍平衡的諧振工作狀態時的諧振電 流Ir ’猎以滿足低輸出電屋紋波和音頻α呆音的更南要求。 在第4圖之實施例中，諧振電容電壓Vcr之中間值Vcrmid 係位於第一電壓值Vcrmax與第二電壓值Vermin之間，但 在某些實施例中諧振電容電壓Vcr之中間值Vcrmid亦可高 於第一電壓值Vcrmax或低於第二電壓值Vermin。 第5圖係為間歇工作模式之實施方式。如圖所示，間 歇模式控制器224包括一驅動脈衝同步電路311、一預設 脈衝寬度電路312、一及閘313、死區電路314與315以及 15 201128912 反相时316。時脈振逢器225用以產生具有預設固定頻率 之振i信號，而滯回比較電路226用以設定誤差放大信號 之門限值。驅動脈衝同步電路311用以使得滞回比較電 路330之輪出信號與驅動脈衝達到同步而預設脈衝寬度 電路312用以藉由RC延遲來設定單次諧振週期的第一個 脈衝（第一調節脈衝）和最後一個脈衝（第二調節脈衝）的寬 度。及閘313係用以控制驅動脈衝的狀態，而死區電路314 與315則用以產生開關元件SW1與SW2導通切換的死區 時間。 籲 第6圖係為諧振轉換器之另一實施例。如圖所示，諧 振轉換器300與第4圖中之諧振轉換器200相似，其差異 在於控制信號217中第一、第二調節脈衝ΔΠ與M2之脈 衝見度並非由前述方程式先計算出來的，而是藉由债測諸 振電路213中之激磁電感電流與諧振電容電壓轉換器進行 及時調整的。為簡化說明，諧振轉換器300與第4圖中之 諧振轉換器200相同的元件與其動作於此不累述。如圖所 示’控制器220”包括電感Ld作為一激磁電感電流監測元 # 件來監測諧振電路210之激磁電感電流ILM，並將所測得的 激磁電感電流ILM送入間歇模式控制器224”。此外，控制 器220”亦會監測諧振電路210之諧振電容電壓Vcr，並將 所測得的諧振電容電壓Vcr送入間歇模式控制器224”。間 歇模式控制器224”係根據所測得之諧振電容電壓Vcr和激 磁電感電流ILM，即時地控制第一、第二調節脈衝的脈衝寬 度’使得在控制信號217之第二脈衝組中每個驅動脈衝的 上升緣來到時，諧振電路214中激磁電感電流ILM大抵上 16 201128912 會相等，譜振電容電壓b也大抵上會相等。 合根輕载或空載情況下，正常模式控制器223 曰根據輸出電谷上的輸出電壓v〇被禁能，而間 則會被致能，用以產生對應的脈衝。在此同7 ㈣：檢:電阻227上的電壓會低於滯回比較電路226 的下限值’沛回比較電路226之輸出信號Vcr{tQ) = vin _ - Vcr max)2 -f ^ · ^(^0) = (nVo + Vcr max- Vin) x cos^O x At2)~ Lr xa〇xhn maxx sin(o,〇x Δ /2) + Vin - nVo where 〇 represents the value of the resonant capacitor, Lr represents the value of the resonant inductor, Lm represents the value of the magnetizing inductance, Vin represents the input voltage 'η represents the turns ratio of one of the high-frequency transformers 214, and the ratio of the turns of the secondary side, Immax Represents the positive peak value of the magnetizing inductor current iLM, Vcrmax represents the voltage value on the spectral capacitance Cr when the corresponding excitation magnetizing current ILM is positive in the normal operating mode. 'Vermin represents the negative peak value corresponding to the magnetizing inductor current Ilm in the normal operating mode. The voltage value 'ωι on the offset capacitor Cr is the oscillation frequency ω0 of the resonance circuit 213 when the first drive pulse (first adjustment pulse) acts as the last drive pulse (second adjustment pulse). The oscillation frequency of the oscillation capacitor Cr is obtained when the Vcr(tO) is the time t0. In this embodiment, since the frequency of the oscillating signal provided by the intermittent mode controller 224 does not change with the wave 14 201128912 of the error amplification signal Vea from time t1 to t2, and the first pulse group has the spectral circuit The magnetizing inductor current Ilm and the resonant capacitor voltage Vcr of 213 are previously adjusted to preset values, so that it is possible to effectively avoid generating a large resonant current in the first few driving pulse periods in the second pulse group. In addition, since the resonant converter 200 operates in a balanced operating state throughout the duty cycle, it can meet output voltage ripple, audio noise, and light load high efficiency requirements. In an embodiment, since the first and second adjustment pulses At1 and At2 are configured, the resonance current in the first few driving pulse periods of the second pulse group is less than the resonance of the 1.8 times balanced resonant operation state. Current Ir, to meet the general requirements of low output voltage ripple and audible noise. In another embodiment, the first and second adjustment pulses Δίΐ and At2 are configured according to the foregoing equation, and the resonant current spikes in the first few driving pulse periods of the second pulse group are less than 1.4 times the balanced resonant operating state. The resonant current Ir is used to meet the high requirements of low output voltage ripple and audible noise. In a preferred embodiment, the first and second adjustment pulses At1 and At2 are further fine-tuned such that the harmonic current peaks in the first few drive pulse periods of the second pulse group are less than 1.2 times the balanced resonant operation state. The resonant current Ir' hunting is to meet the more south requirements of low output electric house ripple and audio alpha dull. In the embodiment of FIG. 4, the intermediate value Vcrmid of the resonant capacitor voltage Vcr is between the first voltage value Vcrmax and the second voltage value Vermin, but in some embodiments, the intermediate value Vcrmid of the resonant capacitor voltage Vcr may also be It is higher than the first voltage value Vcrmax or lower than the second voltage value Vermin. Figure 5 is an implementation of the intermittent mode of operation. As shown, the intermittent mode controller 224 includes a drive pulse synchronizing circuit 311, a predetermined pulse width circuit 312, a sum gate 313, dead band circuits 314 and 315, and 15 201128912 inversion 316. The clock oscillating device 225 is configured to generate a sinus i signal having a predetermined fixed frequency, and the hysteresis comparison circuit 226 is configured to set a threshold value of the error amplifying signal. The driving pulse synchronization circuit 311 is configured to synchronize the wheeling signal of the hysteresis comparison circuit 330 with the driving pulse, and the preset pulse width circuit 312 is configured to set the first pulse of the single resonance period by the RC delay (first adjustment) Pulse) and the width of the last pulse (second adjustment pulse). The gate 313 is used to control the state of the driving pulse, and the dead zone circuits 314 and 315 are used to generate the dead time of the switching of the switching elements SW1 and SW2. Figure 6 is another embodiment of a resonant converter. As shown, the resonant converter 300 is similar to the resonant converter 200 of FIG. 4, with the difference that the first and second adjustment pulses ΔΠ and M2 in the control signal 217 are not first calculated by the aforementioned equation. Rather, it is adjusted in time by the magnetizing inductor current and the resonant capacitor voltage converter in the debt measuring circuit 213. To simplify the description, the same elements of the resonant converter 300 as the resonant converter 200 of Fig. 4 and their actions are not described herein. As shown, the 'controller 220' includes an inductor Ld as a magnetizing inductor current monitoring component to monitor the magnetizing inductor current ILM of the resonant circuit 210 and to feed the measured magnetizing inductor current ILM to the intermittent mode controller 224" . In addition, the controller 220" also monitors the resonant capacitor voltage Vcr of the resonant circuit 210 and feeds the measured resonant capacitor voltage Vcr to the intermittent mode controller 224". The intermittent mode controller 224" instantaneously controls the pulse widths of the first and second adjustment pulses based on the measured resonant capacitor voltage Vcr and the exciting inductor current ILM such that each drive in the second pulse group of the control signal 217 When the rising edge of the pulse comes, the exciting inductor current ILM in the resonant circuit 214 will be equal to 16 201128912, and the spectral capacitor voltage b will be equal to equal. The normal mode controller 223 合 under light load or no load. According to the output voltage v〇 on the output valley, it is disabled, and is enabled to generate the corresponding pulse. Here 7 (4): Check: The voltage on the resistor 227 will be lower than the hysteresis comparison circuit 226 The lower limit value of the output signal of the peek back comparison circuit 226
==模:_ 224”所產生信號= 二?二 半橋驅動器221則根據間歇模式 " 所產生之振盪信號而輸出控制信號217來驅動 半橋轉換③212 ’使得譜振轉換器200之主電路210操作 在間歇模式下。 〃 在間歇模式中，半橋驅動器221根據間歇模式控制器 224所產生之振盪信號而輸出的控制信號Η?包括至少一 第一脈衝、组(例如第4圖中之第一、第二調節脈衝姐與組) 以及至少一第二脈衝組(例如第4圖中時間扒至U的複數 驅動脈衝）’用以控制半橋式轉換器212中之開關元件swi 與SW2。在第二脈衝組之前，半橋驅動器221係根據間歇 模式控制器220”所產生之振盪信號輸出一第一調節脈衝 △tl，使得方波產生器212根據第一調節脈衝Δΐ1導通開關 兀件swi ’用以將諧振電容電壓Vcr和激磁電感電流 分別由中間值Vcrmid與0預先調整到第一電壓值Vcrmax 與正峰值immax，以便使得諧振電路213達到平衡的諧振 工作狀態。在此實施例中’第一調節脈衝Δΐ1的脈衝寬度 係由諧振電容電壓Vcr和激磁電感電流lLM調整至第一電 壓值Vcrmax與正峰值immax所需的時間所決定的。在間 17 201128912 歇模式控制器220”判斷出諧振電容電壓Vcr和激磁電感電 流ILM已調整至第一電壓值Vcrmax與正峰值immax之後， 則會結束第一調節脈衝At 1。 接著，半橋驅動器221係根據間歇模式控制器22〇，，所 產生之振盪信號輸出第二脈衝組，方波產生器212則會根 據控制信號2Π中之第二脈衝組，依序導通開關元件通 與swi。再者，在第二脈衝組結束之後，半橋驅動器22i 係根據間歇模式控卿22『所產生之振餘號而輸出一第 二調節脈衝⑽，使得方波產生器212根據第二調節脈衝 △t2導通開關元件sw卜用以將難電容電壓μ和激磁 電感電流1⑽分別預先調整到中間值Vcrmid與〇。在此實 ，例中’第二調節脈衝組的脈衝寬度係由譜振電容電^ cr調整至中間Vermid所需的時間所決定的。在間歇模式 ::·出諧振電容電壓W已調整至中間值Vcrmid之 節脈衝M2係可以省略的，i且中’第二調 感雷产τ ^…電值Vcrmax或激磁電 Γ 1峰值Immax所f的時間所決定的。 感電、、”盘：Γ第—脈衝組已將諧振電路213之激磁電 :，丨L LM，、咱振電容電壓Vcr預先調整 有效地避免在第二脈衝組 至預5又值故可 报大的譜振電流。此外，由於整個^驅動脈衝週期中產生 皆操作在平衡的工作狀態，因此可皆振轉換器· 音觸音以及輕载高效率的要求了乂滿足輸出電壓波紋、 第7圖係為譜振轉換器操作於間歇模式時的工作波形 201128912 示意圖。如圖所示，控制信號在每個間歇模式工作週期 BMWP中皆包括一第一脈衝組(^ti與“2)以及一第二脈衝 組(PS2)。在此實施例中，第二脈衝組PS2係為具有複數驅 動脈衝之一脈衝序列，而第一脈衝組係由位於第二脈衝組 PS2之前的第一調節脈衝Ati以及接在第二脈衝組pS2之 後的第二調節脈衝At2所構成，但不限定於此。第二調節 脈衝M2用以在前一個第二脈衝組pS2結束之後，將諧振 電容電壓Vcr調整到中間值Vcrmid，而第一調節脈衝 • 用以在後一個第二脈衝組PS2開始之前，將諧振電容電壓 Vcr和激磁電感電流iLM分別由中間值Vcrmid與〇調整到 第一電壓值Vcrmax與正峰值Iinmax。由於諧振電路之激 磁電感電流Ilm與諧振電容電壓vcr已經被調整至第一電 壓值Vcrmax與正峰值immax，故可有效地避免在第二脈 衝組PS2内的前幾個驅動脈衝週期中產生很大的諧振電 流，使得諳振轉換器操作在平衡的工作狀態。因此，在第 二脈衝組PS2中每個驅動脈衝的上升緣時，電感電流Ilm •大抵上會相等，諳振電容電壓Vcr也大抵上會相等。要注 意的是’第一、第二調節脈衝的脈衝寬度係可根據方程式 預先計算出，或者是藉由偵測激磁電感電流Ilm與諧振電 容電壓Vcr即時地調整，但不限定於此。在某些實施例中， 控制信號在每個間歇模式工作週期BMWp中亦可以略去第 二調節脈衝M2’而尸、包括—第一脈衝組以及一第二 脈衝組(PS2)。 第8圖係為諧振轉換器操作於間歇模式時的另一工作 波形不意圖。如圖所示，控制信號在每個間歇模式工作週 201128912 期BMWP中皆包括一第一脈衝组ρ$丨（Δ1;1與At2)以及一第 一脈衝組(PS2)。在此實施例中，第二脈衝組pS2係為具有 複數驅動脈衝之一脈衝序列，而第一脈衝組psl係由位於 第一脈衝組PS2之前的第一調節脈衝與第二調節脈衝 △t2所構成，但不限定於此。在後一個第二脈衝組ps2開 始之刖，第一調節脈衝At2用以將諧振電容電壓Vcr調整 到中間值Vcrmid，而第一調節脈衝Ati接著將諧振電容電 壓Vcr和激磁電感電流Ilm分別由中間值Vcrmid與〇調整 到第電壓值Vcrmax與正峰值immax，使得在第二脈衝 _ 組PS2中每個驅動脈衝的上升緣時，電感電流大抵上 曰相等，諧振電容電壓ycr也大抵上會相等。同樣地，第 一、第二調節脈衝的脈衝寬度係可根據方程式預先計算 出或者是藉由偵測激磁電感電流iLM與諧振電容電壓Vcr 即，地調整，但不限定於此。在某些實施例中，控制信號 在每個間歇模式工作週期BMWp中之第一脈衝組亦可 、略去第一調節脈衝At2，而只包括一第—調節脈衝。 第9圖係為諧振轉換器操作於間歇模式時的另一工作籲 波形不意圖。如圖所示，控制信號在每個間歇模式工作週 d MWP中皆包括一第一脈衝組、△" 2、1盥 » " *** - /、 -2)以及一第二脈衝組(pS2)。在此實施例中，第二脈衝 組PS2係為具有複數驅動脈衝之一脈衝序列，而第一脈衝 、、且係由位於第二脈衝組PS2之前的第一調節脈衝與 以及接在第二脈衝組pS2之後的第二調節脈衝△〇—工 與2所構成，但不限定於此。舉例而言，第一調節脈 衝ΔΠ—1與Atl一2可視為一脈衝序列，而第二調節脈衝 20 201128912 △t2一1與At2_2可視為另一脈衝序列。 第二調節脈衝（1與如、2用以在前-個第二脈衝 組PS2結束之後，將諧振電容電壓Vcr調整到中間值 Vcnnid，而第-調節脈衝礼丨與如―2用以在後一個第 二脈衝組PS2開始之前’將譜振電容電壓—和激磁電感 電抓lLM分別由中間值VCrmid與〇調整到第一電壓值 VCrmax與正峰值。由於諧振電路之激磁電感電流 iLM與譜振電容電壓Ver已經被調整至第一電壓值ν_χ 鲁與正峰值Iinmax，故可有效地避免在第二脈衝組ps2内的 前幾個驅動脈衝週期中產生很大的譜振電流，使得譜振轉 換器操作在平衡的工作狀態。因此，在第二脈衝組ρ§2中 每個驅動脈衝的上升緣時，激磁電感電流^大抵上會相 等’错振電容電壓Vcr也大抵上會相等。要注意的是，第 一、第二調節脈衝的脈衝寬度係可根據方程式預先計算 出，或者是藉由偵測激磁電感電流Ilm與諧振電容電壓Va 即時地調整，但不限定於此。第—脈衝組中第—調節闕 的個數與第二調節脈衝的個數係根據諧振電路中諧振元件 的個數來決定，但不限定於此。在某一實施例中，第一脈 衝組中亦可包括更多的第一調節脈衝與第二調節脈衝。在 某些實施例中，控制信號在每個間歇模式工作週期BMwp 中亦可以略去第二調節脈衝Δί2_ 1與Μ2_2，而只包括第一 脈衝組（^1-1與仏1_2)以及第二脈衝組(PS2)。 雖然本發明已以較佳實施例揭露如上，然其並非用以 限定本發明，任何熟知技藝者，在不脫離本發明之精神和 範圍内，當可作些許更動與潤飾，因此本發明之保護範圍° 21 201128912 當視後附之申請專利範圍所界定者為準。== modulo: _ 224" generated signal = 2-4 1/2 bridge driver 221 outputs control signal 217 according to the oscillating signal generated by the intermittent mode " to drive the half bridge conversion 3212' so that the main circuit of the spectral converter 200 210 operates in the intermittent mode. 〃 In the intermittent mode, the control signal output by the half bridge driver 221 according to the oscillation signal generated by the intermittent mode controller 224 includes at least one first pulse, group (for example, in FIG. 4) The first and second adjustment pulse pairs and the at least one second pulse group (for example, the complex drive pulse of time 扒 to U in FIG. 4) are used to control the switching elements swi and SW2 in the half bridge converter 212. Before the second pulse group, the half bridge driver 221 outputs a first adjustment pulse Δtl according to the oscillation signal generated by the intermittent mode controller 220", so that the square wave generator 212 turns on the switch element according to the first adjustment pulse Δΐ1. Swi ' is used to pre-adjust the resonant capacitor voltage Vcr and the magnetizing inductor current from the intermediate value Vcrmid and 0 to the first voltage value Vcrmax and the positive peak immax, respectively, in order to balance the resonant circuit 213 Resonance working state. In this embodiment, the pulse width of the first adjustment pulse Δΐ1 is determined by the time required for the resonance capacitor voltage Vcr and the magnetizing inductor current lLM to be adjusted to the first voltage value Vcrmax and the positive peak value immax. After the 17 201128912 break mode controller 220" determines that the resonant capacitor voltage Vcr and the magnetizing inductor current ILM have been adjusted to the first voltage value Vcrmax and the positive peak value immax, the first adjusting pulse At 1 is ended. Next, the half bridge driver 221 is based on the intermittent mode controller 22, the generated oscillation signal outputs a second pulse group, and the square wave generator 212 sequentially turns on the switching element to the swi according to the second pulse group of the control signal 2Π. After the end of the second pulse group, the half bridge driver 22i outputs a second adjustment pulse (10) according to the generated residual number of the intermittent mode control 22, so that the square wave generator 212 is based on the second adjustment pulse Δt2. The conduction switching element sw is used to pre-adjust the hard capacitor voltage μ and the magnetizing inductor current 1 (10) to the intermediate values Vcrmid and 分别, respectively. In this example, the pulse width of the second adjustment pulse group is determined by the spectral capacitance. Adjusted to the time required for the intermediate Vermid. In the intermittent mode:: · The resonant capacitor voltage W has been adjusted to the intermediate value Vcrmid. The pulse M2 can be omitted, i and the middle The sensible lightning τ ^...electrical value Vcrmax or the excitation electric Γ 1 peak Immax is determined by the time f. The sense of electricity, "disk: Γ first - pulse group has the excitation circuit 213 of the excitation power:, 丨L LM, The shunt capacitor voltage Vcr is pre-adjusted to effectively avoid the spectral current that can be reported in the second pulse group to the pre-5 value. In addition, since the entire ^ drive pulse period is generated in a balanced operating state, the vibration converter, the sound and the light load and high efficiency are required to meet the output voltage ripple, and the seventh image is the spectral conversion. Schematic diagram of the operating waveform 201128912 when operating in the intermittent mode. As shown, the control signal includes a first pulse group (^ti and "2" and a second pulse group (PS2) in each intermittent mode duty cycle BMWP. In this embodiment, the second pulse group The PS2 is a pulse sequence having a plurality of drive pulses, and the first pulse group is composed of a first adjustment pulse Ati located before the second pulse group PS2 and a second adjustment pulse At2 connected after the second pulse group pS2, However, the second adjustment pulse M2 is used to adjust the resonant capacitor voltage Vcr to the intermediate value Vcrmid after the end of the previous second pulse group pS2, and the first adjustment pulse is used in the latter second pulse group. Before the start of PS2, the resonant capacitor voltage Vcr and the magnetizing inductor current iLM are respectively adjusted from the intermediate values Vcrmid and 〇 to the first voltage value Vcrmax and the positive peak Iinmax. Since the exciting inductor current Ilm and the resonant capacitor voltage vcr of the resonant circuit have been adjusted to The first voltage value Vcrmax and the positive peak value immax can effectively avoid generating a large resonance current in the first several driving pulse periods in the second pulse group PS2, so that the ring-vibrating converter operates In a balanced working state, therefore, in the rising edge of each driving pulse in the second pulse group PS2, the inductor current Ilm • will be substantially equal, and the shunt capacitor voltage Vcr will be equal to equal. 1. The pulse width of the second adjustment pulse may be calculated in advance according to an equation, or may be adjusted instantaneously by detecting the excitation inductor current Ilm and the resonance capacitor voltage Vcr, but is not limited thereto. In some embodiments, the control The signal can also omit the second adjustment pulse M2' in each intermittent mode duty cycle BMWp, and the corpse includes a first pulse group and a second pulse group (PS2). Figure 8 shows the resonant converter operating in the interval. Another working waveform in the mode is not intended. As shown, the control signal includes a first pulse group ρ$丨(Δ1; 1 and At2) and a first pulse in each intermittent mode working week 201128912 period BMWP. Group (PS2). In this embodiment, the second pulse group pS2 is a pulse sequence having a plurality of drive pulses, and the first pulse group ps1 is the first adjustment pulse and the second before the first pulse group PS2. Tune The pulse Δt2 is formed, but is not limited thereto. After the start of the second second pulse group ps2, the first adjustment pulse At2 is used to adjust the resonance capacitance voltage Vcr to the intermediate value Vcrmid, and the first adjustment pulse Ati is then The resonant capacitor voltage Vcr and the exciting inductor current Ilm are adjusted from the intermediate values Vcrmid and 〇 to the first voltage value Vcrmax and the positive peak immax, respectively, so that the inductor current is substantially higher than the rising edge of each driving pulse in the second pulse group PS2. Equally, the resonant capacitor voltage ycr is also substantially equal. Similarly, the pulse widths of the first and second trimming pulses can be pre-calculated according to the equation or by detecting the magnetizing inductor current iLM and the resonant capacitor voltage Vcr. Adjustment, but not limited to this. In some embodiments, the first pulse group of the control signal in each of the intermittent mode duty cycles BMWp may also omit the first adjustment pulse At2 and include only a first adjustment pulse. Fig. 9 is another operation of the resonant converter operating in the intermittent mode. As shown, the control signal includes a first pulse group, Δ" 2, 1盥» " *** - /, -2) and a second pulse group in each intermittent mode working week d MWP (pS2). In this embodiment, the second pulse group PS2 is a pulse sequence having a plurality of drive pulses, and the first pulse, and the first adjustment pulse located before the second pulse group PS2 and the second pulse The second adjustment pulse after the group pS2 is composed of two, but is not limited thereto. For example, the first adjustment pulses ΔΠ-1 and Atl-2 can be regarded as one pulse sequence, and the second adjustment pulse 20 201128912 Δt2_1 and At2_2 can be regarded as another pulse sequence. The second adjustment pulse (1 and 2, for example, after the end of the first-second pulse group PS2, adjusts the resonant capacitor voltage Vcr to the intermediate value Vcnnid, and the first-modulation pulse is used as the "2" Before the start of a second pulse group PS2, 'the spectral capacitor voltage—and the magnetizing inductance is grasped by the intermediate values VCrmid and 〇 respectively to the first voltage value VCrmax and the positive peak. Due to the magnetizing inductance current iLM of the resonant circuit and the spectral excitation The capacitor voltage Ver has been adjusted to the first voltage value ν_χ 鲁 and the positive peak Iinmax, so that it is possible to effectively avoid generating a large spectral current in the first few driving pulse periods in the second pulse group ps2, so that the spectral conversion is performed. The device operates in a balanced operating state. Therefore, in the rising edge of each driving pulse in the second pulse group ρ§2, the magnetizing inductor current ^ is substantially equal to the same. The damping capacitor voltage Vcr is also substantially equal. The pulse widths of the first and second adjustment pulses may be pre-calculated according to an equation, or may be adjusted instantaneously by detecting the magnetizing inductor current Ilm and the resonant capacitor voltage Va, but are not limited thereto. The number of the first adjustment enthalpy and the number of the second adjustment pulses in the first pulse group are determined according to the number of resonance elements in the resonance circuit, but are not limited thereto. In one embodiment, the first pulse The first adjustment pulse and the second adjustment pulse may also be included in the group. In some embodiments, the control signal may also omit the second adjustment pulses Δί2_ 1 and Μ 2_2 in each intermittent mode duty cycle BMwp, and Only the first pulse group (^1-1 and 仏1_2) and the second pulse group (PS2) are included. Although the invention has been disclosed in the preferred embodiments as above, it is not intended to limit the invention, any skilled artisan, The scope of protection of the present invention is determined by the scope of the appended claims, which is defined by the scope of the appended claims.
22 201128912 【圖式簡單說明】 第1圖係為習知諧振轉換器於間歇模式時的工作波形 示意圖。 第2圖係為本發明之諧振轉換器之一電路示意圖。 第3圖係為本發明之諧振轉換器之一實施例。 第4圖係為諧振轉換器之主電路於間歇模式時的工作 波形示意圖。 第5圖係為間歇工作模式之實施方式。 ❿ 第6圖係為諧振轉換器之另一實施例。 第7圖係為諧振轉換器操作於間歇模式時的工作波形 示意圖。 第8圖係為諧振轉換器操作於間歇模式時的另一工作 波形不意圖。 第9圖係為諧振轉換器操作於間歇模式時的另一工作 波形示意圖。 【主要元件符號說明】 100、200、300〜諧振轉換器；110、210〜主電路；120、 220、220”〜控制器；111〜方波產生器；113、213〜諧振電路； 115、215〜輸出整流電路；117、LVG、HVG、217〜控制信 號；121、223〜正常模式控制器；123、224、224”〜間歇模 式控制器；125〜驅動器；211〜輸入電容；212〜半橋式轉換 器；214〜高頻變壓器；216〜輸出電容；221〜半橋驅動器； 23 201128912 222〜選擇開關；225〜時脈振盪器；226〜滯回比較電路；227〜 電流檢測電阻；228〜輪出信號；2231〜壓頻轉換電路；2232〜 回授誤差放大電路；311〜驅動脈衝同步電路；312〜預設脈 衝寬度電路；313〜及閘；314、315〜死區電路；316〜反相器； SW1、SW2〜開關元件；DSR1、DSR2〜二極體；Vin〜輸入 電壓；Vo〜輸出電壓；Cr〜諧振電容；Vcr〜諧振電容電壓；22 201128912 [Simplified Schematic] Figure 1 is a schematic diagram of the operating waveform of a conventional resonant converter in intermittent mode. Figure 2 is a circuit diagram of one of the resonant converters of the present invention. Figure 3 is an embodiment of a resonant converter of the present invention. Fig. 4 is a schematic diagram showing the operation waveform of the main circuit of the resonant converter in the intermittent mode. Figure 5 is an implementation of the intermittent mode of operation. ❿ Figure 6 is another embodiment of a resonant converter. Figure 7 is a schematic diagram of the operating waveform of the resonant converter operating in the intermittent mode. Figure 8 is another operational waveform when the resonant converter is operating in the intermittent mode. Figure 9 is a schematic diagram of another working waveform when the resonant converter is operating in the intermittent mode. [Main component symbol description] 100, 200, 300 ~ resonant converter; 110, 210 ~ main circuit; 120, 220, 220" ~ controller; 111 ~ square wave generator; 113, 213 ~ resonant circuit; 115, 215 ~ output rectifier circuit; 117, LVG, HVG, 217 ~ control signal; 121, 223 ~ normal mode controller; 123, 224, 224" ~ intermittent mode controller; 125 ~ driver; 211 ~ input capacitance; 212 ~ half bridge Converter; 214~high frequency transformer; 216~output capacitor; 221~half bridge driver; 23 201128912 222~select switch; 225~clock oscillator; 226~ hysteresis comparison circuit; 227~ current sense resistor; Turn-out signal; 2231~voltage-frequency conversion circuit; 2232~ feedback error amplification circuit; 311~ drive pulse synchronization circuit; 312~preset pulse width circuit; 313~ and gate; 314, 315~ dead zone circuit; Phase device; SW1, SW2~ switching element; DSR1, DSR2~diode; Vin~ input voltage; Vo~ output voltage; Cr~resonant capacitance; Vcr~resonant capacitor voltage;
Ilm〜激磁電感電流；Immax〜正峰值；immin〜負峰值；Ilm~excited inductor current; Immax~positive peak; immin~ negative peak;
Vcrmax〜第一電壓值；Vermin〜第二電壓值；vcrmid〜中間 值；Vgssl、Vgss2〜驅動信號；Lm〜激磁電感；ιΓ〜譜振電流； Isr〜導通電流；PS1〜第一脈衝組；PS2〜第二脈衝組；Vea〜 誤差放大信號；Vrefl〜下限值；Vref2〜上限值；頻率； Ld、Ls〜電感，BMWP〜間歇模式工作週期；μ、Μ 1 △tl_2〜第一調節脈衝；At2、At2 1 ' 〜第二調節脈衝。Vcrmax~first voltage value; Vermin~second voltage value; vcrmid~intermediate value; Vgssl, Vgss2~ drive signal; Lm~exciting inductance; ιΓ~spectral current; Isr~ conduction current; PS1~first pulse group; PS2 ~ second pulse group; Vea ~ error amplification signal; Vrefl ~ lower limit; Vref2 ~ upper limit; frequency; Ld, Ls ~ inductance, BMWP ~ intermittent mode duty cycle; μ, Μ 1 Δtl_2 ~ first adjustment pulse ; At2, At2 1 '~ second adjustment pulse.
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