SU1659890A1 - Measuring converter of active power - Google Patents

Abstract

The invention relates to information-measuring technology and allows the conversion of active AC power in three-phase and single-phase circuits into a proportional analog signal — direct current or constant voltage. The aim of the invention is to improve the accuracy. The goal is achieved by introducing a third measuring element, the second sum. The device relates to information and measuring technology and can be used to measure the active power in three-phase circuits. The purpose of the invention is to improve the accuracy of the measuring converter of the active power. FIG. 1 shows a block diagram of the device; in fig. 2 - functional scheme. direct current inverting amplifier and large-scale resistor, as well as making the source and reference voltages two-polar and forming new connections. In the measuring elements, the input voltages are converted into high-frequency rectangular pulses of constant duration, modulated by the input voltages in frequency, which are transmitted through blocks of galvanic separation of pulse signals. These pulses control the work of the flip keys of the switching output currents of the device input current converters. The output currents of the keys are summed by two summing DC inverting amplifiers; From the resulting total current, the output filter of the constant component selects a constant component proportional to the measured active power. The device also contains resistive adders, integrators, comparators and single-oscillators, amplitude modulators. In the description of the invention, the construction schemes of the device blocks are given. 1 sec. and 6 з.п f-ly. 2 Il. The device contains (Fig. 1) measuring elements 1-3, the source of the reference voltage 4, the first and second summing inverting amplifiers 5. 6, DC, the output filter 7 constant component, the scale resistor 8. The first inputs of the first - third measuring elements 1-3 are connected to the corresponding phase voltages; their combined second inputs are connected to the first output of the source 4 of the reference voltages. (L S ON SL U 00 O O

Description

connected by the second output to the fourth inputs of the measuring elements 1-3.

The third inputs of the measuring elements 1-3 are connected to current sensors of the respective phases. The combined first outputs of the measuring elements 1-3 are connected to the input of the first summing inverting amplifier 5 DC, the output connected to the input of the output filter 7 constant component. The second outputs of the measuring elements 1-3 are connected to the input of the first or second summing integrating amplifiers 5, 6 of direct current. The third outputs of the measuring elements 1-3 are connected to the input of the second summing inverting amplifier 6 of direct current, the output of which is connected to the input of the first summing integrating amplifier 5 of direct current through the scale resistor 8. The composition of each of the measuring elements includes an integrator 9 (10) direct current, a comparator 11 (12, 13), a one-shot 14 (15, 16), an amplitude modulator 17 (18, 19), a block 20 (21, 22) galvanic separation of pulse signals, current converter 23 (24, 25), reversible switch 26 (27, 28) and compensating resistor 29 (30, 31), resistive adder 32 (33.34), DC integrator 35.

The comparator 11 (12, 13) and the one-shot 14 (15, 16) of each measuring element are combined into one circuit (FIG. 2) containing an RS flip-flop 36 (37, 38), an RC circuit 39 (40, 41) of the integrating type and dongle 42 (43, 44). The block 20 (21, 22) of galvanic separation of pulse signals of each measuring element contains a logic inverter 45 (46, 47), a limiting resistor 48 (49, 50), a three-winding differentiating transformer 51 (52, 53) with a primary winding 54 (55 , 56) and two secondary 57 (58, 59) and 60 (61, 62), two diodes 63, 64 (65-68) and output RS-flip-flop 69 (70, 71).

A three-winding transformer 72 (73, 74) with a primary winding 75 (76.77), a winding 78 (79, 80) feedback and an output winding 81 (82.83) are included in the current converter 23 (24, 25) of each measuring element. ), the amplifier 84 (85,86) AC and the load resistor 87 (88,89). Each of the two summing inverting amplifiers 5, 6 of the direct current includes an operational amplifier 90 (91). feedback resistor 92 (93) and two filter capacitors 94, 95 (96, 97).

The source of reference voltages consists of a parametric dc regulator 98 and a constant voltage inverter 99.

All three measuring elements are identical; therefore, we consider one of them, for example, the first one. The first input of the resistive adder 32 (Fig. 1) is connected to phase A, the second input is connected to the first output of the source 4 of reference voltages, the common output of which is connected to the zero phase, the third to the output of the amplitude modulator 17. The output of the resistive adder 32 is connected to the input of the DC integrator 35, to the output of which the input of the comparator 11 is connected. The output of the comparator 11 is connected to the input of a single vibrator 14, the output of which is connected to the control input of the amplitude modulator 17. The signal input of the amplitude modulator is connected to the second output source of opposite polar reference voltages. The input unit 20 of the galvanic separation of pulse signals is connected to the output of the one-shot 14.

A current is applied to the input of the converter 23, which loads the phase A, the output of the current converter is connected to a movable contact of the toggle switch 26, the control input of which is connected to the output of the galvanic separation unit 20 of the pulse signals.

The first fixed contact of the switch 26 is connected to the input of the first summing inverting amplifier 5 DC, the output of which is connected to the output of the output filter 7 of the constant component, the output of which produces the output signal of the device - a direct current of 1st or a constant voltage. A compensating resistor 29 is connected between the converter 23 and the input of one of the summing inverting amplifiers 5 or 6 (depending on the parameters of the circuit).

The second fixed contact of the toggle key 26 is connected to the input of the second summing inverting amplifier 6 dc, the output of which is connected to the input of the first summing inverting dc amplifier through a scale resistor 8.

To a single output RS-flip-flop 36 (Fig. 2). part of the one-shot 14 and the comparator 11, is connected to the RS-circuit 39 of the integrating type, the output of which is connected to the input R of the flip-flop. The trigger input S is connected to the output of the DC integrator 35. Parallel to the capacitor of the RC circuit, the bit switch 42 is turned on, the control input of which is connected in such a way that the switch is open when the trigger state is single. If the key is closed by signal 1, the control input must be connected to the output Q of the RS flip-flop, and if it is not present, to the output of the logic inverter 45 connected to the single output of the flip-flop.

Amplitude modulator 17 contains two CMOS logic repeater (level converter) elements, connected in parallel to the inputs and outputs. The power input of the voltage repeater, which represents the signal input of the amplitude modulator, is connected to the second (positive) output of the source of alternating reference DC voltages, the common point of the voltage repeater is connected to the common output of the source. In parallel, the combined inputs of both elements form the control input of the amplitude modulator, and the parallel combined outputs form its output. Since the CMOS voltage follower contains six identical elements, the four remaining elements, combined in pairs in parallel, form amplitude modulators 18 and 19 of the second and third measuring elements. The signal inputs of these modulators, as well as of the amplitude modulator 17, are the only input of the CMOS voltage follower and in FIG. 2 not shown. Similarly, the common point of the amplitude modulators 18, 19, connected to the common point of the modulator 17 inside the voltage follower, is not shown either.

The matching unit 20 for galvanic separation of pulse signals consists of two serially connected logical inverters, between the outputs of which the primary winding 54 of the differentiating transformer 51 is connected in series with the limiting resistor 48. The two secondary windings 57 and 60 oppositely connected with each other and with a common point the output part of the circuit, the anodes of diodes 63 and 64 are connected to their second ends. The cathodes of the diodes are connected to the inputs S and R of the output RS-flip-flop 69.

The primary winding 75 of the transformer 72, which is part of converter 23, is connected in series to the current circuit IA, which loads the phase A. The output winding 81 of the transformer is connected to the input of amplifier 84, the output of which is connected to a compensating resistor 29 and feedback winding 78, c the second end of which is connected to the load resistor 87.

The second end of the resistor is connected to a moving contact of the toggle key 26. A feedback resistor 92 (93) included in each of the two summing inverting amplifiers 5, 6 of direct current is connected between the output and the inverting input of the operational amplifier 90 (91). Parallel to the resistor 92 (93), a filter capacitor 95 (97) is turned on, and a filter capacitor 94 (96) is connected between the inputs of the operational amplifier.

The source 4 of the alternating reference DC voltages consists of a parametric stabilizer 98 and an inverting follower 99 of a constant voltage connected to its output. The first source output (negative) is the output of the inverting repeater 99, the second (positive) output of the parametric stabilizer 98.

FIG. 1, 2 designate phases A, B, C and zero phase 0 of a three-phase circuit; phase

5 voltages Od, UB, Uc, phase currents IA, IB, ic, output voltages U0ni and Uon2, respectively, taken from the first and second outputs of the source of opposite polar reference voltages; Ri, R2, Hz

0 resistances of resistors included in the resistive adder 32 and connected to its first - third inputs, respectively (exactly the same are the resistances of the corresponding resistors of adders 33 and 34), IRI, IRJ, IRS - currents flowing through these resistors, and and ii is the input current and output voltage of the DC integrators 35, Ci is the capacitor capacitance in the feedback circuit of integrator 35 (the capacitors in the feedback circuit of integrators 9 and 10 are exactly the same), R is the resistance of the resistor 93 feedback (Fig. 2 ) the second summing inverting amplifiers of 5 l 6 dc current, RS resistance of the scale resistor 8, JRS is the current flowing through the scale resistor 8. IAI is the output current of the current converter 23, IAI and IAI are the currents flowing through the first and

0 second fixed contacts, respectively, of key 26 (they correspond to currents IBI, lei. IBI, C1, icb.ici in measuring elements 2 and 3), RG is the resistance of the feedback resistor 92 of the first summing in5 rotating amplifier 5 of direct current, R - the resistances of the same compensating resistors 29-31 of the measuring elements 1-3, Re are the same resistances of the load resistors 87-89 included in the current transducers 23-25. Rg and Ca - Resistor Resistance

and the capacitor capacitance is time for giving the RC circuit 39, UAL UBI, uci - the voltage at the output of the amplifiers 84-86, included in the current transducer 23-25, 1V, and output - the output current and voltage of the device. In addition, point a in FIG. 1 denotes an inverting input of a DC integrator 35,

The device works as follows.

First consider the operation of the measuring element 1. The phase voltage OA in the resistor RI of the resistive adder 32 (Fig. 1) creates a current IRI, the instantaneous value of which, given that point a is potentially grounded, is equal to

° A

iR1 RT

(one)

Through the resistor R2 of the adder 32, a direct current IR2 flows through the voltage Uoni at the first output of the source 4 reference voltages, equal to

 Const

(2)

We assume that the voltage Uoni is negative. Resistor R2 is selected such that the current IR2 exceeds IRI at all possible values of voltage UA.

The current through the resistor Ra of the adder 32 does not flow if the amplitude modulator 17 connects this resistor to the common point of the circuit (the lower key of the amplitude modulator 17 is closed, and the upper one is open). Let's call this tact of operation of the measuring element 1 first.

In the second cycle, the amplitude modulator 17 with the upper key (lower open) connects the resistor Ra to a constant positive voltage Uon2, taken from the second output of the source 4 of the reference voltages. Through the resistor R3 at the same time a current №3 flows, equal to

  const

(3)

The resistor Ra is chosen so that the current of the IRS exceeds the sum of the currents IRI and RZ at all possible values of the voltage UA, i.e.

where UKO is the voltage at the integrator output at.

The voltage increment over time t ti. equal to the duration of the first bar,

di

AI (d-imo | (| «, + 1g) H ;. (6) o

At t ti, the voltage UM (t) reaches

The trigger level of the comparator 11. A pulse appearing at its output triggers a one-shot 14, controlling the operation of the amplitude modulator 17, which from this moment connects the resistor Pz to the voltage + Uon2. The second cycle of the circuit begins. Since inequality (4) is satisfied, the resulting output current 1 and adder 32, now equal to (iRi + lR2), changes

its direction, as a result of which the output voltage and integrator 35 begins to decrease. The duration of the pulse produced by the one-shot 14 is determined by its parameters and is equal to T.I.

The change in voltage at the output of the integrator DI2 during the time 1IA is determined by analogy with (5) and (6) expression. i.H.

all

" WITH

U, "". ()

After the end of the one-shot pulse, the first cycle of operation begins again, the amplitude modulator 17 switches to the initial position, the input current 1I of the integrator 35 takes the initial value 1- (r + fR2 and the output voltage of the integrator starts to increase again in accordance with (5) In the following, the processes in the circuit are repeated, and the TA period of the circuit operation is equal to

 1 + 1A. (8)

The increment of voltage during the first cycle of D1Ln is equal to the decrease in Ai2 during the second. Equating (6) and (7), considering that

"-." "And 4 ,,

one . 1 .t ,,,

lt | 0

llR3t l "Rll + IIR2I. (4)

The resulting output current of the adder 32 is the input current 1 and the DC integrator 35. During the first cycle, the current is 1i, equal to lM iRi + lR2. is directed in the same direction as the current IR2 and the voltage and at the output of the integrator 35, the gain, varies according to the law

. “And. ,,, (5)

and solving the resulting equation for TA 1 / Td 1 / (ti + 1id) we get

(.tg

{t; -i r

Where - i,, „t (9)

 | | in the average for the period TA operation of the circuit value of the current IRL

Substituted in (9) the values IRI, IR2, IRS. from (1) - (3), we find

Sfc / Ie; .- 1--1 (Yu)

where ",., - Ј J (to

the average for the period TA of the circuit, the value of the voltage UA;

Uoni Ra ° U0n2 R2 tMA (12) is a constant component of the frequency fA;

f- Uon2RR3, tnA UACP “;

variable frequency component fA.

The sign of fA in formula (10) changes as the sign of UA changes. During the half-period when UA is negative, the directions of the currents IRI and IR2 coincide, and the frequency THA increases (f A in formula (10) has a plus sign); in the next half-period, when UA is positive, the current IRI changes its direction, and the frequency fA decreases (BA in the formula (10) has a minus sign). The circuit parameters are chosen in such a way that the frequency fA at any moment in time significantly (hundreds of times) exceeds the frequency of the input voltages and currents of the device. As a result, it can be assumed that the mean value of the current circuit for the given period Td of operation of the considered circuit is the value of the voltage UA equal to its instantaneous value, i.e.

UAcp "UA. (13a)

The output pulses of the one-shot 14 with the duration tMA and frequency fA are fed to the input of the galvanic separation unit 20 of the pulse signals, the output pulses of which have the same T.IAA duration and frequency fA, but are formed relative to the common point of the output part of the circuit, galvanically isolated from the input phase stress od, ov, os.

The phase current d is fed to the input of the current converter 23, the output current IAI of which, galvanically isolated from the current 1d, contains in general a constant component of the IAO, not dependent on the current 1d, and a variable 1d proportional to IA, i.e.

iA1 lAO + iA1, (14)

where is IAI (15)

K - coefficient of proportionality. Current IAI flows through the movable and that of the fixed contacts of the cross over key 26, which is currently connected to the movable. The toggle key 26 is controlled by the output pulses of the galvanic separation unit 20 of the pulse signals having a duration tMA and a frequency of following TA (10). During the action of the pulse, the movable contact is connected to the first (upper) fixed contact,

during the pause with the second (bottom). As a result, during a pulse, the current IAI passes to the input of the first summing inverting amplifier 5 DC, and during

pause time to the input of the second amplifier 6. Find the average value of the currents passing through the first and second fixed contacts. The average for the period TA output pulses of the one-shot 14 current value

1d1sr, flowing through the first fixed contact, taking into account (14, 15) is equal to

()

b bijV,

five

Where

,,,,

i "

.one . ,,

I cf

(sixteen)

(M)

average for this interval hid value of the phase current 1d.

The average for the period TA the value of the current iAtcp flowing through the second fixed contact is equal to

Where

; l 1

  ъ

t "1 ((T -t" "ntr" "

oi.l ", to, -;" p,

(18)

(nineteen)

the current average for this interval (TD-Guide), the interval (TD-1id) is adjacent to the previous interval T.I.A.

The current IAI is one of the input currents of the second summing DC inverting amplifier. In the resistor scale 8.

RS, this current creates the average for the TA period the current value of IRScpA. equal to

iRScpA 1D1sr -rG (20)

The iRScpA current together with the iAicp current (16) is summed by the first summing DC inverting amplifier 5, the average for the TA period, the component of its output voltage Ui, due to the operation of measuring element 1 alone, connected to phase A, is equal to

(U,) R6-.-Xup-sfRe. (21)

ift

Substituting in (21) the values of iAicp and 1d1sr from (16) and (18) and taking into account (10), after transformations, we obtain

and | ..т1- о.и (R

 . “$

eleven

 C-

rk, LDP iOi; mR-17) - (22) The currents 1dCr and 1d sr in (22), as can be seen from (17) and (.19), are the average for the two adjacent time intervals tMA and ( TA-T.IAA) input current 1d. Since the frequency fA significantly (by hundreds of times) exceeds the frequency of the current) A, the values of 1dsr and 1d sr can be considered equal to each other and the instantaneous value of the current IA. those. 1dsr 1dsr IA. (23)

An additional error may occur due to the presence of switching emissions that occur when switching the toggle key (26-28), working with a high frequency of TA (fe. Fc). The constant component of these emissions and its instability can cause instability of the output signal of the device. In the scheme, however, switching surges caused by switching a key from one position to another are largely mutually compensated by the presence of summing inverting amplifier 6. Indeed, the constant component of the current caused by emissions from switching the second (lower) fixed contact through the inverting amplifier 6. changes its direction and turns out to be opposite to the constant component of the current, due to the emissions that occur when switching (Top) of the fixed contact, resulting in a mutual cancellation of the two permanent components. In the case of the complete identity of both contacts, the constant components due to the outliers are fully mutually compensated when the condition

R4-R5. (24)

In this case, having passed through the inverting amplifier 6, the constant component of the current, caused by the emissions that occur when switching the second fixed contact, does not change its value in the resistor RS. The keys that are part of one integrated circuit are very close in their parameters to such keys, and each of the toggle keys 26-28 should be built. Substituting (23) and (24) into (22), we obtain

% i-MM (-2twtM) -2RfI ,, e,

 L "LM -" "M4„ 1- (25) ..

Substituted in (25) f0A from (12) and f A from (13) and

taking into account (13a),

find% ,. in, 1 ,, (I) -

2R,

12

. (.1t.1 Mi

(26)

 them

The first term (26) is a constant component of the UL 5 voltage. Due to the instability of the current 1 to this component, the passage through the output filter 7 of the constant component can lead to additional error.

10 The second and third terms, proportional to the input voltage, id and current, d, do not contain a constant component and lead to a change in the output signal of the filter 7, a constant component, which cannot be changed.

Since, however, they have a frequency of input signals, for their suppression it is necessary to significantly increase the time constants, and hence the inertia

20 of the output filter 7 constant. Usually, the current 1d0 is small and the third term is predominant of these two.

The fourth term is proportional to 25 instantaneous power.

RA UAiA, (27)

contains a constant component proportional to the active power

so -p "4jwt, (28)

where T is the period of the input signals, and the variable component suppressed by the filter 7. In the case of purely sinusoidal input signals, the variable component is the second harmonic, much more efficiently supplied by the filter compared to the main one.

Thus, in order to improve the accuracy and speed of the device, the first

40 and the third term of formula (26) must be eliminated, for which it is sufficient to vanish the expression in parentheses for these terms, the same for both terms. To do this, according to (26) and (25), the following condition must be met:

)

or

Wom ra f

aunt ° A1A

R2 2Ra

Wonl Won2

(29a)

The value of the expression on the left-hand side of (29) depends on the ratio of the resistances of precision resistors, which can be maintained with high accuracy without an individual selection of resistors, and on the ratio of output reference voltages Uoni and Uon2, produced by source 4 of the reference voltage. If both voltages are obtained with two different precision-zener diodes, due to the significant variation of the stabilization voltage, the resistors R2 and RS will have to be selected each time to accurately fulfill the condition (29).

To eliminate the need for selecting resistors, the source 4 of the reference voltage is made in the form (Fig. 2) of a parametric stabilizer 98 on a precision zener diode, the output of which produces a positive voltage 11bp2, and a inverter 99 constant voltage, the output of which is a negative voltage Uoni, with a high precision Uon2. Polaga, (26) and (29) rewritten in the form: R

U.cp -R6I4 (-JrHR tMU-iirU "

. ". (G; | -" "nd zo)

t: -.... (3D

Under conditions (29a) and (31), from (26) and (30) we get

"

ultpA-4M ou r - ..

"From KI

(32)

The output signal of the device is proportional to the constant component of the voltage UL allocated by the filter 7, which suppresses the variable component UL Component of the output signal of the filter, due to the voltage uicpA, taking into account (28), as well as the fact that the first term (32 a) constant component does not contain, is equal to

i. «.- s4Ju« f. «dt 2k K CTHu; dt- (33)

where Kf sop81 - coefficient of proportionality, defined by the parameters of the filter 7,

 (33a)

Condition (31) cannot always be fulfilled, since it limits the maximum frequency fAmax to

fAmax 2foA, (34)

at which

TAmln yfAmax 1 and A.

In the general case, when the condition (31) is undesirable or impossible to fulfill for some reason, in order to fully compensate the first and third terms in the formula (30), it is necessary to introduce a compensating resistor 29 (30, 31). It is turned off between the output of a current converter 23 (24, 25), where a voltage signal UAI, (UBI, Uci) proportional to the output current K1d OBI, lei) of the current converter, and the input of one of the summing inverting amplifiers 5 can be obtained. current, depending on the sign of the first and third terms in (25) and (30). If these terms are positive, i.e. if the condition is met

2 2AtnA 1,

(35)

to compensate them completely, it is necessary to obtain a compensating voltage UKA with a negative sign, proportional to the current IAI, and sum up this voltage with the voltage UicpA (30). In order to obtain the required voltage sign UKA, it is sufficient to invert the voltage UAI by supplying it through a compensating resistor 29 to the input of the first summing inverting amplifier 5 DC and appropriately choosing the resistance R of the resistor 29. Let

UA1 K2iA1 K2 (lAo + KllA). (36)

where K2 const is the proportionality coefficient.

At the output of the first summing inverting amplifier 5, the voltage UAI will give the term

U ,, - UA1K / RT-4 (,) (37)

The condition for full compensation of the first and third terms in (30) will be

and .. + (g. "- L1 L-thЈI

Substituting here the UKA value from (37), we find the resistance R of the resistor 29 required for full compensation

R7 1 - (2R3 / R2) (38)

Condition (38) does not depend on the value of the current IAI and can be accurately fulfilled for all values of the input signals.

If the first and third terms of (25) and (30) are negative, i.e. if the condition is met

2 Shch 2 foA 1 and A 1

(39)

five

the sign of the voltage compensating their UKA must change, for which the compensating resistor 29 should be connected to the input of the second summing inverting amplifier 6. Then for UKA with

“Me„ R6. , .. . R

4ft

At

R7r

U

Ai

R7

 K2 (I

Ao HA -fj-- (40)

The condition for full compensation of the first and third terms in (30) will be

and "(and". nI (. (41)

The value of R required from (41) and (40) for full compensation is determined by the modulus of expression (38).

The ratios obtained above were related to the measuring element 1, the inputs of which are supplied with voltage and current of phase A.

Since the measuring elements 2 and 3 are completely identical to element 1 according to their scheme and inclusion, the components SchcrrB and uicpc of the voltage ui at the input of the filter 7 created by them can be found by the formulas similar to (32):

Uupb-srjso

and, (42)

u. ,, .. ()

The output filter 7 of the constant component allocates the sum of the constant components of the voltages uicpA, uicpB and uicpc. By analogy with (33) we have

t-JW t Bbll C f I you 6 + I EX

-K, (4iu., EpBJt4 | u, efedtV

-K (p ° vp a-kpc., - (44)

where K is determined from (ЗЗа), Ra PA + Рв + Рс is the power of a three-phase four-wire circuit.

Thus, the output signal of the device is proportional to the active power of the three-phase circuit.

In the case of a three-wire three-phase circuit, the power measurement is carried out according to a two-element scheme that differs from that considered using only two measuring elements. In this circuit, the common output of the source 4 of the reference voltages is connected to one of the phases of the three-phase circuit, the other two phases are connected to the first inputs of the resistive adders of the remaining measuring elements, and the currents of the same phases are connected to their current transducers.

In the case of power measurement in a single-phase circuit, only one measuring element is used, with the voltage of the measuring circuit being connected between the first input of the resistive adder of this element and the common output of the source of alternating reference DC voltages, and the current of the measuring element is supplied circuit.

As can be seen from the formula (10), the frequency TA (fe, fc) of the output pulses of the one-shot

14 (15, 16) is independent of the capacitance Ci of the capacitor included in the DC integrator 35, and the response voltage of the comparator 11 (12, 13). Voltage

the zero offset and input currents of the operational amplifier included in the integrator circuit 35 affect only the frequency f0A (foB, foe) in expression (10) and do not affect f A (f b, f s). At the same time as

can be seen from the expressed (25), (30), (42), (43), (44), (13), when the condition (29a) (31) or (38) is fulfilled, the output signal of the device (first, second) is determined only f A (f in, fc) and does not depend on foA (foB, foe), as well as on the duration T. IA (giv, tnc) of the output pulses of one-vibrator 14 (15,16). These circumstances make it possible to perform a comparator and a single vibrator according to the simplest scheme, combining them in one block.

A diagram of such a combined block 11, 14 (12, 15; 13, 16) is shown in FIG. 2. It consists of RS-flip-flop 36 (37, 38), to the input S of which the output voltage of the integrator 9 (10), the RC-circuit of the integrating

Type 39 (40, 41), which includes a resistor Rg and a capacitor C2 and the output of which is connected to the input R of flip-flop 36 (37.38) and a key 42 (43, 44) connected in parallel with the capacitor during the driving circuit. The control input of the key is connected in such a way that, with a single trigger state, the key would be open. If the key closes when the control input has an O signal, the control input is connected to a single trigger output, otherwise the control input must be connected to the zero output of the trigger, if available, or to the output of a logic inverter

trigger output if zero trigger output is not available. Output pulses of one-shot C Duration, 1 and C)

formed at the trigger output.

The operation of the circuit will be considered in relation to the measuring element 1.

When the output voltage of the integrator 35 increases in the first cycle of operation of the circuit reaches the trigger voltage of the trigger 36. The latter switches to the one state, which leads to the opening of the key 42 closed before that. As a result, the capacitor C2 starts charging time 39 resistor Ry.

When the rising voltage on the capacitor reaches the trigger level of trigger 36, the latter switches to the zero state, which causes the switch 42 to close and the capacitor to discharge quickly

C2. The circuit goes to its original, stable state.

As soon as the trigger switches to the single state, a fault current in the resistor Ra of the resistive adder 32 appears, the integrator output voltage begins to decrease and becomes less than the trigger trigger level.

Thus, the function of comparator 11 is performed by trigger 36, which is also part of the one-shot 14. The duration of the output pulses of the circuit, is determined by the expression

UCЈ,

1 & A - Rg C2 In (1

U

1 (45J

where Ucp is the trigger voltage of trigger 36 at input R;

U1 is the voltage G at the trigger output.

As already noted, the instability of the duration of the guide (bb, tnc), as well as the instability of the actuation trip on input S, does not change the output signal of the device.

Amplitude modulator 17 (18, 19, Fig. 1) consists of two keys connected in antiphase in series, to which (i.e., a Dona constant voltage is applied to the signal input of the amplitude modulator). If the voltage Uonr is positive, in order to simplify the circuit and increase its accuracy, the output stages of the CMOS digital integrated circuits can be used as such keys, which are two antiphase keys, the first voltage of which is supplied to the first contact, Its second contact is connected to the first contact of another key, forming the output of the circuit, and the second contact of this key is grounded. The functions of the signal input of such an amplitude modulator are made by the output to which the voltage supply of the integrated circuit is applied. The resistance of such switches in CMOS with increased load capacity, for example, in voltage repeaters (level converters), inverters is quite small and can be further reduced by using the parallel connection of several circuits. In the device, in each of the measuring elements, two connected in parallel along the inputs and outputs of the CMOS element of the level converter (voltage follower) are used as amplitude modulators, which contain six such elements with a common voltage Uon2, and the total The output of the integrated circuit is connected to the zero phase of the circuit being measured. Since the power leads are common to all elements, the diagram shows the connection of the signal input and the common point for only one amplitude modulator 17.

The galvanic separation of pulse signals block 20 (21, 22) converts the output pulses of one 0 of the vibrator 14 (15, 16) into the same duration pulses, galvanically isolated from the single vibrator circuit and formed relative to the common point of the output part of the circuit. It consists of two

5 series-connected logical inverters 45 (46, 47). the input of the first of which is connected to the output of a one-shot 14 (15, 16), a limiting resistor 48 (49, 50), a three-winding differentiating transformer 51 (52, 53), two diodes 63, 64 (65, 67, 68) and an output RS -trigger 69 (70. 71). The output of the matching unit produces two antiphase pulse voltage signals,

5 forming a current in the circuit of the series-connected limiting resistor and the primary winding 54 (54, 56) of the differentiating transformer. At the time of the beginning and end of the output impulse of the one-way vibrator, the direction of the current in the primary winding circuit changes and a fast reversal of the small differentiating transformer occurs, which leads to appearance

5 magnetization reversal in two oppositely connected secondary windings 57, 60 (58, 59, 61, 62) of two short opposite-polarity pulses. Positive-polarity pulses pass through diodes 63-68,

0 arriving at the inputs of the RS-flip-flop 69 (70, 71).

The windings of the differentiating transformer are turned on in such a way that the pulse at the input S of the output flip-flop appears at the moment the output pulse of the one-shot begins. the pulse at the input R is at the moment of its termination. As a result, the positive pulses at the single output of the output trigger have the same duration tMA (ti8. TMc). as well as the output pulses of the one-shot, thereby ensuring the independence of the output signal of the device from the pulse duration of the one-shot.

5 Converter 23 (24, 25) current contains (Fig. 2) three-winding transformer 72 (73. 74) with primary winding 75 (7,6, 77) connected to the circuit of the phase current 1d (1c, 1c). winding78 (79.80) feedback and output winding 81 (82, 83), galvanically isolated from the primary amplifier AC 84 (85, 86) and the load resistor 87 (88, 89). The output winding of the transformer is connected to the input of the amplifier, its output is connected to the feedback winding 78 (79, 80) and output winding 81 (82, 83), galvanically isolated from the primary, the amplifier 84 (85:86) AC and the load resistor 87 (88, 89). The output winding of the transformer is connected to the input of the amplifier, its output is connected to the feedback winding, in series with which the load resistor and the moving contact of the toggle key 26 (27, 28) are connected.

The current converter 23 operates as follows. The input current 1d, flowed through the primary winding 75 with the number of turns Wi, creates a voltage field A Wi, which creates a variable magnetic flux Φ in the core. This flux induces a variable emf in all windings of the transformer, including the output emf of the output winding, amplified by an amplifier 84, which forms the variable component UAwero of the output voltage UAI. The voltage UAI creates in the winding 78 of the feedback with the number of turns L / 2, the load resistor 87 having the resistance R, and the contacts of the toggle key 26 closed at the moment, the variable component 1d1l, the current IAL of the Leak 78 The field length L / 2, which, with the appropriate switching on of the winding 78, is directed oppositely to the intensity Hi. Resultant stress

Hp Hi-H2 (46)

Ultimately, in the magnetic core, the resulting variable magnetic flux Fr generates an emf in the output winding.

With a sufficiently large gain Kus amplifier 84 EMF output winding 81, is necessary to obtain any desired voltage UAI, is negligible, indicating the smallness of the resulting variable magnetic flux FR in the core, and hence the smallness of the voltage Hp, creating this flow (46). With large Kus, from (46) we obtain

Wi

lA1i. lA KllA.

(47)

Therefore, at a sufficiently large gain factor of the amplifier 84, the current IAI-H flowing in the feedback winding circuit 78 is directly proportional to

The input current is 1d in magnitude, coincides with it in phase and does not depend on the resistance of this circuit, in particular, the resistance of the load resistor 87, the active resistance of the winding 78 and the resistance Hcl of the currently closed contacts

cross over key 26,

From comparison (47) with (15) it follows that

Kl W2 const

(48)

The voltage UAI at the output of the amplifier 84 generally contains the constant UAO and the variable UAI components, i.e.

UAI UAO + UAI. (49)

Voltage UAO and creates a constant component of IAO current IAI (14),

As can be seen from the diagram, the current d1 and the voltage UAI are related by the relation

- i

"AiOVVkiV

lo olkkt a. (50)

From (36) and (50) finds

 ha + hl, (51)

It is this K2 that should be substituted into formula (38).

The first and second summing inverting amplifiers 5, 6 of the direct current, along with the feedback resistor 92, 93, include capacitors 94-97, through which the amplifiers pre-filter the resulting input current, highlighting its average for the period of high-frequency output voltage. odinovibrator value. This prevents possible overloading of the operational amplifiers 90 and 91 to the current and voltage that can occur if the current pulses coincide in time from the different measuring elements.

The device has the following advantages.

As can be seen from the expressions (44) and (ЗЗа) / proportionality coefficient K, on which the output signal of the converter depends, is determined only by stable resistances of resistors, stable breakdown voltage Uon2 of precision Zener diode, DC gain of the filter, as well as turns of the windings of the transformer included in the current transducers, and does not depend on the capacitance of the capacitors, whose parameters can vary significantly during operation.

The toggle switches that commute the output current of the current transducers are covered by feedback and the output signal of the converter does not depend on their unstable residual resistance, as well as on the unstable active resistance of the feedback winding.

Due to the introduction of the second summing inverting DC amplifier, the parasitic signals due to the switching emissions of the toggle switches are mutually mutually compensated, which significantly (by an order of magnitude and more) increases the switching frequency determined by the frequency of the single-pulse output pulses. It. in turn, it contributes to the reduction of the methodological error inherent in the modulation power converters and due to the discretization of continuous input signals. In addition, the converter may be used to measure the power at an increased frequency of the input signals.

Methodical error is reduced also because the multiplication of input signals is carried out not only during the one-shot pulses. but also during pauses, when a reversible key closes the second fixed (lower) contact, i.e. The converter both current and voltage input signals are processed continuously.

The advantage of the converter is its insensitivity to the constant component of the output current of the current converters, and when using the proposed source circuit of different polar reference DC voltages and fulfilling conditions (29a), (31), the need for compensating resistors 29-31 is eliminated. The condition (31), which requires that the resistance of one of the resistors of the resistance adder be twice the resistance of the second, can be easily and with high accuracy in a wide temperature range. If conditions (29) (31) cannot be fulfilled, i.e. One of the inequalities (35) or (39) is fulfilled, insensitivity to the constant component of the output current of the current converters can be ensured by means of compensating resistors 29-31, the required resistance of which is determined by the expression (38). Since the value of K2, as can be seen from (51), depends on the unstable resistances Ha and r ™, full compensation in a wide temperature range cannot be ensured. Stabilization is possible, but requires a certain complexity of the scheme.

The proposed scheme formed the basis of the active measuring transducer.

Claims (7)

1. A measuring transducer of active power, containing the first and second measuring elements, the output DC filter, the first summing inverting DC amplifier, the reference voltage source, the first inputs of the measuring elements being connected to the corresponding phase voltages, and their second inputs - with the first output of the source of the reference voltage, the third inputs of the measuring elements are connected to current sensors of the corresponding phase, the combined outputs of the first and second measuring The x elements are connected to the input of the first DC inverting amplifier, characterized in that, in order to improve the accuracy, a third measuring element is inserted into the device, the second DC inverting amplifier, a large-scale resistor, and a reference voltage source are made two-pole. its second output is connected to the combined fourth inputs of all measuring elements, the first input of the third measuring element is connected to the third phase voltage, its second input to the first input of the reference voltage source, and the third to the output of the third phase current sensor, the first output of the third the measuring element is connected to the first outputs of the first and second measuring elements, the combined second outputs of all measuring elements are connected to the input of the first or second summing inverting silica gels, the third outputs of all measuring elements are connected to the input of the second summing inverting dc amplifier, the output of which is connected via a scale resistor to the input of the first summing inverting dc amplifier connected to the output of the output component of the constant component. 2. The transducer pop. 1, which differs from the fact that the measuring element contains a current transducer whose input is the third input of the measuring element, a resistive adder with three inputs, a DC integrator, a comparator, a single vibrator, an amplitude modulator, a galvanic separation unit of pulse signals, a reversible switch and a compensating resistor, with the first input of the resistive adder being the first input of the measuring element, the second input the second input of the measuring element, the third input An output with an amplitude modulator output, and an output with a DC integrator input, the output of which is connected to the comparator input, the signal input of the amplitude modulator is the fourth input of the measuring element, and the control input is connected to the one-shot output, whose input is connected to the output the comparator, and the output - with the input of the galvanic separation of pulse signals, the output of which is connected to the control input of the reversing key, the movable contact of which is connected to the output of the current converter, first stationary contact tumbler key is the first output of the measuring element and the second stationary contact tumbler key - the third output of the measuring element, the second output transformation ers compensating current through the resistor connected to the second output of the measuring element. 3. The converter according to claim 1, wherein the reference voltage and the fact that the reference source contains a parametric dc voltage regulator, the output of which is the first output of the reference voltage source, and connected to its the output is an inverting amplifier, the output of which is the second output of the reference voltage source. 4. The converter according to claim 2, characterized in that in the measuring element the resistance of the resistor included in the resistive adder and connected to its second input is equal to twice the product of the resistance of the resistor of the adder connected to it the third input, the ratio of the voltages at the first output of the voltage source to the voltage at its second output, and the resistance of the compensating resistor is assumed to be infinity. 5. The converter according to claim 2, characterized in that the comparator and the one-shot comprise an inverter, an RS trigger, the 5th input of which is connected to the output of the DC integrator, and an integrating type RC circuit, the input of which is connected to the single output of the trigger, and its output is connected to the R-input of the trigger, and in parallel with the capacitor of the RC circuit, a bit switch is connected, the control input of which is connected to the output of the inverter, the input of which is connected to the output of the RS-flip-flop. 6. The converter according to claim 2, characterized in that the amplitude modulator contains a CMOS element of the logical voltage repeater, the power input of which is a signal input of the amplitude modulator, a common point connected to the common output of the source of the reference voltage, the input of the CMOS logic element the repeater is the control input of the amplitude modulator, and the output of the amplitude modulator. 7. The converter according to claim 2, characterized in that the galvanic separation unit of pulse signals contains a logical inverter, a three-winding differentiating transformer, a limiting resistor, two diodes and an RS output trigger, and the input of the galvanic separation unit of pulse signals is connected to the first output of the limiting resistor and the input of the inverter, the primary winding of the differentiating transformer is connected to the second output with the second output of the limiting resistor, and the second output to the output About the inverter, two oppositely connected secondary windings are connected to one common point with the common point of the output part of the device, diodes anodes are connected to their other two terminals, the cathodes of which are connected to the corresponding inputs of the output RS-flip-flop. Jm