SI21003A - Integrated position resolver for hybrid synchronous power drives - Google Patents

Abstract

The submitted invention represents a new considerably improved integrated position resolver for hybrid synchronous power drives, where the magnetic assembly of the electric drive represents the actual magnetic resolver. The solution is achieved by the application of an additional measurement coil (1) inside the drive and by a small modification of the control electronics. The measurement coil (1) measures the variation of the magnetic flux through the permanent magnet (4), while the control electronics also takes care for a mutual phase lagging of PWR pulses in the subsequent drive phases.

Description

Andrej Detela and Uros Platishe

Integrated positional resolver for hybrid synchronous electrical machines

The present invention describes a new method of position control for synchronous electric motors. This new method combines the advantages of resolver and sensorless methods.

Modern electric motors are usually equipped with an additional precision guidance system. One way to achieve this is to use an external position sensor (optical encoder, magnetic resolver, etc.). Another method that is usually more appealing is positional control without the position sensor (ie sensorless or non-sensor method).

Many methods of non-sensory observation of rotor position in synchronous electric motors are already known. Such motors do not require an optical encoder or resolver or other sensor to determine the rotor position. In such non-sensor systems, it is common to determine the rotor position from the electrical voltage in the motor windings. One part of this voltage is the Ohm voltage, which is proportional to the ohmic resistance of the coil and the electrical current in the coil. The second part of this voltage, however, is the voltage due to magnetic induction in the motor winding and is usually greater than the first. This induced voltage is a complex mathematical function of several variables and also includes the position of the impeller. By using powerful Digital Signal Processor (DSP) it is possible to determine the rotor position from the known values of the other variables. These variables are the electrical currents in the windings of all motor phases (eg three currents in three-phase motors) and the voltages in these windings (in the three-phase motor these are three variables again).

This method, however, has at least one drawback. The position of the rotor cannot be determined when the rotor is stationary, since no voltage is induced at zero speed. Therefore, for known non-sensory methods, the rotor must be moved initially (by about ± 60 electrical degrees) and then its initial position is performed. However, this is not in line with soft start, which is one of the basic requirements of many precision applications (such as in robotics, FA, etc.).

On the other hand, we know the position control with sensors. Solutions with traditional resolvers add massive parts to the engine. Often, magnetic resolvers also interfere with motors magnetically, which is a problem for many applications.

This patent offers a solution with a slight change to the construction of a hybrid synchronous motor so that the motor and the magnetic resolver are combined (integrated) together. This change should not introduce any heavy, expensive or complicated parts into the engine. We are particularly interested in such solutions where there are no significant structural changes.

The present invention is a motor which, in comparison with non-sensor control motors, is also of low mass and can be read accurately even when the engine is stationary.

This problem was successfully solved with a small construction additive in a hybrid synchronous motor. Other motor windings have a special measuring coil added. The second part of the solution is the method of generating PWM (Pulse Width Modulation) pulses in the polyphase winding of the motor. Compared to conventional synchronous hybrid sensorless motors, the motor then has a patent-pending measuring coil, special PWM generators and a decoder that calculates the rotor position from the induced voltage in the measuring coil. Synchronous phase regulation is performed by the DSP by accurately determining for each rotor position the appropriate PWM pulse width at all engine stages.

This new method of positional control is the object of the present invention. The essence of the method and the operation of positional control will be explained in the case of hybrid synchronous

-3 engine from patent no. P-200000004 oz. PC / JP01 / 00070 (Synchronous hybrid electric machine with toroidal winding). Of course, this method can be used with other motors in the class of hybrid synchronous electric machines. Hereinafter referred to as PC / JP01 / 00070, the engine will be referred to as a Mukade engine.

Positive control according to the present patent operates at an unmodified resolution even at zero speed, but retains all the advantages of non-sensor methods. The only accessory inside the engine is a small and very simple measuring coil. In addition, there are no other spare parts in the engine.

The invention will now be further explained by means of figures illustrating examples of a machine with an integrated position resolver and electronic functional blocks of the present invention.

• Figure 1 is an axonometric view of a three-phase hybrid synchronous electric machine with a measuring coil of an integrated position resolver according to the invention, in partial section.

• Figure 2 represents the same as Figure 1 with a view of the disassembled parts so that the rotor and the stator are separately visible.

• Figure 3 shows the time patterns of PWM pulses in three phases (U, V, W) of a three-phase synchronous motor with an integrated position resolver according to the invention.

• Figure 4 shows a block diagram of a special PWM generator for the three-phase system of the invention.

• Figure 5 shows a block diagram of a rotor position decoder according to the invention.

The invention will be exemplified by the example of a three-phase synchronous motor with toroidal winding (that is, a Mukade type motor), but the same theory applies to other polyphase hybrid synchronous motors.

-4Figure 1 shows the construction of a three-phase hybrid synchronous electric machine with storoid winding and with the measuring winding of the integrated position resolver of the invention. The same construction is also shown in Figure 2, except that in Figure 2 the rotor (15) and the stator (14) with the ball bearing (8) are shown separately. The active parts of the impeller are four serrated iron rings (9, 10, 11, 12). The active parts of the stator are two serrated iron rings (2, 3), an annular permanent magnet (4) inserted between these iron rings (2, 3) and a toroidal coil (6) wound on the stator ring assembly (2,3,4). According to the invention, a simple measuring winding (1) of an integrated position resolver is wound on the annular permanent magnet (4) so that the winding (1) is coaxial with the motor axis (7). The measuring coil (1) can be made of very thin wire, so it does not take up much space and does not significantly alter the dimensions of other parts of the engine. Both ends (1a, 1b) of the measuring coil (1) are driven from the motor through the stator housing (5) and are connected to the electronics, namely to the rotor position decoder.

The engine design described above is already known, with the exception of the measuring winding (1) of the integrated position resolver. Therefore, only a description of the operation of the measuring coil (1), which is the subject of the invention, will be presented here.

The measuring coil (1) is wound on a ring permanent magnet (4), so it measures changes in the magnetic flux through the ring permanent magnet (4). During the operation of the motor, the magnetic flux through the magnet (4) is constantly changing. These changes are small but measurable. They can be measured e.g. with measuring winding (1). The change in magnetic flux is practically zero if the operating currents in the motor polyphase (6) are well synchronized with the rotor position. The change in flow is, however, increased as soon as the electrical phase of synchronization escapes from the ideal position. According to the mathematical theory of Mukade-type engines and similar hybrid synchronous motors, this flux change is:

Δφ = KI ₀ snry (1) where:

• Δφ is a small change in the magnetic flux through a permanent magnet (4) (relative to the average value) • K is a multiplication constant that takes a value of approximately §.224> _m / I _omax • </ _i > _m average magnetic flux through a permanent magnet ( 4), at the operating point of the magnet • I _o is the amplitude of the electric current in each phase of the winding (6) of the motor • Iomax ^{is the} upper amplitude of this current, at the peak torque of the motor • 7 is the phase offset (electric angle) of the electric current in the main winding ( 6) the motor, according to the phase delay at which the operating current is best synchronized with the position of the rotor.

The above formula applies to the working current in the motor windings (6). This current must be synchronized with the rotor position, but the formula also applies if the current has a different frequency and is not necessarily synchronized with the rotor position. The only condition is that this current is as much a three-phase current as the working current in the winding (6). If the electric frequency does not correspond to the synchronous motor frequency, then the 7 variable is time dependent.

According to the invention, a three-phase power generator (electronic inverter) produces such a signal that the electric current in the three-phase winding system (6) is the sum of two different three-phase currents:

• The first electrical current is the operating current (the main current that produces the torque of the motor), so it has a significant amplitude that is approximately proportional to the torque. The electrical phase of this three-phase current is synchronized with the position of the rotor, so the frequency of this current corresponds to the synchronous frequency of the motor.

• The second current has a much higher frequency but a much smaller amplitude. Otherwise, this is also a three-phase current: the electrical oscillations in the adjacent phases U, V, W are separated by a phase distance of 120 electrical degrees. This second current is called the measuring current.

This second current is out of sync with the rotor position, so the phase delay 7 of the measuring current is constantly changing. This phase shift is the difference between the synchronous phase of the workflow ωί and the phase of the measurement flow ω'ί. So the following applies:

= 70 - (ω '- (2) where:

• ω is the synchronous circular frequency of a workflow defined by:

_ d

Kr * o. Fm Ot where is the mechanical angle of the rotor and K _{r the} number of poles (teeth) on the rotor.

• ω 'is the circular frequency of the measuring current • t is time and • 70 is the phase shift at time t = 0.

When we combine equations (1) and (2), we get a small change in magnetic flux due to the action of the measuring current - so Io is now the amplitude of the measuring current:

Δ, φ = -KIo sin [(u / - u /) f - 70] (3)

-7This is a magnetic flux through a permanent magnet (4), so it also passes through the measuring coil (1). This same flow goes through the stator poles of all three phases (U, V, W), so it contains information about all phases. The induced voltage U in the measuring coil (1) is, according to the law of magnetic induction and according to equations (1), (2) and (3):

o

Ui = Ν'— Δψ = -KN 'I _o (o / - ω) · cos [(w' - ω) ί - γ ₀ ] (4)

We used the following tags here:

• U is the induced voltage in the measuring coil • N 'is the number of wrappers in the measuring coil • I _o is the amplitude of the measuring current in each of the three phases • everything else has the same meaning as before

The electrical phase of the induced voltage U _t in the measuring coil is denoted by φ 'and is given by equation (4):

ψ '= [( ^ω ' - Ji ~ 7ο] (5)

The phase lag between the measuring current in the working coil (6) and the induced voltage in the measuring coil (1) is:

= ω'ί - [(u / - ω) ί - 7ο] - [ut + 70] (6)

Of course, the phase of the measuring current depends on which of the three phases U, V, W is compared. So the phase shift 70 also depends on which of the three phases we compare it with. In terms of theoretical argument, assume that we have an infinite number of stages, or at least many more than three or five. Then we can always find such a winding segment that corresponds to a particular phase of the motor and in which we get 70 = 0. Let's call this selected winding segment a zero segment. For example, this is the segment corresponding to the winding in phase U. So the phase lag between the measuring current in this zero segment and the induced voltage in the measuring coil is:

Δ92 = uit (7)

More generally, ω is time dependent (ω (ί)), so the product (ut) must be replaced by the integral:

Δφ = /

Combining Equations (5) and (6), we get:

Δ92 = ω'ί - φ '(8)

Now, remember that ut from equation (7) is a synchronous electrical phase, ideally when the synchronization is complete. At a time when the rotor rotates by one rotor pole, the operating current describes one complete electrical cycle. During this time, the synchronous phase ut rises exactly by the full electric angle 2π. This means that ut is also a product of the mechanical angle of the rotor and the number of rotor poles:

Δφ = ut = K _r p _m (9) where tp _{m is the} mechanical angle of the rotor. Equation (9) implies a linear relationship between the mechanical phase φ _η and the phase delay Δ </ ?.

The rotor position information is therefore carried by the phase Δ </?. During one engine cycle, the phase shift Δφ increases linearly from 0 to 2π. This is also the phase of the polyphase workflow, which also increases linearly from 0 to 2π and supplies the polyphase winding in the sequence U, V, W. Equation (8) provides the algorithm by which the electronics calculate this phase.

Mathematical analysis of a Mukade-type motor shows that the amplitude of the induced voltage Ui is in the following relation with the amplitude of that voltage induced in the work winding due to the measuring current:

Up N 8α \ ω '(10) where:

• U _io is the voltage amplitude induced in the measuring coil • U _po is the voltage amplitude induced by the working coil due to the measuring current • N 'is the number of wrappers in the measuring coil • N is the number of wraps in one segment of a Mukade-type motor • the coefficient is determined by the geometry of the rotor and stator poles and the width of the air gap; a typical value is 0.033 • ω is the circular frequency of the synchronous workflow • ω ¹ is the circular frequency of the measuring current

An integral part of the patent is also the method by which a single inverter generates a polyphase workflow (typically three-phase) and a polyphase measurement current (typically three-phase again) at the same time. Typically, a workflow sine signal is constructed from PWM pulses of a specific pulse frequency. For example, if this frequency is 17 kHz, then one pulse train period is about 60 microseconds. The PWM pulses in the coil belonging to phase U represent the first train of pulses. According to the invention, the PWM pulses in the phase V winding are phase shifted by one third of the period relative to phase U pulses, which in our case is 20 microseconds. The same is true for phase W, where the pulses are phase-shifted

-1010 for two thirds of the phase U period (40 microseconds in our case). The time patterns of PWM pulses in all three windings (U, V, W) of the three-phase synchronous motor according to the invention are shown in Figure 3 (PWM phase modulation).

The described method of displacement of the PWM pulse centers, separately at each stage, leads to the generation of a new rotating magnetic field in the stator winding (6) of the motor, with a circular frequency ω '.

In each phase (U, V, W), we have a sine workflow with a synchronous circular frequency ω. This workflow consists of PWM pulses whose centers are spaced at a circular frequency ω '. In the example above, ω '= 2vr · 17 kHz = 1.07 · 10 ⁵ .s ^_1 .

The above description was given for the three-phase system. We define this method more generally for any polyphase system. Define an n-phase system with n workflows / j, where j = 0 ... (n - 1). Each Ij current is generated by a PWM generator that can phase shift the PWM pulse centers for a particular phase individually. Define the phase offset of the center PWM of the pulse Cj belonging to phase j with respect to any reference position, which is arbitrary. According to the invention, the following applies:

where k = 0 ... (n - 1). The relationship between the mechanical phase and the induced voltage phase in the measuring coil is linear in the special case where:

which is the default for the three-phase motor described above.

In a sufficiently short time interval, the successive pulses are approximately the same, so the pulse train can be approximated by a periodic function having a circular frequency ω '. This periodic function is expressed by Fourier's evolution. The circular frequency ω 'then corresponds exactly to the circular frequency of the first harmonic component in this development. So there they are

-1111 in phases U, V and W, these first harmonic components are phase shifted by one third of the period. With respect to the first harmonic component in phase U, the first harmonic component in phase V is phase shifted by one third of the period (20 microseconds in our case) and the first harmonic component in phase V is phase shifted by two thirds of the period (40 microseconds). This, however, is exactly what is needed for a three-phase current system. In this way, we can produce three-phase measuring current and three-phase working current with a single inverter. The three-phase measuring current is exactly the first harmonic component of pulse trains in phases U, V and W.

The block diagram of such an inverter for a three-phase system is shown in Figure 4. The clock produces a carrier circular frequency ω 'of the PWM signal, which is equal to 2π / _ρω ^. This clock controls three separate PWM generators through the dock input. From the outside, PWM blocks are synchronized through phase inputs. The variable value parameter determines the current PWM pulse width of each phase individually, thus creating a synchronous workflow.

PWM pulses can be represented as the sum of the first harmonic component with a circular frequency ω 'and higher harmonic components, as well as the low circular frequency ω of the workflow. The relative values of the higher harmonic components can be calculated by Fourier analysis. In practice, these higher harmonic components affect the accuracy of the measurement method described.

We just saw that the first harmonic component meets the rules of three-phase currents. Further analysis shows that the fourth, seventh, ... harmonic component also contributes three-phase currents, of course with the frequency of higher harmonic components. The second, fifth, ... harmonic component contributes three-phase currents, but with an inverted sequence of phases, namely W, V, U instead of U, V, W. The third, sixth, ... harmonic component does not contribute at all to the three-phase current, but simultaneous oscillations in all three phases, which almost cancel each other out in the adjacent segments of the working coils.

-1212

The following table shows the behavior of the higher harmonic components. For each harmonic component, we can see what kind of oscillations contribute to the three-phase system:

harmonic component type of oscillations l, 4, ... (3n + l) three-phase oscillations in the sequence U-V-W-U-V-W -... 2.5, ... (3n - 1) three phase oscillations in the sequence W-V-U-W-V-U ... 3.6, ... (3n) simultaneous oscillations that are magnetically near they nullify

An appropriate method of reducing the effects of unwanted harmonic components is to use a narrow band filter with a central circular frequency of about ω 'and a bandwidth of about 2ω.

Figure 5 shows the block diagram of the position decoder. The voltage from the measuring coil {Ui) is first run through a band-pass filter. Because a narrow band filter was used, signals with a circular frequency ω and higher harmonic components of the circular frequency ω 'are suppressed, and only the first harmonic component ω' is left in the digital phase detectors. The detector compares the phase of the first harmonic component with the reference phase / j, where j is arbitrary (for three-phase systems it may correspond to phase U, V or W).

The main advantage of such an integrated system is the low mass of the motor with the sensor and the high resolution of the rotor position. An important advantage of the present invention is that it gives an absolute position within one half of the rotor division. This means that the engine can run at full power and with full accuracy immediately after starting.

Claims (3)

Patent claims An integrated position resolver for hybrid synchronous electrical machines, characterized in that, in addition to the work winding (6), the hybrid synchronous motor has an additional measuring coil (1) that encloses the magnetic flux of a permanent magnet (4) which magnetizes magnetic fields simultaneously all stages of the motor (in the case of a three-phase motor, these are stages U, V, W). An integrated positional resolver for hybrid synchronous electrical machines according to claim 1, characterized in that the working windings (6) of the corresponding phases of the hybrid motor are fed by a high-frequency polyphase measuring current added to the polyphase operating current. An integrated position resolver for hybrid synchronous electrical machines according to claims 1 and 2, characterized in that the same inverter produces high-frequency polyphase measuring current and polyphase operating current by the relative phase shift of PWM pulses in successive phases.